Frequency Selective Surfaces Based High Performance Microstrip Antenna

SPRINGER BRIEFS IN ELECTRICAL AND COMPUTER
ENGINEERING  COMPUTATIONAL ELECTROMAGNETICS
Shiv Narayan
B. Sangeetha
Rakesh Mohan Jha
Frequency Selective
Surfaces based
High Performance
Microstrip Antenna
SpringerBriefs in Electrical and Computer
Engineering
Computational Electromagnetics
Series editor
Rakesh Mohan Jha, Bangalore, India
More information about this series at http://www.springer.com/series/13885
Shiv Narayan B. Sangeetha
•
Rakesh Mohan Jha
Frequency Selective Surfaces

based High Performance
Microstrip Antenna
123
Shiv Narayan Rakesh Mohan Jha
Centre for Electromagnetics Centre for Electromagnetics
CSIR-National Aerospace Laboratories CSIR-National Aerospace Laboratories
Bangalore, Karnataka Bangalore, Karnataka
India India
B. Sangeetha
Centre for Electromagnetics
CSIR-National Aerospace Laboratories
Bangalore, Karnataka
India
ISSN 2191-8112 ISSN 2191-8120 (electronic)

SpringerBriefs in Electrical and Computer Engineering
ISSN 2365-6239 ISSN 2365-6247 (electronic)
SpringerBriefs in Computational Electromagnetics
ISBN 978-981-287-774-1 ISBN 978-981-287-775-8 (eBook)
DOI 10.1007/978-981-287-775-8
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Dedicated to Dr. Sudhakar Rao
In Memory of Dr. Rakesh Mohan Jha
Great scientist, mentor, and excellent
human being
Dr. Rakesh Mohan Jha was a brilliant contributor to science, a won-

derful human being, and a great mentor and friend to all of us
associated with this book. With a heavy heart we mourn his sudden
and untimely demise and dedicate this book to his memory.
Preface
Frequency selective surface (FSS) technology has been widely used for the design
of high-performance radomes, antennas, radar absorbing structure, reflectors, etc.,
during the past four decades. In such applications, the FSS technology is mainly
employed to enhance the performance of the candidate device/structure, and to
reduce their radar signature.
High-performance low RCS (radar cross section) printed antennas are mostly
preferred in stealth technology. Such printed antennas may be realized by incor-
porating FSS structures, either in its ground plane or as superstrate. In view of this,
the design and analysis of microstrip patch antennas loaded with FSS-based (i) high
impedance surface (HIS) ground plane and (ii) superstrate are presented in this
book.
This brief is organized as follows: Section 1 deals with the introduction of FSS
structure and Sect. 2 describes the characteristics of FSS structures. The design and
analysis of microstrip antenna loaded with FSS-based HIS is discussed in Sect. 3; in
this section, various types of band-stop FSS structures such as Jerusalem cross and
single-square loop are designed to perform as perfect magnetic conductor (PMC),
which is then used as ground plane of microstrip patch antenna (MPA). Further, the
design and analysis of MPA loaded with the superstrate design, using double square
loop-FSS, is studied for directivity enhancement in Sect. 4. Finally, Sect. 5 lists the
conclusions of the work carried out in the book.
Shiv Narayan
B. Sangeetha
Rakesh Mohan Jha
ix
Acknowledgments
We would like to thank Mr. Shyam Chetty, Director, CSIR-National Aerospace

Laboratories, Bangalore for his permission and support to write this SpringerBrief.
We would also like to acknowledge valuable suggestions from our colleagues at
the Centre for Electromagnetics, Dr. R.U. Nair, Dr. Hema Singh, Dr. Balamati
Choudhury, and Mr. K.S. Venu during the course of writing this book. We express
our sincere thanks to Ms. Nimisha S., Ms. Divya K.M., and Ms. Sai Samhitha, the
project staff at the Centre for Electromagnetics, for their consistent support during
the preparation of this book.
But for the concerted support and encouragement from Springer, especially the
efforts of Suvira Srivastav, Associate Director, and Swati Meherishi, Senior Editor,
Applied Sciences and Engineering, it would not have been possible to bring out this
book within such a short span of time. We very much appreciate the continued
support by Ms. Kamiya Khatter and Ms. Aparajita Singh of Springer toward
bringing out this brief.
xi
Contents
Frequency Selective Surfaces-Based High Performance

Microstrip Antenna . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ...... 1
1 Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ...... 1
2 Characteristics of FSS Structures . . . . . . . . . . . . . . . . . . . . . ...... 3
3 Microstrip Antenna Over FSS-Based High Impedance
Ground Plane . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4
3.1 Theoretical Considerations. . . . . . . . . . . . . . . . . . . . . . . . . . . . 5
3.2 EM Design of Microstrip Patch Antenna Over FSS-HIS . . . . . . . 11
3.3 EM Performance Analysis . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12
4 Microstrip Antenna Loaded with FSS-Based Superstrate . . . . . . . . . . . 23
4.1 Theoretical Considerations. . . . . . . . . . . . . . . . . . . . . . . . . . . . 24
4.2 Estimation of Far-Field Radiation Pattern of Antenna . . . . . . . . . 28
4.3 EM Design of Microstrip Patch Antenna Loaded
with FSS Superstrate . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30
4.4 EM Performance Analysis . . . . . . . . . . . . . . . . . . . . . . . . . . . . 33
5 Summary . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 38
References . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 38
About the Book . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 41
Author Index . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 43
Subject Index . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 45
xiii
About the Authors
Dr. Shiv Narayan is with the Centre for Electromagnetics of CSIR-National

Aerospace Laboratories (CSIR-NAL), Bangalore, India as Scientist, since 2008. He
received his Ph.D. in Electronics Engineering from Indian Institute of Technology,
Banaras Hindu University (IIT-BHU), Varanasi, India in 2006. He held the position
of Scientist B between 2007 and 2008, at SAMEER (Society for Applied
Microwave Electronics Engineering and Research), Kolkata, India. His research
interests are broadly in the field of electromagnetics applications; these include
frequency selective surfaces (FSS), metamaterials, numerical methods in electro-
magnetics, EM material characterization, and antennas. Dr. Shiv is the
author/co-author of over 40 technical documents including peer reviewed journal
and conference papers.
Ms. B. Sangeetha obtained her B.E. (ECE) degree from Visvesvaraya
Technological University, Karnataka, India. She is currently a Project Engineer
with the Centre for Electromagnetics of CSIR-National Aerospace Laboratories
(CSIR-NAL), Bangalore, India where she works in the areas of FSS, metamaterials,
and microstrip antennas.
Dr. Rakesh Mohan Jha was Chief Scientist & Head, Centre for Electromagnetics,
CSIR-National Aerospace Laboratories, Bangalore. Dr. Jha obtained a dual degree
in BE (Hons.) EEE and M.Sc. (Hons.) Physics from BITS, Pilani (Raj.) India, in
1982. He obtained his Ph.D. (Engg.) degree from Department of Aerospace
Engineering of Indian Institute of Science, Bangalore in 1989, in the area of
computational electromagnetics for aerospace applications. Dr. Jha was a SERC
(UK) Visiting Post-Doctoral Research Fellow at University of Oxford, Department
of Engineering Science in 1991. He worked as an Alexander von Humboldt Fellow
at the Institute for High-Frequency Techniques and Electronics of the University of
Karlsruhe, Germany (1992–1993, 1997). He was awarded the Sir C.V. Raman
Award for Aerospace Engineering for the Year 1999. Dr. Jha was elected Fellow of
xv
xvi About the Authors
INAE in 2010, for his contributions to the EM Applications to Aerospace

Engineering. He was also the Fellow of IETE and Distinguished Fellow of ICCES.
Dr. Jha has authored or co-authored several books, and more than five hundred
scientific research papers and technical reports. He passed away during the pro-
duction of this book of a cardiac arrest.
Abbreviations
AMC Artificial magnetic conductor

DSL-FSS Double square loop-frequency selective surface
EBG Electromagnetic band gap
ECM Equivalent circuit model
EM Electromagnetics
EMI Electromagnetic interference
FSS Frequency selective surfaces
HIS High impedance surfaces
IWO Invasive weed optimization
JC-FSS Jerusalem crossed FSS
JC-HIS Jerusalem cross-based HIS
MPA Microstrip patch antenna
PEC Perfect electric conductor
PMC Perfect magnetic conductor
RCS Radar cross section
TE Transverse electric
TM Transverse magnetic
xvii
Symbols
ε0 Permittivity of free space

εr Relative permittivity of antenna substrate
η Intrinsic impedance
λ Wavelength
µ0 Permeability of free-space
µr Relative permeability
ΓTE, ΓTM Reflection coefficient for TE and TM mode of incidence
ωr Angular resonance frequency
B(g, t) Capacitive susceptance
BTE, BTM Capacitive susceptance for TE and TM mode of incidence
Ca Capacitance of patch antenna
C1s Capacitance of outer square loop
C2s Capacitance of inner square loop
d Thickness of HIS substrate
ds Thickness of FSS superstrate
fr Resonant frequency
g Separation between adjacent JC-crosses
h Height of microstrip antenna
k0 Wave number
L1 Length of microstrip antenna
La Inductance of antenna
L1s Inductance of outer square loop
L2s Inductance of inner square loop
lg Length of JC-FSS inductive grid
Lg Grid inductance of JC-FSS
N(θ) Refractive index
Q Quality factor of patch antenna
R Resistance of the patch antenna
RD Resistance of the FSS
w Width of inductive grid of JC-FSS
wc Width of capacitive grid of JC-FSS
xix
xx Symbols
W1 Width of patch antenna

Wg Length of capacitive grid JC-FSS
X Grid reactance of JC-FSS
XTE, XTM Inductive reactance for TE and TM mode of incidence
Z0 Characteristic impedance of free-space
Za Input impedance of antenna
Zd Input impedance of grounded dielectric slab
Zs Input impedance of FSS-HIS
ZFSS Input impedance of FSS
List of Figures
Figure 1 Typical FSS types and their frequency response

characteristics; a array of metallic patches shows low-pass
behavior, b array of apertures on conducting screen shows
high-pass behavior, c array of metallic loops shows
band-stop behavior, d array of aperture loops on conducting
screen shows band-pass behavior . . . . . . . . . . . . . . . . . . . .. 4
Figure 2 a Schematic of rectangular microstrip patch antenna over
FSS-based HIS, b unit cell of Jerusalem cross FSS. . . . . . . .. 6
Figure 3 a Equivalent circuit of the rectangular microstrip patch
antenna, b equivalent circuit of the Jerusalem cross FSS,
c equivalent circuit of the FSS-based HIS . . . . . . . . . . . . . .. 7
Figure 4 Input impedance of rectangular microstrip patch antenna.
Red lines show computed results at CEM, CSIR-NAL. Blue
lines show reported results (Volakis 2007) . . . . . . . . . . . . .. 12
Figure 5 a Input impedance of rectangular microstrip patch antenna
designed at 10 GHz. b Return loss of rectangular MPA
versus frequency (designed at 10 GHz) . . . . . . . . . . . . . . . .. 13
Figure 6 Reflection phase of the JC-FSS-based HIS. Solid blue line
shows computed result at CEM based on equivalent circuit
model. Dotted red line shows reported result based on
numerical model (Hosseinipanah and Wu 2009) . . . . . . . . . .. 14
Figure 7 Real and imaginary parts of input impedance
of JC-FSS-based HIS . . . . . . . . . . . . . . . . . . . . . . . . . . . .. 14
Figure 8 Return loss of rectangular MPA microstrip antenna with
PEC ground plane and FSS-HIS ground plane designed at
5.8 GHz. Bullet points show reported results (Monavar and
Komjani 2011). Solid lines show computed results at
CEM . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .. 15
Figure 9 Return loss of the proposed MPA with PEC and FSS-based
HIS ground plane designed at 10 GHz . . . . . . . . . . . . . . . .. 16
xxi
xxii List of Figures
Figure 10 Return loss of rectangular MPA with FSS-HIS ground plane

for different length of the inductive grid (lg) of JC-FSS
element . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .. 17
Figure 11 Return loss of rectangular MPA with FSS-HIS ground plane
for different gap between the adjacent crosses (g) of FSS
element . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .. 17
Figure 12 E-plane radiation pattern of rectangular microstrip antenna
with PEC ground plane and JC-FSS-based HIS ground
plane . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .. 18
Figure 13 a Schematic of square loop FSS-based HIS, b unit cell of
SSL-FSS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .. 19
Figure 14 Equivalent circuit of single square loop FSS-based HIS . . . .. 19
Figure 15 Real and imaginary parts of input impedance of single
square loop FSS-based HIS . . . . . . . . . . . . . . . . . . . . . . . .. 20
Figure 16 Return loss of rectangular MPA with PEC ground plane and
SSL-FSS-based HIS ground plane . . . . . . . . . . . . . . . . . . .. 21
Figure 17 E-plane radiation pattern of rectangular MPA with PEC
ground plane and SSL-FSS-based HIS ground plane. . . . . . .. 21
Figure 18 Real and imaginary parts of input impedance of
SSL-FSS-based HIS (Teflon substrate) . . . . . . . . . . . . . . . .. 22
Figure 19 Return loss of rectangular MPA with PEC ground plane and
SSL-FSS-based HIS (Teflon substrate) ground plane . . . . . .. 22
Figure 20 E-plane radiation pattern of rectangular MPA with PEC
ground plane and SSL-FSS-based HIS (Teflon substrate)
ground plane . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .. 23
Figure 21 a Schematic of rectangular microstrip patch antenna loaded
with FSS superstrate, b geometry of unit cell of
DSL-FSS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .. 25
Figure 22 a Equivalent circuit of rectangular microstrip patch antenna,
b equivalent circuit of DSL-FSS . . . . . . . . . . . . . . . . . . . .. 26
Figure 23 Equivalent circuit of the antenna loaded with square loop
FSS-based superstrate . . . . . . . . . . . . . . . . . . . . . . . . . . . .. 27
Figure 24 Equivalent transmission line of MPA covered with FSS
superstrate . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .. 28
Figure 25 Transmission characteristics of DSL-FSS structure;
a reported (Lou etΓÇÖal. 2005), b computed at CEM . . . . .. 31
Figure 26 Transmission characteristics of proposed DSL-FSS structure
at different incidence angles (0°, 30°, and 45°) for TE
polarizations . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .. 32
Figure 27 Reflection characteristics of proposed DSL-FSS structure at
different incidence angles (0°, 30°, and 45°) for TE
polarizations . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .. 32
Figure 28 EM characteristics of MPA loaded with superstrate; a input
impedance, and b return loss . . . . . . . . . . . . . . . . . . . . . . .. 34
List of Figures xxiii
Figure 29 Validation of a E-plane, and b H-plane pattern of rectan-

gular MPA. Dotted black lines show reported results
(Volakis 2007). Solid blue lines show computed results at
CEM . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 35
Figure 30 Radiation pattern of rectangular MPA and MPA covered
with DSL-FSS superstrate; a E-plane and b H-plane. . . . . . . . 36
Figure 31 Radiation pattern of rectangular microstrip antenna and
MPA covered with FSS-based superstrate by keeping air gap
between them; a E-plane and b H-plane . . . . . . . . . . . . . . . . 37
Frequency Selective Surfaces-Based High
Performance Microstrip Antenna
Abstract In order to fulfill the growing demand high-performance low RCS

antenna in stealth technology, FSS-based antenna is found to be the better candi-
date. In view of this, the design and analysis of microstrip patch antennas
(MPA) loaded with (i) FSS-based HIS (high-impedance surface) ground plane and
(ii) FSS-based superstrate are presented in this book with proper formulations and
graphical presentations.
1 Introduction
Frequency selective surfaces (FSS) technology have potential applications in var-

ious sectors such as aerospace, medical, microwave industry, and real estate. In
medical sector, FSS is used to prevent electromagnetic interference (EMI) in
sophisticated medical instruments such as MRI, implantable medical devices, etc.
(Gupta and Gururaj 2013). In microwave industry, it is used to reduce the leakage
of power through oven door by incorporating the reflection-type FSS structure into
the walls of the oven cavity. In real estate sector, FSS is employed to facilitate better
wireless communications and to reduce EMI/EMC due to unwanted signals
(Philippakis et al. 2004). In aerospace sector, FSS is mainly employed to enhance
the performance and reduce the radar cross section (RCS) of various devices such
as radome, RAS, reflector antennas, antennas, microwave circuits, etc.
Recently, high performance low observable antennas have demanded applica-
tions in strategic area. This can be accomplished by incorporating the FSS structure
to the planar antenna, where FSS is employed either as superstrate or high
impedance ground plane of antenna. The incorporation of FSS to the antenna
mainly enhances its gain, bandwidth, and reduces its out-of-band RCS (Pirhadi et
al. 2012; Li et al. 2010). The planar antennas such as dipoles, microstrip patches,
etc., need a ground plane, which acts as a reflector to enhance the radiation gain.
However, the metallic ground plane is one of the most important scattering com-
ponents of the antenna, because it largely reflects the energy of incident waves
© The Author(s) 2016 1

S. Narayan et al., Frequency Selective Surfaces based High Performance
Microstrip Antenna, SpringerBriefs in Computational Electromagnetics,
DOI 10.1007/978-981-287-775-8_1
2 Frequency Selective Surfaces-Based High Performance …
outside the operating bandwidth. In order to reduce the scattering component

outside the operating bandwidth, the conventional ground plane can be replaced
with a stop-band FSS (Lu et al. 2009) as it works as perfect reflector within the band
and completely pass the signal outside the band.
In view of this, Lu et al. (2009) designed a low RCS antenna by replacing its
ground plane with novel stop-band FSS structure which works as high impedance
ground plane surface. Li et al. (2010) presented a low RCS and high gain
reflect-array antenna backed on FSS structure. The proposed antenna is reported to
exhibit “in-band” improvement of 1.1 dB, reduction in side-lobe level by 6.4 dB,
and significant reduction in out-of-band RCS. Further, a low profile low RCS array
antenna was designed by replacing its PEC ground plane with hybrid FSS structure
(Genovesi et al. 2012) without altering in-band radiation characteristics. The pro-
posed antenna structure was analyzed based on periodic method of moment.
Recently, a microstrip antenna backed by a novel hybrid ground plane consisting of
miniaturized FSS elements with partial metallic plane is presented for out-of-band
RCS reduction over wide frequency range 1–20 GHz (Yang et al. 2013) at oblique
angle of incidence. In order to enhance the bandwidth of the antenna, Yeo et al.
(2002) designed, a novel FSS-based electromagnetic bandgap (EBG) structure
using Genetic algorithm and used it as a ground plane of planar antenna to enhance
the bandwidth of the antenna. Further, the bandwidth of a microstrip antenna has
been enhanced by using Jerusalem crossed FSS (JC-FSS)-based artificial magnetic
conductor (AMC) ground plane (Monavar and Komjani 2011), where invasive
weed optimization (IWO) algorithm were implemented to optimize the FSS
structure and antenna for optimal performance. The proposed JC-FSS-based
antenna was analyzed using full-wave analysis method.
Another application of FSS is the design of superstrate for the antenna to
enhance its radiation characteristics. In this view, Lee et al. (2004) analyzed a
microstrip antenna covered with FSS-based superstrate to enhance the directivity of
the antenna. A significant enhancement in directivity was achieved from 17.29 to
24.92 dBi for three different unit cell dimensions of FSS as compared to the
conventional patch antenna. Pirhadi et al. (2007) demonstrated various types
of FSS-based superstrate designed by single layer and multilayered square loop
structure to enhance the directivity of dual-band EBG resonator antenna. Later, gain
enhancement of various types of antennas were presented in open domain using
FSS-based superstrate (Foroozesh and Shafai 2010; Chiu and Chen 2011; Pirhadi
et al. 2012).
In this brief, the design and analysis of microstrip patch antenna (MPA) is
presented loaded with (i) various types of FSS-based high impedance ground plane
and (ii) FSS-based superstrate. The analysis of composite antenna structure is
carried out using transmission line equivalent circuit model as it requires less
processing time and memory as compared to the methods based on full-wave
analysis. The details of this work have been discussed in the following sections.
2 Characteristics of FSS Structures 3
2 Characteristics of FSS Structures
The FSS is a periodical structure, which has specific reflection and transmission
characteristics for the electromagnetic waves passing through it. The FSS structures
resonate at a designed frequency and attain spectral selectivity (Loui 2006).
Basically, FSS structures can be divided into two categories based on element
geometry. The first type comprises of metallic patches on the substrate, which is
usually referred to as capacitive FSS. Such type of FSS structures exhibit low-pass
filter characteristics. The other type of FSS structure comprises of apertures on a
metallic screen, which is commonly known as an inductive FSS. This type of FSS
configurations shows high-pass filter characteristics. Generally, FSS has two major
applications in aerospace; one application of FSS is to design antenna radomes to
enhance transmission efficiency within the band and sharp roll-off characteristics
outside the operating band. Another is to employ FSS in the design of high per-
formance reflector antenna systems. Apart from these applications, FSS structure is
recently used to design artificial magnetic conductors such as HIS, EBG ground
plane, to enhance the radiation characteristics and reduce the structural RCS of the
antenna.
The transmission type FSS can further be divided into thick or thin, depending
on the thickness of metallic screen (Chen and Stanton 1991). If the thickness of the
FSS (metallic sheet) is less than 0.001λ, the FSS is considered as a “thin”filter,
which is modeled as infinitely thin in the numerical modeling. If the thickness of
metallic sheet is greater than 0.001λ, the FSS is modeled as a “thick” filter.
A thick-metal FSS finds applications, where mechanical strength and power han-
dling are important factors. Further, the FSS characteristics can be divided into four
categories namely; low-pass, high-pass, band-pass, and band-stop, based on their
frequency responses as shown in Fig. 1.
From Fig. 1a, b, it is evident that the low-pass and high-pass FSS structures are
complementary surfaces. It means that they cover the entire surface jointly
(Gustafsson et al. 2005). Moreover, the transmission and reflection properties of
low-pass and high-pass FSS structures are complimentary to each other as per
Babinate’s principle. This concept is also applicable to the band-pass and band-stop
FSS structure as shown in Fig. 1c, d. However, such type of complementary
relationship is applicable to only “thin FSS” structures that do not have dielectric
backing layer. With the dielectric backing, the resonance frequency of FSS structure
pffiffiffiffi
is shifted to lower side by an amount 1= er , where εr is the relative permittivity of
dielectric.
Fig. 1 Typical FSS types and (a) (b)

their frequency response
characteristics; a array of
metallic patches shows
low-pass behavior, b array of
apertures on conducting
screen shows high-pass
behavior, c array of metallic
loops shows band-stop
Power transmission
Power transmission
behavior, d array of aperture
loops on conducting screen
shows band-pass behavior
Frequency Frequency
(c) (d)
Power transmission
Power transmission
Frequency Frequency
3 Microstrip Antenna Over FSS-Based High Impedance

Ground Plane
The band-pass FSS is severally used as radome for antennas to enhance its per-
formance and reduce the RCS of the antenna (Narayan et al. 2012; Costa; and
Monarchio 2012). In contrast, stop-band FSS has been used as high impedance
ground plane in planar antennas to enhance its gain, bandwidth, and out-of-band
RCS reduction (Lu et al. 2009; Genovesi et al. 2012). Antennas such as dipoles,
microstrip patches, etc., need a ground plane, which works as a reflector to enhance
the radiation gain. But, the metallic ground plane is one of the most important
scattering components of the antenna because it largely reflects the energy of
incident waves. In order to reduce the scattering component and hence to enhance
3 Microstrip Antenna Over FSS-Based High Impedance Ground Plane 5
the radiation gain of the microstrip antenna, the conventional ground plane can be
replaced with a stop-band FSS.
Further, it was reported that the bandwidth of conventional MPA can be
enhanced by removing its PEC ground plane with perfect magnetic conductor
(PMC) (Monavar and Komjani 2011). In this case, the image of electric current is
in-phase and parallel to the original current distribution in contrast to PEC ground
plane. So the antenna impedance matching would be possible over a relatively wide
frequency range. Artificial magnetic conductors (AMC) exhibit the behavior of a
PMC at resonance, which are also called as high impedance surfaces (HIS) or EBG
ground planes. Such types of structures are used to enhance the radiation charac-
teristics of antenna and to reduce the effect of surface waves (Hosseini and Hakkak
2008). Generally, the artificial magnetic conductors are designed with FSS backed
by a grounded dielectric (Hosseinipath and Wu 2009).
In this section, the analysis of a rectangular MPA is presented over HIS substrate
designed using different types of FSS structure such as on Jerusalem crossed FSS,
single square loop FSS, etc. Basically, FSS-based HIS acts as PMC ground plane
for the antenna. The EM analysis of proposed microstrip antenna is carried out
based on equivalent circuit model as it requires less memory and CPU time as
compared to full-wave analysis method (Monavar and Komjani 2011). The pro-
posed antenna exhibits a significant enhancement of impedance bandwidth (19.8 %)
as compared with the conventional patch antenna (10.2 %) at 10 GHz.
3.1 Theoretical Considerations
In this work, a rectangular MPA is considered for the theoretical simulation. The
side view of a rectangular patch antenna loaded with HIS ground plane is shown in
Fig. 2, where Jerusalem crossed FSS (stop-band) backed by grounded dielectric acts
as high impedance ground plane for the antenna.
A general microstrip antenna has a ground plane on one side of a dielectric
substrate and a metallic radiating patch on the other side of it. The patch can be fed
through a coaxial line or microstrip line to excite the antenna. According to
modal-expansion cavity model, MPA is considered as a thin TMz-mode cavity
having magnetic walls around the peripheral of the patch and electric walls at the
top and bottom of the patch (Carver and Mink 1981). As antenna is excited, the
fringing field is formed between the ground plane and periphery of the patch that
leads to the radiation from the patch antenna. This is due to the fact that the
dimensions of the patch are finite along its length and width. As a result of fringing
phenomenon, the electrically length of the patch increases and hence its physical
dimensions increases. Let us consider the extension of the length on each side
represented by ΔL. A practical approximation for the normalized extension of
length (ΔL/h) is given by Balanis (1997)
(a)
L1
Ground plane Patch
Feed point
W1
JC-FSS location
FSS substrate
Antenna substrate
(b) g
lg
wg
w
a
wc
Fig. 2 a Schematic of rectangular microstrip patch antenna over FSS-based HIS, b unit cell of
Jerusalem cross FSS

DL ðee þ 0:3Þ Wh1 þ 0:264
¼ 0:412 ð1Þ
h ðee 0:258Þ Wh1 þ 0:8
where, h and W1 are the height of the substrate and width of the antenna, respec-
tively. εe is the effective dielectric constant of the antenna substrate, expressed as
1
er þ 1 er 1 h 2
ee ¼ þ 1 þ 12 ð2Þ
2 2 W1
The width W1 of the patch antenna is determined by

sffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffi
c 2
W1 ¼ ð3Þ
2fr ð e e þ 1Þ
where, c represents the velocity of light in free-space and fr denotes the resonant
frequency of the microstrip antenna. Since the length of the patch is extended by
ΔL on each side, the effective length of the patch is
Leff ¼ L1 þ 2DL ð4Þ
where, L1 represents the actual length of the rectangular MPA.

The analysis of proposed antenna is carried out based on cavity model in
combination with equivalent circuit model. Accordingly, a rectangular MPA can be
represented by a parallel RLC resonant circuit as shown in Fig. 3a, where, Ra
represents the resistance due to the ohmic losses in the metallic parts of the patch.
La and Ca represent the inductance and capacitance due to the magnetic and electric
energy stored in the antenna, respectively. The proposed microstrip antenna is
excited with coaxial probe (50 Ω) feed, which offers a series inductance Lp to the
RLC circuit of patch antenna (Fig. 3a). The expression for Lp is given by (Kanaujia
and Viswakarma 2006)

g0 h 4c
Lp ¼ ln pffiffiffiffi ð5Þ
2pc fx d er
(a)
Lp
(b)
RD Lg Cg
Za
Ra La Ca Zs
Zd
(c)
Za Zs
Z
Fig. 3 a Equivalent circuit of the rectangular microstrip patch antenna, b equivalent circuit of the
Jerusalem cross FSS, c equivalent circuit of the FSS-based HIS
where, ζ = 1.781072…, i = 1 is determined from Euler’s constant. d is the diameter

of coaxial probe and c represents the velocity of light in free-space. η0 is the
free-space impedance.
Now from Fig. 3a, the input impedance of rectangular MPA can be determined
as
1
Za ¼ þ jxLp ð6Þ
1
Ra þ jxCa þ jxL
1
a
where, Ra is the equivalent resistance due to ohmic losses in the metallic parts of the
patch. La and Ca are the equivalent inductance and capacitance, respectively, cor-
responding to magnetic and electric energy stored within the patch antenna. The
expressions for Ra, La, and Ca is given as (Bahl and Bhartia 1980)

ee e0 L1 W1 2 p y0
Ca ¼ cos ð7Þ
2h L1
where, y0 represents the length of the feed-point along the length of the patch
antenna,
1
La ¼ ; ð8Þ
Ca x2r
and
Q
Ra ¼ ð9Þ
xr Ca
where, Q is the total quality factor of the microstrip antenna (Derneryed and Lind
1979) and ωr is the angular resonance frequency of MPA.
In this work, the Jerusalem crossed FSS (JC-FSS) is used to design the high
impedance ground plane for the proposed antenna. The unit cell of the Jerusalem
cross FSS is comprised of capacitive and inductive elements (Fig. 2b) and it
exhibits band-stop characteristics. For this structure, the surface impedance plays a
vital role in determining the resonant frequency and the phase of reflection coef-
ficient (Hosseinipanah and Wu 2009). The equivalent circuit of Jerusalem crossed
FSS backed by grounded dielectric is shown in Fig. 3b, where RD, Lg, and Cg
represent the resistance, inductance, and capacitance, respectively associated with
dielectric backed Jerusalem cross FSS. Zd is the impedance offered by the grounded
dielectric.
The inductive reactance of the Jerusalem cross FSS is given by Hosseini and
Hakkak (2008) as

klg
Xg ¼ Z0 tan ð10Þ
2
where, lg is the length of the inductive grid and Z0 is the characteristic impedance of
Jerusalem cross strip. k is the wave number, expressed as
pffiffiffiffiffiffiffiffiffipffiffiffiffiffiffiffiffi
k ¼ x l0 e0 ereff ð11Þ
where, εreff is the effective relative permittivity of HIS substrate and is determined as
1
er þ 1 er 1 d 2
ereff ¼ þ 1 þ 10 ð12Þ
2 2 Wg
where, Wg is the length of the capacitive grid and d is the height of the HIS
substrate.
The capacitance offered by Jerusalem cross array can be determined, based on
the capacitance between two parallel patches placed apart on a dielectric slab as

2Wg 1 a
Cg ¼ e0 ereff cosh ð13Þ
p g
Since, the electric field lines associated with a lossy medium surrounding the
FSS structure lead to the dielectric loss. Such loss component can be expressed as a
series resistor in parallel with lossless capacitor (between the adjacent elements) and
is given by Costa and Monarchio (2012)
2e00r
RD ¼ 2 ð14Þ
xC0 e0r þ 1
where, ε′r and ε′′r are the real and imaginary parts of the complex relative permit-
tivity, respectively. C0 represents the capacitance of the FSS structure in
free-standing configuration, is given as

2Wg 1 a
C0 ¼ e0 cosh ð15Þ
p g
The lumped impedance offered by JC-FSS printed on lossy substrate can be

expressed as
ZFSS ¼ Rs þ jXs ð16Þ
where, Rs and Xs is the resistance and reactance, respectively offered by FSS

structure.
The input impedance of a PEC-backed dielectric slab of thickness d is given as

(Costa and Monarchio 2012)
g0 pffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffi
Zd ¼ j pffiffiffiffiffiffiffiffiffiffiffiffiffiffiffi
0 00
tan k0 e0r þ je00r d ð17Þ
er þ jer
where, η0 is the free-space impedance and k0 is the free-space wave number. From
Eq. (17), the real and imaginary parts of the input impedance of the grounded
dielectric can be expressed as
" ! #
f0 e00r pffiffiffiffi e00r pffiffiffiffi
RefZd g ffi pffiffiffiffi0 tan k 0 d er k0 d pffiffiffiffi0 1 þ tan k0 d er
0 2 0 ð18Þ
er 2e0r 2 er
and
f pffiffiffiffi
ImfZd g ffi p0ffiffiffiffi0 tan k0 d e0r ð19Þ

er
The input impedance of the FSS-HIS structure is determined by the parallel

combination of impedance of JC-FSS and surface impedance of grounded dielectric
slab (Fig. 3b) is expressed as
Zd ZFSS
Zs ¼ Zd jjZFSS ¼ ð20Þ
Zd þ ZFSS
Now, the input impedance of proposed microstrip antenna designed over HIS
substrate can be estimated by equivalent circuit modelas shown in Fig. 3c. It is
apparent that the input impedance of proposed microstrip antenna can be deter-
mined by the parallel combination of HIS impedance and the MPA impedance. In
order to calculate the input impedance of MPA, the height of the HIS substrate is
also added to the height of the conventional MPA. Thus, the input impedance of the
HIS-based antenna can be expressed as
Za Zs
Z ¼ Za jjZs ¼ ð21Þ
Za þ Zs
Using Eq. (21), the return loss of the proposed antenna can be computed as
R ¼ 20 logjCj ð22Þ
where, Γ is the reflection coefficient of the proposed antenna given as
Z Zc
C¼ ð23Þ
Z þ Zc
where, Zc is the characteristic impedance of coaxial feed line (50 Ω).

3.2 EM Design of Microstrip Patch Antenna Over FSS-HIS
Since the proposed antenna comprises of a rectangular MPA whose ground plane is
replaced by FSS-based HIS. The EM design considerations of conventional rect-
angular patch antenna and FSS-based HIS is discussed separately in the following
subsections.
3.2.1 EM Design of Microstrip Patch Antenna
In this work, a rectangular MPA is designed at the operating frequency of 10 GHz.

The design parameters of the antenna are determined based on cavity model. The
height of the antenna substrate and its dielectric constant is considered to be
1.588 mm and 2.2, respectively. The length and width of the patch are calculated to
be 9.06 and 11.86 mm, respectively.
The rectangular MPA is excited with a coaxial probe (50 Ω) located at
x0 = 0.312 mm (along the length of MPA) and y0 = W/2. For efficacy of cavity
model and equivalent circuit approach, the input impedance of a rectangular MPA
is compared with that of reported result at 1.575 GHz (Volakis 2007). The reported
rectangular MPA was designed at resonant frequency 1.575 GHz and its designed
parameters are; length L = 62.55 mm, width W = 93.83 mm, dielectric constant of
the substrate εr = 2.2, substrate height h = 1.524 mm. The reported antenna was
excited with a coaxial probe of radius 0.635 mm and located at x0 = 18.5 mm and
y0 = W/2. A good agreement is achieved between computed and reported results as
shown in Fig. 4. Further, the input impedance and return loss of proposed rect-
angular MPA is computed and their frequency response is also studied as shown in
Fig. 5a, b. It is observed that the proposed rectangular MPA antenna resonates at
10 GHz with perfect feed matching.
3.2.2 EM Design of FSS-Based HIS
The HIS structure consists of Jerusalem cross FSS backed by grounded dielectric
substrate in this work. The Jerusalem cross FSS is intended to design at the same
frequency as that of rectangular MPA (10 GHz), for band-stop characteristics.
The designed parameters of the JC-FSS-based HIS are; width of the inductive
component, w = 0.1 mm, length of the inductive component, lg = 4.0 mm, length of
the capacitive component, Wg = 3.43 mm, width of the capacitive segment,
wc = 0.29 mm, and separation between the adjacent crosses, g = 0.38 mm. The
height of the HIS substrate is considered to be, d = 0.34 mm.
The impedance of FSS-based HIS is also determined based on equivalent circuit
model. In order to validate the approach, the reflection phase of the Jerusalem
crossed FSS-based HIS is computed based on equivalent circuit model as discussed
in the previous section for the designed parameters of Wg = 3.5 mm, wc = 0.1 mm,
g = 0.4 mm, lg = 4 mm, w = 0.1 mm, d = 1 mm, and εr = 2.2. The reflection phase of
50
40
30
Input impedance (Ohm)
20
10
0
Re [Za]: Reported
-10 Im [Za]: Reported
Re [Za]: Computed
-20 Im [Za]: Computed
-30
1.53 1.54 1.55 1.56 1.57 1.58 1.59 1.6 1.61 1.62
Frequency (GHz)
Fig. 4 Input impedance of rectangular microstrip patch antenna. Red lines show computed results
at CEM, CSIR-NAL. Blue lines show reported results (Volakis 2007)
the JC-FSS-based HIS is studied with respect to operating frequency and compared
with reported result estimated based on numerical simulation as shown in Fig. 6. It
is observed that excellent agreement is obtained between computed and reported
result (Hosseinipanah and Wu 2009).
Further, the impedance of the proposed FSS-based HIS is determined using
equivalent circuit model and studied its frequency response as shown in Fig. 7. It is
observed that the proposed FSS-based HIS resonates at 10 GHz and exhibits very
high impedance at resonance. Thus, the Jerusalem crossed FSS backed by grounded
dielectric behaves as HIS at resonance and hence it can be used as ground plane for
microstrip antenna.
3.3 EM Performance Analysis
The EM analysis of microstrip antenna loaded with various FSS-based HIS such as
Jerusalem cross FSS and square loop FSS are carried out based on equivalent
transmission line approach. The details are discussed in the following subsections.
3.3.1 Jerusalem Cross FSS-Based Microstrip Antenna
This subsection deals with the EM performance analysis of rectangular MPA over
JC-FSS-based HIS followed by validation of equivalent circuit approach.
(a) 50
40
Re[Za]
Im[Za]
30
Input Impedence (Ohm)
20
10
-10
-20
-30
8 9 10 11 12
Frequency (GHz)
(b) 0
-10
Return Loss (dB)
-20
-30
-40
-50
8 9 10 11 12
Frequency (GHz)
Fig. 5 a Input impedance of rectangular microstrip patch antenna designed at 10 GHz. b Return
loss of rectangular MPA versus frequency (designed at 10 GHz)
Validation of equivalent circuit approach: The EM analysis of microstrip

antenna loaded with FSS-based HIS is carried out based on equivalent circuit
model. For efficacy of this novel approach, a rectangular MPA loaded with
Jerusalem cross FSS-based HIS is investigated based on equivalent circuit model
200
150
100
Reflection phase (degree)
50 Computed
Reported
0
-50
-100
-150
-200
0 3 6 9 12 15
Frequency (GHz)
Fig. 6 Reflection phase of the JC-FSS-based HIS. Solid blue line shows computed result at CEM
based on equivalent circuit model. Dotted red line shows reported result based on numerical model
(Hosseinipanah and Wu 2009)
10000
Re[Zs]
8000
Im[Zs]
6000
Impedance (Ohm)
4000
2000
-2000
-4000
-6000
8 9 10 11 12
Frequency (GHz)
Fig. 7 Real and imaginary parts of input impedance of JC-FSS-based HIS
and its result is compared with that of reported result (Monavar and Komjani 2011)
for the designed parameters; length of the microstrip antenna, L1 = 12.5 mm, width
of the patch, W1 = 17.5 mm, height of antenna substrate, h = 1.58 mm, width of
inductive grid, w = 0.55 mm, length of capacitive grid, wg = 8.5 mm, length of the
inductive grid, lg = 11.64 mm, distance between two inductive grids, a = 4.31 mm,
and height of FSS-HIS substrate = 3.16 mm, at designed frequency of 5.8 GHz.
The reported antenna which was simulated based on full-wave method, exhibited
a bandwidth of 3.44 % with PEC ground plane and 10.41 % on FSS-based HIS
ground plane as shown in Fig. 8. While the rectangular MPA which is simulated
based on equivalent circuit approach, exhibits a bandwidth of 4.3 % with PEC
ground plane and enhanced bandwidth of 11.89 % on FSS-based HIS ground plane
(Fig. 8). It is observed that both approaches show almost similar bandwidth
enhancement of *7 % as compared to the conventional rectangular MPA.
However, in equivalent circuit approach, the resonance frequency of HIS-based
antenna shifted to 5.71 GHz instead of 5.8 GHz, which may be due to the
approximation of lumped parameters considered in this approach.
Performance analysis of JC-FSS-based antenna: Finally, the EM performance
analysis of the proposed antenna carried out in this work is based on equivalent
circuit approach. For the analysis, both rectangular MPA and FSS-based HIS is
designed at the center frequency of 10 GHz. The return loss of proposed microstrip
antenna is computed and compared with that of conventional rectangular MPA as
shown in Fig. 9. It is noted that the proposed microstrip antenna exhibits an
impedance bandwidth (−10 dB) from 8.65 to 10.63 GHz, i.e., 19.8 % with HIS
substrate, while it shows 10.2 % for PEC ground plane. It is obvious that the
impedance bandwidth of the proposed antenna with HIS ground plane is enhanced
-10
Return loss (dB)
-20
PEC (Reported)
-30 FSS-HIS (Reported)
PEC (Computed)
FSS-HIS (Computed)
-40
-50
4.5 5 5.5 6 6.5
Frequency (GHz)
Fig. 8 Return loss of rectangular MPA microstrip antenna with PEC ground plane and FSS-HIS
ground plane designed at 5.8 GHz. Bullet points show reported results (Monavar and Komjani
2011). Solid lines show computed results at CEM
-10
Return loss (dB)
-20
-30
With PEC ground plane

-40
With HIS ground plane
-50
8 9 10 11 12
Frequency (GHz)
Fig. 9 Return loss of the proposed MPA with PEC and FSS-based HIS ground plane designed at
10 GHz
by 9.6 % as compared to PEC ground plane. Thus, the Jerusalem crossed

FSS-based HIS enhances the bandwidth of a rectangular MPA.
Further, the effect of geometrical parameters of FSS elements on return loss of
antenna is studied by varying the length of the inductive grid (lg) and gap between
the adjacent crosses (g) of JC-FSS. It is observed that as the length of the inductive
grid increases, the bandwidth and resonance frequency of the proposed antenna
decreases as shown in Fig. 10. While, the bandwidth and resonance frequency of
proposed antenna increases with the increase in the gap between the adjacent
crosses of JC-FSS (Fig. 11). Thus, the bandwidth of the proposed HIS-based
antenna can be tuned by varying the geometrical parameters of JC-FSS elements
such as lg and g.
The far-field pattern of proposed antenna is estimated using the principles of
reciprocity theorem and equivalent transmission line analogy (Volakis 2007). The
details of this approach are discussed in Sect. 4. Accordingly, the proposed antenna
problem reduces to the scattering of plane wave on the grounded multilayered
structure and its reflection coefficient is determined based on equivalent transmis-
sion line method as discussed in Sect. 3.1.
The far-field radiation pattern of proposed antenna is determined in E-plane and
compared with the rectangular MPA having PEC ground plane (Fig. 12). It is
observed that the FSS-HIS-based antenna shows a significant enhancement of
beamwidth (13.18°) as compared to that of conventional microstrip antenna.
lg = 3.9 mm
-5
lg = 4.0 mm
lg = 4.1 mm
-10
Return loss (dB)
-15
-20
-25
-30
-35
8.0 8.5 9.0 9.5 10.0 10.5 11.0 11.5 12.0
Frequency (GHz)
Fig. 10 Return loss of rectangular MPA with FSS-HIS ground plane for different length of the
inductive grid (lg) of JC-FSS element
-5 g = 0.36 mm
g = 0.38 mm
g = 0.40 mm
-10
Return loss (dB)
-15
-20
-25
-30
-35
8 9 10 11 12
Frequency (GHz)
Fig. 11 Return loss of rectangular MPA with FSS-HIS ground plane for different gap between the
adjacent crosses (g) of FSS element
90
0
120 60
-5
150 -10 30
PEC ground plane

FSS-HIS ground plane
-15
180 -20 0
0 -5 -10 -15 -20 -15 -10 -5 0
-15
210 -10 330
-5
240 300
0
270
Fig. 12 E-plane radiation pattern of rectangular microstrip antenna with PEC ground plane and
JC-FSS-based HIS ground plane
3.3.2 Square Loop FSS-Based Microstrip Antenna
The performance analysis of proposed microstrip antenna is further studied by

using single square loop FSS (SSL-FSS)-based HIS ground plane. The schematic of
square loop FSS-based HIS is shown in Fig. 13, where p, d, and s represent the
periodicity, side length of square loop, and thickness of square loop grid, respec-
tively. The equivalent circuit of square loop FSS backed by grounded dielectric
which forms the HIS is shown in Fig. 14. Here, RD, Ls, and Cs are associated with
the dielectric loss around the square loop, magnetic current, and gap between the
square loops, respectively. Zd represents the impedance offered by grounded
dielectric beneath the SL-FSS. The numerical value of RD, Ls, Cs, and Zd are
determined using the same concept as considered for JC-FSS HIS. Similar to the
JC-FSS case, the single square loop FSS-based HIS is also designed at 10 GHz.
The designed dimensions of square loop FSS-based HIS are given in Table 1.
The input impedance of SSL-FSS-based HIS are computed based on equivalent
circuit model. From Fig. 15, it is evident that the square loop FSS backed by
Fig. 13 a Schematic of
square loop FSS-based HIS,
b unit cell of SSL-FSS (a)
Ground plane
FSS substrate
Square loop-FSS
(b)
p d
Fig. 14 Equivalent circuit of RD Ls Cs

single square loop FSS-based
HIS
Z sq
Zd
grounded dielectric (εr = 2.2 and tan δ = 0.0009) reveals very high impedance
(7134 Ω) at resonance. Thus, it can be used as HIS ground plane for microstrip
antenna.
Further, the MPA is loaded over SSL-FSS-based HIS and is analyzed based on
transmission line equivalent circuit model. The return loss of SSL-FSS-based
antenna is studied with respect to operating frequency and compared with that of
Table 1 Designed Designed parameters Optimized value

parameters of single square
loop FSS-based HIS Periodicity of square loop (p) 7.22 mm
Thickness of square loop grid (s) 0.22 mm
Length of square grid (d) 6.0 mm
Distance between two inductive grids (a) 1.66 mm
Height of HIS substrate 0.36 mm
Dielectric constant of HIS substrate 2.2
8000
6000
Re [Zs]
Im [Zs]
4000
Impedance (Ohm)
2000
-2000
-4000
-6000
8 9 10 11 12
Frequency (Ghz)
Fig. 15 Real and imaginary parts of input impedance of single square loop FSS-based HIS
conventional rectangular MPA as shown in Fig. 16. It is observed that the micro-
strip antenna designed over SL-FSS-based HIS ground plane shows a bandwidth
(10 dB) enhancement of 6.06 % as compared with that of conventional MPA.
The radiation characteristic of proposed antenna is determined based on reci-
procity theorem and transmission line theory. The computed radiation characteristic
is compared with that of conventional rectangular MPA. It is found that over the
single square loop FSS-based HIS ground plane, MPA exhibits a significant
enhancement of beamwidth (11.4°) in E-plane as compared to that of conventional
MPA. This is evident from Fig. 17. However, the resonance frequency of FSS-HIS
loaded microstrip antenna slightly shifted to 9.75 GHz apart from 10 GHz, which
may be due to the approximate considerations of lumped parameters in equivalent
circuit model.
In order to further study the effect of dielectric constant of HIS substrate on the
EM performance of proposed microstrip antenna, the dielectric material of proposed
SSL-FSS-based HIS is replaced with Teflon (εr = 2.08 and tan δ = 0.001) and then
composite antenna structure is analyzed based on equivalent circuit model.
-10
Return loss (dB)
-20

PEC ground plane
-30
-40
-50
8 9 10 11 12
Frequency (Ghz)
Fig. 16 Return loss of rectangular MPA with PEC ground plane and SSL-FSS-based HIS ground
plane
Fig. 17 E-plane radiation 90

0
pattern of rectangular MPA 120 60
with PEC ground plane and
SSL-FSS-based HIS ground -5
plane
150 -10 30

PEC ground plane
-15
180 -20 0
0 -5 -10 -15 -20 -15 -10 -5 0
-15
210 -10 330
-5
240 300
0
270
The input impedance of SSL-FSS-based HIS is studied with respect to operating

frequency. It is evident from Fig. 18 that the Teflon-based HIS exhibits very high
impedance (7364 Ω) at resonance as compared to that of RT-Duroid-based
8000
6000
Re [Zs]
Im [Zs]
4000
Impedance (Ohm)
2000
-2000
-4000
-6000
8 9 10 11 12
Frequency (GHz)
Fig. 18 Real and imaginary parts of input impedance of SSL-FSS-based HIS (Teflon substrate)
-10
Return loss (dB)
-20
PEC groung plane
-30
-40
-50
8 9 10 11 12
Frequency (Ghz)
Fig. 19 Return loss of rectangular MPA with PEC ground plane and SSL-FSS-based HIS (Teflon
substrate) ground plane
SSL-FSS-based HIS. Further, the return loss of composite antenna is studied with
respect to operating frequency and compared with that of conventional MPA as
shown in Fig. 19. It is observed that the MPA over Teflon substrate HIS reveals a
90
0
120 60
-5
150 -10 30
-15
PEC ground plane
180 -20 0
0 -5 -10 -15 -20 -15 -10 -5 0
-15
210 -10 330
-5
240 300
0
270
Fig. 20 E-plane radiation pattern of rectangular MPA with PEC ground plane and SSL-FSS-based
HIS (Teflon substrate) ground plane
significant enhancement of bandwidth (8.4 %) as compared to the conventional

MPA. This result concludes that the bandwidth of proposed antenna can further be
enhanced by changing the dielectric properties of HIS substrate. The radiation
characteristics of composite antenna loaded with Teflon-based HIS is also studied
as shown in Fig. 20. This shows a significant enhancement in beamwidth (12.6°) as
compared to conventional MPA.
4 Microstrip Antenna Loaded with FSS-Based

Superstrate
A superstrate layer is generally used to enhance the directivity as well as bandwidth

of the microstrip antenna (Jackson and Alexopoulos 1985; Pirhadi et al. 2006). In
order to design low profile low RCS antenna, the superstrate must be compact and
easily fabricated. For the design of such antenna, frequency selective surfaces are
found to be the best suitable candidate for the design of superstrate (Foroozesh and
Shafai 2010; Pirhadi et al. 2012) as it exhibits the filter characteristics for the EM
wave impinging on it. The bandwidth, polarization, and radiation characteristics
(e.g., side lobe level, directivity etc.) of antenna can be controlled using FSS
superstrate or its combination with reactive surfaces (Foroozesh and Shafai 2006;
Rodes et al. 2007). In addition, the FSS superstrate can also be used as a polarizer
(Pirhadi et al. 2012).
Several methods have been proposed in open domain for the analysis of
microstrip antenna loaded with superstrate such as FEM, MoM, etc., (Alexopoulos
and Jackson 1984; Pirhadi et al. 2007). However, the analyzes based on full-wave
methods are computationally complex and require large CPU time and memory to
converge the solution. In this endeavor, the EM analysis of a MPA covered with
FSS-superstrate is presented based on transmission line equivalent circuit model
(ECM) as it is computationally less complex and requires less CPU time and
memory to handle such problems.
The radiation characteristic of proposed antenna is estimated using transmission
line analogy and reciprocity theorem. Here, double square loop-frequency selective
surface (DSL-FSS) has been utilized to design the FSS-based superstrate layer.
Since, the DSL-FSS is basically a double resonant structure that provides a trans-
mission band formed due to combination of two reflection bands. This means that
the DSL-FSS provides both transmission and reflection bands, which are insensitive
to the angle of incidence (Luo et al. 2005). The proposed antenna exhibits a
directivity enhancement of 3.85 dB in E-plane and 4.06 dB in H-plane as compared
to that of conventional microstrip antenna.
4.1 Theoretical Considerations
A rectangular MPA is considered for the theoretical simulation in this work. The
side view of rectangular MPA covered with FSS-based superstrate is shown in
Fig. 21, where the DSL-FSS backed by a dielectric substrate is used to design the
superstrate. The superstrate loaded antenna is fed through a 50 Ω coaxial cable to
excite the field within the antenna. According to cavity model theory, a rectangular
MPA structure can be analyzed by solving a parallel RLC resonant circuit as shown
in Fig. 22a, where Ra represents the resistance due to the ohmic losses in the
metallic parts of the patch. La and Ca represent the inductance and capacitance due
to the magnetic and electric energy stored in the antenna, respectively. The input
impedance of antenna is determined using Eq. (6).
In this work, the DSL-FSS structure is used to design the superstrate for the
proposed microstrip antenna. The geometry of DSL-FSS is shown in the Fig. 21b,
which consists of concentric inner and outer rings separated by a gap in between.
According to the equivalent circuit model, a DSL-FSS structure can be represented
as parallel combination of two series LC resonant circuits as shown in Fig. 22b,
4 Microstrip Antenna Loaded with FSS-Based Superstrate 25
(a)
FSS substrate
Ground plane L1
DSL FSS
Feed point
location
W1
Patch
Antenna
substrate
(b)
p d1
d2
t2
t1
g1 g2
Fig. 21 a Schematic of rectangular microstrip patch antenna loaded with FSS superstrate,
b geometry of unit cell of DSL-FSS
where L1s and C1s are the inductance and capacitance, respectively offered by the
outer square ring. L2s and C2s represent the inductance and capacitance, respectively
offered by the inner square ring.
The numerical value of L1s, C1s, L2s, and C2s can be determined by the
expressions (Luo et al. 2005), given below

X1s X2s d1
X1 ¼ xL 1s ¼ 2:0 ð24Þ
X1s þ X2s p
Fig. 22 a Equivalent circuit (a)

of rectangular microstrip
patch antenna, b equivalent
circuit of DSL-FSS Lp
Za
Ra La Ca
(b)
L1s L2s
C1s C2s
where, X1s ¼ F ðp; t1 ; k Þ and X2s ¼ F ðp; t2 ; k Þ

d2
X2 ¼ xL2s ¼ F ðp; 2t2 ; kÞ ð25Þ
p
where, F ðp; 2t2 ; kÞ is determined from (Lee and Langley 1985)

d1
B1 ¼ xC1s ¼ 0:75 B1s ð26Þ
p
where, B1s ¼ 4:0 eeff F ðp; g1 ; kÞ:

Here, eeff is the effective dielectric constant and it is given by Costa et al. (2012)

1
eeff ¼ er h þ ðer h 1Þ ð27Þ
expN ð xÞ
where,
er þ 1
er h ¼ ; ð28Þ
2
10 d
x¼ ; ð29Þ
p
and N is an exponential factor in which numerical value is 1.3 for ring-like structure
and 1.8 for the cross-like structure. d and p represent the thickness of the FSS
substrate and periodicity of the FSS unit cell, respectively.

B 1s B 2s d2
B2 ¼ xC2s ¼ ð30Þ
B 1s þ B 2s p
where, B2s ¼ 4:0 eeff F ðp; g2 ; kÞ:

Since in practice, the inductive and capacitive reactances offered by FSS
structure are different for TE and TM mode of incidence wave. These reactances
can be determined separately for TE and TM polarizations (Lee and Langley 1985).
Now the admittance of DSL-FSS is determined as

B1 B2
Y ¼ j þ ð31Þ
1 X1 B1 1 X2 B2
The impedance offered by DSL-FSS-based superstrate can be computed by
1
Zs ¼ ð32Þ
Y
Finally, the input impedance of proposed antenna (i.e., microstrip antenna

covered with FSS superstrate) shown in the Fig. 23 is determined by the parallel
Fig. 23 Equivalent circuit of

the antenna loaded with
square loop FSS-based
superstrate
ZS Za
Z
combination of the impedance of MPA and the impedance of FSS superstrate

(DSL-FSS) expressed as
Za Zs
Z ¼ Za jjZs ¼ ð33Þ
Za þ Zs
Using Eq. (33), the input impedance and return loss of MPA loaded with FSS
superstrate can be determined. This is to be noted that the above expression will
only be used to determine the input impedance and return loss of composite antenna
when there would not be any gap between the superstrate and antenna. For the air
gap between the antenna superstrate, the additional impedance offered by gap will
be used in parallel to the antenna impedance.
4.2 Estimation of Far-Field Radiation Pattern of Antenna
The radiation pattern of antenna is determined based on transmission line theory

and reciprocity theorem. Here, the MPA loaded with FSS superstrate is considered
as a transmission line. According to the transmission line theory, the impedance of
the whole structure can be determined by considering the antenna and FSS as a
section of the transmission line as show in Fig. 24. Each section of transmission line
is analyzed by its characteristic impedance and propagation constant that basically
depends on the dielectric properties of the layer and angle of incidence (Jackson and
Alexopoulos 1985). Thus, the input impedance of each layer is estimated and used
as the load for the preceding layer is explained below.
In Fig. 24, the first section represents the microstrip antenna, where the shorted
load denotes the ground plane of the antenna and Zc1 is the characteristic impedance
FSS Antenna
Antenna ground
Z c0 Zs Z c1
plane
Z2 = l Z1 = h
Fig. 24 Equivalent transmission line of MPA covered with FSS superstrate

of the antenna. The expressions of Zc1 for both TE and TM polarization is given by
Jackson and Alexopoulos (1985) as
g0 N 1 ð hÞ
Zc1 ¼ ; For TE-polarization ð34Þ
e1
g0 l 1
Zc1 ¼ ; For TM-polarization ð35Þ
N1 ðhÞ
The impedance at the terminal plane Z ¼ h can be determined by transmission

line theory as
Z1 ¼ j Zc1 tanðk0 N1 ðhÞ hÞ ð36Þ
where, k0 and N1 ðhÞ are the propagation constant and refractive index corre-
sponding to the antenna substrate, respectively. Similarly, the input impedance at
the terminal plane Z ¼ l can be computed by
Z1 þ j Zs tan ðb lÞ
Z2 ¼ Zs ð37Þ
Zs þ j Z1 tan ðb lÞ
where, Zs is the characteristic impedance of FSS superstrate which can be computed

by Eq. (32). l represents the height of the FSS substrate. Finally, the reflection
coefficient of the whole structure can be determined as
Z2 Zc0
C¼ ð38Þ
Z2 þ Zc0
where, Zc0 represents the characteristic impedance of free-space, which can be

expressed for both TE and TM polarizations (Jackson and Alexopoulos 1985) as
Zc0 ¼ g0 cosðhÞ; For TE-polarization ð39Þ
Zc0 ¼ g0 secðhÞ; For TM-polarization ð40Þ
Thus, the reflection coefficient of proposed antenna is determined using Eq. (38),
which will be used for the estimation of far-field radiation pattern of antenna both in
E-plane and H-plane as discussed below.
The far-field pattern of proposed antenna is estimated using the principles of
reciprocity theorem. Accordingly, the far-field Ei¼ h; / ðr; h; /Þ of antenna can be
determined by placing a unit-amplitude testing dipole at the far-field distance in the
direction of interest (h or /). Thus, the proposed antenna problem reduces to the
scattering of plane wave on the grounded multilayered structure and its reflection
coefficient is determined based on transmission line analogy as discussed above.
The far-field radiation pattern of antenna loaded with superstrate for both E- and
H-plane can be computed by Volakis (2007)

E0 L
Ehpatch ðr; h; /Þ ¼ 2Wh cos / 1 C TM
ðhÞ cos kx
g0 2
ð41Þ
L
sin c ky tan cðkZ1 hÞ
2

E0 L
Eupatch ðr; h; / Þ ¼ 2Wh ðcos h sin /Þ 1 C ðhÞ cos kx
TE
g0 2
ð42Þ
W
sin c ky tan cðkZ1 hÞ
2
where,

jxl0 jk0 R
E0 ¼ e ð43Þ
4p R
and
kx ¼ k0 sin h cos / ð44Þ
ky ¼ k0 sin h sin / ð45Þ
kz1 ¼ k0 N1 ðhÞ ð46Þ

qffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffi
N 1 ð hÞ ¼ er lr sin2 h ð47Þ
where, the term CTE and CTM denotes the reflection coefficients in TE and TM
mode, respectively, g0 is the free-space impedance, R is the far-field distance
measured from the center of the patch. N1 ðhÞ is the refractive index of the medium.
4.3 EM Design of Microstrip Patch Antenna Loaded

with FSS Superstrate
A rectangular MPA is designed in this work at the operating frequency of 10 GHz.

The design parameters of the antenna are determined based on cavity model. The
height of the antenna substrate and its dielectric constant is considered to be
1.588 mm and 2.2, respectively. The length and width of the patch are calculated to
be 9.06 and 11.86 mm, respectively. The microstrip antenna is excited with a
coaxial probe of 50 Ω located at the feed length, y = 3.126 mm. The equivalent
circuit method is first employed to calculate the input impedance and then the return
loss of the MPA.
Further, the FSS-based superstrate is designed at the same operating frequency

(10 GHz) using transmission line equivalent circuit approach. In this work, DSL has
been used to design the FSS-based superstrate, which shows the pass-band char-
acteristic at the desired frequency for both TE and TM polarizations. The optimized
design parameters of DSL-FSS structure are: periodicity of array, p = 8.102 mm,
width of the outer square loop, w1 = 0.48 mm, width of the inner loop,
w2 = 0.48 mm, gap between two outer square loop, g1 = 0.162 mm, gap between
inner and outer square loop, g2 = 0.58 mm, length of the outer square loop,
d1 = 7.939 mm, and length of the inner square loop, d2 = 6.779 mm.
For efficacy of the equivalent circuit approach, the transmission characteristics of
proposed DSL-FSS is studied and compared with the reported measured results at
normal incidence (Luo et al. 2005). A good agreement is obtained between the
computed and reported result as shown in the Fig. 25. Further, the transmission
characteristic of proposed DSL-FSS is studied at various angles of incidence (0°,
30°, and 45°) for TE polarizations as shown in Fig. 26. It is observed that the
designed FSS reveals excellent transmission efficiency of 10 GHz for both
polarizations.
This is also corroborated from reflection data as shown in Fig. 27 for TE
polarizations, which shows very low (almost zero) reflection efficiency at different
incidence angles corresponding to the desired frequency. Thus, the proposed
DSL-FSS is found to be best suitable candidate for the design of superstrate to
enhance the directivity of MPA.
-10
Transmission coefficient (dB)
-20
-30 Reported (measured)

Computed
-40
-50
2 4 6 8 10 12 14 16 18
Frequency (GHz)
Fig. 25 Transmission characteristics of DSL-FSS structure; a reported (Luo et al. 2005),

b computed at CEM
-10
Transmission coefficient (dB)
-20
-30 00
300
450
-40
-50
-60
4 6 8 10 12 14 16
Frequency (GHz)
Fig. 26 Transmission characteristics of proposed DSL-FSS structure at different incidence angles

(0°, 30°, and 45°) for TE polarizations
-10
Refllection coefficient (dB)
-20
00
300
-30 450
-40
-50
-60
4 6 8 10 12 14 16
Frequency (dB)
Fig. 27 Reflection characteristics of proposed DSL-FSS structure at different incidence angles

(0°, 30°, and 45°) for TE polarizations
4.4 EM Performance Analysis
The input impedance and return loss of superstrate-based antenna is determined

using equivalent circuit approach. For the analysis, both microstrip antenna and the
FSS are designed at the frequency of 10 GHz. The input impedance of MPA loaded
with superstrate is estimated using Eq. (33). The variation of real as well as
imaginary parts of input impedance is studied with respect to operating frequency
as shown in Fig. 28a. It is observed that the MPA loaded with FSS-based super-
strate resonates at 10 GHz as it shows maximum value (48 Ω) of real parts of input
impedance and minimum value (zero) of imaginary parts of input impedance at the
resonance. Further, the return loss of the proposed antenna is determined using
Eq. (38) and its response is studied with respect to frequency as shown in Fig. 28b.
It shows minimum return loss (−11.8 dB) at the resonance frequency of 10 GHz.
The radiation characteristic of proposed antenna is estimated based on reci-
procity theorem and transmission line analogy. For efficacy of this approach, the
far-field radiation pattern of a rectangular MPA is validated with that of the reported
results (Volakis 2007), for the antenna design parameters; height of the substrate,
h = 0.1588 cm, ratio of height to wavelength, h/λ = 0.02, and patch aspect ratio
(width to length ratio), W/L = 1.5. Excellent agreement is obtained between com-
puted and reported results for both E- and H-plane pattern as shown in the Fig. 29a,
b, respectively.
In order to compute the radiation characteristics of proposed antenna, the
reflection characteristics of antenna is first determined based on transmission line
theory for the plane wave impinging on it using Eq. (38). The radiation charac-
teristics (E- and H-plane) of the antenna loaded with FSS-superstrate is then
computed using the Eqs. (41) and (42), respectively at the frequency of 10 GHz.
The radiation characteristics of proposed antenna in E- and H-plane are com-
pared with the conventional rectangular MPA as shown in Fig. 30a, b. It is evident
that the directivity of proposed antenna is enhanced by 3.85 dB in E-plane and
4.06 dB in H-plane as compared to that of conventional MPA.
4.4.1 Performance Analysis with Air Gap Between Antenna

and Superstrate
The radiation characteristics of proposed antenna is also analyzed by keeping air

gap between the MPA and FSS-superstrate. This air gap is considered to be half of
the operating wavelength. The directivity of the antenna is computed for E-plane as
well as H-plane by using the same transmission line analogy and reciprocity the-
orem as discussed in the previous section. The E-plane and H-plane pattern of
proposed antenna with air gap is compared with the conventional rectangular MPA
as shown in Fig. 31a, b.
(a) 60
40
Impedence (Ohm)
20
-20
Re (Z)
Im (Z)
-40
8 9 10 11 12
Frequency (GHz)
(b) 0
-2
-4
Return loss (dB)
-6
-8
-10
-12
-14
8 9 10 11 12
Frequency (GHz)
Fig. 28 EM characteristics of MPA loaded with superstrate; a input impedance, and b return loss
The directivity enhancement of antenna is observed to be 4.7 dB in E-plane and

4.06 in H-plane as compared to conventional rectangular MPA. This implies that
the directivity of antenna in E-plane is further enhanced with air gap, while H-plane
remained the same as that of without air gap.
90
0
120 60
(a) -10
-20
150 30
Computed
Reported -30
-40
180 -50 0
0 -10 -20 -3 0 -40 --50
50 -40 -30 -20 -10 0
-40
-30
210 330
-20
-10
240 300
0
270
90
(b) 120
0
60
-10
-20
150 Computed 30
Reported
-30
-40
180 -50 0
0 -10 -20 -3 0 -40 -50
-50 -40 -30 -20 -10 0
-40
-30
210 330
-20
-10
240 300
0
270
Fig. 29 Validation of a E-plane, and b H-plane pattern of rectangular MPA. Dotted black lines
show reported results (Volakis 2007). Solid blue lines show computed results at CEM
(a) 90
10
120 60
150 0 30
-5
With FSS Superstrate
MPA
180 -10 0
10 5 0 -5 -10 -5 0 5 10
-5
210 0 330
240 300
10
270
(b) 90
10
120 60
150 0 30
-5
MPA
180 -10 0
10 5 0 -5 -10 -5 0 5 10
-5
210 0 330
240 300
10
270
Fig. 30 Radiation pattern of rectangular MPA and MPA covered with DSL-FSS superstrate;
a E-plane and b H-plane
90
10
(a) 120 60
150 0 30
-5
MPA
180 -10 0
10 5 0 -5 --10
10 -5 0 5 10
-5
210 0 330
240 300
10
270
90
10
(b) 120 60
150 0 30
-5
MPA
180 -10 0
10 5 0 -5 -10 -5 0 5 10
-5
210 0 330
240 300
10
270
Fig. 31 Radiation pattern of rectangular microstrip antenna and MPA covered with FSS-based
superstrate by keeping air gap between them; a E-plane and b H-plane
5 Summary
The present book dealt with the design and analysis of FSS-based high performance
MPA. The FSS structures were used as high impedance ground plane and superstrate
for the antenna. The EM analysis of a microstrip antenna loaded with FSS-based HIS
has been carried out using cavity model in combination with equivalent circuit
approach. The microstrip antenna was designed using cavity model, while FSS-based
HIS was designed based on equivalent circuit model. For efficacy of the equivalent
circuit approach, the computed results of HIS-based antenna is validated with
reported results, which was obtained based on full-wave analysis method. Further,
the EM performance characteristics of rectangular MPA over FSS-based HIS is
studied based on equivalent circuit model. It is found that the impedance bandwidth
(−10 dB) of proposed antenna is enhanced to 19.8 % as compared with that of
conventional rectangular MPA (10.2 %) by loading the FSS-based HIS as ground
plane. It is also revealed that the impedance bandwidth of the FSS-HIS-based antenna
can be tuned by varying the geometrical parameters of FSS elements. The EM
performance of proposed MPA over single square loop FSS-based HIS is also studied
for different dielectric material of HIS substrate. It is found that the bandwidth of
HIS-based antenna can be tuned by changing the dielectric material of HIS substrate.
Further, the EM analysis of a microstrip antenna loaded with FSS superstrate has
been carried out in this book by using transmission line equivalent circuit model
and reciprocity theorem. For efficacy of the approach, the far-field pattern (E- and
H-plane) of a rectangular MPA is validated with that of reported results. Excellent
agreement is obtained between computed and reported results. The MPA covered
with DSL-FSS superstrate exhibited directivity enhancement of 3.85 dB in E-plane
and 4.06 dB in H-plane as compared to that of conventional MPA. Further, the
directivity of proposed antenna has been enhanced by keeping air-gap between
antenna and FSS layer, which is observed to be 4.7 dB in E-plane and 4.06 in
H-plane as compared to the conventional rectangular MPA. Since superstrate has
been designed using DSL-FSS which will reject the impinging signal on antenna
structure outside the operating band, and hence it will reduce out-of-band RCS of
proposed antenna. However, estimation of out-of-band RCS has not been discussed
in this context. Thus, the proposed FSS-superstrate-based antenna may find
potential applications at low observable platforms. To conclude, the topics dis-
cussed in this book would serve as the building block for designing the FSS-based
low observable antennas.
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About the Book
This book focuses on performance enhancement of printed antennas using fre-

quency selective surfaces (FSS) technology. The growing demand of stealth tech-
nology in strategic areas requires high-performance low-RCS (radar cross section)
antennas. Such requirements may be accomplished by incorporating FSS into the
antenna structure either in its ground plane or as the superstrate, due to the filter
characteristics of FSS structure. In view of this, a novel approach based on FSS
technology is presented in this book to enhance the performance of printed antennas
including out-of-band structural RCS reduction. In this endeavor, the EM design of
microstrip patch antennas (MPA) loaded with FSS-based (i) high impedance sur-
face (HIS) ground plane, and (ii) the superstrates are discussed in detail. The EM
analysis of proposed FSS-based antenna structures have been carried out using
transmission line analogy, in combination with the reciprocity theorem. Further,
various types of novel FSS structures are considered in designing the HIS ground
plane and superstrate for enhancing the MPA bandwidth and directivity. The EM
design and performance analyses of FSS-based antennas are explained here with the
appropriate expressions and illustrations.

DOI 10.1007/978-981-287-775-8
Author Index
A H
Alexopoulos, N.G., 23–24, 29 Hakkak, M., 2, 5, 8, 23
Allan, R., 1 Hosseini, M., 5, 8
Appleton, S., 1 Hosseinipanah, M., 5, 8, 12, 14
Arnaud, E., 24
J
B Jackson, D.R., 23–24, 28, 29
Bahl, I.J., 8 Jecko, B., 24
Bahrami, H., 1, 2, 23–24
Balanis, C.A., 5 K
Bhartia, P., 8 Kanaujia, B.K., 7
Burton, C., 1 Karlsson, A., 3
Kemp, D., 1
Keshmiri, F., 2, 23
C Komjani, M., 2, 5, 14
Cao, X.-Y., 2
Carver, K.R., 5 L
Chahavany, S., 2 Langley, R.J., 26–27
Chen, J.C., 3 Lee, C.K., 24, 26–27
Chen, S.Y., 2 Lee, Y.J., 2
Chiu, S.C., 2 Li, H., 1, 2
Clift, M., 1 Li, W.-Q., 2
Costa, F., 4, 9–10, 26 Lind, A.G., 8
Ling, J., 2
D Loui, H., 3
Damerell, W., 1 Lu, B., 2, 4
Derneryed, A.G., 8 Luo, X.F., 24, 25, 31
Diblanc, M., 24
M
F Ma, J.-J., 2
Foroozesh, A., 2, 23 Manara, G., 4, 9–10
Martel, C., 1
G Massey, S., 1
Genovesi, S., 2, 4 Mink, J.S., 5
Gong, X., 2 Mittra, R., 2
Gupta, G., 1 Monarchio, A., 4, 9–10
Gururaj, K., 1 Monavar, F.M., 2, 5, 14
Gustafsson, M., 3 Monédière, T., 24

DOI 10.1007/978-981-287-775-8
44 Author Index
N T
Nair, R.U., 4 Tayarani, M., 2
Nasri, J., 1, 24 Teo, P.T., 24
P V
Park, W.S., 2 Viswakarma, B.R., 7
Parker, E.A., 1 Volakis, J.L., 12, 16, 29, 33
Philippakis, M., 1
Pirhadi, A., 1, 2, 23–24 W
Prasad, K., 4 Wang, B.-Z., 1
Widenberg, B., 3
Q Wu, Q., 5, 8, 12–14
Qing, A., 24
Y
R Yang, H.-H., 2
Rebelo, A.P.P., 3 Yeo, J., 2
Rodes, E., 24 Yuan, H.W., 2
S Z
Shafai, L., 2, 24 Zheng, G., 1–2
Shao, W., 1 Zheng, Q.-R., 2
Stanton, P.H., 3
Subject Index
A J
Antennas, 1, 2, 4 Jerusalem crossed FSS (JC-FSS), 2, 5, 8–9, 11,
Artificial magnetic conductor, 2, 3, 5 16
B M
Bandwidth, 1–2, 4–5, 15, 16, 20, 23, 38 Microstrip patch antenna, 2, 5, 11, 19, 24, 25,
Beamwidth, 16, 20, 23 28, 38
C P
Capacitive FSS, 3 Perfect magnetic conductor (PMC), 5
Cavity model, 5, 7, 11, 24, 30, 38 Planar antennas, 1, 4
Complex relative permittivity, 9 Propagation constant, 28–29
D R
Directivity, 2, 23–24, 31, 33–38 Radar cross section (RCS), 1, 2, 4, 23, 38
Radiation characteristics, 2, 3, 23, 33
E Radiation pattern, 18, 28, 29, 33
Effective length, 6 Reciprocity theorem, 16, 20, 24, 28, 29, 33
Electromagnetic interference (EMI), 1 Rectangular MPA, 7, 8, 11, 15, 16, 20, 24, 30,
Equivalent circuit model, 2, 5, 10, 11–13, 20, 38
38 Reflection coefficient, 8, 10, 16, 29–30
Refractive index, 29–30
F Return loss, 10, 11, 15, 21, 22, 28, 30
Frequency selective surfaces (FSS), 1, 23
Fringing phenomenon, 5 S
Full-wave analysis method, 2, 5, 38 Single square loop, 5, 18, 20, 38
Superstrate, 1–2, 23–25, 27–31, 33–34
G
Ground plane, 1–5, 8, 11, 12, 15–19, 21, 28, 38 T
Thick filter, 3
H Thin filter, 3
High impedance ground plane, 1–2, 4, 8, 38 Transmission efficiency, 3, 31
High impedance surface (HIS), 9–11 Transmission line analogy, 16, 24, 29, 33
I
Inductive FSS, 3

DOI 10.1007/978-981-287-775-8

Frequency Selective Surfaces Based High Performance Microstrip Antenna

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What is the book about?

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What technology is discussed in the book?

SPRINGER BRIEFS IN ELECTRICAL AND COMPUTER

ENGINEERING  COMPUTATIONAL ELECTROMAGNETICS

Rakesh Mohan Jha

Frequency Selective Surfaces

ISSN 2191-8112 ISSN 2191-8120 (electronic)

Library of Congress Control Number: 2015947811

Springer Singapore Heidelberg New York Dordrecht London

Printed on acid-free paper

Dr. Rakesh Mohan Jha was a brilliant contributor to science, a won-

We would like to thank Mr. Shyam Chetty, Director, CSIR-National Aerospace

Frequency Selective Surfaces-Based High Performance

About the Book . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 41

Dr. Shiv Narayan is with the Centre for Electromagnetics of CSIR-National

INAE in 2010, for his contributions to the EM Applications to Aerospace

AMC Artiﬁcial magnetic conductor

ε0 Permittivity of free space

W1 Width of patch antenna

Figure 1 Typical FSS types and their frequency response

Figure 10 Return loss of rectangular MPA with FSS-HIS ground plane

Figure 29 Validation of a E-plane, and b H-plane pattern of rectan-

Abstract In order to fulﬁll the growing demand high-performance low RCS

Frequency selective surfaces (FSS) technology have potential applications in var-

© The Author(s) 2016 1

outside the operating bandwidth. In order to reduce the scattering component

2 Characteristics of FSS Structures

Fig. 1 Typical FSS types and (a) (b)

3 Microstrip Antenna Over FSS-Based High Impedance

3.1 Theoretical Considerations

The width W1 of the patch antenna is determined by

Leff ¼ L1 þ 2DL ð4Þ

where, L1 represents the actual length of the rectangular MPA.

where, ζ = 1.781072…, i = 1 is determined from Euler’s constant. d is the diameter

The lumped impedance offered by JC-FSS printed on lossy substrate can be

ZFSS ¼ Rs þ jXs ð16Þ

where, Rs and Xs is the resistance and reactance, respectively offered by FSS

The input impedance of a PEC-backed dielectric slab of thickness d is given as

ImfZd g ffi p0ffiffiffiffi0 tan k0 d e0r ð19Þ

The input impedance of the FSS-HIS structure is determined by the parallel

where, Γ is the reflection coefﬁcient of the proposed antenna given as

where, Zc is the characteristic impedance of coaxial feed line (50 Ω).

3.2 EM Design of Microstrip Patch Antenna Over FSS-HIS

3.2.1 EM Design of Microstrip Patch Antenna

In this work, a rectangular MPA is designed at the operating frequency of 10 GHz.

3.2.2 EM Design of FSS-Based HIS

3.3 EM Performance Analysis

3.3.1 Jerusalem Cross FSS-Based Microstrip Antenna

Validation of equivalent circuit approach: The EM analysis of microstrip

Fig. 7 Real and imaginary parts of input impedance of JC-FSS-based HIS

With PEC ground plane

by 9.6 % as compared to PEC ground plane. Thus, the Jerusalem crossed

PEC ground plane

210 -10 330

3.3.2 Square Loop FSS-Based Microstrip Antenna

The performance analysis of proposed microstrip antenna is further studied by

Fig. 14 Equivalent circuit of RD Ls Cs