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WO2015056491A1 - Power conversion device and power conversion method - Google Patents

Power conversion device and power conversion method Download PDF

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Publication number
WO2015056491A1
WO2015056491A1 PCT/JP2014/073171 JP2014073171W WO2015056491A1 WO 2015056491 A1 WO2015056491 A1 WO 2015056491A1 JP 2014073171 W JP2014073171 W JP 2014073171W WO 2015056491 A1 WO2015056491 A1 WO 2015056491A1
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WO
WIPO (PCT)
Prior art keywords
voltage
power
circuit
phase
voltage conversion
Prior art date
Application number
PCT/JP2014/073171
Other languages
French (fr)
Japanese (ja)
Inventor
祐輔 図子
ジョン シー クレア
アラン ワトソン
Original Assignee
日産自動車株式会社
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Application filed by 日産自動車株式会社 filed Critical 日産自動車株式会社
Priority to JP2015542538A priority Critical patent/JP6168155B2/en
Publication of WO2015056491A1 publication Critical patent/WO2015056491A1/en

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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/483—Converters with outputs that each can have more than two voltages levels
    • H02M7/49—Combination of the output voltage waveforms of a plurality of converters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/4807—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode having a high frequency intermediate AC stage
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/4815—Resonant converters
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes

Definitions

  • the present invention relates to a power conversion device and a power conversion method for converting DC power output from a DC power source into AC power.
  • a distributed module power supply has been proposed as a power supply device for driving a load such as an AC motor (see Patent Document 1).
  • the distributed module power supply includes a plurality of DC power supplies, and converts the DC voltage output from each DC power supply into an AC voltage using an inverter. And the alternating voltage output from each inverter is added in series, the alternating voltage of a desired level is produced
  • a conventional power conversion device requires a large-capacitance capacitor as a smoothing capacitor, which increases the overall size of the device.
  • the present invention can cancel the ripple currents of the respective phases, so that the smoothing capacitor can be reduced and the apparatus can be reduced in size and power conversion method.
  • the purpose is to provide.
  • the power conversion device is a power conversion device that converts a DC voltage into an AC voltage and supplies the converted AC voltage to a load having a plurality of phases, the plurality of DC power supplies, A smoothing capacitor connected in parallel to each of a plurality of DC power supplies and a number of phases corresponding to each of the plurality of DC power supplies so as to cancel each other out ripple currents generated when converting DC voltage to AC voltage
  • a plurality of voltage conversion means for converting a DC voltage from the connected DC power supply to an AC voltage; a control means for controlling the plurality of voltage conversion means; and an output of the voltage conversion means connected to a different DC power supply.
  • a plurality of output terminals that are connected in series, add the AC voltage from each voltage conversion means, and output to each phase of the load, respectively.
  • a smoothing capacitor is connected in parallel to each of a plurality of DC power supplies, and a plurality of voltage conversion means are connected in parallel to each of the plurality of DC power supplies by the number of the plurality of phases.
  • the output of the voltage conversion means connected to the different DC power supply is connected in series, the DC voltage is converted into an AC voltage, and the power converter that supplies the converted AC voltage to a load having a plurality of phases is used.
  • a power conversion method in which a plurality of voltage conversion means converts a DC voltage from a plurality of DC power supplies into an AC voltage, and an AC voltage from a voltage conversion means connected to a different DC power supply is added in series. Output to each phase of the load, and for each voltage conversion means connected in parallel, cancel each other ripple currents generated when the DC voltage is converted into the AC voltage. And wherein the door.
  • FIGS. 7A to 7C are timing charts showing an example of the ripple current waveform of each phase, and FIG.
  • FIG. 7D shows an example of the waveform obtained by adding the ripple current of each phase. It is a timing chart. It is a graph showing the relationship between the frequency and gain which concern on embodiment of this invention. It is the schematic for demonstrating the feedforward control of the DC / DC converter which concerns on embodiment of this invention. It is a timing chart which shows an example of the signal waveform in PWM of the level shift system which concerns on the 1st modification of embodiment of this invention. It is a circuit diagram which shows an example of the power converter device which concerns on the 2nd modification of embodiment of this invention. It is a circuit diagram which shows an example of the DC / DC converter which concerns on the 3rd modification of embodiment of this invention. It is a circuit diagram which shows an example of the DC / DC converter which concerns on the 4th modification of embodiment of this invention. It is a circuit diagram which shows an example of the DC / DC converter which concerns on the 5th modification of embodiment of this invention.
  • FIG. 1 a device that drives a motor M1 using a load as a three-phase AC motor (hereinafter simply referred to as “motor”) M1 as a load.
  • motor a three-phase AC motor
  • the power conversion device according to the embodiment of the present invention drives motor M1 by supplying AC voltages that are 120 ° different from each other to each of a plurality of phases (U phase, V phase, W phase) of motor M1. .
  • the power conversion device is arranged in a row direction and a column direction with a plurality (n; n is a natural number of 2 or more) of DC power supplies VB1, VB2, VB3.
  • a control device (control means) 31 for controlling the operation.
  • a plurality (n) of voltage conversion modules 11-1, 11-2,... 11-n arranged in the same column supply an AC voltage to the U phase.
  • a plurality (n) of voltage conversion modules 12-1, 12-2,... 12-n arranged in the same column supply an AC voltage to the V phase.
  • a plurality (n) of voltage conversion modules 13-1, 13-2,... 13-n arranged in the same column supply an AC voltage to the W phase.
  • the voltage conversion modules 11-1, 12-1, and 13-1 arranged in the same row are connected in parallel to the DC power supply VB1 and commonly use the DC power supply VB1.
  • the voltage conversion modules 11-2, 12-2, and 13-2 arranged in the same row are connected in parallel to the DC power supply VB2, and use the DC power supply VB2 in common.
  • the voltage conversion modules 11-n, 12-n, 13-n arranged in the same row are connected in parallel to the DC power source VBn, and commonly use the DC power source VBn.
  • the uppermost voltage conversion modules 11-1, 12-1, and 13-1 are set to the high potential side, and the lowermost voltage conversion modules 11-n, 12-n, and 13-n are set to the low potential side.
  • the potential is set step by step.
  • the output terminals of the lowest voltage conversion modules 11-n, 12-n, 13-n are connected to the reference potential.
  • the outputs of the voltage conversion modules 11-1 to 11-n arranged in the same column are connected in series.
  • the AC voltages output from the voltage conversion modules 11-1 to 11-n are added in series and output to the U phase via the output terminal N1.
  • the outputs of the voltage conversion modules 12-1 to 12-n arranged in the same column are connected in series.
  • the AC voltages output from the voltage conversion modules 12-1 to 12-n are added in series and output to the V phase via the output terminal N2.
  • the outputs of the voltage conversion modules 13-1 to 13-n arranged in the same column are connected in series.
  • the AC voltages output from the voltage conversion modules 13-1 to 13-n are added in series and output to the W phase via the output terminal N3.
  • one voltage conversion module 11-1 among the voltage conversion modules 11-1 to 11-n, 12-1 to 12-n, and 13-1 to 13-n will be described as a representative.
  • the other voltage conversion modules 11-2 to 11-n, 12-1 to 12-n, and 13-1 to 13-n have the same configuration as that of the voltage conversion module 11-1.
  • the voltage conversion module 11-1 includes a DC / DC converter (transformer means) 21 whose input side is connected to the DC power source VB1, an inverter circuit 22 whose input side is connected to the DC / DC converter 21, and a positive electrode of the DC power source VB1.
  • a harmonic removing capacitor C1 connected between the negative electrodes and a smoothing capacitor C2 connected between the DC / DC converter 21 and the inverter circuit 22 are provided.
  • the harmonic removing capacitor C1 removes harmonics of the DC voltage from the DC power supply VB1.
  • the smoothing capacitor C ⁇ b> 2 suppresses voltage fluctuations that occur due to the switching operation of the inverter circuit 22 and smoothes the output voltage of the DC / DC converter 21.
  • the smoothing capacitor C2 may not be included in the voltage conversion module 11-1, but may be provided separately from the voltage conversion module 11-1.
  • the DC / DC converter 21 boosts or steps down the DC voltage from the DC power supply VB1.
  • the DC / DC converter 21 includes a primary circuit (full bridge (H bridge) circuit) 21a having four switching elements Q11 to Q14 and four switching elements Q21 to Q24, as shown in FIG.
  • Such a DC / DC converter 21 is also called a dual active bridge (DAB) circuit.
  • DAB dual active bridge
  • the switching elements Q11 to Q14 and Q21 to Q24 are constituted by, for example, insulated gate bipolar transistors (IGBT).
  • IGBT insulated gate bipolar transistors
  • the transformer TR1 is an insulating transformer that boosts or steps down the DC voltage from the primary circuit 21a and transmits it to the secondary circuit 21b, and insulates the primary circuit 21a and the secondary circuit 21b.
  • the transformer TR1 has a transformation ratio of m: n (m and n are integers).
  • the transformer TR1 may have a transformation ratio of 1: 1. In other words, the DC voltage may be transmitted as it is without being transformed.
  • the DC / DC converter 21 is an insulation type converter having an insulation transformer TR1
  • the control device 31 includes a primary side voltage detection unit 33, a secondary side voltage detection unit 34, a secondary side current detection unit 36, a main control unit 32, and a drive circuit 35.
  • the primary side voltage detector 33 detects the voltage V1 on the primary circuit 21a side and outputs it to the main controller 32.
  • the secondary side voltage detector 34 detects the voltage V ⁇ b> 2 on the secondary circuit 21 b side and outputs it to the main controller 32.
  • the main control unit 32 is configured as an integrated computer including a central processing unit (CPU) and storage means such as a RAM, a ROM, and a hard disk.
  • CPU central processing unit
  • storage means such as a RAM, a ROM, and a hard disk.
  • the main control unit 32 observes the input voltage V1 and the output voltage V2 so as to follow the command voltage value (target output voltage value) Vref by the master controller 41, which is the host device, and switches the switching elements Q11 to Q14, Q21 to Q24.
  • the on / off command signal is output to the drive circuit 35.
  • the drive circuit 35 outputs drive signals to the control terminals (bases) of the switching elements Q11 to Q14 and Q21 to Q24 based on the on / off command signal from the main control unit 32.
  • the switching elements Q11 and Q14 on the primary circuit 21a side of the DC / DC converter 21 and the switching elements Q12 and Q13 are alternately turned on and off periodically based on a drive signal from the drive circuit 35. Further, the switching elements Q21 and Q24 on the secondary circuit 21b side of the DC / DC converter 21 and the switching elements Q22 and Q23 are alternately turned on and off periodically based on a drive signal from the drive circuit 35.
  • the inverter circuit 22 is a full bridge (H bridge) circuit having four switching elements Q31 to Q34.
  • the switching elements Q31 to Q34 are composed of, for example, an IGBT.
  • a diode is connected between the two terminals of the switching elements Q31 to Q34.
  • the output of the inverter circuit 22 of the voltage conversion module 11-1 is connected in series with the output of the inverter circuits of the voltage conversion modules 11-2 to 11-n arranged in the same column as the voltage conversion module 11-1. .
  • the inverter circuit 22 is a PWM circuit that turns on / off the switching elements Q31 to Q34 by performing phase shift type pulse width modulation (PWM).
  • PWM phase shift type pulse width modulation
  • the drive signals are compared by comparing the signal values of the carrier waves W1 to W6 and the signal wave (sine wave) W0 while shifting the phases of the carrier waves (triangular waves) W1 to W6. Is generated.
  • the switching elements Q31 and Q34 are turned on when the carrier waves W1 to W6 are smaller than the signal wave W0, and the switching elements Q32 and Q33 are turned on when the carrier waves W1 to W6 are larger than the signal wave W0.
  • the switching frequency of the DC / DC converter 21 is set higher than the switching frequency of the inverter circuit 22.
  • the ripple current Iu As shown in FIG. 4, in the voltage conversion module 11-1, voltage fluctuation occurs due to the switching operation of the inverter circuit 22 and the like, and a ripple current Iu corresponding to the voltage fluctuation that cannot be absorbed by the smoothing capacitor C2 flows.
  • the ripple current Iu is transmitted from the secondary circuit 21b side to the primary circuit 21a side via the transformer TR1.
  • the ripple current Iu has a pulse waveform as shown in FIG.
  • the fundamental frequency of the ripple waveform of the ripple current is the switching frequency of the inverter circuit 22.
  • the amplitude of the ripple waveform of the ripple current oscillates (changes) at the phase current frequency.
  • the maximum value of the amplitude of the ripple waveform of the ripple current is the maximum value Imax of the phase current.
  • the ripple currents Iu, Iv, and Iw are transmitted from the secondary circuit 21b side to the transformer TR1. Are respectively transmitted to the primary circuit 21a side.
  • the periods are 120 ° out of phase with each other.
  • the DC power supply VB1 is shared by the U-phase, V-phase, and W-phase voltage conversion modules 11-1, 12-1, and 13-1, as shown in FIG.
  • the phase current fundamental wave components of the ripple currents Iu, Iv, and Iw flowing in the respective phases are added and canceled.
  • the switching frequency fs of the inverter circuit 22 is always higher than the maximum value fp of the phase current frequency.
  • the ripple currents Iu, Iv, and Iw themselves contain a higher order frequency than the switching frequency of the inverter circuit 22, but transmit the phase current frequency component from the secondary circuit 21b side of the DC / DC converter 21 to the primary circuit 21a side. Includes only the fundamental wave component of the switching frequency fs of the inverter circuit 22. Therefore, the cut-off frequency of the DC / DC converter 21 may be equal to or higher than the switching frequency fs of the inverter circuit 22. Therefore, the voltage control band of the DC / DC converter 21 is set to a frequency higher than the switching frequency fs of the inverter circuit 22.
  • the final output of the main control unit 32 is a PWM signal.
  • the PWM signal is a square wave with a duty ratio of approximately 50%, the phase of which is shifted by ⁇ on the primary circuit 21a side and the secondary circuit 21b side.
  • the PI controller included in the main control unit 32 is configured so that the voltage V2 on the secondary circuit 21b side follows the command voltage value Vref while referring to the voltage V2 on the secondary circuit 21b side and the voltage V1 on the primary circuit 21a side.
  • the phase difference ⁇ is controlled.
  • the average value of current i 2 of the secondary circuit 21b side can be obtained by the following equation (1).
  • phase difference (shift amount) ⁇ for achieving the actual current i 2 on the secondary circuit 21b side can be obtained by the following equation (2).
  • the nominal value i 2 (nom) of the current on the secondary circuit 21b side is used as in equations (3) and (4), and the phase shift amount ⁇ (nom) is set.
  • a PWM signal is generated by adding the phase shift amount ⁇ (nom) obtained by the FF control term to the output of the PI controller.
  • the plurality of voltage conversion modules 11-1 to 11-n, 12-1 to 12-n, and 13-1 to 13-n shown in FIG. 1 convert the DC voltages from the plurality of DC power sources VB1 to VBn to AC. Convert to voltage. Specifically, in each of the voltage conversion modules 11-1 to 11-n, 12-1 to 12-n, and 13-1 to 13-n, the DC / DC converter 21 is based on a control signal from the control device 31. By switching the switching elements Q11 to Q14 and Q21 to Q24, the DC voltage from the DC power sources VB1 to VBn is transformed. Further, the inverter circuit 22 converts the DC voltage output from the DC / DC converter 21 into an AC voltage by switching the switching elements Q31 to Q34 based on a control signal from the control device 31.
  • AC voltages output from the voltage conversion modules 11-1 to 11-n are added in series and output to the U phase. Also, the AC voltages output from the voltage conversion modules 12-1 to 12-n are added in series and output to the V phase. Further, AC voltages output from the voltage conversion modules 13-1 to 13-n are added in series and output to the U phase.
  • the U-phase, V-phase, and W-phase voltage conversion modules 11-1, 12-1, and 13-1 arranged in the same row are connected in common to the DC power supply VB1, so that the ripple current of each phase is obtained.
  • the phase current fundamental wave components of Iu, Iv, and Iw can be added together and canceled.
  • the phase current fundamental wave components of the ripple currents Iu, Iv, and Iw of each phase can be added and canceled.
  • the U-phase, V-phase, and W-phase voltage conversion modules 11-1, 12-1, and 13-1 arranged in the same row are connected to the DC power supply VB1.
  • the phase current fundamental wave components of the ripple currents Iu, Iv, Iw of each phase can be added together and canceled. Therefore, in each of the voltage conversion modules 11-1 to 11-n, 12-1 to 12-n, and 13-1 to 13-n, the smoothing capacitor C2 can be reduced, and the overall size of the device can be reduced. be able to.
  • the U-phase, V-phase, and W-phase voltage conversion modules 11-1, 12-1, and 13-1 arranged in the same row are commonly connected to the DC power supply VB1, thereby reducing the power between the phases. Even if the balance occurs, the energy consumption of the DC power supply VB1 can be made uniform.
  • the voltage conversion modules 11-1, 12-1, 13-1 of each phase can be insulated from each other, and the voltage conversion modules 11-1, 12-1 and 13-1 can be connected to a common DC power supply VB1.
  • the use of DAB as the DC / DC converter 21 can reduce the size and increase the efficiency of the apparatus. Further, by making the switching frequency of DAB higher than the switching frequency of the inverter circuit 22, the responsiveness of DAB can be improved. Further, by making the DAB cutoff frequency higher than the switching frequency of the inverter circuit 22, the responsiveness of the DAB can be improved. Further, DAB responsiveness can be improved by feedforward control of DAB. Further, by controlling the inverter circuit 22 with the phase shift type PWM, the energy consumption of all the batteries can be made uniform.
  • the inverter circuit 22 according to the first modification is controlled by level shift type PWM.
  • level shift method as shown in FIG. 10, voltage levels of carrier waves (triangular waves) W1 to W6 are modulated.
  • the magnitudes of the carrier waves W1 to W6 and the signal wave (sine wave) W0 on / off of the switching elements Q31 to Q34 of the inverter circuit 22 is controlled.
  • the first modification high efficiency can be achieved by controlling the inverter circuit 22 with level shift PWM. Further, since the number of turn-offs and ons is small even at the same switching frequency as compared with the phase shift method, the same effect as the embodiment of the present invention can be obtained even with a low response DC / DC converter (DAB).
  • DAB low response DC / DC converter
  • the inverter circuit 22 (Second modification) Another example of the inverter circuit 22 will be described as a second modification. As shown in FIG. 11, the inverter circuit 22 according to the second modification is different in that it includes a half bridge circuit having two switching elements Q31 and Q32 instead of the H bridge circuit. Further, in order to generate a negative voltage, DC power sources VB (n ⁇ 1) and VBn on the reference potential side are connected in the reverse direction. Also in the second modification, for example, phase shift type PWM control is possible as in the embodiment of the present invention.
  • each of the voltage conversion modules 11-1 to 11-n, 12-1 to 12-n is compared with the case of the H-bridge circuit. , 13-1 to 13-n, the number of switching elements can be reduced by two, and the circuit configuration can be simplified.
  • the DC / DC converter 21 according to the third modified example uses a metal oxide semiconductor field effect transistor (MOSFET) instead of the IGBT as the switching elements Q11 to Q14 and Q21 to Q24.
  • MOSFET metal oxide semiconductor field effect transistor
  • the MOSFET has a built-in output capacitance.
  • soft switching can be performed using the built-in output capacitance of the MOSFET without providing capacitors at both ends of the IGBT as shown in FIG. 1, and noise generated during switching can be suppressed. Can do.
  • the DC / DC converter 21 according to the fourth modification is different in that a capacitor is connected in parallel to the IGBT in each of the switching elements Q11 to Q14 and Q21 to Q24.
  • soft switching can be performed by using the capacitors of the switching elements Q11 to Q14 and Q21 to Q24, and noise generated during switching can be suppressed.
  • the DC / DC converter 21 may have an LC series resonance type primary circuit 21a.
  • the DC / DC converter 21 may include an LLC series resonance type primary circuit 21a.
  • the DC / DC converter 21 may have the LCC series resonance type primary circuit 21a.
  • the DC / DC converter 21 may have the LC parallel resonance type primary circuit 21a.
  • the present invention is not limited to this, and a single-phase AC voltage is generated. It is also applicable to.
  • the configuration in which the power conversion apparatus includes the DC / DC converter 21 has been described.
  • the DC / DC converter 21 may not necessarily be included.
  • the harmonic component removing capacitor C1 is not necessarily provided.
  • the present invention can be used to reduce the smoothing capacitor of the power conversion device and to reduce the size of the device.

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  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Inverter Devices (AREA)
  • Dc-Dc Converters (AREA)

Abstract

Provided is a power conversion device in which the size of a smoothing capacitor can be reduced since ripple currents in the respective phases can be compensated each other, so that the size of the device can be reduced. The power conversion device comprises: a plurality of DC power supplies (VB1 to VBn); a smoothing capacitor (C2) connected to each of the DC power supplies (VB1 to VBn) in parallel; a plurality of inverter circuits (22) connected to each of the DC power supplies (VB1 to VBn) by the number of phases in parallel so that the ripple currents occurring upon DC to AC voltage conversion are compensated each other and converting the DC voltages supplied from the connected DC power supplies (VB1 to VBn) to AC voltages; and a control means (31) for controlling the inverter circuits (22). In the power conversion device, the outputs of the inverter circuits (22) connected to the different DC power supplies (VB1 to VBn) are connected in series and output to each phase of a load.

Description

電力変換装置及び電力変換方法Power conversion device and power conversion method
　本発明は、直流電源より出力される直流電力を交流電力に変換する電力変換装置及び電力変換方法に関する。 The present invention relates to a power conversion device and a power conversion method for converting DC power output from a DC power source into AC power.
　交流モータ等の負荷を駆動するための電源装置として、分散モジュール電源が提案されている（特許文献１参照。）。分散モジュール電源では、複数の直流電源を備え、各直流電源から出力される直流電圧をインバータを用いて交流電圧に変換する。そして、各インバータから出力される交流電圧を直列的に加算することにより所望のレベルの交流電圧を生成し、負荷に供給する。 A distributed module power supply has been proposed as a power supply device for driving a load such as an AC motor (see Patent Document 1). The distributed module power supply includes a plurality of DC power supplies, and converts the DC voltage output from each DC power supply into an AC voltage using an inverter. And the alternating voltage output from each inverter is added in series, the alternating voltage of a desired level is produced | generated, and it supplies to load.
特開２０１１－１５５７８６号公報JP 2011-155786 A
　従来の電力変換装置を三相交流モータの駆動回路として用いる場合には、駆動回路の動作に伴って生じる電圧変動を抑制し、出力電圧を平滑化するための平滑用コンデンサを設ける必要がある。従来の電力変換装置では、平滑用コンデンサとして大容量のコンデンサが必要であり、装置全体のサイズが大きくなる。 When using a conventional power converter as a drive circuit for a three-phase AC motor, it is necessary to provide a smoothing capacitor for smoothing the output voltage by suppressing voltage fluctuations caused by the operation of the drive circuit. A conventional power conversion device requires a large-capacitance capacitor as a smoothing capacitor, which increases the overall size of the device.
　上記問題点を鑑み、本発明は、各相のリプル電流を互いに相殺することができるので、平滑用コンデンサを小さくすることができ、装置の小型化を図ることができる電力変換装置及び電力変換方法を提供することを目的とする。 In view of the above problems, the present invention can cancel the ripple currents of the respective phases, so that the smoothing capacitor can be reduced and the apparatus can be reduced in size and power conversion method. The purpose is to provide.
　本発明の第一の態様に係る電力変換装置は、直流電圧を交流電圧に変換し、変換された交流電圧を複数相を有する負荷に供給する電力変換装置であって、複数の直流電源と、複数の直流電源のそれぞれに並列に接続された平滑用コンデンサと、直流電圧を交流電圧に変換する際に生じるリプル電流を互いに相殺するように複数の直流電源のそれぞれに複数相の数だけ並列に接続され、接続された直流電源からの直流電圧を交流電圧に変換する複数の電圧変換手段と、複数の電圧変換手段を制御する制御手段と、異なる直流電源に接続された電圧変換手段の出力を直列に接続し、各電圧変換手段からの交流電圧を加算して負荷の各相へそれぞれ出力する複数の出力端子とを備えることを特徴とする。 The power conversion device according to the first aspect of the present invention is a power conversion device that converts a DC voltage into an AC voltage and supplies the converted AC voltage to a load having a plurality of phases, the plurality of DC power supplies, A smoothing capacitor connected in parallel to each of a plurality of DC power supplies and a number of phases corresponding to each of the plurality of DC power supplies so as to cancel each other out ripple currents generated when converting DC voltage to AC voltage A plurality of voltage conversion means for converting a DC voltage from the connected DC power supply to an AC voltage; a control means for controlling the plurality of voltage conversion means; and an output of the voltage conversion means connected to a different DC power supply. A plurality of output terminals that are connected in series, add the AC voltage from each voltage conversion means, and output to each phase of the load, respectively.
　本発明の第二の態様に係る電力変換方法は、複数の直流電源のそれぞれに平滑用コンデンサが並列に接続され、複数の直流電源のそれぞれに複数の電圧変換手段が複数相の数だけ並列に接続され、異なる直流電源に接続された電圧変換手段の出力が直列に接続され、直流電圧を交流電圧に変換し、変換された交流電圧を複数相を有する負荷に供給する電力変換装置を用いた電力変換方法であって、複数の電圧変換手段が、複数の直流電源からの直流電圧を交流電圧に変換するステップと、異なる直流電源に接続された電圧変換手段からの交流電圧を直列的に加算して、負荷の各相へそれぞれ出力するステップと、並列に接続された電圧変換手段毎に、直流電圧を交流電圧に変換する際に生じるリプル電流を互いに相殺するステップとを含むことを特徴とする。 In the power conversion method according to the second aspect of the present invention, a smoothing capacitor is connected in parallel to each of a plurality of DC power supplies, and a plurality of voltage conversion means are connected in parallel to each of the plurality of DC power supplies by the number of the plurality of phases. The output of the voltage conversion means connected to the different DC power supply is connected in series, the DC voltage is converted into an AC voltage, and the power converter that supplies the converted AC voltage to a load having a plurality of phases is used. A power conversion method in which a plurality of voltage conversion means converts a DC voltage from a plurality of DC power supplies into an AC voltage, and an AC voltage from a voltage conversion means connected to a different DC power supply is added in series. Output to each phase of the load, and for each voltage conversion means connected in parallel, cancel each other ripple currents generated when the DC voltage is converted into the AC voltage. And wherein the door.
本発明の実施の形態に係る電力変換装置の一例を示す回路図である。It is a circuit diagram which shows an example of the power converter device which concerns on embodiment of this invention. 本発明の実施の形態に係るＤＣ／ＤＣコンバータ及び制御装置の一例を示す回路図である。It is a circuit diagram showing an example of a DC / DC converter and a control device concerning an embodiment of the invention. 本発明の実施の形態に係る電力変換装置の位相シフト方式のＰＷＭにおける信号波形の一例を示すタイミングチャートである。It is a timing chart which shows an example of the signal waveform in PWM of the phase shift system of the power converter device which concerns on embodiment of this invention. 本発明の実施の形態に係るリプル電流について説明するための回路図である。It is a circuit diagram for demonstrating the ripple current which concerns on embodiment of this invention. 本発明の実施の形態に係るリプル電流の波形の一例を示すタイミングチャートである。It is a timing chart which shows an example of the waveform of the ripple current concerning an embodiment of the invention. 本発明の実施の形態に係るリプル電流について説明するための概略図である。It is the schematic for demonstrating the ripple current which concerns on embodiment of this invention. 図７（ａ）～図７（ｃ）は、各相のリプル電流の波形の一例を示すタイミングチャートであり、図７（ｄ）は、各相のリプル電流を足し合わせた波形の一例を示すタイミングチャートである。FIGS. 7A to 7C are timing charts showing an example of the ripple current waveform of each phase, and FIG. 7D shows an example of the waveform obtained by adding the ripple current of each phase. It is a timing chart. 本発明の実施の形態に係る周波数とゲインとの関係を表すグラフである。It is a graph showing the relationship between the frequency and gain which concern on embodiment of this invention. 本発明の実施の形態に係るＤＣ／ＤＣコンバータのフィードフォワード制御を説明するための概略図である。It is the schematic for demonstrating the feedforward control of the DC / DC converter which concerns on embodiment of this invention. 本発明の実施の形態の第１の変形例に係るレベルシフト方式のＰＷＭにおける信号波形の一例を示すタイミングチャートである。It is a timing chart which shows an example of the signal waveform in PWM of the level shift system which concerns on the 1st modification of embodiment of this invention. 本発明の実施の形態の第２の変形例に係る電力変換装置の一例を示す回路図である。It is a circuit diagram which shows an example of the power converter device which concerns on the 2nd modification of embodiment of this invention. 本発明の実施の形態の第３の変形例に係るＤＣ／ＤＣコンバータの一例を示す回路図である。It is a circuit diagram which shows an example of the DC / DC converter which concerns on the 3rd modification of embodiment of this invention. 本発明の実施の形態の第４の変形例に係るＤＣ／ＤＣコンバータの一例を示す回路図である。It is a circuit diagram which shows an example of the DC / DC converter which concerns on the 4th modification of embodiment of this invention. 本発明の実施の形態の第５の変形例に係るＤＣ／ＤＣコンバータの一例を示す回路図である。It is a circuit diagram which shows an example of the DC / DC converter which concerns on the 5th modification of embodiment of this invention.
　次に、図面を参照して、本発明の実施の形態を説明する。以下の図面の記載において、同一又は類似の部分には同一又は類似の符号を付している。 Next, an embodiment of the present invention will be described with reference to the drawings. In the following description of the drawings, the same or similar parts are denoted by the same or similar reference numerals.
　本発明の実施の形態に係る電力変換装置として、図１に示すように、負荷である三相交流モータ（以下、単に「モータ」という。）Ｍ１を負荷として、モータＭ１を駆動する装置を一例として説明する。本発明の実施の形態に係る電力変換装置は、モータＭ１の複数相（Ｕ相、Ｖ相、Ｗ相）のそれぞれに、位相が互いに１２０°異なる交流電圧を供給することによりモータＭ１を駆動する。 As an example of the power conversion device according to the embodiment of the present invention, as shown in FIG. 1, a device that drives a motor M1 using a load as a three-phase AC motor (hereinafter simply referred to as “motor”) M1 as a load. Will be described. The power conversion device according to the embodiment of the present invention drives motor M1 by supplying AC voltages that are 120 ° different from each other to each of a plurality of phases (U phase, V phase, W phase) of motor M1. .
　本発明の実施の形態に係る電力変換装置は、図１に示すように、複数（ｎ個；ｎは２以上の自然数）の直流電源ＶＢ１，ＶＢ２，ＶＢ３と、行方向及び列方向に配列され、モータＭ１に供給するための交流電圧をそれぞれ出力する複数（３×ｎ個）の電圧変換モジュール（電圧変換手段）１１－１，１１－２，…，１１－ｎ，１２－１，１２－２，…，１２－ｎ，１３－１，１３－２，…，１３－ｎと、電圧変換モジュール１１－１～１１－ｎ，１２－１～１２－ｎ，１３－１～１３－ｎの動作を制御する制御装置（制御手段）３１とを備える。 As shown in FIG. 1, the power conversion device according to the embodiment of the present invention is arranged in a row direction and a column direction with a plurality (n; n is a natural number of 2 or more) of DC power supplies VB1, VB2, VB3. A plurality of (3 × n) voltage conversion modules (voltage conversion means) 11-1, 11-2,..., 11-n, 12-1, 12- that respectively output AC voltages to be supplied to the motor M1. 2, ..., 12-n, 13-1, 13-2, ..., 13-n and voltage conversion modules 11-1 to 11-n, 12-1 to 12-n, 13-1 to 13-n And a control device (control means) 31 for controlling the operation.
　同一の列に配列された複数（ｎ個）の電圧変換モジュール１１－１，１１－２，…１１－ｎは、Ｕ相に交流電圧を供給する。また、同一の列に配列された複数（ｎ個）の電圧変換モジュール１２－１，１２－２，…１２－ｎは、Ｖ相に交流電圧を供給する。同一の列に配列された複数（ｎ個）の電圧変換モジュール１３－１，１３－２，…１３－ｎは、Ｗ相に交流電圧を供給する。 A plurality (n) of voltage conversion modules 11-1, 11-2,... 11-n arranged in the same column supply an AC voltage to the U phase. In addition, a plurality (n) of voltage conversion modules 12-1, 12-2,... 12-n arranged in the same column supply an AC voltage to the V phase. A plurality (n) of voltage conversion modules 13-1, 13-2,... 13-n arranged in the same column supply an AC voltage to the W phase.
　同一の行に配列された電圧変換モジュール１１－１，１２－１，１３－１は、直流電源ＶＢ１に並列に接続され、直流電源ＶＢ１を共通で使用する。また、同一の行に配列された電圧変換モジュール１１－２，１２－２，１３－２は、直流電源ＶＢ２に並列に接続され、直流電源ＶＢ２を共通で使用する。また、同一の行に配列された電圧変換モジュール１１－ｎ，１２－ｎ，１３－ｎは、直流電源ＶＢｎに並列に接続され、直流電源ＶＢｎを共通で使用する。 The voltage conversion modules 11-1, 12-1, and 13-1 arranged in the same row are connected in parallel to the DC power supply VB1 and commonly use the DC power supply VB1. In addition, the voltage conversion modules 11-2, 12-2, and 13-2 arranged in the same row are connected in parallel to the DC power supply VB2, and use the DC power supply VB2 in common. Further, the voltage conversion modules 11-n, 12-n, 13-n arranged in the same row are connected in parallel to the DC power source VBn, and commonly use the DC power source VBn.
　行方向において最上段の電圧変換モジュール１１－１，１２－１，１３－１側を高電位側とし、最下段の電圧変換モジュール１１－ｎ，１２－ｎ，１３－ｎ側を低電位側として、段階的に電位が設定されている。最下段の電圧変換モジュール１１－ｎ，１２－ｎ，１３－ｎの出力端子が基準電位に接続される。 In the row direction, the uppermost voltage conversion modules 11-1, 12-1, and 13-1 are set to the high potential side, and the lowermost voltage conversion modules 11-n, 12-n, and 13-n are set to the low potential side. The potential is set step by step. The output terminals of the lowest voltage conversion modules 11-n, 12-n, 13-n are connected to the reference potential.
　同一の列に配列された電圧変換モジュール１１－１～１１－ｎの出力は直列に接続されている。電圧変換モジュール１１－１～１１－ｎから出力される交流電圧は直列的に加算されて、出力端子Ｎ１を介してＵ相へ出力される。また、同一の列に配列された電圧変換モジュール１２－１～１２－ｎの出力は直列に接続されている。電圧変換モジュール１２－１～１２－ｎから出力される交流電圧は直列的に加算されて、出力端子Ｎ２を介してＶ相へ出力される。また、同一の列に配列された電圧変換モジュール１３－１～１３－ｎの出力は直列に接続されている。電圧変換モジュール１３－１～１３－ｎから出力される交流電圧は直列的に加算されて、出力端子Ｎ３を介してＷ相へ出力される。 The outputs of the voltage conversion modules 11-1 to 11-n arranged in the same column are connected in series. The AC voltages output from the voltage conversion modules 11-1 to 11-n are added in series and output to the U phase via the output terminal N1. The outputs of the voltage conversion modules 12-1 to 12-n arranged in the same column are connected in series. The AC voltages output from the voltage conversion modules 12-1 to 12-n are added in series and output to the V phase via the output terminal N2. The outputs of the voltage conversion modules 13-1 to 13-n arranged in the same column are connected in series. The AC voltages output from the voltage conversion modules 13-1 to 13-n are added in series and output to the W phase via the output terminal N3.
　以下では、電圧変換モジュール１１－１～１１－ｎ，１２－１～１２－ｎ，１３－１～１３－ｎのうち、１つの電圧変換モジュール１１－１の構成を代表して説明する。なお、他の電圧変換モジュール１１－２～１１－ｎ，１２－１～１２－ｎ，１３－１～１３－ｎも、電圧変換モジュール１１－１の構成と同様の構成を有する。 Hereinafter, the configuration of one voltage conversion module 11-1 among the voltage conversion modules 11-1 to 11-n, 12-1 to 12-n, and 13-1 to 13-n will be described as a representative. The other voltage conversion modules 11-2 to 11-n, 12-1 to 12-n, and 13-1 to 13-n have the same configuration as that of the voltage conversion module 11-1.
　電圧変換モジュール１１－１は、直流電源ＶＢ１に入力側が接続されたＤＣ／ＤＣコンバータ（変圧手段）２１と、ＤＣ／ＤＣコンバータ２１に入力側が接続されたインバータ回路２２と、直流電源ＶＢ１の正極と負極の間に接続された高調波除去用コンデンサＣ１と、ＤＣ／ＤＣコンバータ２１とインバータ回路２２との間に接続された平滑用コンデンサＣ２とを備える。高調波除去用コンデンサＣ１は、直流電源ＶＢ１からの直流電圧の高調波を除去する。平滑用コンデンサＣ２は、インバータ回路２２のスイッチング動作等に伴い発生する電圧変動を抑制し、ＤＣ／ＤＣコンバータ２１の出力電圧を平滑化する。なお、平滑用コンデンサＣ２は、電圧変換モジュール１１－１には含まれずに電圧変換モジュール１１－１とは個別に設けられていてもよい。 The voltage conversion module 11-1 includes a DC / DC converter (transformer means) 21 whose input side is connected to the DC power source VB1, an inverter circuit 22 whose input side is connected to the DC / DC converter 21, and a positive electrode of the DC power source VB1. A harmonic removing capacitor C1 connected between the negative electrodes and a smoothing capacitor C2 connected between the DC / DC converter 21 and the inverter circuit 22 are provided. The harmonic removing capacitor C1 removes harmonics of the DC voltage from the DC power supply VB1. The smoothing capacitor C <b> 2 suppresses voltage fluctuations that occur due to the switching operation of the inverter circuit 22 and smoothes the output voltage of the DC / DC converter 21. The smoothing capacitor C2 may not be included in the voltage conversion module 11-1, but may be provided separately from the voltage conversion module 11-1.
　ＤＣ／ＤＣコンバータ２１は、直流電源ＶＢ１からの直流電圧を昇圧又は降圧する。ＤＣ／ＤＣコンバータ２１は、図１及び詳細には図２に示すように、４つのスイッチング素子Ｑ１１～Ｑ１４を有する一次回路（フルブリッジ（Ｈブリッジ）回路）２１ａと、４つのスイッチング素子Ｑ２１～Ｑ２４を有する二次回路（Ｈブリッジ回路）２１ｂと、一次回路２１ａと二次回路２１ｂとを結合するトランスＴＲ１とを有する。このようなＤＣ／ＤＣコンバータ２１は、デュアルアクティブブリッジ（ＤＡＢ）回路とも呼ばれる。 The DC / DC converter 21 boosts or steps down the DC voltage from the DC power supply VB1. The DC / DC converter 21 includes a primary circuit (full bridge (H bridge) circuit) 21a having four switching elements Q11 to Q14 and four switching elements Q21 to Q24, as shown in FIG. A secondary circuit (H bridge circuit) 21b, and a transformer TR1 that couples the primary circuit 21a and the secondary circuit 21b. Such a DC / DC converter 21 is also called a dual active bridge (DAB) circuit.
　スイッチング素子Ｑ１１～Ｑ１４，Ｑ２１～Ｑ２４は、例えば絶縁ゲートバイポーラトランジスタ（ＩＧＢＴ）で構成されている。スイッチング素子Ｑ１１～Ｑ１４，Ｑ２１～Ｑ２４の２つの端子間にはダイオードが接続されている。 The switching elements Q11 to Q14 and Q21 to Q24 are constituted by, for example, insulated gate bipolar transistors (IGBT). A diode is connected between the two terminals of the switching elements Q11 to Q14 and Q21 to Q24.
　トランスＴＲ１は、一次回路２１ａからの直流電圧を昇圧又は降圧して二次回路２１ｂに伝達するとともに、一次回路２１ａと二次回路２１ｂとを絶縁する絶縁型トランスである。トランスＴＲ１は、ｍ：ｎ（ｍ、ｎは整数）の変圧比を有する。なお、トランスＴＲ１の変圧比は１：１であってもよく、換言すれば、変圧せずにそのまま直流電圧を伝達してもよい。 The transformer TR1 is an insulating transformer that boosts or steps down the DC voltage from the primary circuit 21a and transmits it to the secondary circuit 21b, and insulates the primary circuit 21a and the secondary circuit 21b. The transformer TR1 has a transformation ratio of m: n (m and n are integers). The transformer TR1 may have a transformation ratio of 1: 1. In other words, the DC voltage may be transmitted as it is without being transformed.
　ＤＣ／ＤＣコンバータ２１は、絶縁トランスＴＲ１を有する絶縁型の変換器であるので、直流電源ＶＢ１と各相の電圧変換モジュール１１－１，１２－１，１３－１とを絶縁することができる。このため、各相の電圧変換モジュール１１－１，１２－１，１３－１を共通の直流電源ＶＢ１に接続することができる。 Since the DC / DC converter 21 is an insulation type converter having an insulation transformer TR1, it is possible to insulate the DC power supply VB1 from the voltage conversion modules 11-1, 12-1, and 13-1 of each phase. Therefore, the voltage conversion modules 11-1, 12-1, and 13-1 for each phase can be connected to the common DC power supply VB1.
　図２に示すように、ＤＣ／ＤＣコンバータ２１のスイッチング動作は、制御装置３１により制御される。制御装置３１は、一次側電圧検出部３３、二次側電圧検出部３４、二次側電流検出部３６、主制御部３２及び駆動回路３５を備える。一次側電圧検出部３３は、一次回路２１ａ側の電圧Ｖ１を検出し、主制御部３２に出力する。二次側電圧検出部３４は、二次回路２１ｂ側の電圧Ｖ２を検出し、主制御部３２に出力する。主制御部３２は、例えば、中央演算装置（ＣＰＵ）や、ＲＡＭ、ＲＯＭ、ハードディスク等の記憶手段からなる一体型のコンピュータとして構成される。主制御部３２は、上位機器であるマスターコントローラ４１による指令電圧値（目標出力電圧値）Ｖｒｅｆに追従するように、入力電圧Ｖ１及び出力電圧Ｖ２を観測しながらスイッチング素子Ｑ１１～Ｑ１４，Ｑ２１～Ｑ２４のオン、オフ指令信号を駆動回路３５へ出力する。駆動回路３５は、主制御部３２からのオン、オフ指令信号に基づいて、各スイッチング素子Ｑ１１～Ｑ１４，Ｑ２１～Ｑ２４の制御端子（ベース）に駆動信号を出力する。 As shown in FIG. 2, the switching operation of the DC / DC converter 21 is controlled by the control device 31. The control device 31 includes a primary side voltage detection unit 33, a secondary side voltage detection unit 34, a secondary side current detection unit 36, a main control unit 32, and a drive circuit 35. The primary side voltage detector 33 detects the voltage V1 on the primary circuit 21a side and outputs it to the main controller 32. The secondary side voltage detector 34 detects the voltage V <b> 2 on the secondary circuit 21 b side and outputs it to the main controller 32. The main control unit 32 is configured as an integrated computer including a central processing unit (CPU) and storage means such as a RAM, a ROM, and a hard disk. The main control unit 32 observes the input voltage V1 and the output voltage V2 so as to follow the command voltage value (target output voltage value) Vref by the master controller 41, which is the host device, and switches the switching elements Q11 to Q14, Q21 to Q24. The on / off command signal is output to the drive circuit 35. The drive circuit 35 outputs drive signals to the control terminals (bases) of the switching elements Q11 to Q14 and Q21 to Q24 based on the on / off command signal from the main control unit 32.
　ＤＣ／ＤＣコンバータ２１の一次回路２１ａ側のスイッチング素子Ｑ１１，Ｑ１４と、スイッチング素子Ｑ１２，Ｑ１３とは、駆動回路３５からの駆動信号に基づいて、交互に周期的にオン・オフする。また、ＤＣ／ＤＣコンバータ２１の二次回路２１ｂ側のスイッチング素子Ｑ２１，Ｑ２４と、スイッチング素子Ｑ２２，Ｑ２３とは、駆動回路３５からの駆動信号に基づいて、交互に周期的にオン・オフする。 The switching elements Q11 and Q14 on the primary circuit 21a side of the DC / DC converter 21 and the switching elements Q12 and Q13 are alternately turned on and off periodically based on a drive signal from the drive circuit 35. Further, the switching elements Q21 and Q24 on the secondary circuit 21b side of the DC / DC converter 21 and the switching elements Q22 and Q23 are alternately turned on and off periodically based on a drive signal from the drive circuit 35.
　図１に示したインバータ回路２２は、ＤＣ／ＤＣコンバータ２１からの直流電圧を交流電圧に変換する。インバータ回路２２は、４つのスイッチング素子Ｑ３１～Ｑ３４を有するフルブリッジ（Ｈブリッジ）回路である。スイッチング素子Ｑ３１～Ｑ３４は、例えばＩＧＢＴで構成されている。スイッチング素子Ｑ３１～Ｑ３４の２つの端子間にはダイオードが接続されている。電圧変換モジュール１１－１のインバータ回路２２の出力は、電圧変換モジュール１１－１と同一の列に配列された電圧変換モジュール１１－２～１１－ｎのインバータ回路の出力と直列に接続されている。 1 converts the DC voltage from the DC / DC converter 21 into an AC voltage. The inverter circuit 22 is a full bridge (H bridge) circuit having four switching elements Q31 to Q34. The switching elements Q31 to Q34 are composed of, for example, an IGBT. A diode is connected between the two terminals of the switching elements Q31 to Q34. The output of the inverter circuit 22 of the voltage conversion module 11-1 is connected in series with the output of the inverter circuits of the voltage conversion modules 11-2 to 11-n arranged in the same column as the voltage conversion module 11-1. .
　インバータ回路２２は、位相シフト方式のパルス幅変調（ＰＷＭ）を行うことにより、スイッチング素子Ｑ３１～Ｑ３４をオン・オフするＰＷＭ回路である。位相シフト方式のＰＷＭでは、図３に示すように、搬送波（三角波）Ｗ１～Ｗ６の位相をずらしつつ、搬送波Ｗ１～Ｗ６と信号波（正弦波）Ｗ０の信号値の大小を比較して駆動信号が生成される。例えば、搬送波Ｗ１～Ｗ６が信号波Ｗ０よりも小さい場合にスイッチング素子Ｑ３１，Ｑ３４をオンし、搬送波Ｗ１～Ｗ６が信号波Ｗ０よりも大きい場合にスイッチング素子Ｑ３２，Ｑ３３をオンする。ここで、ＤＣ／ＤＣコンバータ２１のスイッチング周波数が、インバータ回路２２のスイッチング周波数よりも高く設定されている。 The inverter circuit 22 is a PWM circuit that turns on / off the switching elements Q31 to Q34 by performing phase shift type pulse width modulation (PWM). In the phase shift type PWM, as shown in FIG. 3, the drive signals are compared by comparing the signal values of the carrier waves W1 to W6 and the signal wave (sine wave) W0 while shifting the phases of the carrier waves (triangular waves) W1 to W6. Is generated. For example, the switching elements Q31 and Q34 are turned on when the carrier waves W1 to W6 are smaller than the signal wave W0, and the switching elements Q32 and Q33 are turned on when the carrier waves W1 to W6 are larger than the signal wave W0. Here, the switching frequency of the DC / DC converter 21 is set higher than the switching frequency of the inverter circuit 22.
　図４に示すように、電圧変換モジュール１１－１において、インバータ回路２２のスイッチング動作等に伴い電圧変動が発生し、平滑用コンデンサＣ２により吸収しきれない電圧変動に応じたリプル電流Ｉｕが流れる。リプル電流Ｉｕは、二次回路２１ｂ側からトランスＴＲ１を介して一次回路２１ａ側へ伝達される。リプル電流Ｉｕは、図５に示すように、パルス状の波形である。リプル電流のパルス波形の基本周波数は、インバータ回路２２のスイッチング周波数となる。一方、リプル電流のパルス波形の振幅は、相電流周波数で振動（変化）する。リプル電流のパルス波形の振幅の最大値が相電流の最大値Ｉｍａｘとなる。 As shown in FIG. 4, in the voltage conversion module 11-1, voltage fluctuation occurs due to the switching operation of the inverter circuit 22 and the like, and a ripple current Iu corresponding to the voltage fluctuation that cannot be absorbed by the smoothing capacitor C2 flows. The ripple current Iu is transmitted from the secondary circuit 21b side to the primary circuit 21a side via the transformer TR1. The ripple current Iu has a pulse waveform as shown in FIG. The fundamental frequency of the ripple waveform of the ripple current is the switching frequency of the inverter circuit 22. On the other hand, the amplitude of the ripple waveform of the ripple current oscillates (changes) at the phase current frequency. The maximum value of the amplitude of the ripple waveform of the ripple current is the maximum value Imax of the phase current.
　本発明の実施の形態では、図６に示すように、各相の電圧変換モジュール１１－１，１２－１，１３－１において、リプル電流Ｉｕ，Ｉｖ，Ｉｗが二次回路２１ｂ側からトランスＴＲ１を介して一次回路２１ａ側へそれぞれ伝達される。 In the embodiment of the present invention, as shown in FIG. 6, in the voltage conversion modules 11-1, 12-1, and 13-1 for each phase, the ripple currents Iu, Iv, and Iw are transmitted from the secondary circuit 21b side to the transformer TR1. Are respectively transmitted to the primary circuit 21a side.
　図７（ａ）～図７（ｃ）に、各相のリプル電流Ｉｕ，Ｉｖ，Ｉｗの波形を示す。リプル電流のパルス波形の振幅は、相電流周波数に応じて振動（変化）するため、図７（ａ）～図７（ｃ）に示すように、各相のリプル電流Ｉｕ，Ｉｖ，Ｉｗは、互いに１２０°位相がずれた周期を有する。本発明の実施の形態では、Ｕ相、Ｖ相、Ｗ相の電圧変換モジュール１１－１，１２－１，１３－１で直流電源ＶＢ１を共通化しているため、図７（ｄ）に示すように、各相に流れるリプル電流Ｉｕ，Ｉｖ，Ｉｗの相電流基本波成分が足し合わされて相殺される。 7A to 7C show the waveforms of the ripple currents Iu, Iv, and Iw for each phase. Since the amplitude of the pulse waveform of the ripple current oscillates (changes) according to the phase current frequency, the ripple currents Iu, Iv, and Iw of each phase are expressed as shown in FIGS. 7 (a) to 7 (c). The periods are 120 ° out of phase with each other. In the embodiment of the present invention, since the DC power supply VB1 is shared by the U-phase, V-phase, and W-phase voltage conversion modules 11-1, 12-1, and 13-1, as shown in FIG. In addition, the phase current fundamental wave components of the ripple currents Iu, Iv, and Iw flowing in the respective phases are added and canceled.
　図８に示すように、インバータ回路２２のスイッチング周波数ｆｓは、必ず相電流周波数の最大値ｆｐよりも高い。リプル電流Ｉｕ，Ｉｖ，Ｉｗ自体はインバータ回路２２のスイッチング周波数よりも高次の周波数を多く含むが、ＤＣ／ＤＣコンバータ２１の二次回路２１ｂ側から一次回路２１ａ側に相電流周波数成分を伝達するには、インバータ回路２２のスイッチング周波数ｆｓの基本波成分まで含まれていればよい。そのため、ＤＣ／ＤＣコンバータ２１のカットオフ周波数が、インバータ回路２２のスイッチング周波数ｆｓ以上であればよい。したがって、ＤＣ／ＤＣコンバータ２１の電圧制御帯域が、インバータ回路２２のスイッチング周波数ｆｓより高い周波数に設定される。 As shown in FIG. 8, the switching frequency fs of the inverter circuit 22 is always higher than the maximum value fp of the phase current frequency. The ripple currents Iu, Iv, and Iw themselves contain a higher order frequency than the switching frequency of the inverter circuit 22, but transmit the phase current frequency component from the secondary circuit 21b side of the DC / DC converter 21 to the primary circuit 21a side. Includes only the fundamental wave component of the switching frequency fs of the inverter circuit 22. Therefore, the cut-off frequency of the DC / DC converter 21 may be equal to or higher than the switching frequency fs of the inverter circuit 22. Therefore, the voltage control band of the DC / DC converter 21 is set to a frequency higher than the switching frequency fs of the inverter circuit 22.
　次に、図９を用いて、ＤＣ／ＤＣコンバータ２１のフィードフォワード制御について説明する。最終的な主制御部３２の出力はＰＷＭ信号である。ＰＷＭ信号は、一次回路２１ａ側と二次回路２１ｂ側で位相がδだけシフトした、デューティ比が略５０％の方形波である。主制御部３２に含まれるＰＩコントローラは、二次回路２１ｂ側の電圧Ｖ２と一次回路２１ａ側の電圧Ｖ１を参照しながら、二次回路２１ｂ側の電圧Ｖ２が指令電圧値Ｖｒｅｆに追従するよう位相差δを制御する。二次回路２１ｂ側の電流ｉ２の平均値は、以下の式（１）で求めることができる。
Figure JPOXMLDOC01-appb-M000001
 Next, feedforward control of the DC / DC converter 21 will be described with reference to FIG. The final output of the main control unit 32 is a PWM signal. The PWM signal is a square wave with a duty ratio of approximately 50%, the phase of which is shifted by δ on the primary circuit 21a side and the secondary circuit 21b side. The PI controller included in the main control unit 32 is configured so that the voltage V2 on the secondary circuit 21b side follows the command voltage value Vref while referring to the voltage V2 on the secondary circuit 21b side and the voltage V1 on the primary circuit 21a side. The phase difference δ is controlled. The average value of current i 2 of the secondary circuit 21b side can be obtained by the following equation (1).
Figure JPOXMLDOC01-appb-M000001
　ここで、ｆｐは相電流の周波数であり、ＬｓはトランスＴＲ１の漏れインダクタンスである。この式（１）から、実際の二次回路２１ｂ側の電流ｉ２を達成するための位相差（シフト量）δを、以下の式（２）により求めることができる。
Figure JPOXMLDOC01-appb-M000002
 Here, fp is the frequency of the phase current, and Ls is the leakage inductance of the transformer TR1. From this equation (1), the phase difference (shift amount) δ for achieving the actual current i 2 on the secondary circuit 21b side can be obtained by the following equation (2).
Figure JPOXMLDOC01-appb-M000002
　主制御部３２のＦ－Ｆ制御項のパラメータには、式（３）及び（４）のように、二次回路２１ｂ側の電流の公称値ｉ２（ｎｏｍ）を用いて、位相のシフト量δ（ｎｏｍ）を設定する。
Figure JPOXMLDOC01-appb-M000003
Figure JPOXMLDOC01-appb-M000004
 As a parameter of the FF control term of the main control unit 32, the nominal value i 2 (nom) of the current on the secondary circuit 21b side is used as in equations (3) and (4), and the phase shift amount δ (nom) is set.
Figure JPOXMLDOC01-appb-M000003
Figure JPOXMLDOC01-appb-M000004
　ＰＩコントローラの出力に、Ｆ－Ｆ制御項で求まった位相のシフト量δ（ｎｏｍ）を加算することによりＰＷＭ信号を生成する。この結果、二次回路２１ｂ側の電流ｉ２が急峻に変化しても、二次回路２１ｂ側の電圧Ｖ２が指令電圧値Ｖｒｅｆに追従する制御が可能となる。 A PWM signal is generated by adding the phase shift amount δ (nom) obtained by the FF control term to the output of the PI controller. As a result, even when the current i 2 of the secondary circuit 21b side abruptly changes, it is possible to control the voltage V2 of the secondary circuit 21b side to follow the command voltage value Vref.
　次に、本発明の実施の形態に係る電力変換装置を用いた電力変換方法の一例を説明する。 Next, an example of a power conversion method using the power conversion device according to the embodiment of the present invention will be described.
　まず、図１に示した複数の電圧変換モジュール１１－１～１１－ｎ，１２－１～１２－ｎ，１３－１～１３－ｎが、複数の直流電源ＶＢ１～ＶＢｎからの直流電圧を交流電圧に変換する。具体的には、各電圧変換モジュール１１－１～１１－ｎ，１２－１～１２－ｎ，１３－１～１３－ｎにおいて、ＤＣ／ＤＣコンバータ２１が、制御装置３１からの制御信号に基づいてスイッチング素子Ｑ１１～Ｑ１４，Ｑ２１～Ｑ２４をスイッチング動作させることにより、直流電源ＶＢ１～ＶＢｎからの直流電圧を変圧する。また、インバータ回路２２が、制御装置３１からの制御信号に基づいてスイッチング素子Ｑ３１～Ｑ３４をスイッチング動作させることにより、ＤＣ／ＤＣコンバータ２１から出力された直流電圧を交流電圧に変換する。 First, the plurality of voltage conversion modules 11-1 to 11-n, 12-1 to 12-n, and 13-1 to 13-n shown in FIG. 1 convert the DC voltages from the plurality of DC power sources VB1 to VBn to AC. Convert to voltage. Specifically, in each of the voltage conversion modules 11-1 to 11-n, 12-1 to 12-n, and 13-1 to 13-n, the DC / DC converter 21 is based on a control signal from the control device 31. By switching the switching elements Q11 to Q14 and Q21 to Q24, the DC voltage from the DC power sources VB1 to VBn is transformed. Further, the inverter circuit 22 converts the DC voltage output from the DC / DC converter 21 into an AC voltage by switching the switching elements Q31 to Q34 based on a control signal from the control device 31.
　更に、電圧変換モジュール１１－１～１１－ｎから出力された交流電圧を直列的に加算して、Ｕ相へ出力する。また、電圧変換モジュール１２－１～１２－ｎから出力された交流電圧を直列的に加算して、Ｖ相へ出力する。また、電圧変換モジュール１３－１～１３－ｎから出力された交流電圧を直列的に加算して、Ｕ相へ出力する。 Further, the AC voltages output from the voltage conversion modules 11-1 to 11-n are added in series and output to the U phase. Also, the AC voltages output from the voltage conversion modules 12-1 to 12-n are added in series and output to the V phase. Further, AC voltages output from the voltage conversion modules 13-1 to 13-n are added in series and output to the U phase.
　この際、同一の行に配列されたＵ相、Ｖ相、Ｗ相の電圧変換モジュール１１－１，１２－１，１３－１を直流電源ＶＢ１に共通に接続することにより、各相のリプル電流Ｉｕ，Ｉｖ，Ｉｗの相電流基本波成分を足し合わせて相殺することができる。なお、同一の行に配列されたＵ相、Ｖ相、Ｗ相の電圧変換モジュール１１－２，１２－２，１３－２及び電圧変換モジュール１１－ｎ，１２－ｎ，１３－ｎでも同様に、各相のリプル電流Ｉｕ，Ｉｖ，Ｉｗの相電流基本波成分を足し合わせて相殺することができる。 At this time, the U-phase, V-phase, and W-phase voltage conversion modules 11-1, 12-1, and 13-1 arranged in the same row are connected in common to the DC power supply VB1, so that the ripple current of each phase is obtained. The phase current fundamental wave components of Iu, Iv, and Iw can be added together and canceled. The same applies to U-phase, V-phase, and W-phase voltage conversion modules 11-2, 12-2, and 13-2 and voltage conversion modules 11-n, 12-n, and 13-n arranged in the same row. The phase current fundamental wave components of the ripple currents Iu, Iv, and Iw of each phase can be added and canceled.
　以上説明したように、本発明の実施の形態によれば、同一の行に配列されたＵ相、Ｖ相、Ｗ相の電圧変換モジュール１１－１，１２－１，１３－１を直流電源ＶＢ１に共通に接続することにより、各相のリプル電流Ｉｕ，Ｉｖ，Ｉｗの相電流基本波成分を足し合わせて相殺することができる。したがって、各電圧変換モジュール１１－１～１１－ｎ，１２－１～１２－ｎ，１３－１～１３－ｎにおいて、平滑用コンデンサＣ２をそれぞれ小さくすることができ、装置全体のサイズを小さくすることができる。更に、同一の行に配列されたＵ相、Ｖ相、Ｗ相の電圧変換モジュール１１－１，１２－１，１３－１を直流電源ＶＢ１に共通に接続することにより、各相間の電力のアンバランスが生じても、直流電源ＶＢ１のエネルギー消費を均一化することができる。 As described above, according to the embodiment of the present invention, the U-phase, V-phase, and W-phase voltage conversion modules 11-1, 12-1, and 13-1 arranged in the same row are connected to the DC power supply VB1. By connecting them in common, the phase current fundamental wave components of the ripple currents Iu, Iv, Iw of each phase can be added together and canceled. Therefore, in each of the voltage conversion modules 11-1 to 11-n, 12-1 to 12-n, and 13-1 to 13-n, the smoothing capacitor C2 can be reduced, and the overall size of the device can be reduced. be able to. Further, the U-phase, V-phase, and W-phase voltage conversion modules 11-1, 12-1, and 13-1 arranged in the same row are commonly connected to the DC power supply VB1, thereby reducing the power between the phases. Even if the balance occurs, the energy consumption of the DC power supply VB1 can be made uniform.
　更に、絶縁型のＤＣ／ＤＣコンバータ２１を有することにより、各相の電圧変換モジュール１１－１，１２－１，１３－１を互いに絶縁することができ、各相の電圧変換モジュール１１－１，１２－１，１３－１を共通の直流電源ＶＢ１に接続することができる。 Further, by including the insulation type DC / DC converter 21, the voltage conversion modules 11-1, 12-1, 13-1 of each phase can be insulated from each other, and the voltage conversion modules 11-1, 12-1 and 13-1 can be connected to a common DC power supply VB1.
　更に、本発明の実施の形態によれば、ＤＣ／ＤＣコンバータ２１としてＤＡＢを用いることにより、装置の小型化及び高効率化を図ることができる。また、ＤＡＢのスイッチング周波数をインバータ回路２２のスイッチング周波数よりも高周波とすることにより、ＤＡＢの応答性を向上させることができる。また、ＤＡＢのカットオフ周波数をインバータ回路２２のスイッチング周波数よりも高周波とすることにより、ＤＡＢの応答性を向上させることができる。また、ＤＡＢをフィードフォワード制御することにより、ＤＡＢの応答性を向上させることができる。また、インバータ回路２２を位相シフト方式のＰＷＭで制御することにより、全バッテリのエネルギー消費を均一化することができる。 Furthermore, according to the embodiment of the present invention, the use of DAB as the DC / DC converter 21 can reduce the size and increase the efficiency of the apparatus. Further, by making the switching frequency of DAB higher than the switching frequency of the inverter circuit 22, the responsiveness of DAB can be improved. Further, by making the DAB cutoff frequency higher than the switching frequency of the inverter circuit 22, the responsiveness of the DAB can be improved. Further, DAB responsiveness can be improved by feedforward control of DAB. Further, by controlling the inverter circuit 22 with the phase shift type PWM, the energy consumption of all the batteries can be made uniform.
　（第１の変形例）
　第１の変形例として、インバータ回路２２の他の制御方法を説明する。第１の変形例に係るインバータ回路２２は、レベルシフト方式のＰＷＭで制御される。レベルシフト方式では、図１０に示すように、搬送波（三角波）Ｗ１～Ｗ６の電圧レベルが変調される。この搬送波Ｗ１～Ｗ６と信号波（正弦波）Ｗ０との大小を比較して、インバータ回路２２のスイッチング素子Ｑ３１～Ｑ３４のオン・オフが制御される。 (First modification)
As a first modification, another control method of the inverter circuit 22 will be described. The inverter circuit 22 according to the first modification is controlled by level shift type PWM. In the level shift method, as shown in FIG. 10, voltage levels of carrier waves (triangular waves) W1 to W6 are modulated. By comparing the magnitudes of the carrier waves W1 to W6 and the signal wave (sine wave) W0, on / off of the switching elements Q31 to Q34 of the inverter circuit 22 is controlled.
　第１の変形例によれば、インバータ回路２２をレベルシフト方式のＰＷＭで制御することにより、高効率化を図ることができる。更に、位相シフト方式と比較して、同じスイッチング周波数でもターンオフ・オン回数が少ないため、低応答のＤＣ／ＤＣコンバータ（ＤＡＢ）であっても本発明の実施の形態と同様の効果が得られる。 According to the first modification, high efficiency can be achieved by controlling the inverter circuit 22 with level shift PWM. Further, since the number of turn-offs and ons is small even at the same switching frequency as compared with the phase shift method, the same effect as the embodiment of the present invention can be obtained even with a low response DC / DC converter (DAB).
　（第２の変形例）
　第２の変形例として、インバータ回路２２の他の一例を説明する。第２の変形例に係るインバータ回路２２は、図１１に示すように、Ｈブリッジ回路の代わりに、２個のスイッチング素子Ｑ３１，Ｑ３２を有するハーフブリッジ回路を有する点が異なる。また、負電圧を生成するため、基準電位側の直流電源ＶＢ（ｎ－１），ＶＢｎが逆方向に接続されている。第２の変形例においても、例えば本発明の実施の形態と同様に位相シフト方式のＰＷＭ制御が可能である。 (Second modification)
Another example of the inverter circuit 22 will be described as a second modification. As shown in FIG. 11, the inverter circuit 22 according to the second modification is different in that it includes a half bridge circuit having two switching elements Q31 and Q32 instead of the H bridge circuit. Further, in order to generate a negative voltage, DC power sources VB (n−1) and VBn on the reference potential side are connected in the reverse direction. Also in the second modification, for example, phase shift type PWM control is possible as in the embodiment of the present invention.
　第２の変形例によれば、インバータ回路２２がハーフブリッジ回路であることにより、Ｈブリッジ回路の場合と比較して、各電圧変換モジュール１１－１～１１－ｎ，１２－１～１２－ｎ，１３－１～１３－ｎでスイッチング素子を２つずつ減らすことができ、回路構成を簡素化することが可能となる。 According to the second modification, since the inverter circuit 22 is a half-bridge circuit, each of the voltage conversion modules 11-1 to 11-n, 12-1 to 12-n is compared with the case of the H-bridge circuit. , 13-1 to 13-n, the number of switching elements can be reduced by two, and the circuit configuration can be simplified.
　（第３の変形例）
　第３の変形例として、ＤＣ／ＤＣコンバータ２１の他の一例を説明する。第３の変形例に係るＤＣ／ＤＣコンバータ２１は、図１２に示すように、スイッチング素子Ｑ１１～Ｑ１４，Ｑ２１～Ｑ２４として、ＩＧＢＴの代わりに金属酸化物半導体電界効果トランジスタ（ＭＯＳＦＥＴ）を用いる点が異なる。ＭＯＳＦＥＴは内蔵出力容量を有する。 (Third Modification)
As a third modification, another example of the DC / DC converter 21 will be described. As shown in FIG. 12, the DC / DC converter 21 according to the third modified example uses a metal oxide semiconductor field effect transistor (MOSFET) instead of the IGBT as the switching elements Q11 to Q14 and Q21 to Q24. Different. The MOSFET has a built-in output capacitance.
　第３の変形例によれば、図１に示すようにＩＧＢＴの両端にコンデンサを設けることなく、ＭＯＳＦＥＴの内蔵出力容量を用いてソフトスイッチングを行うことが可能となり、スイッチング時に生じるノイズを抑制することができる。 According to the third modification, soft switching can be performed using the built-in output capacitance of the MOSFET without providing capacitors at both ends of the IGBT as shown in FIG. 1, and noise generated during switching can be suppressed. Can do.
　（第４の変形例）
　第４の変形例として、ＤＣ／ＤＣコンバータ２１の他の一例を説明する。第４の変形例に係るＤＣ／ＤＣコンバータ２１は、図１３に示すように、各スイッチング素子Ｑ１１～Ｑ１４，Ｑ２１～Ｑ２４においてＩＧＢＴにコンデンサが並列に接続されている点が異なる。 (Fourth modification)
As a fourth modification, another example of the DC / DC converter 21 will be described. As shown in FIG. 13, the DC / DC converter 21 according to the fourth modification is different in that a capacitor is connected in parallel to the IGBT in each of the switching elements Q11 to Q14 and Q21 to Q24.
　第４の変形例によれば、各スイッチング素子Ｑ１１～Ｑ１４，Ｑ２１～Ｑ２４のコンデンサを用いることにより、ソフトスイッチングを行うことが可能となり、スイッチング時に生じるノイズを抑制することができる。 According to the fourth modification, soft switching can be performed by using the capacitors of the switching elements Q11 to Q14 and Q21 to Q24, and noise generated during switching can be suppressed.
　（第５の変形例）
　第５の変形例として、ＤＣ／ＤＣコンバータ２１の他の一例を図１４（ａ）～図１４（ｄ）を用いて説明する。図１４（ａ）に示すように、ＤＣ／ＤＣコンバータ２１が、ＬＣ直列共振型の一次回路２１ａを有していてもよい。また、図１４（ｂ）に示すように、ＤＣ／ＤＣコンバータ２１が、ＬＬＣ直列共振型の一次回路２１ａを有していてもよい。また、図１４（ｃ）に示すように、ＤＣ／ＤＣコンバータ２１が、ＬＣＣ直列共振型の一次回路２１ａを有していてもよい。また、図１４（ｄ）に示すように、ＤＣ／ＤＣコンバータ２１が、ＬＣ並列共振型の一次回路２１ａを有していてもよい。 (Fifth modification)
As a fifth modification, another example of the DC / DC converter 21 will be described with reference to FIGS. 14 (a) to 14 (d). As shown in FIG. 14A, the DC / DC converter 21 may have an LC series resonance type primary circuit 21a. As shown in FIG. 14B, the DC / DC converter 21 may include an LLC series resonance type primary circuit 21a. Moreover, as shown in FIG.14 (c), the DC / DC converter 21 may have the LCC series resonance type primary circuit 21a. Moreover, as shown in FIG.14 (d), the DC / DC converter 21 may have the LC parallel resonance type primary circuit 21a.
　第５の変形例によれば、図１４（ａ）～図１４（ｄ）に示した構成の場合でも、それぞれソフトスイッチングを実現しながら、スナバコンデンサと同様の機能を果たすことが可能となる。 According to the fifth modification, even in the case of the configuration shown in FIGS. 14A to 14D, it is possible to perform the same function as the snubber capacitor while realizing soft switching.
　（その他の実施の形態）
　上記のように、本発明は実施の形態によって記載したが、この開示の一部をなす論述及び図面はこの発明を限定するものであると理解すべきではない。この開示から当業者には様々な代替実施の形態、実施例及び運用技術が明らかとなろう。 (Other embodiments)
As described above, the present invention has been described according to the embodiment. However, it should not be understood that the description and drawings constituting a part of this disclosure limit the present invention. From this disclosure, various alternative embodiments, examples and operational techniques will be apparent to those skilled in the art.
　例えば、本発明の実施の形態においては、三相交流モータＭ１を駆動させるための三相の交流電圧を生成する場合を説明したが、これに限定されず、単相の交流電圧を生成することにも適用可能である。 For example, in the embodiment of the present invention, the case of generating a three-phase AC voltage for driving the three-phase AC motor M1 has been described. However, the present invention is not limited to this, and a single-phase AC voltage is generated. It is also applicable to.
　また、本発明の実施の形態では、電力変換装置がＤＣ／ＤＣコンバータ２１を備える構成を説明したが、ＤＣ／ＤＣコンバータ２１は必ずしも備えていなくてもよい。また、高調波成分除去用コンデンサＣ１も必ずしも備えていなくてもよい。 In the embodiment of the present invention, the configuration in which the power conversion apparatus includes the DC / DC converter 21 has been described. However, the DC / DC converter 21 may not necessarily be included. Further, the harmonic component removing capacitor C1 is not necessarily provided.
　このように、本発明はここでは記載していない様々な実施の形態等を含むことは勿論である。したがって、本発明の技術的範囲は上記の説明から妥当な特許請求の範囲に係る発明特定事項によってのみ定められるものである。 Thus, it goes without saying that the present invention includes various embodiments not described herein. Therefore, the technical scope of the present invention is defined only by the invention specifying matters according to the scope of claims reasonable from the above description.
特願２０１３－２１６０９２号（出願日：２０１３年１０月１７日）の全内容は、ここに援用される。 The entire contents of Japanese Patent Application No. 2013-216092 (filing date: October 17, 2013) are incorporated herein by reference.
　以上、実施の形態に沿って本発明の内容を説明したが、本発明はこれらの記載に限定されるものではなく、種々の変形及び改良が可能であることは、当業者には自明である。 The contents of the present invention have been described above according to the embodiments, but the present invention is not limited to these descriptions, and it is obvious to those skilled in the art that various modifications and improvements are possible. .
　本発明は、電力変換装置の平滑用コンデンサを小さくすることができ、装置の小型化を図ることに利用することができる。 The present invention can be used to reduce the smoothing capacitor of the power conversion device and to reduce the size of the device.
　Ｃ１　高調波除去用コンデンサ
　Ｃ２　平滑用コンデンサ
　Ｍ１　三相交流モータ
　Ｎ１，Ｎ２，Ｎ３　出力端子
　Ｑ１１～Ｑ１４，Ｑ２１～Ｑ２４，Ｑ３１～Ｑ３４　スイッチング素子
　ＶＢ１，ＶＢ２，ＶＢｎ　直流電源
　ＴＲ１　絶縁トランス
　１１－１～１１－ｎ，１２－１～１２－ｎ，１３－１～１３－ｎ　電圧変換モジュール（電圧変換手段）
　２１　ＤＣ／ＤＣコンバータ（変圧手段）
　２１ａ　一次回路
　２１ｂ　二次回路
　２２　インバータ回路
　３１　制御装置（制御手段）
　３２　主制御部
　３３　一次側電圧検出部
　３４　二次側電圧検出部
　３５　駆動回路
　３６　二次側電流検出部
　４１　マスターコントローラ
  C1 Harmonic elimination capacitor C2 Smoothing capacitor M1 Three-phase AC motor N1, N2, N3 Output terminals Q11 to Q14, Q21 to Q24, Q31 to Q34 Switching element VB1, VB2, VBn DC power supply TR1 Insulation transformer 11-1 to 11 -N, 12-1 to 12-n, 13-1 to 13-n Voltage conversion module (voltage conversion means)
21 DC / DC converter (transformer)
21a Primary circuit 21b Secondary circuit 22 Inverter circuit 31 Control device (control means)
32 Main control unit 33 Primary voltage detection unit 34 Secondary voltage detection unit 35 Drive circuit 36 Secondary current detection unit 41 Master controller

Claims (9)

  1. 　直流電圧を交流電圧に変換し、変換された交流電圧を複数相を有する負荷に供給する電力変換装置であって、
    　複数の直流電源と、
    　前記複数の直流電源のそれぞれに並列に接続された平滑用コンデンサと、
    　前記直流電圧を交流電圧に変換する際に生じるリプル電流を互いに相殺するように前記複数の直流電源のそれぞれに前記複数相の数だけ並列に接続され、接続された前記直流電源からの直流電圧を交流電圧に変換する複数の電圧変換手段と、
    　前記複数の電圧変換手段を制御する制御手段と、
    　異なる前記直流電源に接続された前記電圧変換手段の出力を直列に接続し、各電圧変換手段からの交流電圧を加算して前記負荷の各相へそれぞれ出力する複数の出力端子
    　とを備えることを特徴とする電力変換装置。     A power conversion device that converts a DC voltage into an AC voltage and supplies the converted AC voltage to a load having a plurality of phases,
    Multiple DC power supplies,
    A smoothing capacitor connected in parallel to each of the plurality of DC power sources;
    The plurality of DC power supplies are connected in parallel to each of the plurality of phases so as to cancel ripple currents generated when the DC voltage is converted into AC voltage, and the DC voltages from the connected DC power supplies are A plurality of voltage conversion means for converting to AC voltage;
    Control means for controlling the plurality of voltage conversion means;
    A plurality of output terminals for connecting the outputs of the voltage conversion means connected to the different DC power sources in series, adding the AC voltages from the voltage conversion means and outputting them to each phase of the load, respectively. A power conversion device.
  2. 　前記複数の電圧変換手段のそれぞれが、
    　前記直流電源に並列に接続され、前記直流電源からの直流電圧を変圧する絶縁型のＤＣ／ＤＣコンバータと、
    　前記直流電源に並列に接続され、前記ＤＣ／ＤＣコンバータからの直流電圧を交流電圧に変換するインバータ回路
    　とを備えることを特徴とする請求項１に記載の電力変換装置。     Each of the plurality of voltage conversion means
    An insulated DC / DC converter connected in parallel to the DC power source and transforming a DC voltage from the DC power source;
    The power converter according to claim 1, further comprising: an inverter circuit connected in parallel to the DC power source and converting a DC voltage from the DC / DC converter into an AC voltage.
  3. 　前記ＤＣ／ＤＣコンバータは、一次回路と二次回路とが絶縁トランスを介して結合されたデュアルアクティブブリッジ回路であることを特徴とする請求項２に記載の電力変換装置。 The power converter according to claim 2, wherein the DC / DC converter is a dual active bridge circuit in which a primary circuit and a secondary circuit are coupled via an insulating transformer.
  4. 　前記デュアルアクティブブリッジ回路のスイッチング周波数が、前記インバータ回路のスイッチング周波数よりも高いことを特徴とする請求項３に記載の電力変換装置。 The power conversion device according to claim 3, wherein a switching frequency of the dual active bridge circuit is higher than a switching frequency of the inverter circuit.
  5. 　前記デュアルアクティブブリッジ回路のカットオフ周波数が、前記インバータ回路のスイッチング周波数よりも高いことを特徴とする請求項３に記載の電力変換装置。 The power conversion device according to claim 3, wherein a cutoff frequency of the dual active bridge circuit is higher than a switching frequency of the inverter circuit.
  6. 　前記デュアルアクティブブリッジ回路をフィードフォワード制御することを特徴とする請求項３～５のいずれか１項に記載の電力変換装置。 The power converter according to any one of claims 3 to 5, wherein the dual active bridge circuit is feedforward controlled.
  7. 　前記インバータ回路は、レベルシフト方式でパルス幅変調を行うパルス幅変調回路であることを特徴とする請求項２に記載の電力変換装置。 The power converter according to claim 2, wherein the inverter circuit is a pulse width modulation circuit that performs pulse width modulation by a level shift method.
  8. 　前記インバータ回路は、位相シフト方式でパルス幅変調を行うパルス幅変調回路であることを特徴とする請求項２に記載の電力変換装置。 The power converter according to claim 2, wherein the inverter circuit is a pulse width modulation circuit that performs pulse width modulation by a phase shift method.
  9. 　複数の直流電源のそれぞれに平滑用コンデンサが並列に接続され、前記複数の直流電源のそれぞれに複数の電圧変換手段が前記複数相の数だけ並列に接続され、異なる前記直流電源に接続された前記電圧変換手段の出力が直列に接続され、直流電圧を交流電圧に変換し、変換された交流電圧を複数相を有する負荷に供給する電力変換装置を用いた電力変換方法であって、
    　前記複数の電圧変換手段が、複数の直流電源からの直流電圧を交流電圧に変換するステップと、
    　異なる前記直流電源に接続された前記電圧変換手段からの交流電圧を直列的に加算して、前記負荷の各相へそれぞれ出力するステップと、
    　前記並列に接続された電圧変換手段毎に、前記直流電圧を交流電圧に変換する際に生じるリプル電流を互いに相殺するステップ
    　とを含むことを特徴とする電力変換方法。     A smoothing capacitor is connected in parallel to each of a plurality of DC power supplies, a plurality of voltage conversion means are connected in parallel to each of the plurality of DC power supplies by the number of the plurality of phases, and the DC power supplies are connected to different DC power supplies. A power conversion method using a power conversion device in which the output of the voltage conversion means is connected in series, converts a DC voltage into an AC voltage, and supplies the converted AC voltage to a load having a plurality of phases,
    The plurality of voltage converting means converting a DC voltage from a plurality of DC power sources into an AC voltage;
    Adding AC voltage from the voltage conversion means connected to the different DC power supply in series, and outputting each to each phase of the load;
    Canceling ripple currents generated when the DC voltage is converted into an AC voltage for each of the voltage conversion means connected in parallel with each other.
PCT/JP2014/073171 2013-10-17 2014-09-03 Power conversion device and power conversion method WO2015056491A1 (en)

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Cited By (7)

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JP2019154142A (en) * 2018-03-02 2019-09-12 株式会社豊田中央研究所 Electric power conversion device
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