JP7003294B2 - Motor and air conditioner equipped with it - Google Patents

Description

The present invention relates to an electric motor provided with a rotor and an air conditioner equipped with the motor.

Conventionally, an electric motor using an inverter for driving the electric motor is known (see, for example, Patent Document 1). Further, an electric motor using a planar MOSFET (Metal Oxide Semiconductor Field Effect Transistor) as a switching element of an inverter is known (see, for example, Patent Document 2). Patent Document 2 proposes using a super junction MOSFET instead of a planar MOSFET in order to reduce the loss of the switching element.

Japanese Unexamined Patent Publication No. 2010-17404 Japanese Unexamined Patent Publication No. 2014-87199

By using a super junction MOSFET instead of the planar MOSFET as the switching element of the inverter, the steady loss of the switch is reduced. However, since the super junction MOSFET has a larger switching loss than the planar MOSFET, there is a possibility that the power loss of the entire circuit will be large.

The present invention has been made to solve the above-mentioned problems, and provides an electric motor using a super junction MOSFET in which the power loss of the entire circuit is suppressed, and an air conditioner equipped with the electric motor.

The motor according to the present invention includes a rotor into which a rotating shaft is inserted, a stator provided on the outer periphery of the rotor and having three-phase windings, and three sets of upper arm switches corresponding to the three phases. And an inverter including a lower arm switch, a freewheeling diode connected in parallel to each of the three sets of upper arm switch and lower arm switch, and the three sets of upper arm switch and lower arm switch at a constant carrier frequency. It has a controller that performs pulse width modulation by non-complementary switching, each of the three sets of upper arm switch and lower arm switch is a superjunction electric field effect transistor, and the controller is the upper arm switch and the lower arm switch. The pulse width modulation is performed with a duty ratio of an on-time longer than the on-time of the upper arm switch when the arm switch operates by complementary switching.

The air conditioner according to the present invention is provided as a drive source for at least one of an indoor unit including a load-side blower, an outdoor unit including a heat source-side blower, and the load-side blower and the heat-source-side blower. It has an electric motor.

According to the present invention, non-complementary switching is performed on the upper arm switch and the lower arm switch to suppress the occurrence of switching loss, and the pulse width modulation of the upper arm switch at a high duty ratio allows current to flow through the freewheeling diode. The time is shortened. Therefore, the power loss of the entire circuit can be suppressed.

It is an external view which shows one structural example of the electric motor which concerns on Embodiment 1 of this invention. It is an external view which shows one configuration example of the built-in board shown in FIG. FIG. 3 is an external view showing another configuration example of the built-in substrate shown in FIG. 1. It is a block diagram which shows one configuration example of the controller and the power IC shown in FIG. It is a figure which shows one configuration example of the power MOSFET used for the upper arm switch and the lower arm switch shown in FIG. It is a figure which shows the structural example of the MOSFET of the comparative example. It is a block diagram which shows an example of the case where the magnetic sensor shown in FIG. 1 is a Hall IC. It is a timing chart which shows the case of performing complementary switching for the upper arm switch and the lower arm switch of each phase as PWM control of a comparative example. It is a figure which shows the operation procedure of the upper arm switch and the lower arm switch of U phase and V phase in the range sandwiched by the two broken lines shown in FIG. It is a figure which shows the operation procedure of the upper arm switch and the lower arm switch of U phase and V phase in the range sandwiched by the two broken lines shown in FIG. It is a figure which shows the operation procedure of the upper arm switch and the lower arm switch of U phase and V phase in the range sandwiched by the two broken lines shown in FIG. 9 is an enlarged view showing the operation of the U-phase upper arm switch and the lower arm switch in FIGS. 9 to 11. It is a timing chart which shows the case where the controller shown in FIG. 4 performs non-complementary switching for the upper arm switch and the lower arm switch of each phase. It is a figure which shows the operation procedure of the upper arm switch and the lower arm switch of U phase and V phase in the range sandwiched by the two broken lines shown in FIG. It is a figure which shows the operation procedure of the upper arm switch and the lower arm switch of U phase and V phase in the range sandwiched by the two broken lines shown in FIG. 14 is an enlarged view showing the operation of the U-phase upper arm switch and the lower arm switch in FIGS. 14 and 15. It is a figure for comparing the duty ratio in the PWM control of the comparative example, and the duty ratio in the PWM control of the first embodiment. It is an external view which shows one structural example of the air conditioner which concerns on Embodiment 2 of this invention. It is a refrigerant circuit diagram which shows one configuration example of the air conditioner shown in FIG. It is a side perspective view which shows one configuration example of the indoor unit shown in FIG. It is a side perspective view which shows one configuration example of the indoor unit of the comparative example.

Embodiment 1.
The configuration of the electric motor of the first embodiment will be described. In the first embodiment, the electric motor is a brushless DC motor. FIG. 1 is an external view showing a configuration example of an electric motor according to the first embodiment of the present invention. In FIG. 1, a part is shown in a cross-sectional structure for the purpose of explaining the configuration of the electric motor. FIG. 1 shows a radial gap type brushless DC motor, but the electric motor of the first embodiment is not limited to the radial gap type. FIG. 2 is an external view showing an example of the configuration of the built-in substrate shown in FIG.

As shown in FIG. 1, the electric motor 1 is equipped with a rotor 30 into which a rotating shaft 31 is inserted, a stator 20 provided on the outer periphery of the rotor 30, and a circuit for controlling the drive of the rotor 30. It has a built-in substrate 11. The stator 20 and the built-in substrate 11 are integrally molded by the mold stator 10. The mold stator 10 has a mold resin 12 having a recess formed therein, and the rotor 30 is housed in the recess.

At one end of the rotating shaft 31, an output side bearing 33 that supports the rotating rotating shaft 31 is provided. At the other end of the rotating shaft 31, a counter-output side bearing 34 for supporting the rotating rotating shaft 31 is provided. The counter-output side bearing 34 is covered with a conductive bracket 60. The outer ring of the counter-output side bearing 34 is fitted inside the bracket 60. The press-fitting portion 61 of the bracket 60 is fitted into the inner peripheral portion of the mold stator 10 so that the bracket 60 closes the opening of the recess of the mold stator 10.

The stator 20 has a plurality of stator cores 21 radially arranged around a rotation shaft 31, and an insulator 23 integrally molded with the plurality of stator cores 21. Each stator core 21 has a structure in which a plurality of electromagnetic steel sheets are laminated. A winding 22 for generating magnetic flux is wound around each stator core 21. The winding 22 is made of a conductive wire rod such as copper and aluminum. The insulator 23 serves to insulate the stator core 21 and the winding 22 from each other.

As shown in FIG. 1, the built-in substrate 11 is arranged between the output side bearing 33 and the stator 20, and is fixed to the insulator 23. As shown in FIG. 2, the shape of the built-in substrate 11 is a disk having a through hole 35 formed in the center. The rotation shaft 31 is arranged so as to pass through the through hole 35. The built-in substrate 11 is arranged inside the motor 1 so that the plane parallel to the disk is perpendicular to the axial direction (Z-axis arrow direction) of the rotation axis 31. As shown in FIG. 2, the built-in substrate 11 includes a power IC (Integrated Circuit) 80 that supplies electric power to the winding 22, a controller 70 that controls the power IC 80, and a magnetic sensor 50 that detects the magnetic pole positions of the rotor 30. And have. In the configuration example shown in FIG. 2, three magnetic sensors 50 are provided on the built-in substrate 11. The magnetic sensor 50 is, for example, a Hall IC.

As shown in FIG. 1, the rotor 30 has a rotating shaft 31 and a rotor magnet 40 fixed to the rotating shaft 31. The rotor magnet 40 is composed of a columnar permanent magnet. It is arranged so as to face the plurality of stator cores 21 arranged inside the mold stator 10. The rotor magnet 40 is manufactured, for example, by injection molding a bonded magnet manufactured by mixing a ferrite magnet or a rare earth magnet with a thermoplastic resin material. A magnet is incorporated in the injection molding mold of the rotor magnet 40, and the molding of the rotor magnet 40 is performed while applying orientation.

In the configuration shown in FIG. 1, the rotor magnet 40 has a sensor magnet portion which is a portion close to the magnetic sensor 50 and a main magnet portion which is a portion other than the sensor magnet portion in the axial direction of the rotating shaft 31. The sensor magnet portion serves to cause the magnetic sensor 50 to detect the position of the rotor 30. The main magnet portion serves to generate a rotational force in the rotor 30 according to the magnetic flux generated by the winding 22. Considering the rotor magnet 40 as a cylinder with the rotation shaft 31 as the central axis, the diameter of the sensor magnet portion is smaller than the diameter of the main magnet portion. With this configuration, magnetic flux easily flows into the magnetic sensor 50 from the magnetic pole of the sensor magnet portion. In the configuration example shown in FIG. 1, the sensor magnet portion and the main magnet portion can be distinguished by the step provided on the rotor magnet 40.

The magnetic sensor 50 is arranged at a position away from the winding 22 shown in FIG. 1 on the built-in substrate 11 in order to minimize the influence of the magnetic flux generated by the winding 22 of the stator 20. That is, the three magnetic sensors 50 are arranged at positions close to the rotation axis 31 shown in FIG. 1 in the built-in substrate 11 shown in FIG.

The power IC 80 of the built-in board 11 and the winding 22 are connected by wiring via a winding terminal (not shown in the figure). The built-in board 11 is provided with a lead opening portion 14 for taking a lead wire 13 connected to a higher-level device on which the motor 1 is mounted into the inside of the motor 1. The host device on which the motor 1 is mounted is, for example, an air conditioner. When the host device is an air conditioner, it is electrically connected to the controller 70 via the lead wire 13 with the control device of the air conditioner.

A controller 70, a magnetic sensor 50, and passive components such as resistors and capacitors (not shown in the figure) are arranged on the stator surface of the built-in substrate 11. When the power IC 80 is a lead type, only the power IC 80 is arranged on the anti-stator surface. In this case, when the terminals of the electronic components are soldered to the printed wiring of the built-in board 11 in the manufacturing process of the built-in board 11, the power IC 80 can be mounted on the built-in board 11 in the one-sided flow process. When the power IC 80 is a surface mount type, the power IC 80 is also arranged on the stator surface. In this case, when the terminals of the electronic components are soldered to the printed wiring of the built-in board 11, the power IC 80 can be mounted on the built-in board 11 in the one-sided reflow process.

Although FIG. 2 shows a case where the controller 70 and the power IC 80 are composed of separate ICs, the controller 70 and the power IC 80 may be composed of one IC. FIG. 3 is an external view showing another configuration example of the built-in substrate shown in FIG. The built-in board 11a shown in FIG. 3 corresponds to a portion of the built-in board 11 shown in FIG. 2 having a central angle of about 90 degrees when the built-in board 11 is a disk. The cutout portion 36 of the built-in substrate 11a corresponds to a part of the circumference of the through hole 35. Module 79 includes electronic circuits that perform the respective functions of the controller 70 and the power IC 80.

In the built-in substrate 11a shown in FIG. 3, the number of electronic components mounted on the substrate is smaller than that in the configuration shown in FIG. 2, so that the mounting area of the electronic components is reduced. As a result, the substrate area of the built-in substrate 11a can be reduced. As in the built-in board 11a shown in FIG. 3, the board area can be reduced by devising the layout of the electronic components on the board and effectively using a part of the shape of the through hole 35.

Further, regarding the rotor magnet 40, the case where the main magnet portion and the sensor magnet portion are configured as one magnet has been described with reference to FIG. 1, but the main magnet portion and the sensor magnet portion are configured by separate magnets. You may.

Next, the configurations of the controller 70 and the power IC 80 shown in FIG. 2 will be described. FIG. 4 is a block diagram showing a configuration example of the controller and the power IC shown in FIG. The power IC 80 includes an inverter 81, a gate drive circuit 82, and a protection circuit 83. An overcurrent detection resistor 71 is connected between the controller 70 and the power IC 80 and the ground (earth).

The inverter 81 converts the input DC voltage E into a three-phase AC voltage including a U phase, a V phase, and a W phase. The inverter 81 has a U-phase upper arm switch 84t and a lower arm switch 94t, a V-phase upper arm switch 85t and a lower arm switch 95t, and a W-phase upper arm switch 86t and a lower arm switch 96t. A freewheeling diode 84d is connected in parallel to the upper arm switch 84t. A freewheeling diode 94d is connected in parallel to the lower arm switch 94t. A freewheeling diode 85d is connected in parallel to the upper arm switch 85t. A freewheeling diode 95d is connected in parallel to the lower arm switch 95t. A freewheeling diode 86d is connected in parallel to the upper arm switch 86t. A freewheeling diode 96d is connected in parallel to the lower arm switch 96t.

The motor 1 shown in FIG. 1 has a U-phase winding 22u, a V-phase winding 22v, and a W-phase winding 22w as the winding 22 as shown in FIG. The U-phase winding 22u is connected between the upper arm switch 84t and the lower arm switch 94t. The V-phase winding 22v is connected between the upper arm switch 85t and the lower arm switch 95t. The W-phase winding 22w is connected between the upper arm switch 86t and the lower arm switch 96t.

The gate drive circuit 82 controls on and off of the upper arm switches 84t to 86t and the lower arm switches 94t to 96t according to the switching signal received from the controller 70. Specifically, the gate drive circuit 82 applies a voltage High higher than the threshold voltage to the gate electrode when the switch is turned on, and gates a voltage Low lower than the threshold voltage when the switch is turned off. Apply to the electrode. The protection circuit 83 serves to protect the inverter 81 and the gate drive circuit 82. For example, the protection circuit 83 prevents a high current from flowing back to the gate drive circuit 82 from the ground side. Further, when the temperature of the inverter 81 and the gate drive circuit 82 becomes high, the protection circuit 83 turns off all the transistors of the inverter 81 to suppress element destruction due to the high temperature.

The controller 70 is, for example, a microcomputer. The controller 70 may be a dedicated IC such as an ASIC (Application Specific Integrated Circuit). The controller 70 may be configured to have a memory for storing a program and a CPU (Central Processing Unit) for executing processing according to the program.

The controller 70 controls on and off of the upper arm switches 84t to 86t and the lower arm switches 94t to 96t at a constant carrier frequency according to the speed command signal received from the host device on which the motor 1 is mounted. To generate. The controller 70 outputs a switching signal to the gate drive circuit 82 to perform pulse width modulation (PWM) control for the upper arm switches 84t to 86t and the lower arm switches 94t to 96t. The controller 70 estimates the magnetic pole position of the rotor 30 based on the magnetic pole position signal input from the magnetic sensor 50, and calculates the rotation speed of the rotor 30 from the estimated magnetic position. The controller 70 outputs a rotation speed signal indicating the calculated rotation speed to the host device.

When the voltage across the overcurrent detection resistor 71 becomes a constant voltage or higher, the controller 70 forcibly turns off the upper arm switches 84t to 86t and the lower arm switches 94t to 96t. This prevents an overcurrent from flowing through the winding 22. The voltage across the overcurrent detection resistor 71 is equal to or higher than a constant voltage, which corresponds to an overcurrent detection signal input from the overcurrent detection resistor 71 to the controller 70. Further, a temperature sensitive element (not shown in the figure) may be provided on the built-in substrate 11, for example. In this case, when the controller 70 receives a signal indicating that the temperature is abnormally high from the temperature sensitive element, the controller 70 forcibly turns off the upper arm switches 84t to 86t and the lower arm switches 94t to 96t. In this way, it may be possible to prevent an overcurrent from flowing in the winding 22.

Next, the upper arm switches 84t to 86t and the lower arm switches 94t to 96t shown in FIG. 4 will be described. The upper arm switches 84t to 86t and the lower arm switches 94t to 96t are, for example, power MOSFETs. In the first embodiment, the power MOSFET used for these switches is a super junction MOSFET. In the following, the super junction MOSFET will be referred to as SJ-PWM.

FIG. 5 is a diagram showing a configuration example of a power MOSFET used for the upper arm switch and the lower arm switch shown in FIG. The SJ-HPLC 120 has a gate electrode 121, a drain electrode 122, and a source electrode 123. The SJ-HPLC 120 has a configuration in which an oxide film 127, an n + diffusion layer 126, and a p diffusion layer 124 are formed on an n-semiconductor substrate 125. The n-semiconductor substrate 125 is a semiconductor substrate in which N-type conductive impurities are diffused to a low concentration. The n + diffusion layer 126 is a region in which N-type conductive impurities are diffused at a high concentration. The p-diffusion layer 124 is a region in which P-type conductive impurities are diffused. The bottom surface of the p-diffusion layer 124 extends to a position close to n + the diffusion layer 126 of the drain electrode 122.

FIG. 6 is a diagram showing a configuration example of a MOSFET of a comparative example. As shown in FIG. 6, the planar MOSFET 130 has a gate electrode 121, a drain electrode 122, and a source electrode 123. The planar MOSFET 130 has a configuration in which an oxide film 127, an n + diffusion layer 126, and a p diffusion layer 131 are formed on an n-semiconductor substrate 125. The p-diffusion layer 131 is a region in which P-type conductive impurities are diffused. The bottom surface of the p-diffusion layer 131 is located deeper than the bottom surface of the n + diffusion layer 126, but is located close to the top surface of the n-semiconductor substrate 125.

Comparing the SJ-HPLC 120 shown in FIG. 5 and the planar MOSFET 130 shown in FIG. 6, the bottom surface of the p-diffusion layer 124 extends to a position closer to the n + diffusion layer 126 of the drain electrode 122 than the bottom surface of the p-diffusion layer 131. ing. With the configuration shown in FIG. 5, the SJ-HPLC 120 has a smaller on-resistance than the planar MOSFET 130. Therefore, the SJ-HPLC 120 has a smaller switch steady loss Pi [J] than the planar MOSFET 130. As a result, it is possible to improve the efficiency of the electric power of the electric motor 1.

However, since the SJ-HPLC 120 has a large area of the PN junction, which is the junction of the p-diffusion layer 124 with respect to the n-semiconductor substrate 125, the switching loss Plsw [J] tends to be large. Therefore, when the switching frequency is high, such as when the carrier frequency is high, the effect of improving the loss (steady state loss + switching loss) of the SJ-HPLC 120 may not be expected. In the motor 1 of the first embodiment, the switching loss Plsw is improved by performing the control described later.

The electronic components such as the six switches, the gate drive circuit 82, and the six freewheeling diodes shown in FIG. 4 may be formed on one semiconductor chip or may be composed of separate components. good.

Further, in the first embodiment, the case where the controller 70 controls the rotation of the electric motor 1 according to the magnetic pole position signal detected by the magnetic sensor 50 will be described, but the control is not limited to the control using the magnetic sensor 50. The controller 70 performs sensorless control to control the rotation of the motor 1 by estimating the magnetic pole position of the rotor magnet 40 based on the current flowing through the winding 22, the voltage applied to the winding 22, and the voltage generated in the winding 22. You may.

Further, in the first embodiment, the case where the electronic component including the controller 70 is mounted on the built-in substrate 11 provided inside the electric motor 1 has been described, but the present invention is not limited to such a configuration. For example, the magnetic sensor 50 and passive components such as resistors and capacitors may be mounted on the built-in board 11, and the controller 70 and the power IC 80 may be arranged outside the mold stator 10.

Further, the magnetic sensor 50 may be a Hall IC whose output signal is a digital signal, or may be a Hall element whose output signal is an analog signal. FIG. 7 is a block diagram showing an example of a case where the magnetic sensor shown in FIG. 1 is a Hall IC.

As shown in FIG. 7, the magnetic sensor 50 has a sensor unit 51 and an amplification unit 52. The amplification unit 52 includes an amplifier 53, a transistor 54, and a resistance element 55. The output terminal 56 is connected between the collector electrode of the transistor 54 and the resistance element 55. The sensor unit 51 outputs the reference voltage v0 and the detection voltage v corresponding to the magnetic pole to be detected to the amplifier 53. The amplifier 53 amplifies the voltage difference vs between the reference voltage v0 input from the sensor unit 51 and the detection voltage v, and outputs the voltage difference vs. to the base electrode of the transistor 54. The resistance element 55 applies a constant voltage to the collector electrode of the transistor 54. When the amplified voltage difference vs. is input to the base electrode of the transistor 54, the magnetic position signal is output from the output terminal 56.

In the magnetic sensor 50 shown in FIG. 7, a case where the sensor unit 51 and the amplification unit 52 are composed of separate semiconductor chips will be described. The sensor unit 51 is formed of a non-silicon semiconductor which is a semiconductor other than silicon, and the amplification unit 52 is formed of a silicon semiconductor. Such a magnetic sensor 50 is referred to as a non-silicon type Hall IC. Since the non-silicon Hall IC has two semiconductor chips, the sensor center position is arranged at a position different from the center of the IC body. A non-silicon semiconductor such as indium antimonide (InSb) is used for the substrate of the sensor unit 51 of the non-silicon type Hall IC. Such non-silicon semiconductors have advantages such as higher sensitivity and smaller offset due to stress strain than silicon semiconductors. Here, the case where the magnetic sensor 50 is a non-silicon Hall IC has been described, but even if the sensor unit 51 and the amplification unit 52 are formed of a silicon semiconductor and the sensor unit 51 and the amplification unit 52 are composed of one semiconductor chip. good.

Next, the operation of the electric motor 1 of the first embodiment will be described. First, the rotation control of the rotor 30 based on the magnetic position of the rotor 30 will be described. The magnetic sensor 50 outputs a magnetic pole position signal to the controller 70. The controller 70 estimates the magnetic pole position of the rotor 30 based on the magnetic pole position signal input from the magnetic sensor 50. The controller 70 generates a switching signal corresponding to the estimated magnetic pole position and the speed command signal received from the host device (not shown in the figure). The controller 70 outputs the generated switching signal to the power IC 80. The gate drive circuit 82 controls on and off of the upper arm switches 84t to 86t and the lower arm switches 94t to 96t according to the switching signal received from the controller 70. In this way, the controller 70 switches the six power MOSFETs in the inverter 81 at appropriate timings according to the magnetic pole position of the rotor magnet 40 of the rotor 30, so that the rotor 30 obtains a driving force and rotates. do.

Next, the PWM control performed by the controller 70 shown in FIG. 4 on the inverter 81 will be described. The controller 70 performs switching corresponding to the speed command signal by changing the duty ratio Dr of the PWM control. The duty ratio Dr means the energization rate, which is the ratio of the on-time to the period T of the carrier frequency. Assuming that the motor voltage, which is the voltage applied to the winding 22, is Vm, the larger the duty ratio Dr, the larger the motor voltage Vm.

Here, in order to explain the PWM control executed by the controller 70 of the first embodiment in an easy-to-understand manner, the PWM control of the comparative example will be described. It is assumed that the PWM control is a 120-degree energization method and the carrier frequency is equal to or higher than the audible frequency. The audible frequency is, for example, a frequency of 16 kHz.

FIG. 8 is a timing chart showing a case where complementary switching is performed for the upper arm switch and the lower arm switch of each phase as PWM control of a comparative example. The vertical axis of FIG. 8 shows the on and off of the gate electrode of each switch, and the horizontal axis of FIG. 8 is the electric angle. The section Int1 shown in FIG. 8 is a portion that switches according to the PWM control of the duty ratio Dr0. The section Int2 shown in FIG. 8 is a portion that is switched by the inverting signal of the PWM control of the section Int1.

Next, the relationship between the on / off timing of some switches and the current flowing through the winding and the switch in the range between the two broken lines shown in FIG. 8 will be described. In order to make the explanation easy to understand, a case where a current is passed from the U-phase winding 22u to the V-phase winding 22v will be described.

9 to 11 are diagrams showing the operation procedure of the upper arm switch and the lower arm switch of the U phase and the V phase in the range sandwiched by the two broken lines shown in FIG. FIG. 12 is an enlarged view showing the operation of the U-phase upper arm switch and the lower arm switch in FIGS. 9 to 11.

As shown in FIG. 9, the U-phase upper arm switch 84t is turned on, and the V-phase lower arm switch 95t is turned on. As a result, a current flows from the inverter 81 to the V-phase winding 22v via the U-phase winding 22u.

As shown in FIG. 10, according to the duty ratio Dr0 corresponding to the switching signal, the U-phase upper arm switch 84t is switched from the on state to the off state during the dead time td. At this time, since the current tends to continue to flow due to the inductor component of the winding 22, the current flows through the freewheeling diode 94d of the U-phase lower arm switch 94t. The dead time td is a time for preventing the upper arm switch 84t and the lower arm switch 94t from being short-circuited. Further, in the period T of the carrier frequency, the off time of the upper arm switch 84t is set to toff.

After the dead time td has elapsed, as shown in FIG. 11, the U-phase lower arm switch 94t is turned on. As shown in FIG. 12, after the dead time dt elapses after the upper arm switch 84t is switched from the on state to the off state, the lower arm switch 94t is turned on. As a result, the current flowing through the U-phase freewheeling diode 94d flows through the lower arm switch 94t. The switch steady-state loss caused by the current flowing while the lower arm switch 94t is on is defined as Pi [J]. The switch steady-state loss Pi is represented by Pi = wi × (toff-td) [J], where wi [W] is the switch steady-state loss power.

Further, the steady loss caused by the current flowing through the freewheeling diode 94d is defined as Pd [J]. Assuming that the steady loss power of the freewheeling diode is wd [W], the steady loss Pd of the freewheeling diode 94d is represented by Pd = wd × (toff-td) [J]. Since the lower arm switch 94t is the SJ-HPLC120, the switch steady loss Pi of the lower arm switch 94t is smaller than the steady loss Pd of the freewheeling diode 94d. Therefore, the power loss of the entire circuit can be suppressed to a low level.

On the other hand, during the operation shown in FIG. 11, a switching loss Plsw occurs. The switching loss Plsw is a value proportional to the voltage E × the current I × Δt, where Δt is the switching time. The current I is the current flowing through the switching switch. Since the SJ-HPLC 120 has a larger switching loss than the planar MOSFET 130, the power loss may be large. The operations shown in FIGS. 9 to 11 are performed every cycle T of the carrier frequency.

Next, the PWM control executed by the controller 70 of the first embodiment will be described. It is assumed that the PWM control is a 120-degree energization method and the carrier frequency is equal to or higher than the audible frequency. FIG. 13 is a timing chart showing a case where the controller shown in FIG. 4 performs non-complementary switching with respect to the upper arm switch and the lower arm switch of each phase. The vertical axis of FIG. 13 shows the on and off of the gate electrode of each switch, and the horizontal axis of FIG. 13 is the electric angle. The section Int1 shown in FIG. 13 is a portion that switches according to the PWM control of the duty ratio Dr1.

Next, the relationship between the on / off timing of some switches and the current flowing through the winding and the switch in the range between the two broken lines shown in FIG. 13 will be described. In order to make the explanation easy to understand, a case where a current is passed from the U-phase winding 22u to the V-phase winding 22v will be described.

14 and 15 are diagrams showing the operation procedure of the U-phase and V-phase upper arm switches and lower arm switches in the range sandwiched by the two broken lines shown in FIG. FIG. 16 is an enlarged view showing the operation of the U-phase upper arm switch and lower arm switch in FIGS. 14 and 15.

As shown in FIG. 14, the controller 70 turns on the U-phase upper arm switch and turns on the V-phase lower arm switch. As a result, a current flows from the inverter 81 to the V-phase winding 22v via the U-phase winding 22u.

As shown in FIG. 15, the controller 70 switches the U-phase upper arm switch 84t from the on state to the off state according to the duty ratio Dr1. The duty ratio Dr1 will be described in detail later. In the first embodiment, as shown in FIG. 16, the controller 70 does not turn on the lower arm switch 94t. In the operation shown in FIG. 15, since the current tends to continue to flow due to the inductor component of the winding 22, the current flows through the freewheeling diode 94d of the U-phase lower arm switch 94t. The loss generated at this time is the steady loss Pd of the freewheeling diode 94d. The operations shown in FIGS. 14 and 15 are performed every cycle T of the carrier frequency.

In the control shown in FIGS. 14 and 15, the controller 70 does not switch the U-phase lower arm switch 94t, so that the switching loss Plsw does not occur. When each switch is an SJ-HPLC 120 having a large switching loss, the control of the first embodiment has a smaller power loss of the entire circuit than the PWM control of the comparative example.

FIG. 17 is a diagram for comparing the duty ratio in the PWM control of the comparative example and the duty ratio in the PWM control of the first embodiment. The on-time of the upper arm switch 84t of the complementary switching of the comparative example is ton0, and the on-time of the upper arm switch 84t of the non-complementary switching of the first embodiment is ton1. The duty ratio Dr0 is represented by Dr0 = (ton0 / T). The duty ratio Dr1 is represented by Dr1 = (ton1 / T). From FIG. 17, it can be seen that the relationship is Dr0 <Dr1.

In FIG. 17, in the U-phase upper arm switch 84t of the comparative example, a current flows through the V-phase freewheeling diode 95d during the on-time ton0, so that the steady-state loss is reduced. However, when the upper arm switch 84t is switched from the off state to the on state, a switching loss Plsw occurs. On the other hand, in the U-phase upper arm switch 84t of the first embodiment, the on-time ton1 is longer than the on-time ton0, and the switch is not switched. Therefore, switching loss Plsw does not occur. Further, in the U-phase upper arm switch 84t of the first embodiment, the off time is shortened by the lengthening of the on-time ton1, so that the time for the current to flow through the freewheeling diode 94d is shortened. Therefore, the power loss is suppressed.

In the PWM control of the comparative example described with reference to FIGS. 9 to 11, the power loss in the period T of the carrier frequency is Plsw + Pi. On the other hand, in the PWM control described with reference to FIGS. 13 and 14, the power loss in the period T of the carrier frequency is Pd. From these contents, if the condition of Plsw + Pi> Pd is not satisfied, the effect of reducing the power loss by applying the SJ-HPLC 120 to each switch cannot be obtained. Therefore, in the first embodiment, the controller 70 performs PWM control with a duty ratio Dr1 satisfying the condition of Plsw + Pi> Pd. Assuming that the loss difference between the steady-state loss Pd of the freewheeling diode and the switch steady-state loss Pi is Pdi, the loss difference Pdi is calculated by Pdi = Pd-Pi. That is, the controller 70 performs PWM control with a duty ratio Dr1 that satisfies the condition of Plsw> Pdi.

The steady loss Pi of the switch is proportional to (on resistance x current), and the steady loss Pd of the freewheeling diode is proportional to (forward voltage x current). It is smaller than the loss Pd. Therefore, in the case of complementary switching, the loss difference is large between the on-time and the off-time of the upper arm switch. In particular, since the SJ-HPLC 120 has a small on-resistance, the loss difference between the on-state and the off-state of the upper arm switch becomes remarkable in complementary switching. Therefore, when the switch is SJ-HPLC120, it is better to switch the upper arm switch non-complementarily with a high duty ratio than to control the upper arm switch and the lower arm switch by complementary switching, even if the output of the motor is the same, the power loss. Is suppressed. As a result, power efficiency is improved.

The induced voltage constant, which is the coefficient of the induced voltage generated in the winding 22 by the rotation of the rotor 30, is Ke, the rotation speed is N, the voltage generated by the winding resistance is Vr, and the voltage generated by the winding inductance is Vl. Then, the motor voltage Vm is expressed by the following equation (1).
Vm = Ke × N + Vr + Vl ・ ・ ・ (1)

Since the motor voltage Vm is a value proportional to (voltage E × duty ratio Dr), it is proportional to the duty ratio Dr. From the equation (1), some parameters for increasing the duty ratio Dr can be considered. The induced voltage constant Ke of the first term on the right side is one of the parameters affecting the duty ratio Dr. The larger the induced voltage constant Ke of the motor 1, the larger the duty ratio Dr, and the power loss of the entire circuit can be reduced.

Further, the induced voltage constant Ke is proportional to the number of turns of the winding 22. Therefore, by increasing the number of turns of the winding 22, the power efficiency is improved. Further, the induced voltage constant Ke is proportional to the magnetic force of the main magnet portion of the rotor 30. Therefore, by increasing the magnetic force of the main magnet portion, the power efficiency is improved. By increasing the number of turns of the winding 22, the induced voltage constant Ke becomes large. As a result, the motor voltage Vm becomes large and the duty ratio Dr also becomes large.

In particular, by increasing the number of turns of the winding 22 and reducing the wire diameter of the winding, it is possible to further improve the power efficiency with the same winding weight. If the winding weight is the same, the cost for the winding having a high material unit price does not change, so that it is possible to suppress the increase in the manufacturing cost of the motor 1.

In the first embodiment, the case where the carrier frequency is equal to or higher than the audible frequency has been described, but the frequency may be lower than the audible frequency. When the electric motor 1 is a fan motor of an air conditioner, since the fan motor and the fan are installed in the air passage, it is difficult to take measures against the noise generated by driving the motor. Therefore, it is desirable that the carrier frequency is equal to or higher than the audible frequency. However, when the motor 1 can be covered with a soundproof material or the like, such as when the motor 1 is used for a compressor, the carrier frequency may be set to a value smaller than the audible frequency. In this case, the switching loss Plsw can be reduced.

The motor 1 of the first embodiment is non-complementary to the rotor 30, the stator 20, three sets of upper arm switch and lower arm switch, and three sets of upper arm switch and lower arm switch at a constant carrier frequency. It has a controller 70 that performs pulse width modulation by switching. The three sets of upper arm switch and lower arm switch are SJ- MOSFETs. The controller 70 performs pulse width modulation with a duty ratio of an on-time longer than the on-time of the upper arm switch when the upper arm switch and the lower arm switch operate by complementary switching.

In the first embodiment, since the controller 70 operates the upper arm switch and the lower arm switch of the inverter 81 by non-complementary switching, the switching loss Plsw due to the switching operation of the lower arm switch does not occur. Further, the controller 70 performs pulse width modulation with a high duty ratio Dr1 in which the on-time of the upper arm switch is longer than in the case of complementary switching, so that a current flows through a freewheeling diode having a larger loss than the switch steady loss Pi. The time will be shorter. Therefore, even if the SJ-HPLC 120 is used for the upper arm switch and the lower arm switch, the power loss of the entire circuit including the inverter 81 can be suppressed.

In the first embodiment, the controller 70 may perform PWM control with a duty ratio Dr1 satisfying the condition of Plsw> Pdi. In this case, the power loss of the entire circuit is reduced and the power efficiency is improved.

Embodiment 2.
The second embodiment is an air conditioner equipped with the electric motor described in the first embodiment. In the second embodiment, the same configurations are designated by the same reference numerals to the configurations described in the first embodiment, and detailed description thereof will be omitted.

The configuration of the air conditioner according to the second embodiment of the present invention will be described. FIG. 18 is an external view showing a configuration example of an air conditioner according to the second embodiment of the present invention. The air conditioner 200 has an indoor unit 210 and an outdoor unit 220 connected to the indoor unit 210 by a refrigerant pipe 251. The outdoor unit 220 has a heat source side blower 223.

FIG. 19 is a refrigerant circuit diagram showing a configuration example of the air conditioner shown in FIG. The outdoor unit 220 includes a compressor 221 that compresses and discharges the refrigerant, a four-way valve 226 that switches the flow direction of the refrigerant, a heat source side heat exchanger 222 that exchanges heat with the outside air, and a heat source side heat exchanger that exchanges the outside air with the outside air. It has a heat source side blower 223 that supplies the 222. A motor 224 is connected to the heat source side blower 223 as a drive source for the fan.

The indoor unit 210 includes a throttle device 211 that decompresses and expands a high-pressure refrigerant, a load-side heat exchanger 212 that exchanges heat with the air in the air-conditioned space, and a load-side heat exchanger 212 that exchanges heat with the air in the air-conditioned space. It has a load side blower 213 to supply to. A motor 214 is connected to the load-side blower 213 as a drive source for the fan. As the electric motors 214 and 224, the electric motors 1 described in the first embodiment are used.

The compressor 221 and the heat source side heat exchanger 222, the throttle device 211 and the load side heat exchanger 212 are connected by a refrigerant pipe 251 to form a refrigerant circuit 250 in which the refrigerant circulates. In the configuration example shown in FIG. 19, the indoor unit 210 has a control device 215 that controls the refrigeration cycle of the air conditioner 200, but the control device 215 may be provided in the outdoor unit 220. The control device 215 is connected to the built-in board 11 by the lead wire 13 shown in FIG.

FIG. 20 is a side perspective view showing an example of the configuration of the indoor unit shown in FIG. FIG. 21 is a side perspective view showing a configuration example of the indoor unit of the comparative example. In FIGS. 20 and 21, for the sake of explanation, the main configuration of the indoor unit is shown, and the other configurations are omitted from the drawings.

FIG. 20 shows a case where the indoor unit 210 is attached to the wall 300. When the load-side blower 213 rotates, the air in the room, which is the space to be air-conditioned, is sucked into the indoor unit 210, and after heat exchange with the refrigerant in the load-side heat exchanger 212, it is blown out into the room from the outlet 230. To.

As shown in FIG. 21, the indoor unit 310 of the comparative example is attached to the wall 300. The indoor unit 310 has a load side heat exchanger 312 and a load side blower 313. By rotating the load-side blower 313, the air in the room is sucked into the indoor unit 310, heat is exchanged with the refrigerant in the load-side heat exchanger 312, and then blown out into the room from the outlet 330.

As shown in FIGS. 20 and 21, the indoor unit 210 of the air conditioner 200 of the second embodiment has a larger housing and fan diameter than the indoor unit 310 of the comparative example. Therefore, the load-side blower 213 can blow out a larger amount of air at a low rotation speed. As a result, the load-side blower 213 has a lower maximum rotation speed than the load-side blower 313 of the indoor unit 310 of the comparative example.

Referring to the equation (1) described in the first embodiment, since the rotation speed N is included in the first term on the right side, the maximum rotation speed decreases as the induced voltage constant Ke increases. On the other hand, in the case of the load side blower 213 having a low maximum rotation speed as in the second embodiment, the influence of this demerit is small. By increasing the diameter of the housing of the indoor unit 210 and the fan, it is possible to improve the power efficiency of the air conditioner 200 as a whole. Further, by rotating the load-side blower 213, even if there is a sliding portion where two or more objects rub against each other, the lower the rotation speed of the blower, the smaller the noise generated in the sliding portion. As a result, there is an advantage that the noise generated in proportion to the magnitude of the air volume is reduced.

Further, in the air conditioner, the rated operation rotation speed is generally set separately from the maximum rotation speed. In the rated operation, the rotation speed may differ in each operation mode of heating operation, cooling operation, and dehumidification operation. For example, different rotation speeds are set for each operation mode of heating operation, cooling operation, and dehumidification operation. Since the temperature is proportional to the air density, the torque with respect to the rotation speed of the blower may be different even if the rotation speed of the blower is the same. In such a case, it is necessary to control the inverter of the electric motor of the blower with a different duty ratio Dr in response to the required torque. In particular, in the case of an outdoor unit, the difference in required torque is remarkable because the temperature difference between the outside air temperature during heating operation and the outside air temperature during cooling operation is large. The efficiency of rated operation affects the power efficiency when air conditioners are used in the market, and as a result, the electricity bill is also greatly affected.

In recent years, since air conditioners have increased the diameter of fans to improve aerodynamic efficiency, blowers tend to operate at low rotation speeds and high torque. In the second embodiment, by applying the electric motor 1 described in the first embodiment to the blower motor, it is possible to realize an air conditioner in which the load on the fan is reduced and the power consumption efficiency is improved.

Further, if SJ- MOSFET is used for the switching element of the inverter of a motor having a high carrier frequency and a small current flowing in a winding such as a blower motor, the switching loss becomes large and the power loss of the entire circuit can be reduced. do not have. On the other hand, in the second embodiment, even if the electric motor 1 is used as the drive source of the fan motor, the power loss of the entire circuit can be reduced as compared with the planar MOSFET as described above.

In the second embodiment, the case where the motor 1 is mounted on the air conditioner 200 has been described, but the device on which the motor 1 is mounted is not limited to the air conditioner 200. The motor 1 may be mounted on other devices such as ventilation fans, home appliances and machine tools. Even if the electric motor 1 is mounted on these devices, the same effect as that of the second embodiment can be obtained.

1 Motor, 10 Molded resistor, 11, 11a Built-in board, 12 Molded resin, 13 Lead wire, 14 Lead outlet, 20 Controller, 21 Transformer core, 22 winding, 22u U-phase winding, 22v V-phase winding Wire, 22w W-phase winding, 23 insulator, 30 rotor, 31 rotating shaft, 33 output side bearing, 34 non-output side bearing, 35 through hole, 36 notch, 40 rotor magnet, 50 magnetic sensor, 51 sensor part , 52 Amplifier, 53 Amplifier, 54 Transistor, 55 Resistor, 56 Output Terminal, 60 Bracket, 61 Press Fit, 70 Controller, 71 Overcurrent Detection Resistor, 79 Module, 80 Power IC, 81 Inverter, 82 Gate Drive Circuit, 83 protection circuit, 84d-86d recirculation diode, 84t-86t upper arm switch, 94d-96d recirculation diode, 94t-96t lower arm switch, 120 SJ-PWM, 121 gate electrode, 122 drain electrode, 123 source electrode, 124 p diffusion Layer, 125 n-semiconductor substrate, 126 n + diffusion layer, 127 oxide film, 130 planar MOSFET, 131 p diffusion layer, 200 air conditioner, 210 indoor unit, 211 throttle device, 212 load side heat exchanger, 213 load side blower , 214 motor, 215 controller, 220 outdoor unit, 221 compressor, 222 heat source side heat exchanger, 223 heat source side blower, 224 motor, 226 four-way valve, 230 outlet, 250 refrigerant circuit, 251 refrigerant pipe, 300 wall, 310 Indoor unit, 312 Load side heat exchanger, 313 Load side blower, 330 Outlet.

Claims (7)

With the rotor with the axis of rotation inserted,
A stator provided on the outer circumference of the rotor and provided with three-phase windings,
An inverter including three sets of upper arm switches and lower arm switches corresponding to the three phases and a freewheeling diode connected in parallel to each of the three sets of upper arm switches and lower arm switches.
It has a controller that performs pulse width modulation of the three sets of upper arm switch and lower arm switch by non-complementary switching at a constant carrier frequency.
Each of the three sets of upper arm switch and lower arm switch is a super junction field effect transistor.
The controller is an electric motor that performs pulse width modulation with a duty ratio of an on-time longer than the on-time of the upper arm switch when the upper arm switch and the lower arm switch operate by complementary switching. When the switching loss when the upper arm switch and the lower arm switch operate in the complementary switching is Plsw, the steady loss of the freewheeling diode is Pd, and the steady loss of the lower arm switch is Pi.
The motor according to claim 1, wherein the controller performs the pulse width modulation with a duty ratio satisfying the condition of Plsw> Pd-Pi. The steady loss power of the freewheeling diode is wd, the steady loss power of the lower arm switch is wi, and the off time of the upper arm switch of the complementary switching with respect to the period of the carrier frequency is toff. When the dead time from when the upper arm switch is turned off to when the lower arm switch is turned on is td.
The steady loss Pd = wd × (toff-td) of the freewheeling diode.
The steady loss Pi = wi × (toff-td) of the lower arm switch.
The motor according to claim 2. The motor according to claim 2 or 3, wherein the duty ratio satisfying the above conditions is proportional to an induced voltage constant which is a coefficient of an induced voltage generated in the winding by the rotation of the rotor. The motor according to claim 4, wherein the winding is the number of winding turns that becomes the induced voltage constant. The motor according to any one of claims 1 to 5, wherein the controller performs the pulse width modulation at the carrier frequency equal to or higher than the audible frequency. Indoor units including load side blowers and
An outdoor unit including a blower on the heat source side,
The motor according to any one of claims 1 to 6, which is provided as a drive source for at least one of the load-side blower and the heat-source-side blower.
Air conditioner with.

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