JP4265354B2 - Bidirectional DC-DC converter - Google Patents

Description

The present invention relates to a bidirectional DC-DC converter, and more particularly to loss reduction in a bidirectional DC-DC converter.

  As shown in FIG. 13, in the chopper circuit 260 disclosed in Patent Document 1, when the motor 110 is at a high output, the voltage of the battery 130 is boosted to supply power to the motor 110, and when the motor 110 is regenerated, the motor power generation voltage is increased. A circuit configuration for charging the battery 130 by lowering the voltage is disclosed. Here, the motor 110 is an induction motor used for a traveling motor such as an electric vehicle, and shows a case where it is driven by an inverter circuit 150.

  When rotating the motor 110 at a high rotational speed, the control circuit 420 turns on the transistor 280 and causes the current from the battery 130 to flow through the reactor 320 to accumulate electromagnetic energy. Thereafter, the transistor 280 is turned off, and the electromagnetic energy stored in the reactor 320 is stored in the capacitor 220 via the flywheel diode 290. The voltage between the terminals of the capacitor 220 becomes higher than the voltage of the battery 130, and the chopper circuit 260 functions as a step-up chopper.

  Moreover, when the rotation speed of the motor 110 falls, the motor 110 becomes regenerative braking. The control circuit 420 turns on the transistor 270 and charges the battery 130 with the regenerative current from the motor 110 via the reactor 320. Thereafter, the transistor 270 is turned off, and the electromagnetic energy stored in the reactor 320 is stored in the battery 130. The motor power generation voltage is stepped down to charge the battery 130. The chopper circuit 260 functions as a step-down chopper.

  In addition, there exists a DC-DC converter currently disclosed by patent document 2 as another related technique.

JP-A-8-214592 (paragraphs 0048 to 0051, FIG. 1) JP 2003-33013 A

  However, in the chopper circuit 260 of Patent Document 1, the transistors 270 and 280 are switched on / off from an off state in which a voltage is applied between the transistor terminals to an on state, and a current flows through the transistor. When switching from the on-state to the off-state, the transition of the conductive state of the transistors 270 and 280 takes a predetermined transition time. Therefore, in the transient state during the transition period, a voltage is applied between the terminals of the transistor. However, current flows and power is consumed. So-called switching loss occurs. This switching loss is dominant as a loss in switching control of transistors such as the chopper circuit 260 or the DC-DC converter.

  Due to the switching loss, the ratio of the driving power of the motor 110 to the input power from the battery 130, that is, the power efficiency is lowered, which is a problem. In addition, the power consumed unnecessarily due to the switching loss is converted into heat energy and causes heat generation of the device. When the amount of heat generated is limited due to the usage environment of the device, the cooling device such as a heat sink is increased in size. This is a problem because it may increase the size and weight of the equipment.

The present invention has been made to solve at least one of the problems of the prior art, and the switching operation is performed with a small voltage applied between the terminals of the switching element, thereby reducing the switching loss. Thus, it is an object of the present invention to provide a bidirectional DC-DC converter capable of improving power efficiency and reducing the amount of heat generation and capable of high-frequency operation.

In order to achieve the object, a bidirectional DC-DC converter according to claim 1 is connected in series between a first inductor having one terminal connected to a primary power supply terminal, and a secondary power supply terminal and a reference terminal. A bidirectional DC-DC converter comprising an upper switching element and a lower switching element, wherein the other terminal of the first inductor is connected to a connection point of the upper and lower switching elements. When electromagnetic energy is stored in the first inductor by conduction between the capacitor connected in parallel between at least one of the current path terminals and the upper switching element, the current path terminals between the upper switching elements are conducted prior to conduction of the upper switching element. In order to minimize the applied voltage, a larger current than the current flowing through the first inductor must be connected. When storing electromagnetic energy in the first inductor by conduction of the first auxiliary switching element that forms the first auxiliary current path flowing into the first switching element and the lower switching element, the current path terminal of the lower switching element prior to conduction of the lower switching element And a second auxiliary switching element that forms a second auxiliary current path for flowing a current larger than the current flowing through the first inductor from the connection point. 2 Auxiliary current path has one winding of the transformer, the other winding of the transformer has an intermediate tap, one of the intermediate tap and both side terminals is connected to the reference terminal, the other is the secondary power supply terminal or primary The first and second terminals connected to the power supply terminal are connected with the path from the reference terminal to the secondary power supply terminal or the primary power supply terminal as the forward direction. Characterized in that it comprises a second diode.

In the bidirectional DC-DC converter according to claim 1, when current flows from the secondary power supply terminal to the first inductor due to conduction of the upper switching element, and the electromagnetic energy is accumulated, the accumulated electromagnetic energy is stored in the lower switching element. It is discharged to the primary power supply terminal with conduction. At this time, prior to the conduction of the upper and lower switching elements, the lower and upper switching elements are rendered non-conductive so that there is no period in which the upper switching element and the lower switching element are both conducted.
When the upper switching element is non-conductive, if the capacitor is connected in parallel between the current path terminals of the upper switching element, the capacitor is in a discharged state, and the capacitor is connected in parallel between the current path terminals of the lower switching element. If connected, the capacitor is in a charged state, so that the voltage change at the connection point is performed slowly. For this reason, the upper switching element is rendered non-conductive with a small voltage difference between the current path terminals. Prior to the subsequent re-conduction of the upper switching element, the conduction of the first auxiliary switching element causes a larger current to flow from the first auxiliary current path to the connection point than the current when electromagnetic energy is stored in the first inductor. Therefore, the upper switching element is conducted with a small voltage difference between the current path terminals. Here, due to the current flowing through the first auxiliary current path, a part of the electromagnetic energy of the first inductor is returned to the secondary power supply terminal or the primary power supply terminal via the transformer and the first or second diode.
The conduction of the lower switching element is charged by a capacitor connected in parallel between the current path terminals of the upper switching element by electromagnetic energy accumulated in the first inductor in response to the non-conduction of the upper switching element. In addition, if a capacitor is connected in parallel between the current path terminals of the lower switching element, the capacitor is discharged and then performed. The lower switching element conducts in a state where the voltage difference between the current path terminals is small. Further, the non-conduction of the lower switching element means that the capacitor is in a charged state when the capacitor is connected in parallel between the current path terminals of the upper switching element, and the capacitor is connected between the current path terminals of the lower switching element. If the capacitors are connected in parallel, the capacitor is in a discharged state, so that the voltage at the connection point is gradually changed. For this reason, the lower switching element is rendered non-conductive with a small voltage difference between the current path terminals.
Further, when current flows from the primary power supply terminal to the first inductor due to the conduction of the lower switching element, and the electromagnetic energy is accumulated, the accumulated electromagnetic energy is released to the secondary power supply terminal with the conduction of the upper switching element. . At this time, prior to the conduction of the upper and lower switching elements, the lower and upper switching elements are rendered non-conductive so that there is no period in which the upper switching element and the lower switching element are both conducted.
The non-conducting state of the lower switching element means that when the capacitor is connected in parallel between the current path terminals of the upper switching element, the capacitor is in a charged state and the capacitor is connected in parallel between the current path terminals of the lower switching element. If the capacitor is connected, the capacitor is in a discharged state, so that the voltage change at the connection point is performed slowly. For this reason, the lower switching element is rendered non-conductive with a small voltage difference between the current path terminals. Prior to the subsequent re-conduction of the lower switching element, a large current flows from the connection point into the second auxiliary current path due to the conduction of the second auxiliary switching element, compared to the current when electromagnetic energy is accumulated in the first inductor. Therefore, the lower switching element is conducted with a small voltage difference between the current path terminals. Here, due to the current flowing through the second auxiliary current path, a part of the electromagnetic energy of the main inductor is returned to the secondary power supply terminal or the primary power supply terminal via the transformer and the first or second diode.
The conduction of the upper switching element is caused by the electromagnetic energy accumulated in the first inductor in response to the non-conduction of the lower switching element, if the capacitor is connected in parallel between the current path terminals of the upper switching element. In addition, if a capacitor is connected in parallel between the current path terminals of the lower switching element, the capacitor is charged and then performed. The upper switching element conducts in a state where the voltage difference between the current path terminals is small. Further, when the upper switching element is non-conductive, if the capacitor is connected in parallel between the current path terminals of the upper switching element at the time of previous conduction, the capacitor is in a discharged state, and the capacitor between the current path terminals of the lower switching element is Are connected in parallel, the capacitor is in a charged state, so that the voltage difference between the current path terminals of the upper switching element is made non-conductive in a state where the voltage difference is small.

Thereby, in the bidirectional DC-DC converter, the switching operation can be performed with a small voltage difference between the current path terminals in the upper and lower switching elements. Switching loss consumed by the upper and lower switching elements during switching can be reduced.

With the reduction of switching loss, it is possible to improve the power efficiency in voltage conversion of the bidirectional DC-DC converter. Heat generation due to switching loss in the upper and lower switching elements can be reduced, and a cooling device such as a heat sink can be reduced in size and weight.

The switching operation in the bidirectional DC-DC converter can be performed at a high frequency, and the switching operation can be performed at a frequency higher than the audible frequency band. The vibration of the first inductor or the like accompanying the electromagnetic energy during operation can be shifted from the audible frequency band, and abnormal noise during operation can be prevented.

Further, the first and second auxiliary current paths formed to conduct the first switching element are provided with one winding of the transformer. The transformer usually includes a leakage inductance connected in series with one winding as an equivalent circuit, and is considered to have an inductance value smaller than the inductance value of the first inductor. This inductance value is combined with the voltage across the terminals caused by the voltage value of one winding and the voltage value of the connection point induced according to the winding ratio of the transformer, and the time slope of the current input to the first inductor In contrast, a current having a steeper time gradient is caused to flow through the first and second auxiliary current paths. The timing at which the magnitude relation of the current value is reversed and the time width of the reversed state can be adjusted by the time gradient adjusted according to the leakage inductance value and the winding ratio of the transformer, and the conduction of the first and second switching elements can be adjusted. The timing can be set stably within a sufficient time margin.

Also, the current flowing through the first and second auxiliary current paths can be discharged from the other winding to the low-voltage power supply terminal or the high-voltage power supply terminal via the transformer, and the power efficiency in the voltage conversion of the bidirectional DC-DC converter. Can be improved.
Thereby, a bidirectional converter with reduced switching loss can be configured.

Furthermore, the bidirectional DC-DC converter according to claim 2, in the bidirectional DC-DC converter according to claim 1, between one winding connection points and the transformer, characterized in that it comprises a second inductor And

In the bidirectional DC-DC converter according to claim 2, the time gradient of the current flowing through the auxiliary current path is determined by the second inductor instead of or together with the leakage inductance of the transformer. Thereby, the time inclination of the current flowing through the first and second auxiliary current paths can be adjusted by adjusting the inductance value of the second inductor.

A bidirectional DC-DC converter according to claim 3 is applied to the other winding when the first and second auxiliary current paths are formed in the bidirectional DC-DC converter according to claim 1 or 2. The leakage inductance component of the transformer, the second inductor, or the leakage existing between the connection point and the one winding due to the voltage induced in the one winding in accordance with the winding ratio of the transformer The voltage applied to the combined inductor of the inductance component and the second inductor is approximately half of the voltage at the secondary power supply terminal.

  As a result, the time slope of each increase / decrease in current can be made substantially equal for the current flowing through the leakage inductance component of the transformer, the second inductor, or the combined inductor of the leakage inductance component and the second inductor.

Furthermore, the bidirectional DC-DC converter according to claim 4, in the bidirectional DC-DC converter according to claim 1, between the upper and current path terminal of the lower switching element, storing electromagnetic energy in the first inductor And an anti-parallel diode connected with the direction opposite to the current direction of the current flowing as a forward direction.

In the bidirectional DC-DC converter according to claim 4, in comparison with the current flowing when electromagnetic energy is accumulated in the first inductor in the first and second auxiliary current paths prior to re-conduction of the upper and lower switching elements. When a large current flows, a part of it flows through the antiparallel diode. Thereby, the voltage difference between the current path terminals in the upper and lower switching elements can be clamped to the forward voltage of the antiparallel diode.

According to the present invention, it is possible to provide a bidirectional DC-DC converter in which a switching operation can be performed with a small voltage applied between the terminals of the switching element, and switching loss is reduced.

Hereinafter, embodiments of the bidirectional DC-DC converter of the present invention will be described in detail with reference to the drawings based on FIGS. In FIGS. 3 to 6 and FIGS. 8 to 11, the circuit elements in the non-conduction state are omitted for easy understanding of the conduction state.

FIG. 1 is a circuit diagram of the bidirectional DC-DC converter of the first embodiment. It has a circuit configuration in which the auxiliary circuit unit 1 which is an invention part is added to the bidirectional converter.

The bidirectional converter boosts the input voltage VIN and supplies it to the load LD. When the load LD is a motor or the like, the bidirectional converter has a function of reducing the regenerative energy and recharging the input voltage VIN. The bidirectional converter shown in FIG. 1 is a so-called non-insulated DC-DC converter in which a reference terminal is commonly connected between an input voltage VIN and a load LD. The high voltage side terminal of the input voltage VIN is an input terminal, and the high voltage side terminal of the load LD is an output terminal. In the transistors Q1 and Q2, the emitter terminal of the transistor Q1 and the collector terminal of the transistor Q2 are connected at the connection point X, the collector terminal of the transistor Q1 is connected to the output terminal, and the emitter terminal of the transistor Q2 is connected to the reference voltage. The output terminal and the reference voltage are connected in series. The base terminals of the transistors Q1 and Q2 are exclusively controlled by a controller (not shown). Further, anti-parallel diodes D1 and D2 are connected to the transistors Q1 and Q2 in the forward direction from the emitter terminal to the collector terminal. An inductor L1 is connected between the connection point X and the input terminal. Capacitors CIN and COUT are connected in parallel with the input voltage VIN and the load LD between the input terminal, the output terminal, and the reference terminal. Here, the transistor Q1 is an upper switching element, and the transistor Q2 is a lower switching element. The input terminal is a primary power supply terminal, and the output terminal is a secondary power supply terminal. The input terminal to which the input voltage VIN is connected is a low voltage power supply terminal, and the output terminal connected to the load LD is a high voltage power supply terminal. Although the load LD is not specifically illustrated, for example, an induction motor driven through an inverter circuit or the like can be considered. An example is a case where the present invention is applied to a hybrid vehicle that travels by switching between a gasoline engine and motor drive, an electric vehicle that travels only by motor drive, and the like. For example, 300 V is supplied to the input voltage VIN, and 500 V is supplied to the output voltage VOUT to be supplied to the load LD.

  When operating as a boost converter that boosts the input voltage VIN to the output voltage VOUT, the electromagnetic energy accumulated in the inductor L1 due to the conduction of the transistor Q2 is supplied to the load LD via the transistor Q1 and the antiparallel diode D1. Done. When operating as a step-down converter that steps down the output voltage VOUT to the input voltage VIN, the electromagnetic energy accumulated in the inductor L1 due to the conduction of the transistor Q1 is supplied to the input voltage VIN via the transistor Q2 and the antiparallel diode D2. Is done.

  Here, the capacitors CIN and COUT are smoothing capacitors. Transistors such as IGBT, MOS, and bipolar can be used as the transistors Q1 and Q2. In this case, the anti-parallel diodes D1 and D2 can be separately connected to a diode element in addition to the case where they are built in the transistors Q1 and Q2.

  The auxiliary circuit unit 1 includes capacitors C1 and C2 connected between the collectors and emitters of the transistors Q1 and Q2. Further, first and second auxiliary current paths are formed between the connection point X of the transistors Q1 and Q2, and the output terminal and the reference terminal.

  From the connection point X to the one winding of the inductor L2 and the transformer T1, it is common to the first and second auxiliary current paths. In the first auxiliary current path, a path is formed from one winding to the output terminal via the transistor Q3. In the second auxiliary current path, a path is formed from one winding to the reference terminal via the transistor Q4. Here, the transistors Q3 and Q4 are first and second auxiliary transistors.

  The other winding of the transformer T1 is provided with an intermediate tap connected to the reference terminal, both end terminals are connected to the anodes of the diodes D3 and D4, and the cathodes of the diodes D3 and D4 are connected to the output terminal. ing.

First, with reference to FIG. 2 and FIGS. 3 to 6, the boosting operation in the bidirectional converter of the first embodiment will be described. FIG. 2 shows a timing chart, and FIGS. 3 to 6 show circuit operation states in each operation. In the following description, the timing chart (FIG. 2) of the boosting operation will be described with reference to the operation state on the circuit (FIGS. 3 to 6) as appropriate. In FIG. 2, VGQ1, VGQ2, and VGQ4 are voltages applied to the base terminals GQ1, GQ2, and GQ4 of the transistors Q1, Q2, and Q4. IL1 and IL2 are currents flowing through the inductors L1 and L2 having a positive direction from the input voltage VIN to the connection point X and the current from the connection point X to the transformer T1, ID4 is a current flowing through the diode D4, and ID1 and ID2 are , Currents flowing in the anti-parallel diodes D1 and D2, IQ1 and IQ2 indicate currents flowing in the transistors Q1 and Q2. VQ2 indicates a voltage at the connection point X. VL2 is a voltage applied between the terminals of the inductor L2.

  In FIG. 2, (1) and (2), and FIG. 3 are accumulation periods of electromagnetic energy in the inductor L1. In the period (1) in FIG. 2, electromagnetic energy is accumulated in the inductor L1. FIG. 3 (1) shows the operating state during this period. The gate voltage VGQ2 applied to the gate terminal GQ2 of the transistor Q2 is at a high level, and the transistor Q2 is conductive. A current path is established from the input voltage VIN back to the reference terminal via the inductor L1 and transistor Q2. An input voltage VIN is applied between the terminals of the inductor L1, and an inductor current IL1 having a predetermined positive time gradient flows in a direction from the input voltage VIN toward the connection point X (this direction is a positive direction). Inductor L1 stores electromagnetic energy corresponding to inductor current IL1.

  After a predetermined time elapses, the process proceeds to (2) in FIG. When the gate voltage VGQ2 applied to the gate terminal GQ2 transitions to a low level, the transistor Q2 becomes non-conductive. The voltage VQ2 at the connection point X at this time has a voltage value substantially equal to the reference voltage which is the voltage at the reference terminal because the transistor Q2 is conductive until just before. For this reason, the capacitor C1 is in a charged state and the capacitor C2 is in a discharged state. Since the inductor current IL1 flowing through the inductor L1 after the transistor Q2 is turned off is consumed for discharging the capacitor C1 and charging C2 ((2) in FIG. 3), the voltage value of the voltage VQ2 at the connection point X is The rise rises after the non-conduction of the transistor Q2. Therefore, switching of the transistor Q2 to the non-conducting state is performed in a state where a slight voltage is applied between the collector and the emitter. 0V switching is performed and the switching loss to the non-conducting state of the transistor Q2 can be reduced.

  In FIG. 2, (3) to (5) and FIG. 4 are the electromagnetic energy emission periods from the inductor L1. In the period (3) in FIG. 2, the gate voltage VGQ1 applied to the gate terminal GQ1 of the transistor Q1 becomes high level, and the transistor Q1 becomes conductive. The conducting transistor Q1 flows the inductor current IL1 from the inductor L1 toward the output terminal together with the antiparallel diode D1. As a result, electromagnetic energy is released to the output terminal, and the boosted output voltage VOUT is supplied to the load LD ((3) in FIG. 4). The connection point X is substantially equal to the output voltage VOUT, and the difference voltage between the output voltage VOUT and the input voltage VIN is between the terminals of the inductor L1 in the direction from the connection point X to the input voltage VIN (this direction is a negative direction). The inductor current IL1 having a predetermined negative time slope flows through the inductor L1. Note that when the gate voltage VGQ1 transitions to a high level and the transistor Q1 transitions to a conducting state, the capacitor C1 is in a discharging state, so that the switching of the transistor Q1 to the conducting state is slightly between the collector and the emitter. This is performed in a state where a voltage is applied. 0V switching is performed, and the switching loss to the conduction state of the transistor Q1 can be reduced.

  From this state, a high level gate voltage VGQ4 is applied to the gate terminal GQ4 of the transistor Q4 ((4) in FIG. 2). An auxiliary current path is formed from the connection point X to the reference terminal via the inductor L2, the one winding of the transformer T1, and the transistor Q4, and the inductor current IL2 begins to flow ((4) in FIG. 4). At the same time, electromagnetic energy is supplied to the output terminal from the other winding from the intermediate tap of the transformer T1 to the output terminal via the diode D4. In the initial stage of forming the auxiliary current path, the electromagnetic energy of the inductor L1 supplied to the output terminal via the transistor Q1 and the antiparallel diode D1 is partly divided into the inductor current IL2. Then, it is supplied to the output terminal via the transformer T1. An increase in loss associated with the formation of the auxiliary current path can be suppressed.

  Here, in the transformer T1, if the winding ratio of one winding to the other winding is 1: 2, a voltage approximately half of the output voltage VOUT is induced between the one winding and the inductor L2. It will be.

  2 and 4, during the period of (4) and (5), the difference voltage between the output voltage VOUT and the input voltage VIN is applied in the negative direction of the inductor L1, and the inductor current IL1 is stored as electromagnetic energy. From the current value according to, it decreases with a predetermined negative slope. On the other hand, a difference voltage that is approximately half of the output voltage VOUT is applied to the inductor L2 having an inductance value smaller than the inductance value of the inductor L1 in a direction that increases the inductor current IL2. After the elapse, the inductor current IL2 increases beyond the inductor current IL1 ((5) in FIGS. 2 and 4). Of the inductor current IL2, the current increased beyond the inductor current IL1 is supplied from the output terminal via the transistor Q1 in the conductive state.

  From this state, the gate voltage VGQ1 is changed to a low level, and the transistor Q1 is changed to a non-conductive state ((6) in FIGS. 2 and 5). This and (7) in FIG. 2 and FIG. 5 are periods in which the electromagnetic energy from the inductor L1 is shifted to the re-accumulation.

  In FIG. 2 (6), when the gate voltage VGQ1 applied to the gate terminal GQ1 transitions to a low level, the transistor Q1 becomes non-conductive. The voltage VQ2 at the connection point X at this time has a voltage value substantially equal to the output voltage VOUT because the transistor Q1 is conductive until just before. Capacitor C1 is in a discharged state, and capacitor C2 is in a charged state. After the transistor Q1 is turned off, the current exceeding the inductor current IL1 in the inductor current IL2 is consumed for charging the capacitor C1 and discharging the capacitor C2 ((6) in FIG. 5), so that the voltage VQ2 at the connection point X The voltage value falls after the non-conduction of the transistor Q1. Therefore, switching of the transistor Q1 to the non-conducting state is performed in a state where a slight voltage is applied between the collector and the emitter. 0V switching is performed and the switching loss to the non-conducting state of the transistor Q1 can be reduced.

  After the capacitor C1 is charged and the capacitor C2 is discharged and the voltage VQ2 at the node X drops to a voltage value near the reference voltage, the supply of the inductor current IL2 is continued through the antiparallel diode D2 of the transistor Q2. The Therefore, the voltage VQ2 at the connection point X is maintained at the reference voltage or a voltage that is lowered from the reference voltage by the forward voltage of the diode, and a voltage is applied to the inductor L2 in a direction that decreases the inductor current IL2, and the inductor current IL2 Has a negative time slope ((7) in FIGS. 2 and 5).

  Note that the voltage applied to the inductor L2 is fixed at substantially half of the output voltage VOUT at the connection point with the transformer T1, and the voltage VQ2 at the connection point X is equal to the output voltage VOUT during the electromagnetic energy release period. The voltage value is substantially equal, and the voltage value is substantially equal to the reference voltage during the electromagnetic energy accumulation period. That is, in the energy release period and the energy storage period, the applied voltage value is applied while the direction of application is reversed while the magnitude of the applied voltage value is maintained at approximately half the output voltage VOUT. The inductor current IL2 increased at a predetermined time slope during the energy release period decreases at a substantially equal time slope during the energy storage period.

  The timing when the transistor Q2 is turned on again after the transistor Q1 is turned off is shown in FIG. 2 and (8) in FIG. As described above, when the transistor Q1 becomes non-conductive and the capacitors C1 and C2 are completely charged and discharged, the inductor current IL2 starts to decrease. In a period in which the inductor current IL2 is larger than the inductor current IL1, the inductor current IL2 is replenished via the antiparallel diode D2, and the current path of the inductor current IL1 after the inductor current IL2 falls below the inductor current IL1. It is necessary to ensure. Therefore, the conduction timing of the transistor Q2 is set in a period in which the inductor current IL2 is larger than the inductor current IL1. At this time, since the antiparallel diode D2 is conducting, the transistor Q2 is switched to the conducting state while a slight voltage is applied between the collector and the emitter. 0V switching is performed, and the switching loss to the conduction state of the transistor Q2 can be reduced. Then, the gate terminal VGQ4 applied to the gate terminal GQ4 of the transistor Q4 becomes low level, and the transistor Q4 becomes non-conductive.

  Thereafter, returning to (1) in FIG. 2 and FIG. 3, the above operation is repeated to perform the boosting operation.

Next, the step-down operation in the bidirectional converter of the first embodiment will be described with reference to FIGS. 7 and 8 to 11. FIG. 7 shows a timing chart, and FIGS. 8 to 11 show circuit operation states in each operation. In the following description, the timing chart (FIG. 7) of the step-down operation will be described with reference to the operation state on the circuit (FIGS. 8 to 11) as appropriate. In FIG. 7, VGQ1, VGQ2, and VGQ3 are voltages applied to the base terminals GQ1, GQ2, and GQ3 of the transistors Q1, Q2, and Q3. IL1 and IL2 are currents flowing through the inductors L1 and L2 having a positive direction from the input voltage VIN to the connection point X and the current from the connection point X to the transformer T1, ID3 is a current flowing through the diode D3, and ID1 and ID2 are , Currents flowing in the anti-parallel diodes D1 and D2, IQ1 and IQ2 indicate currents flowing in the transistors Q1 and Q2. VQ2 indicates a voltage at the connection point X. VL2 is a voltage applied between the terminals of the inductor L2.

  In FIG. 7, (1) and (2), and FIG. 8 are accumulation periods of electromagnetic energy in the inductor L1. In the period (1) in FIG. 7, electromagnetic energy is accumulated in the inductor L1. FIG. 8 (1) shows the operating state during this period. The gate voltage VGQ1 applied to the gate terminal GQ1 of the transistor Q1 is at a high level, and the transistor Q1 is conductive. A current path is established from the output voltage VOUT regenerated from the load LD connected to the output terminal to the input terminal via the transistor Q1 and the inductor L1. A differential voltage between the output voltage VOUT and the input voltage VIN is applied between the terminals of the inductor L1, and a predetermined positive time slope is formed in a direction from the connection point X toward the input voltage VIN (this direction is a negative direction). An inductor current IL1 having Inductor L1 stores electromagnetic energy corresponding to inductor current IL1.

  After a lapse of a predetermined time, the process proceeds to (2) in FIG. When the gate voltage VGQ1 applied to the gate terminal GQ1 transitions to a low level, the transistor Q1 becomes non-conductive. The voltage VQ2 at the connection point X at this time has a voltage value substantially equal to the output voltage VOUT because the transistor Q1 is conductive until just before. Therefore, the capacitor C1 is in a discharged state and the capacitor C2 is in a charged state. Since the inductor current IL1 flowing through the inductor L1 after the transistor Q1 is turned off is consumed for charging the capacitor C1 and discharging C2 ((2) in FIG. 8), the voltage value of the voltage VQ2 at the connection point X is The drop is delayed from the non-conduction of the transistor Q1. Therefore, switching of the transistor Q1 to the non-conducting state is performed in a state where a slight voltage is applied between the collector and the emitter. 0V switching is performed and the switching loss to the non-conducting state of the transistor Q1 can be reduced.

  In FIG. 7, (3) to (5) and FIG. 9 are electromagnetic energy emission periods from the inductor L1. In the period (3) in FIG. 7, the gate voltage VGQ2 applied to the gate terminal GQ2 of the transistor Q2 becomes high level, and the transistor Q2 becomes conductive. The conducting transistor Q2 flows the inductor current IL1 from the inductor L1 to the input terminal together with the antiparallel diode D2. As a result, electromagnetic energy is released to the input terminal, and the input voltage VIN that is stepped down from the output voltage VOUT is supplied ((3) in FIG. 9). The connection point X is substantially equal to the reference voltage, and the difference voltage between the input voltage VIN and the reference voltage is between the terminals of the inductor L1 from the input terminal toward the connection point X (this direction is defined as a positive direction). And an inductor current IL1 having a predetermined negative time slope flows through the inductor L1. Note that when the gate voltage VGQ2 transitions to a high level and the transistor Q2 transitions to a conductive state, the capacitor C2 is in a discharged state. Therefore, the switching of the transistor Q2 to the conductive state is slightly between the collector and the emitter. This is performed in a state where a voltage is applied. 0V switching is performed, and the switching loss to the conduction state of the transistor Q2 can be reduced.

  From this state, a high level gate voltage VGQ3 is applied to the gate terminal GQ3 of the transistor Q3 ((4) in FIG. 7). An auxiliary current path is formed from the output terminal to the transistor Q3, the one winding of the transformer T1, the inductor L2, and the connection point X, and the inductor current IL2 begins to flow ((4) in FIG. 9). At the same time, electromagnetic energy is supplied to the output terminal from the other winding from the intermediate tap of the transformer T1 to the output terminal via the diode D3. A part of the inductor L1 that has been supplied to the input terminal via the transistor Q2 and the anti-parallel diode D2 before the formation of the auxiliary current path is supplied by the inductor current IL2 due to the formation of the auxiliary current path. . The electromagnetic energy accumulated in the inductor L1 is returned to the output terminal via the transformer T1 by the inductor current IL2. An increase in loss due to the formation of the auxiliary current path can be suppressed.

  Here, in the transformer T1, if the winding ratio of one winding to the other winding is 1: 2, a voltage approximately half of the output voltage VOUT is induced between the one winding and the inductor L2. It will be.

  7 and 9, during the period of (4) and (5), the difference voltage between the input voltage VIN and the reference voltage is applied in the positive direction of the inductor L1, and the inductor current IL1 is stored in the accumulated electromagnetic energy. The current value decreases from the corresponding current value with a predetermined positive slope. On the other hand, since a difference voltage approximately half of the output voltage VOUT is applied to the inductor L2 having an inductance value smaller than the inductance value of the inductor L1 so as to increase the inductor current IL2 in the negative direction, After the elapse of the predetermined time, the inductor current IL2 increases beyond the inductor current IL1 ((5) in FIGS. 7 and 9). Of the inductor current IL2, the current increased beyond the inductor current IL1 flows to the reference terminal via the transistor Q2 in the conductive state.

  From this state, the gate voltage VGQ2 is changed to a low level, and the transistor Q2 is changed to a non-conductive state ((6) in FIGS. 7 and 10). This and (7) in FIG. 7 and FIG. 10 is a period in which the electromagnetic energy is released from the inductor L1 and re-accumulated.

  In FIG. 7 (6), the gate voltage VGQ2 applied to the gate terminal GQ2 transitions to a low level, whereby the transistor Q2 becomes non-conductive. The voltage VQ2 at the connection point X at this time has a voltage value substantially equal to the reference voltage because the transistor Q2 is conductive until just before. Capacitor C2 is in a discharged state, and capacitor C1 is in a charged state. Since the current exceeding the inductor current IL1 among the inductor current IL2 after the transistor Q2 is turned off is consumed for charging the capacitor C2 and discharging the capacitor C1 ((6) in FIG. 10), the voltage VQ2 at the connection point X The voltage value rises after the non-conduction of the transistor Q2. Therefore, switching of the transistor Q2 to the non-conducting state is performed in a state where a slight voltage is applied between the collector and the emitter. 0V switching is performed and the switching loss to the non-conducting state of the transistor Q2 can be reduced.

  After the capacitor C2 is charged and the capacitor C1 is discharged and the voltage VQ2 at the node X rises to a voltage value near the output voltage VOUT, the inductor current IL2 is continued through the antiparallel diode D1 of the transistor Q1. . Therefore, the voltage VQ2 at the connection point X is maintained at the output voltage VOUT, and a voltage is applied to the inductor L2 in a direction to decrease the inductor current IL2, and the inductor current IL2 has a positive time slope (see FIGS. 7 and 7). 10 (7)).

  Note that the voltage applied to the inductor L2 is fixed at approximately half of the output voltage VOUT at the connection point with the transformer T1, and the voltage VQ2 at the connection point X is approximately equal to the reference voltage during the electromagnetic energy release period. The voltage values are equal, and the voltage value is substantially equal to the output voltage VOUT during the electromagnetic energy accumulation period. That is, in the energy release period and the energy storage period, the applied voltage value is applied while the direction of application is reversed while the magnitude of the applied voltage value is maintained at approximately half the output voltage VOUT. The inductor current IL2 that has increased in the negative direction with a predetermined time slope during the energy release period decreases with a substantially equal time slope during the energy storage period.

  The timing at which the transistor Q1 is turned on again after the transistor Q2 is turned off is shown in (8) of FIGS. As described above, when the transistor Q2 is turned off and charging and discharging of the capacitors C2 and C1 are completed, the inductor current IL2 starts to decrease. When the inductor current IL2 is larger than the inductor current IL1, the inductor current IL2 is maintained via the antiparallel diode D1, and the current path of the inductor current IL1 after the inductor current IL2 falls below the inductor current IL1. It is necessary to ensure. Therefore, the conduction timing of the transistor Q1 is set in a period in which the inductor current IL2 is larger than the inductor current IL1. At this time, since the antiparallel diode D1 is conducting, the transistor Q1 is switched to the conducting state while a slight voltage is applied between the collector and the emitter. 0V switching is performed, and the switching loss to the conduction state of the transistor Q1 can be reduced. Then, the gate terminal VGQ3 applied to the gate terminal GQ3 of the transistor Q3 becomes low level, and the transistor Q3 becomes non-conductive.

  Thereafter, returning to (1) in FIG. 7 and FIG. 8, the above operation is repeated to perform the step-down operation.

FIG. 12 shows a circuit diagram of the bidirectional DC-DC converter of the second embodiment. It is an embodiment about a bidirectional converter similar to the first embodiment. Instead of the auxiliary circuit unit 1 in the first embodiment, an auxiliary circuit unit 1A is provided.

  In the auxiliary circuit unit 1A, the other winding of the transformer T1 is connected to the input terminal via diodes D3 and D4. In this configuration, the energy accompanying the inductor current IL2 is returned to the input voltage VIN when the auxiliary current path is formed. Since other circuit configurations, operations and effects are the same as those in the first embodiment, description thereof is omitted here.

As described above in detail, according to the bidirectional DC-DC converter according to the present embodiment, the transistors Q1 and Q2 that perform the switching operation when the electromagnetic energy is stored and discharged from the inductor L1 are connected between the collector and the emitter at the time of switching. The switching loss can be reduced by performing the 0V switching operation with a small voltage difference of.

With the reduction of switching loss, it is possible to improve the power efficiency in voltage conversion of the bidirectional DC-DC converter. Heat generation due to switching loss in the transistors Q1 and Q2 can be reduced, and a cooling device such as a heat sink can be reduced in size and weight.

The switching operation in the bidirectional DC-DC converter can be performed at a high frequency, and the switching operation can be performed at a frequency higher than the audible frequency band. Vibrations of the inductor L1 and the like accompanying electromagnetic energy during operation can be shifted from the audible frequency band, and noise during operation can be prevented.

  The auxiliary current path is provided with one winding of a transformer T1. The transformer T1 normally includes a leakage inductance connected in series with one winding as an equivalent circuit, and is considered to have an inductance value smaller than the inductance value of the inductor L1. This inductance value is combined with the voltage between the terminals induced by the voltage value of one winding and the voltage value of the connection point X induced according to the winding ratio of the transformer T1, and the time of the inductor current IL1 as the input current A current having a time gradient steeper than the gradient flows in the auxiliary current path. The timing at which the magnitude relation of the current value is reversed and the time width of the reversed state can be adjusted by the time gradient adjusted according to the leakage inductance value and the winding ratio of the transformer T1, and the conduction timing of the transistor Q1 or Q2 can be adjusted. It can be set stably within a sufficient time margin.

  Here, by increasing the winding ratio of the transformer T1 to substantially half of the output voltage VOUT, the leakage inductance component of the transformer T1, the inductor L2, or the inductor current IL2 flowing through these combined inductors is increased or decreased. Can be made substantially equal to each other.

Further, the inductor current IL2 flowing in the auxiliary current path can be sent from the other winding to the output terminal via the transformer T1, or returned to the input terminal. It is possible to improve the power efficiency in the voltage conversion of the bidirectional DC-DC converter.

The present invention is not limited to the above-described embodiment, and it goes without saying that various improvements and modifications can be made without departing from the spirit of the present invention.
For example, in the present embodiment, the case where the present invention is used for a bidirectional converter has been described, but the present invention is not limited to this, and can be similarly applied to a boost converter or a step-down converter. Needless to say.
In addition, although the inductor L2 having a predetermined inductance value is used and the winding ratio of one winding to the other winding of the transformer T1 is 1: 2, the present invention is not limited to this. By adjusting these values, the time gradient of the inductor current IL2 can be adjusted, and the conduction timing of the transistor Q2 or Q1 can be adjusted.
In this embodiment, the case where the capacitors C1 and C2 are connected in parallel between the current path terminals of the transistors Q1 and Q2 has been described as an example. However, the voltage at the connection point X determined by charging and discharging of the capacitors is described. In the case where the capacitance value of the capacitor can be secured so that the change gradually changes to the extent that 0V switching of the transistors Q1 and Q2 is possible, one of the capacitors C1 and C2 may be provided.
In the present embodiment, with respect to the other winding of the transformer T1, the intermediate tap is connected to the reference terminal, both side terminals are connected to the anodes of the diodes D3 and D4, and the cathodes of the diodes D3 and D4 are connected to the output terminal. Although the case has been described as an example, it is needless to say that connection in the reverse direction is also possible. That is, even when the intermediate tap is connected to the output terminal, both side terminals are connected to the cathodes of the diodes D3 and D4, and the anodes of the diodes D3 and D4 are connected to the reference terminal, the electromagnetic energy of the inductor L1 is supplied to the output terminal. can do. It is also conceivable to return electromagnetic energy to the input terminal by connecting to the input terminal instead of the output terminal.
In the present embodiment, the case where the input voltage VIN is connected to the primary power supply terminal, the load is connected to the secondary power supply terminal, and energy when braking the load such as a motor is regenerated to the input voltage VIN has been described. The present invention is not limited to this. Needless to say, the present invention can also be applied to the case where a voltage source such as a battery having a higher voltage than the input voltage VIN is connected to the secondary power supply terminal.

1 is a circuit diagram of a bidirectional DC-DC converter of a first embodiment. It is a timing chart which shows the pressure | voltage rise operation | movement in the bidirectional | two - way DC-DC converter of 1st Embodiment. It is a figure which shows the accumulation | storage period of the electromagnetic energy to an inductor among pressure | voltage rise operations. It is a figure which shows the discharge | release period of the electromagnetic energy from an inductor among pressure | voltage rise operations. It is a figure which shows the period which changes from discharge | release of electromagnetic energy to accumulation | storage among boosting operations. It is a figure which shows the conduction | electrical_connection timing of the transistor at the time of changing to the accumulation state of electromagnetic energy among pressure | voltage rise operations. 3 is a timing chart illustrating a step-down operation in the bidirectional DC-DC converter of the first embodiment. It is a figure which shows the accumulation | storage period of the electromagnetic energy to an inductor among step-down operations. It is a figure which shows the discharge | release period of the electromagnetic energy from an inductor among step-down operations. It is a figure which shows the period which changes from discharge | release of electromagnetic energy to accumulation | storage in step-down operation. It is a figure which shows the conduction | electrical_connection timing of the transistor at the time of changing to the accumulation | storage state of electromagnetic energy among step-down operations. It is a circuit diagram of the bidirectional DC-DC converter of 2nd Embodiment. It is a circuit diagram of a motor drive device provided with a chopper circuit in the prior art.

Explanation of symbols

1, 1A Auxiliary circuit C1, C2 Capacitors D1, D2 Anti-parallel diodes IL1, IL2 Inductor current L1, L2 Inductor LD Load Q1-Q4 Transistor T1 Transformer VIN Input voltage

Claims (4)

  A first inductor having one terminal connected to the primary power supply terminal; and an upper switching element and a lower switching element connected in series between the secondary power supply terminal and the reference terminal, wherein the upper and lower switching elements are connected. A bidirectional DC-DC converter in which the other terminal of the first inductor is connected to a point;
  A capacitor connected in parallel between the current path terminals of at least one of the upper and lower switching elements;
  When the electromagnetic energy is accumulated in the first inductor due to the conduction of the upper switching element, the first inductor is reduced in order to minimize the applied voltage between the current path terminals of the upper switching element before the conduction of the upper switching element. A first auxiliary switching element that forms a first auxiliary current path for flowing a current larger than a current flowing through the connection point;
  When the electromagnetic energy is stored in the first inductor due to the conduction of the lower switching element, the first inductor is reduced in order to reduce the applied voltage between the current path terminals of the lower switching element before the conduction of the lower switching element. A second auxiliary switching element forming a second auxiliary current path for flowing a current larger than a current flowing through the connection point,
  The first and second auxiliary current paths include one winding of a transformer,
  The other winding of the transformer includes an intermediate tap, one of the intermediate tap and both side terminals is connected to a reference terminal, the other is connected to the secondary power supply terminal or the primary power supply terminal,
  The bidirectional DC-DC converter, wherein the both-side terminals include first and second diodes connected with a path from the reference terminal toward the secondary power supply terminal or the primary power supply terminal as a forward direction. The bidirectional DC-DC converter according to claim 1, further comprising a second inductor between the connection point and one winding of the transformer. When the first or second auxiliary current path is formed, with respect to the voltage applied to the other winding, the connection is caused by the voltage induced in the one winding according to the winding ratio of the transformer. The voltage applied to the leakage inductance component of the transformer, the second inductor, or the combined inductor of the leakage inductance component and the second inductor, which exists between the point and the one winding, is the secondary power source. 3. The bidirectional DC-DC converter according to claim 1, wherein the bidirectional DC-DC converter is approximately half of a voltage at a terminal. Between the current path terminals of the upper and lower switching elements, an anti-parallel diode connected with the direction opposite to the current direction of the current flowing when electromagnetic energy is stored in the first inductor as a forward direction is provided. The bidirectional DC-DC converter according to claim 1.

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