JP4053968B2 - Synchronous motor driving device, refrigerator and air conditioner - Google Patents

Description

  The present invention relates to a synchronous motor driving device that drives a synchronous motor, a control method for the synchronous motor driving device, a refrigerator-freezer, and an air conditioner.

  In recent years, social interest in energy conservation has increased as part of global environmental problems. Under such circumstances, in the electric motor field, a permanent magnet synchronous motor (PMSM: Permanent Magnet Synchronous Motor) that is more efficient than a conventional induction motor is used, and surface motors are also used in PM motors in terms of efficiency. An embedded permanent magnet synchronous motor (IPMSM) is being used from a permanent permanent magnet synchronous motor (SPMSM). Further, in the field of electric motor driving, a sine wave energization method that is more efficient than a conventional rectangular wave energization method is being used.

  A conventional synchronous motor drive device includes a plurality of energization drive means for obtaining output voltage data, selects one of the plurality of drive means according to the efficiency of the motor, and outputs PWM ( (Pulse Width Modulation) signal is generated (see, for example, Patent Document 1).

Further, another conventional synchronous motor driving apparatus has a sine wave approximate PWM method and a uniform width PWM method, and drives both of these methods in combination (for example, see Patent Document 2).
JP 2001-45789 A (first page, FIG. 1) JP 2002-191185 A (first page, FIG. 1)

  In the synchronous motor drive device in Patent Document 1, in order to realize a plurality of energization widths, the same number of voltage data calculation means as the energization widths are required. For this reason, there is a problem that the configuration of software is complicated, and a high-performance microprocessor is required because of the large amount of resources such as the memory and calculation capability of the microprocessor used for calculation, leading to an increase in cost. When the arithmetic means is realized by hardware, since there are a plurality of arithmetic methods, there is a problem that the hardware configuration becomes complicated, leading to a decrease in reliability and an increase in cost.

  Moreover, although 120 degree energization drive and 180 degree energization drive are disclosed as an example of a plurality of energization widths, in this case, voltage calculation means for 120 degree energization drive and voltage calculation means for 180 degree energization drive are required. Become. For this reason, there is a problem that the configuration of software is complicated, and a high-performance microprocessor is required because of the large amount of resources such as the memory and calculation capability of the microprocessor used for calculation, leading to an increase in cost.

  In terms of hardware, as a control method at 120 energization driving, a method of detecting an induced voltage generated in a non-energized section, detecting a rotor position of an electric motor, and applying a voltage according to rotor position information is used. Yes. Further, during 180-degree energization driving, a method is used in which the current phase is detected and the voltage is controlled so that the current phase becomes a predetermined value. For this reason, an induced voltage detection circuit for detecting an induced voltage generated in a non-energized section is necessary during 120-degree energization driving, and a current detection circuit for detecting a current phase is necessary during 180-degree energization. . For this reason, there is a problem that the hardware configuration becomes complicated, leading to a decrease in reliability and an increase in cost.

  Further, in this control method during 120 energization drive, the non-energization section decreases as the energization width approaches 180 degrees, and therefore rotor position detection using induced voltage due to the influence of current circulation and the like that occurs during non-energization. It becomes impossible and control cannot be continued. For this reason, it is impossible to drive the synchronous motor using an arbitrary energization width in the range of 120 to 180 degrees.

  In addition, since the energization drive method is simply switched when the energization method is switched, the magnitude and phase of the output voltage of the inverter device changes suddenly when the energization drive method is switched, and the control becomes unstable. There is. In some cases, an overcurrent flows at the time of switching, causing the inverter device to stop, and there is a problem that it is difficult to switch the energization driving method in a state where the stability of control is maintained.

  In the synchronous motor driving device disclosed in Patent Document 2, in order to realize the sine wave approximate PWM method and the equal width PWM method, the sine wave energization method requires hardware for current detection, In the equal width PWM method, hardware for detecting the induced voltage is required.

  In addition, when performing drive control of a synchronous motor, the sine wave approximate PWM method requires a voltage control method based on the current detection value, and the equi-width PWM method requires a voltage control method based on the induced voltage detection value. Each is required. For this reason, both hardware and software require at least two types of configurations corresponding to the respective systems, which complicates the configuration and causes a problem of reducing reliability and increasing costs.

  Further, a method of detecting an induced voltage generated in a non-energized section, detecting a rotor position of an electric motor, and applying a voltage according to rotor position information is used as a control method of the equal width PWM method. However, in this method, as the energization width approaches 180 degrees, the non-energization section decreases, and therefore the rotor position cannot be detected using the induced voltage due to the influence of current recirculation that occurs during non-energization. It cannot be sustained. For this reason, it is impossible to drive the synchronous motor using an arbitrary energization width in the range of 120 to 180 degrees.

  In addition, since the two PWM systems have different hardware and control methods at the time of operation, it is difficult to easily switch between the respective systems, and there is a problem that it is not possible to instantaneously shift to the required PWM system. . When switching instantaneously, voltage fluctuation occurs, control becomes unstable, and control may fail in the worst case.

  The present invention has been made to solve the above-described problems. In one circuit configuration, rectangular wave energization capable of selecting an arbitrary energization width between 120 degrees and 180 degrees, sinusoidal energization, and the like. A synchronous motor drive device and a synchronous motor drive capable of switching energized waveforms in an extremely short time according to the drive conditions (high efficiency drive condition, low noise drive condition) required for the synchronous motor to be driven It aims at providing the control method of an apparatus, a refrigerator-freezer, and an air conditioner.

  The synchronous motor drive device according to the present invention is an synchronous motor drive device that drives a synchronous motor by an inverter device, an output voltage vector calculation means for obtaining an output voltage vector output from the inverter device according to a rotation speed command value, and an output A rectangular wave energizing means for outputting a voltage with an arbitrary energization width between 120 degrees and 180 degrees in a rectangular wave shape according to the voltage vector, and a sine wave energizing means for outputting a voltage with a sinusoidal energization according to the output voltage vector And an energization waveform selection means for selecting an energization method and an energization width according to conditions required for driving the synchronous motor.

  The synchronous motor drive device according to the present invention comprises an output voltage vector calculation means for obtaining an output voltage vector output from the inverter device in accordance with a rotational speed command value, and an output voltage in the synchronous motor drive device that drives the synchronous motor by the inverter device. A rectangular wave energizing means for outputting a voltage with an arbitrary energization width between 120 degrees and 180 degrees in a rectangular wave shape according to a vector; a sine wave energizing means for outputting a voltage with a sinusoidal energization according to an output voltage vector; Since it has a configuration including an energization waveform selection unit that selects an energization method and an energization width according to the conditions required for driving the synchronous motor, any energization between 120 degrees and 180 degrees can be achieved with one circuit configuration. A rectangular wave with selectable width and sine wave energization can be realized, and the drive conditions (high efficiency drive condition, low noise drive condition) required for the synchronous motor to be driven can be met. Te, thereby obtaining a synchronous motor drive apparatus capable of conduction waveform switching in a short time. Further, this makes it possible to set the total efficiency when the synchronous motor is driven to a predetermined value or more, or to reduce the noise when the synchronous motor is driven to a predetermined value or less.

Embodiment 1 FIG.
FIG. 1 is a diagram showing the first embodiment and shows a synchronous motor driving device. In the figure, a DC power supply unit 1 is connected to an inverter device 2 and supplies DC power to the inverter device 2. The synchronous motor 6 is connected to the inverter device 2, and electric power for driving the synchronous motor 6 is supplied from the inverter device 2. Here, the DC power source unit 1 is a unit obtained by converting an AC power source into DC using a rectifier circuit and smoothing, a unit using a booster circuit or a step-down circuit, a battery, or the like.

  The inverter device 2 includes a U-phase upper switching element 3a, a V-phase upper switching element 3b, a W-phase upper switching element 3c, a U-phase lower switching element 3d, a V-phase lower switching element 3e, and a W-phase lower switching element 3f. An inverter main circuit 5 including a plurality of switching elements 3 and a free-wheeling diode 4 connected in parallel with the plurality of switching elements 3, a current detection means 7a for detecting a current for one phase flowing into the synchronous motor 6, and a current detection means Current detection means 7 comprising current detection means 7b for detecting a current of a phase different from 7a, and inverter control for generating PWM signals for on / off control of the plurality of switching elements 3 based on the current detected by the current detection means 7. A plurality of switching elements based on the PWM signal obtained from the means 8 and the inverter control means 8 A gate drive circuit 9 for turning on / off driving, and a DC voltage detection unit 10.

The inverter control means 8 includes a phase current calculation means 11 that converts the current signal obtained from the current detection means 7 into a three-phase current flowing into the synchronous motor 6, and a current signal obtained from the phase current calculation means 11 and an external signal. Output voltage vector calculation means 12 for obtaining an output voltage vector command value based on a given rotation speed command value, and output based on a DC voltage signal obtained from the DC voltage detection means 10 and an output voltage vector obtained from the output voltage vector calculation means 12 The modulation rate / phase calculation means 13 for obtaining the modulation rate / phase angle of the voltage vector, the current signal obtained from the phase current calculation means 11, the output voltage vector calculation means 12, and the rotation speed command value given from the outside The energization waveform selection means 14 for selecting (energization method and energization width) and the output voltage level obtained from the modulation factor / phase calculation means 13. A rectangular wave PWM generating means 15 for obtaining a PWM signal for energizing a rectangular wave based on a toll modulation rate and an energization width command value obtained from the energizing waveform selecting means 14; and an output voltage vector modulation rate obtained from the modulation rate / phase calculating means 13. And a sine wave PWM generation means 16 for obtaining a PWM signal for sine wave energization based on the energization width command value obtained from the energization waveform selection means 14, and a rectangular wave PWM generation based on the energization method signal obtained from the energization waveform selection means 14. Means 15 and PWM output means 17 for selectively outputting the PWM of the sine wave PWM generation means 16 are provided.
The energization waveform selection means corresponds to the energization waveform selection means 14.
The rectangular wave energizing means corresponds to the rectangular wave PWM generating means 15.
The sine wave energizing means corresponds to the sine wave PWM generating means 16.

The operation of the synchronous motor driving apparatus configured as described above will be described with reference to FIG. In the figure, the inverter device 2 detects currents for two phases among the phase currents flowing into the synchronous motor 6 by means of current detection means 7a, 7b. The inverter control means 8 uses the current for two phases detected by the current detection means 7a and 7b, for example, the U-phase current Iu and the V-phase current Iv, and the DC voltage signal Vdc obtained from the DC voltage detection means 10, A PWM signal is output to satisfy the rotational speed command value ω ^* given from the outside as the rotational speed of the synchronous motor 6. Based on the PWM signal obtained by the inverter control means 8, the gate drive circuit 9 drives on / off a plurality of switching elements 3 provided in the inverter main circuit 5. By the on / off operation of the plurality of switching elements 3, the DC power supplied from the DC power supply unit 1 is converted into three-phase AC power, which is supplied from the inverter device 2 to the synchronous motor 6 to drive the synchronous motor 6. Here, the rotational speed command value ω ^* is not shown in a host microprocessor (not shown) or other interface (man machine interface etc.) for controlling the entire system of a system in which the inverter device 2 is installed, for example, the inverter device. Z).

The operation of the inverter control means 8 will be described.
In the inverter control means 8, the three-phase currents Iu, Iv flowing into the synchronous motor 6 in the phase current calculation means 11 using the current signals Iu, Iv for the two phases detected by the current detection means 7a, 7b. Iw is required. The three-phase currents Iu, Iv, Iw are given to the output voltage vector calculation means 12, and the output voltage vector calculation means 12 rotates the rotation speed of the synchronous motor 6 from the three-phase currents Iu, Iv, Iw and the rotation speed command value ω ^*. determined by the output voltage computing a vector command value Vo ^* for driving so as to satisfy the speed command value omega ^*.

The modulation factor / phase calculation means 13 is a ratio (modulation factor) of the output voltage vector to the DC voltage based on the DC voltage Vdc obtained from the DC voltage detection means 10 and the output voltage vector command value Vo ^* obtained from the output voltage vector calculation means 12. Vk ^* and the phase angle θv of the output voltage vector are obtained. The phase relationship of the output voltage vector on the three-phase coordinates is as shown in FIG. The modulation factor Vk ^* is obtained by the following equation where the output voltage vector command value is Vo ^* and the DC voltage is Vdc.
Vk ^* = √2 × | Vo ^* | / Vdc (1)

The energization waveform selection means 14 includes three-phase currents Iu, Iv, Iw obtained from the phase current calculation means 11, output voltage vector modulation rate Vk ^* obtained from the modulation factor / phase calculation means 13, rotation speed command value ω ^* , external Based on the synchronous motor drive condition Jk ^* given by (1), the most suitable energization method Md and energization width θon for realizing the synchronous motor drive condition Jk ^* are obtained and output. Here, the synchronous motor drive condition Jk ^* is given from the host device of the inverter device 2 in the same manner as the rotational speed command value ω ^* . The condition required for driving the synchronous motor is, for example, a high efficiency drive request or a low noise drive request.

The rectangular wave PWM generation means 15 sets the energization width to θon from the output voltage vector modulation rate Vk ^* obtained from the modulation factor / phase calculation means 13, the output voltage vector phase angle θv, and the energization width θon obtained from the energization waveform selection means 14. A rectangular wave energizing PWM signal is generated.

The sine wave PWM generating means 16 generates a PWM signal for energizing a sine wave from the output voltage vector modulation rate Vk ^* obtained from the modulation rate / phase calculation means 13 and the output voltage vector phase angle θv.

  The PWM output means 17 selects and outputs one of the rectangular wave PWM generation means 15 and the sine wave PWM generation means 16 in accordance with the energization method signal Md output by the energization waveform selection means 14.

  The inverter control means 8 described above is constituted by a microprocessor, and the constituent elements in the inverter control means 8 are executed by processing by software in the microprocessor. Here, the inverter control means 8 may be configured entirely or partially by hardware.

The PWM signal generation method of the sine wave PWM generation means 16 will be described below. FIG. 2 is a vector diagram showing the relationship of the UVW phase on the three-phase coordinates of the output voltage vector. The output voltage vector command value Vo ^* obtained from the output voltage vector calculation means 12 is as shown in the figure. At this time, the phase based on the U phase is θv, θv is divided every 60 degrees, and a stage is provided as shown in the figure. As an example of the sine wave PWM generation method, a case where space vector modulation is used will be described. In the space vector modulation, a plurality of switching elements 3 are driven so as to appropriately switch the output voltage vector for each carrier period so that the output voltage vector draws a circular (broken line) locus at the rotational speed frequency ω ^* . PWM signal is generated.

The ratio (hereinafter referred to as on-duty) of turning on the upper switching element of each UVW phase within a predetermined time (carrier cycle) is obtained from the modulation factor Vk ^*, and the on-duty of each phase is obtained so that the output voltage vector draws a circular locus. FIG. 3 shows the voltage phase angle θv on the horizontal axis. In FIG. 3, the signal indicated by (a) indicates the on-duty signals Duty_U, Duty_V, and Duty_W of U, V, and W phases, and is a carrier that is a triangular wave signal having an amplitude of 1 at a frequency fc (hereinafter, carrier frequency). Signal Sc is shown. FIG. 3B shows the stage of the vector diagram shown in FIG. As shown in FIG. 3C, the PWM signal compares the on-duty signal Duty_U, Duty_V, and Duty_W for each phase with the carrier signal Sc, and when the on-duty signal becomes larger than the carrier signal, High is output. When the on-duty signal becomes smaller than the carrier signal, Low is output. The signal obtained by inverting this signal High and Low becomes the PWM signal of the lower switching element of each phase. Finally, the PWM signal of each phase obtained here is provided with a short-circuit prevention time Td for preventing a short circuit between the upper switching element and the lower switching element, and is output as a sine wave PWM signal.

  The PWM signal of each phase at this time is shown in FIG. In FIG. 3C, the PWM signal for the U-phase upper switching element 3a is Up, the PWM signal for the V-phase upper switching element 3b is Vp, the PWM signal for the W-phase upper switching element 3c is Wp, and the U-phase lower switching element 3d. The PWM signal for the V-phase lower switching element 3e is Vn, and the PWM signal for the W-phase lower switching element 3f is Wn. In this way, in the sine wave PWM, the upper and lower switching elements of each phase are repeatedly turned on / off at the carrier frequency, so that all the six switching elements 3 operate.

  Next, a PWM signal generation method of the rectangular wave PWM generation means 15 will be described below. In the rectangular wave PWM generation means 15, the rectangular wave PWM is generated based on the PWM generation method of the sine wave PWM generation means 16. The purpose of the rectangular wave PWM is to reduce the circuit loss of the inverter device 2 caused by switching by reducing the number of operation of the switching element compared to the sine wave PWM. Therefore, the PWM signal is generated as follows so that the locus of the output voltage vector draws a circular (broken line) locus while reducing the number of switching operation elements.

The number of output voltage vectors when the rectangular wave is energized is reduced so that the locus of the output voltage vector becomes substantially circular with reference to the generation of PWM when the sine wave is energized. When the energization width θon is 120 degrees, as shown in FIG. 4, the output voltage vectors are six of V ₀₁ , V ₁₁ , V ₂₁ , V ₃₁ , V ₄₁ , and V ₅₁ . When the energization width θon is greater than 120 degrees and less than 180 degrees, the output voltage vectors are V ₀₀ , V ₀₁ , V ₁₀ , V ₁₁ , V ₂₀ , V ₂₁ , There are twelve of V ₃₀ , V ₃₁ , V ₄₀ , V ₄₁ , V ₅₀ , and V ₅₁ . Further, when the energization width θon is 180 degrees, the voltage vectors output as shown in FIG. 6 are six V ₀₀ , V ₁₀ , V ₂₀ , V ₃₀ , V ₄₀ , and V ₅₀ . In sine wave energization, the number of generated voltage vectors during one rotation differs depending on the carrier frequency fc. When the rotational speed on the electrical angle at which the voltage vector makes one revolution is ωe, the number Nv of voltage vectors generated while the voltage vector makes one revolution can be obtained by the following equation.
Nv = 2π × fc / ωe (2)

  4 to 6, when a rectangular wave is energized, a circular locus is drawn using only a voltage vector having a specific phase among voltage vectors generated when a sine wave is energized.

  Here, the generation method of the voltage vector shown in FIGS. 4 to 6 will be described with reference to FIG. The waveform shown in FIG. 7A is the same as that shown in FIG. 3, and shows the voltage phase angle θv when the sine wave is energized and the ON duty signals Duty_U, Duty_V, and Duty_W of each phase of UVW. The rectangular wave energization is based on this signal, the stage where the on-duty signal crosses 0.5 for each phase is set as the non-energized section, and the other two consecutive stages are energized at 120 degrees, which is the basis for energizing the rectangular wave. Interval. When the energization width θon exceeds 120 degrees, the energization width is symmetrically expanded from the basic 120-degree section to the front and rear non-energization sections, and rectangular wave energization of 120 to 180 degrees is obtained. When energized 180 degrees, there is no non-energized section.

  An energization section exceeding 120 degrees, which is the base, is defined as an overlap section, and the energization width is defined as θov. The voltage waveform of each phase when the rectangular wave is energized with the overlap section θov is the waveform shown in FIG. Here, in each stage, an overlap section and a non-energized section are defined as stage modes as shown in FIG. In each stage, modes 0 and 2 are overlap sections, and mode 1 is a non-energized section. Here, there is no overlap section (mode 0 and mode 2) when energized at 120 degrees, and there is no non-energized section (mode 1) when energized at 180 degrees.

  In each stage, mode 0 and mode 2 are states in which a voltage is applied from any one phase of the UVW phase to the other two phases, or a voltage is applied from any two phases to the other one phase. Is a state in which a voltage is applied from one predetermined phase to another phase, and the other phase is in a non-energized section. In FIG. 7, the voltage vector generated at each stage-mode condition is as shown in FIG. The vector names described in (d) of FIG. 7 are the same as the vector names of FIGS. However, the direction of the voltage vector is the positive voltage direction.

For example, the voltage vector V ₀₀ generated in the stage 0-mode 0 is a state in which a voltage is applied from the U phase to the V and W phases. At this time, the positive side of the DC voltage is applied to the U phase and the DC is applied to the V and W phases. The negative side of the voltage is applied. The voltage vector V ₀₁ generated in the stage 0-mode 1 is a state in which a voltage is applied from the U phase to the W phase, the positive side of the DC voltage is applied to the U phase, and the negative side of the DC voltage is applied to the W phase side. To be. _{Similarly, V 10} is applied to the W-phase U, _{V-phase, V 11} is applied to the W-phase from the _{V-phase, V 20} is applied from V phase W, the _{U-phase, V 21} is U from the V-phase V ₃₀ is applied from the V and W phases to the U phase, V ₃₁ is applied from the W phase to the U phase, V ₄₀ is applied from the W phase to the U and V phases, and V ₄₁ is the W phase. To V phase, V ₅₀ is applied from W, U phase to V phase, and V ₅₁ is applied from U phase to V phase.

  The PWM signal generation method for outputting each of the voltage vectors described above is as follows. The purpose of the rectangular wave energization is to reduce the on / off operation of the sine wave energization switching element 3, and therefore the operation of the switching element 3 is suppressed as much as possible. FIG. 7E shows a PWM signal for the operation of the six switching elements obtained at this time.

For example, in the operation in the stage 0-mode 0, the output voltage vector is V ₀₀ and is a state in which voltage is applied from the U phase to the V and W phases. At this time, the U phase to which the positive side of the DC voltage is applied is only the U phase upper switching element 3a among the U phase upper switching element 3a and the U phase lower switching element 3d connected to the U phase of the inverter main circuit 5. The U-phase lower switching element 3d is always turned off. Further, since the negative side of the DC voltage is applied in the V and W phases, only the V phase lower switching element 3e and the W phase lower switching element 3f are operated in the V and W phases, and the V phase upper switching element 3b is operated. The W-phase upper switching element 3c is always off under this stage-mode condition. Further, as shown in FIG. 7E, the U-phase upper switching element 3a is always turned on between the same stage and the mode, and the operation of the switching element is further reduced. At this time, three of the six switching elements are always off, one is always on, and the other two carrier periods are turned on / off. As a result, the number of operation of the switching element in the stage 0-mode 0 becomes about 1/3 compared with the sine wave PWM.

Further, in the operation in the stage 0-mode 1, the output voltage vector is V ₀₁ , and the voltage is applied from the U phase to the W phase. At this time, the U phase to which the positive side of the DC voltage is applied operates only the U phase upper switching element 3a, and the U phase lower switching element 3d is always off. In the W phase, since the negative side of the DC voltage is applied, only the W phase lower switching element 3f is operated, and the W phase upper switching element 3c is always turned off. Further, in the V phase in which no energization is performed, both the V phase upper switching element 3b and the V phase lower switching element 3e are always off. Further, as shown in FIG. 7E, the U-phase upper switching element 3a is always turned on between the same stage and the mode, and the operation of the switching element is further reduced. At this time, four of the six switching elements are always off, one is always on, and the other is an on / off operation in the carrier cycle. As a result, the number of operation of the switching element in the stage 0-mode 1 becomes about 1/6 compared with the sine wave PWM.

  The same applies to the other stages. For each stage-mode, a PWM signal as shown in FIG. 7E is generated to operate the switching element. In the PWM signal of each switching element in FIG. 7 (e), a section indicated by A indicates a state in which the stage-mode is always on, and a shaded portion indicates a section in which on / off is repeated according to the on-duty. ing. As shown in FIG. 7 (e), in mode 0 and mode 2 of each stage, a total of three switching elements, that is, a positive phase upper switching element and a negative phase lower switching element when energized are operated. Furthermore, any one of the switching elements is fixed to be on, and the number of operations is about 1/3 when the sine wave is energized. Further, in mode 1 of each stage, a total of two switching elements, that is, the upper switching element of the positive phase when energized and the lower switching element of the negative phase are operated, and any one switching element Is fixed to ON, and the number of operations is about 1/6 when the sine wave is energized. Here, in FIG. 7E, the section that is always on is set to 60 degrees in the latter half of the reference 120-degree section, but may be changed to an arbitrary section.

  Here, mode 0 of each stage and mode 2 of the preceding stage are conditions for outputting the same voltage vector, but in FIG. 7, the PWM generation method is changed. This is because in this way, the voltage fluctuation of the phase that starts the on / off operation in mode 2 can be reduced. Of course, the same PWM may be output.

Next, when the rectangular wave PWM generating means 15 generates the PWM signal as described above, the on / off on-duty setting method for the switching element that performs the on / off operation among the six switching elements 3 will be described. This will be described below. The sine wave PWM generation means 16 obtains the on-duty as shown in FIG. 3A based on the modulation rate Vk ^* of the output voltage vector obtained from the modulation rate / phase calculation means 13. The rectangular wave PWM is obtained from the modulation factor Vk ^* as in the sine wave PWM.

For example, the condition of stage 0-mode 1 is shown here in the case where a voltage is applied from the U phase to the W phase, the U phase upper switching element 3a is fixed to ON, and the W phase lower switching element 3f. May be driven on / off. The voltage vector V ₀₁ generated at this time is a voltage vector Vuw generated when a voltage is applied between the UWs as shown in FIG. In order for this voltage vector V ₀₁ to generate a voltage having the same magnitude as the voltage vector command value | Vo ^* | between the UW phases, the W-phase lower switching element is used by using the modulation factor Vk ^* at this time. The on-duty of 3f may be Vk ^* . Similarly, in mode 1 of each stage, if the on-duty of the PWM signal of the switching element that performs the on / off operation is the modulation factor Vk ^* , the average output voltage magnitude in the carrier period is the voltage vector command value. Is equal to | Vo ^* |.

Further, the condition of stage 0-mode 0 is shown here, in which the voltage is applied from the U phase to the V and W phases, the U phase upper switching element 3a is fixed on, and the V phase lower switching The element 3e and the W-phase lower switching element 3f may be driven on / off. As shown in FIG. 8B, the voltage vector V ₀₀ generated at this time is a voltage vector Vuv generated when a voltage is applied between UV and a voltage vector Vuw generated when applied between UWs. It is a composite vector. In order for this combined vector to generate a voltage having the same magnitude as the voltage vector command value | Vo ^* |, using the modulation factor Vk ^* at this time, the V-phase lower switching element 3e and the W-phase lower The on-duty of the side switching element 3f may be set to 1 / √3 of Vk ^* . Similarly, in modes 0 and 2 of each stage, if the on-duty of the PWM signal of the switching element that performs the on / off operation is 1 / √3 of the modulation factor Vk ^* , the average output voltage in the carrier period is The magnitude is equal to the magnitude | Vo ^* | of the voltage vector command value.

As described above, in the rectangular wave PWM, if the PWM signal is obtained as described above in the modes 0, 2 and mode 1 of each stage, the average output voltage in the carrier cycle for generating the PWM is the output voltage vector command value. It becomes equal to the magnitude | Vo ^* |. However, considering an electrical angle cycle, that is, a case where the voltage vector makes one rotation, a difference occurs in the effective value of the voltage in the electrical angle cycle depending on the energization width θon. For this reason, when switching from sine wave energization to rectangular wave energization, switching from rectangular wave energization to sine wave energization, or switching the energization width at the time of rectangular wave energization, the output voltage changes sharply. . This is not preferable in terms of control, and the output torque of the synchronous motor 6 changes abruptly, causing problems such as stoppage due to overcurrent or unstable control.

  In order to switch the energization method and the energization width while maintaining control stability, the output torque of the synchronous motor 6 may be made constant before and after switching. Since the output torque of the synchronous motor 6 is proportional to the primary frequency component of the current flowing into the synchronous motor 6, the magnitude and phase of the primary frequency component included in the output voltage are made constant before and after switching. The on-duty may be adjusted.

With reference to the primary frequency component of the output voltage between the phases generated by the inverter device 2 by the PWM signal from the sine wave PWM generation means 16 described above, the phase between the phases generated by the inverter device 2 by the PWM signal from the rectangular wave PWM generation means 15 described above is used as a reference. FIG. 9 shows a comparison of primary frequency components included in the output voltage. At this time, considering the voltage for each phase, in the non-energized section (mode 1) in the rectangular wave energization, the voltage is not applied from the inverter device 2 side in the non-energized phase. However, when the synchronous motor 6 is rotating, an induced voltage is generated by rotation of a magnet in the rotor of the synchronous motor 6. For this reason, in the non-energized section, a difference occurs between the voltage vector applied by the inverter device 2 and the voltage generated at the terminal of the actual synchronous motor 6. Therefore, in the non-energized section during energization of the rectangular wave, it is assumed that an induced voltage generated from the rotor of the synchronous motor 6 is generated in the primary frequency component of the output voltage. For example, regarding the induced voltage Vemf in the non-energized section, the following equation is used assuming that the rotor is rotating at the same rotation speed as the primary rotation speed ωe of the output voltage. However, φf is an induced voltage constant of the synchronous motor 6.
Vemf = ωe × φf (3)

Based on FIG. 9, the correction coefficient obtained from the reciprocal of FIG. 9 is used to make the magnitude of the primary frequency component of the interphase output voltage constant so as to make the output torque of the synchronous motor 6 constant regardless of the energization method and the energization width. Using K ₀ , the modulation factor Vk ^* is corrected by the following formula, and a rectangular wave energization PWM signal is generated using the corrected modulation factor Vk ^* ′.
K ₀ = 0.0078 × θon + 0.0284 (4)
Vk ^* ′ = K ₀ × Vk ^* (5)

By using the correction coefficient K ₀ above, even if switching of switching and conducting width θon the energization method Md by conduction waveform selecting means 14, the output torque is maintained substantially constant, stable control during switching Therefore, it is possible to secure the characteristics and prevent overcurrent, and instantly switch to the energization waveform output by the energization waveform selection means 14.

  Here, the above equation (5) is used as an example of an arithmetic expression for making the primary component of the interphase output voltage constant during waveform switching. However, in actuality, due to the error between the modulation factor in the inverter device 2 and the voltage actually output from the inverter device 2, the distortion component included in the induced voltage generated by the synchronous motor 6, the influence of the armature reaction, etc. In the equation ()), an error occurs and the switching may not go smoothly. Such a case may be dealt with by calculating a correction coefficient in consideration of these error factors.

  On the other hand, when it is not necessary to instantaneously switch to the energization waveform such as the energization method Md or the energization width θon selected by the energization waveform selection means 14, or when switching instantaneously is not preferable in terms of sound and vibration, etc. The energization waveform may be switched gently. For example, when the operating conditions of the synchronous motor 6 change during driving with sine wave energization and the energization waveform selection means 14 selects a rectangular wave energization waveform, the rectangular wave is energized 180 degrees first, and then changes at a predetermined rate of change. It is only necessary to finally shift to the energization width θon selected by the energization waveform selection means. Conversely, when the operating condition of the synchronous motor 6 changes during driving with rectangular wave energization, and the energization waveform selection means 14 selects a sine wave energization waveform, first, 180 degrees energization is performed in the rectangular wave energization state. The energization method may be switched to sinusoidal energization when the energization is changed at a predetermined change rate and energization is 180 degrees.

The energization waveform selection means 14 will be described in detail. The energization waveform selection means 14 obtains and outputs the most suitable energization method Md and energization width θon for realizing the synchronous motor drive condition Jk ^* . Here, when the energization waveform selection means 14 selects the sine wave energization as the energization method, the energization width θon is basically 180 degrees because the energization is always 180 degrees energization during the sine wave energization. When the sine wave is energized, the energization width θon is irrelevant. Here, sine wave energization and 180 degree rectangular wave energization output different PWM signals.

Ideally, it is desirable that a single energization waveform can achieve the maximum effect in both high efficiency operation and low noise. However, depending on the specifications of the synchronous motor driving device and the synchronous motor, it may be difficult to achieve both. In such a case, the required specification changes mainly depending on which condition is required. For example, a certain amount of noise is allowed and high-efficiency driving is emphasized, or a certain degree of efficiency reduction is allowed and low-noise driving is emphasized. Corresponding to such a case, the synchronous motor drive device according to the present invention sets the high efficiency drive request and the low noise drive request as the synchronous motor drive condition Jk ^{*, and} is configured as follows.

For example, when high-efficiency driving is requested as the synchronous motor drive condition Jk ^* , the energization waveform selection unit 14 uses the three-phase currents Iu, Iv, Iw obtained from the phase current calculation unit 11 and the output voltage vector calculation unit 12. The energization method Md and the energization width θon maximizing the efficiency are obtained according to the output voltage vector command value V _o ^* obtained and the operation state of the synchronous motor 6 obtained from the rotation speed command value ω ^* . At this time, the operating state of the synchronous motor 6 is obtained in the energization waveform selection means 14 from the rotational speed command value ω ^* and the three-phase currents Iu, Iv, Iw.

At this time, in the energization waveform selection means 14, a table or a table that receives the rotation speed command value ω ^* and the three-phase currents Iu, Iv, and Iw and outputs the energization method Md and the energization width θon for realizing the maximum efficiency. Select the energization waveform using the function. If the three-phase currents Iu, Iv, and Iw are simply input, the amount of variables in the table and function increases, and the calculation becomes complicated. Therefore, the number of variables can be changed by using means such as converting the three-phase current to the current effective value Irms. May be reduced. Further, the output of the synchronous motor 6 is obtained from the rotational speed command value ω ^* , the three-phase currents Iu, Iv, Iw and the motor constant of the synchronous motor 6, or the motor output information is received from the outside, etc. A table or function may be used.

In the energization waveform selection means 14, the modulation rate Vk ^* of the output voltage vector is monitored, and the energization waveform is switched according to the modulation rate Vk ^* . For example, when the modulation rate Vk ^* becomes Vk ^* > 1, that is, a voltage saturation state where the magnitude | Vo ^* | of the output voltage vector command value exceeds the DC voltage that can be output by the inverter device. At this time, the inverter device 2 cannot output the required voltage obtained by the inverter control means 8, and it becomes difficult to drive the rotational speed of the synchronous motor 6 to satisfy the rotational speed command value ω ^* . In the voltage saturation state, there are methods such as weak excitation control and overmodulation operation as a method for expanding the operation range of the synchronous motor 6, but the sine wave energization method is more advantageous in realizing these methods. For this reason, the energization waveform selection means 14 may be switched to the sine wave energization method when the voltage is saturated, that is, when Vk ^* > 1.

  The optimum conditions for the energization waveform at the time of voltage saturation are different depending on the configuration of the inverter device 2 and the specification of the synchronous motor 6 used in the energization waveform selection means 14. For this reason, what is necessary is just to obtain | require the optimal conditions of the inverter apparatus 2 and the synchronous motor 6 by experiment and simulation, and to reflect in a table or a function. When the rectangular wave energization is superior in terms of efficiency or stability as the drivability at the time of voltage saturation due to the configuration of the inverter device 2 and the specification of the synchronous motor 6, the rectangular wave energization may be selected. The specifications of the inverter device 2 are switching element characteristics, short-circuit prevention time, and carrier frequency. The specifications of the synchronous motor 6 include the number of poles of the motor, the rotor shape such as IPMSM and SPMSM, the stator shape such as concentrated winding and distributed winding, the magnetized shape of the rotor magnet, and the induced voltage waveform.

  Here, the experimentally obtained results of the circuit efficiency, the motor efficiency, and the overall efficiency with respect to the energization waveform will be described. The circuit efficiency is obtained by (circuit output power) / (circuit input power), the motor efficiency is obtained by (motor output) / (motor input = circuit output power), and the overall efficiency is (motor output) / (circuit input power). ) = (Circuit efficiency) × (motor efficiency). A combination of a certain inverter device and a synchronous motor having the structure shown in FIG. 10, under the rotational speed and load torque conditions that are typical operating points of this synchronous motor, the circuit efficiency of the inverter device with respect to the energized waveform, the motor efficiency, The results of experimentally determining the relationship with the overall efficiency are shown in FIGS. 11, 12, and 13, respectively.

  In the synchronous motor of FIG. 10, the synchronous motor includes a stator M1, a stator winding M2, a rotor M3, and a permanent magnet M4 in the rotor. Further, in FIG. 11, FIG. 12, and FIG. 13, each result is shown as a ratio of efficiency in each energization width, assuming that each efficiency during sine wave energization is 1. Waveform A shows the efficiency when the rotational speed is ω1 and the load torque is τ1, waveform B shows the efficiency when the rotational speed is ω2 and the load torque is τ1, and waveform C shows the rotation. Each efficiency is shown when the speed is ω3 and the load torque is τ2, and the waveform D shows each efficiency when the rotation speed is ω4 and the load torque is τ2. Here, ω1 <ω2 <ω3 <ω4 and τ1 <τ2.

  The circuit loss of the inverter device 2 is divided into a switching loss mainly caused by the on / off operation of the switching element 3 and a conduction loss caused by a current flowing through the inverter main circuit 5. Since the switching loss decreases as the number of operations per unit time of the switching element 3 that performs the on / off operation decreases, the switching loss decreases as the carrier frequency decreases and the energization width decreases. On the other hand, the conduction loss is heat generated by the resistance component of the circuit, and increases in proportion to the square of the current flowing through the circuit. Therefore, the conduction loss increases as the current becomes a higher load condition. The relationship between the energization waveform (energization method and energization width) of the inverter device 2 used in the experiment and the inverter efficiency is as shown in FIG.

  In order to reduce the switching loss, the number of on / off times of the switching element per unit time may be reduced. Therefore, when the carrier frequency is constant, as shown in FIG. 11, if the load is light, the sine wave energization with the highest number of switching times has the worst circuit efficiency and the energization with the rectangular wave energization with the lower number of switching times. The narrower the condition, the better the circuit efficiency.

  In order to reduce the switching loss, it is effective to narrow the energization width. However, when the energization width is narrowed, the current flowing into the synchronous motor 6 is distorted and the included harmonic current increases, so the current value increases. . For this reason, the conduction loss of the inverter circuit tends to increase. For this reason, under heavy load conditions (conditions with a large current), the conduction loss is more dominant than the switching loss, and the conduction width with the best circuit efficiency is shifted in the direction of the wider conduction width.

  Further, the electrical loss in the synchronous motor 6 can be mainly separated into copper loss and iron loss. Copper loss is Joule heat generated by the current flowing through the winding of the synchronous motor 6 and the winding resistance, and is proportional to the square of the current. The iron loss is a loss caused by magnetic hysteresis and eddy current in the iron core portion of the motor, and is generated depending on the frequency and magnitude of the current. The frequency components of the main current involved in the iron loss are the primary frequency component (synchronized with the rotation speed of the output voltage vector), the carrier frequency component generated by the on / off operation of the plurality of switching elements 3 by the PWM signal, and rectangular wave energization This is a high-frequency component generated by converting the output voltage into a rectangular wave.

  Here, the relationship between the energization waveform (energization method and energization width) of the synchronous motor 6 shown in FIG. 10 used in the experiment and the motor efficiency is as shown in FIG. As long as the carrier frequency is constant, the rotational speed is the same, and the load torque is the same, there is almost no difference due to the difference in the energization waveform in the primary component of the current required to obtain the torque. However, in the harmonic component, in the rectangular wave energization, compared to the sine wave energization, a decrease in the carrier frequency component due to a decrease in the number of switching operations and an increase in the harmonic component due to an increase in current distortion due to the rectangular wave energization can be considered. For this reason, between sine wave energization and rectangular wave energization, the amount of iron loss and copper loss changes due to the difference in harmonic components contained in the current, resulting in a difference in motor efficiency. However, in this result, since the current harmonics of the carrier frequency component occupy a small percentage of the total current, even if the energization width is reduced and the number of switchings is reduced, the carrier frequency component is not reduced so much. For this reason, in the electric motor shown here, as shown in FIG. 12, the efficiency of the electric motor decreases due to the increase in the iron loss and the copper loss as the energization width is narrow and the energizing width is narrower. On the contrary, the motor efficiency is improved under the condition that the energization width is wide by sine wave energization or rectangular wave energization with less harmonic components.

  The total efficiency shown in FIG. 13 is obtained by multiplying the results of FIGS. 11 and 12 because it is given by the product of the circuit efficiency and the motor efficiency. As shown in FIG. 13, the energization waveform that maximizes the overall efficiency varies depending on the rotational speed and load conditions. In this way, switching the energization waveform based only on the motor efficiency does not necessarily maximize the overall efficiency. Thus, by providing the energization waveform selection means 14 with the information based on the experimental results and the analysis results based on the simulation as a table or function, the energization waveform that maximizes the overall efficiency is selected according to the operating state of the synchronous motor 6. To.

Based on the results obtained in FIG. 13, FIG. 14 shows an example of an energization waveform selection table provided in the energization waveform selection means 14 when the synchronous motor drive condition Jk ^* is requested to be highly efficient. FIG. 14 is a table setting example when the synchronous motor used here is used. In FIG. 14, the reference value for selecting the energization waveform is the speed command value ω ^* and the current effective value Irms, and a table for outputting the energization waveform for bringing the total efficiency close to the maximum is shown.

  This will be described with reference to FIG. The line indicated by ωA is the maximum rotational speed of the synchronous motor, and is determined by the specifications of the synchronous motor 6 and the load driven by the synchronous motor. The line indicated by I1 is the maximum current value of the inverter device 2 or the synchronous motor 6, and is determined by the allowable current value of the semiconductor element used in the inverter device 2 or the allowable current value of the synchronous motor (demagnetization limit current, etc.). . A line indicated by P <b> 1 is a line indicating the maximum output of the synchronous motor 6 and is determined by the specifications of the synchronous motor 6. A line indicated by P2 indicates a region where the output voltage of the inverter device 2 is saturated, and an overmodulation state is obtained on the right side of the line. P2 is determined by the specifications of the synchronous motor 6 and the DC bus voltage Vdc of the inverter device 2. The above ωA, I1, P1, and P2 are conditions determined by the inverter device 2, the synchronous motor 6, and the load driven by the synchronous motor 6, and the other conditions of I2, P3, P4, and P5 are shown in FIG. The decision will be based on efficiency data.

  I2 indicates a limit current value of rectangular wave energization. When this current effective value is exceeded during energization of the rectangular wave, the current is forcibly switched to sine wave energization. If the load torque increases during the rectangular wave energization and the current effective value approaches the current limit value I1, the difference between the sine wave energization and the rectangular wave energization decreases in terms of overall efficiency. Further, in the rectangular wave energization, no voltage is applied from the inverter device 2 in the non-energized section, and thus current control cannot be performed in this region. However, in the non-energized zone, a circulating current is generated that continues to flow until the current flowing in the energized zone immediately before the non-energized zone becomes zero. Since the circulating current flows in the non-energized section, it cannot be controlled. For this reason, when the current value increases, the circulating current also increases, and the control may become unstable due to the influence of the circulating current. Therefore, the current limit value I2 for energizing the rectangular wave is set with a current value that is not affected by the circulating current. When the current effective value exceeds the predetermined value I2, control stability can be maintained by switching to sine wave energization.

  P3, P4, and P5 indicate lines where the output of the synchronous motor 6 is constant. The output of the synchronous motor 6 so as to realize an energization waveform that maximizes the overall efficiency for each of the operating conditions A, B, C, and D in which the synchronous motor 6 shown in FIGS. 11, 12, and 13 is mainly used. This is a table in the case where a boundary line for switching the energization waveform is determined using a line in which is constant. Here, the rectangular wave energization region is divided into 150 ° energization, 165 ° energization, and 180 ° energization. In such a case, the operating condition of the synchronous motor 6 is in the vicinity of the energization waveform switching region. In this case, the energization waveform may be frequently switched due to minute fluctuations in the operating conditions. This phenomenon is not particularly problematic for control, but it may cause noise. To avoid this phenomenon, a hysteresis characteristic is provided at the switching boundary of the energized waveform, and the switching of the energized waveform does not occur frequently. You can do that. Further, the energization waveform boundary as shown in FIG. 14 may be eliminated, and the energization waveform may be switched continuously.

  Here, these tables need to be set in accordance with the specifications of the inverter device 2 and the synchronous motor 6 that are actually configured as the synchronous motor driving device. The specifications of the inverter device 2 are switching element characteristics, short-circuit prevention time, and carrier frequency. The specifications of the synchronous motor 6 include the number of poles of the motor, the rotor shape such as IPMSM and SPMSM, the stator shape such as concentrated winding and distributed winding, the magnetized shape of the rotor magnet, and the induced voltage waveform. The efficiency improvement effect obtained by switching the energization method according to the rotation speed and load condition shown here is particularly effective for an electric motor having a low output (about several kW output or less) and a large stator winding impedance.

The condition for switching the energization waveform shown in FIG. 14 is such that the energization waveform is obtained by referring to a two-dimensional table of the rotational speed command value ω ^* and the current effective value Irms. In order to achieve this, the parameters to be referred to may be reduced. For example, the output power of the synchronous motor 6 may be obtained from the voltage value and current value output from the inverter device, or the rotation speed and current value, and the energization waveform may be switched according to the output power value. In addition, the energization waveform may be switched with reference to one parameter such as a rotation speed, a current effective value, and a modulation rate.

  FIG. 15 shows an example of a table that is simplified and configured to use one rotation speed command value as a reference parameter in the energization waveform selection table used in the energization waveform selection means 14. In FIG. 15, the switching of the energization waveform is an example in which the energization width is smoothly changed without providing a boundary. Similarly, FIG. 16 shows an example in which the rotational speed command value is divided for each speed range and an energization waveform is assigned to each. Here, FIG. 15 or FIG. 16 is a table that increases the energization width as the speed command value increases. However, depending on the combination of the synchronous motor 6 and the inverter device 2, the energization width may be reduced in the middle. In some cases, it is better to configure a simple table. This varies depending on the combination of the inverter device 2 and the synchronous motor 6. 15 and 16, the table reference parameter is the rotation speed command value, but the same applies even when the effective current value or the modulation factor is used as the horizontal axis in FIGS.

  Furthermore, it is possible to further simplify the operation conditions that do not require much accuracy of maximizing the overall efficiency by the energization waveform conditions. In the synchronous motor shown in FIG. 13, the overall efficiency is better when the rectangular wave energization is performed on the low speed side. Therefore, the rectangular wave energization is used in the region below the predetermined rotation speed, and the sine wave energization is performed at a higher rotation speed. May be used. For example, up to about 40% of the maximum number of rotations of the synchronous motor is 150-degree rectangular wave energization, and the region beyond that is sine wave energization. If it is sine wave energization, when the modulation factor becomes 1 or more, it is possible to smoothly shift to the overmodulation operation.

  On the other hand, in the synchronous motor 6 to be driven by the synchronous motor driving device, the relationship between the parameters (rotation speed, load current, modulation factor) and the energization waveform for maximizing the overall efficiency as shown in FIG. 14 is tested. In some cases, it cannot be obtained in advance by simulation or simulation. As another method that can be used in such a case, the energization waveform selection means 14 is configured to automatically select an energization waveform that achieves the maximum efficiency as follows. The flow of automatic search at this time is shown in FIG.

  The flow of automatic search in FIG. 17 will be described. In FIG. 17, in the energization waveform changing step ST11, the energization waveform is switched from a 120-degree rectangular wave energization at a predetermined energization width increase rate. At this time, when the previous energization waveform is 180-degree rectangular wave energization, sine wave energization is performed. In a predetermined time waiting step ST12, after switching the energization waveform, the same energization waveform is maintained until the driving state of the synchronous motor is stabilized. In a predetermined time waiting step ST12, the process proceeds to the next step after the stability of the driving state is detected or after a predetermined time has elapsed. In the current detection step ST13, a current flowing into the inverter device 2 is detected. In the energization waveform / current recording step ST14, the energization waveform information changed in the energization waveform changing step ST11 and the current signal detected in the current detection step ST13 are recorded in the memory of the microprocessor. In the energization waveform round detection step ST15, it is determined whether or not the energization waveform has made a complete cycle (for example, whether or not switching from the 120-degree rectangular wave energization to the 180-degree rectangular wave energization followed by switching to the sine wave energization) If not, the process proceeds to the minimum current energization waveform selection step ST16. If not, the process returns to the energization waveform change step ST11. In the minimum current energization step ST16, an energization waveform having the minimum current value is selected from the relationship between the energization waveform recorded on the memory of the microprocessor and the current value, and the energization waveform is output to be executed. In this way, the energization waveform is changed as appropriate, and the energization waveform that minimizes the energization waveform is selected, and the energization waveform that automatically minimizes the current, that is, the more efficient energization waveform is searched.

In this automatic energization waveform search, the optimum energization waveform is searched again when the rotation speed command value ω ^* changes, when the current flowing into the synchronous motor 6 changes, or after a predetermined time has elapsed. Here, since the current detected after changing the energization waveform may be a current flowing into the inverter device 2, it is a direct current in the example shown in FIG. As another current detection element, an alternating current that is input to a rectifying means (not shown) provided to create the direct current power supply unit 1 of FIG. 1 may be used.

For example, when low noise driving is requested as the synchronous motor drive condition Jk ^* , the energization waveform selection unit 14 obtains the three-phase currents Iu, Iv, Iw obtained from the phase current calculation unit 11 and the output voltage vector calculation unit. In accordance with the operating state of the synchronous motor 6 obtained from the output voltage vector command value Vk ^* and the rotational speed command value ω ^* , the energization method and the energization width that minimize the noise are obtained.

At this time, in the energization waveform selection means 14, a table or a table that receives the rotation speed command value ω ^* and the three-phase currents Iu, Iv, Iw and outputs the energization method Md and the energization width θon for realizing noise reduction. Select the energization waveform using the function. If the three-phase currents Iu, Iv, and Iw are simply input, the amount of variables in the table and function increases, and the calculation becomes complicated. Therefore, the number of variables can be changed by using means such as converting the three-phase current to the current effective value Irms. May be reduced. Further, the output of the synchronous motor 6 is obtained from the rotational speed command value ω ^* , the three-phase currents Iu, Iv, Iw and the motor constant of the synchronous motor 6, or the motor output information is received from the outside, etc. A table or function may be used.

In the energization waveform selection means 14, the modulation rate Vk ^* of the output voltage vector is monitored, and the energization waveform is switched according to the modulation rate Vk ^* . For example, when the modulation rate Vk ^* becomes Vk ^* > 1, that is, a voltage saturation state where the magnitude | Vo ^* | of the output voltage vector command value exceeds the DC voltage that can be output by the inverter device. At this time, the inverter device 2 cannot output the required voltage obtained by the inverter control means 8, and it becomes difficult to drive the rotational speed of the synchronous motor 6 to satisfy the rotational speed command value ω ^* . In the voltage saturation state, there are methods such as weak excitation control and overmodulation operation as a method for expanding the operation range of the synchronous motor 6, but the sine wave energization method is more advantageous in realizing these methods. For this reason, the energization waveform selection means 14 may be switched to the sine wave energization method when the voltage is saturated, that is, when Vk ^* > 1.

  Regarding the relationship between the energization waveform and the noise, when comparing the sine wave energization and the rectangular wave energization, the sine wave energization has lower noise, and in the rectangular wave energization, the wider the energization width, the lower the noise. This is because the harmonic component contained in the current flowing into the synchronous motor 6 causes fluctuations in the output torque of the synchronous motor, and thus ripples are generated in the output torque of the synchronous motor, and vibration and sound are generated. However, in the sine wave energization, although the ripple component of the output torque of the synchronous motor 6 is small, the high frequency sound of the carrier frequency component is likely to be generated because the plurality of switching elements 3 of the inverter device 2 are frequently turned on / off. . For this reason, when the carrier frequency is used as a human audible frequency, an electromagnetic sound of the carrier frequency component is generated, and in some cases, the noise may be higher than the rectangular wave energization.

  In particular, with respect to noise, the situation in which noise is generated varies depending on the specifications of the synchronous motor 6, the environment in which the synchronous motor driving device is used, and the system having the synchronous motor driving device as one component. Even in an energization waveform condition that does not cause noise in the combination of the inverter device 2 and the synchronous motor 6, there may be noise that coincides with the mechanical resonance point of the system having the synchronous motor drive device as one component. In consideration of these, depending on the state in which the synchronous motor driving device is used, the relationship between the energization waveform realizing low noise and the operating condition of the synchronous motor is obtained by experiment or simulation, and this data is obtained as the energization waveform selecting means 14. It is good to provide as a table or function.

FIG. 18 shows an example of an energization waveform selection table in the energization waveform selection means 14 used when a low noise condition is required as the synchronous motor drive condition Jk ^* . In the table shown in FIG. 18, a rotational speed command value is used as an input parameter for switching the energization waveform. When the speed command value is in the range of ω1 ″ to ω2 ″ and in the range of ω3 ″ to ω4 ″, the distortion component of the voltage or current waveform matches the mechanical resonance point in the rectangular wave energization, and noise is generated. . In these examples, sine wave energization is used in these regions, and an energization method that reduces noise in other rotation speed regions is selected.

Even when high-efficiency driving or low-noise driving is required as the synchronous motor driving condition Jk ^* , if the energization method is switched for control, the control itself becomes unstable or eventually stops or stops due to overcurrent In order to ensure controllability regardless of the synchronous motor drive condition Jk ^* , it may be forcibly switched to the energized waveform with the most stable control. For example, as described above, since the number of voltage vectors to be output is smaller in the rectangular wave energization than in the sine wave energization, the controllability is deteriorated compared to the sine wave energization. As a result, when the rectangular wave is energized, the control may not be continued during acceleration / deceleration, extremely low speed, and extremely low load (effective current value is not more than a predetermined position). In such a case, sine wave energization is forcibly set regardless of the table or function shown in FIG.

Further, when the DC voltage of the DC power supply unit 1 decreases due to fluctuations in the power supply voltage, the maximum value of the magnitude of the voltage vector that can be output from the inverter device decreases due to voltage saturation. Even in such a case, the modulation rate Vk ^* of the output voltage vector is monitored in the energization waveform selection means 14, and when the modulation rate Vk ^* becomes Vk ^* > 1, a sine wave energization which is a stable energization method in terms of control. You may make it switch to. At this time, since the voltage is saturated, the synchronous motor 6 is driven by overmodulation operation with sine wave energization or weak excitation operation. Here, when the threshold for switching the energization waveform is set at one point, the energization method may be frequently switched in the vicinity of the switching threshold. This phenomenon is not particularly problematic from the viewpoint of controllability, but noise may occur. In this case, a hysteresis characteristic may be provided in the switching threshold value to suppress frequent switching of the energization method when operating near the switching threshold value.

The output voltage vector calculation means 12 will be described in detail. Here, based on the current signal obtained from the current detection means 7, an output composed of the magnitude and phase angle of the output voltage for driving the synchronous motor 6 given from the outside so as to satisfy the rotational speed command value ω ^*. Obtain the voltage vector command value. Here, as a control method for obtaining the output voltage vector command value, any control method may be used as long as the method can finally obtain the magnitude and phase angle of the output voltage.

  As an example of the control method, there is a vector control method in which the coordinate axis for control is converted into the magnetic flux direction (d axis) and the output torque direction (q axis) with respect to the synchronous motor 6, and the voltage command value of each axis is obtained. . This is because the output voltage vector calculation means 12 detects a phase current flowing into the synchronous motor 6 and converts the phase current obtained by the current detection means 7 into an excitation current component and a torque current component 3. An energization method selected by an energization waveform selection means, comprising: a phase 2-phase conversion means; and a voltage command value calculation means for obtaining an output voltage command value based on an excitation current component and a torque current component obtained by the 3-phase 2-phase conversion means Alternatively, regardless of the energization width, the same output voltage vector command value calculation means is used so that one piece of hardware is required for each of the rectangular wave energization and the sine wave energization, and further the output voltage vector command value calculation is performed. The software can also be one, and the configuration of the inverter device can be simplified, reduced in size, and reduced in cost. Furthermore, the required amount of RAM (Random Access Memory) and ROM (Read Only Memory) of the microprocessor can be reduced, and the computational load is not increased significantly, so a lower cost microprocessor can be used. In addition, the cost of the inverter device can be reduced.

  Alternatively, depending on the rotor position of the synchronous motor 6, a control method for applying a voltage at a predetermined advance phase angle, or a phase of a voltage vector applied to the synchronous motor 6 and a phase of a current vector flowing into the synchronous motor 6 are predetermined. A control method that controls the phase angle, or a control method that controls the output voltage so as to eliminate an error between the control coordinate axis and the actual rotor coordinate axis, or the amount of magnetic flux generated on the control coordinate axis is constant. For example, there is a control method for controlling the output voltage.

  Each control method uses a method of directly detecting the rotor position of the synchronous motor 6 using a position sensor or the like, and at least one of a current detected without using the position sensor or an output voltage command value. There are methods to estimate the rotor position by calculation using the induced voltage and saliency of the motor, or to detect the rotor position by detecting the potential change at the neutral point of the motor. Also good.

  When the output voltage vector calculation means 12 is a position sensor for detecting the rotor position or speed of the motor or a position / speed sensorless control means that does not use a speed sensor, the reliability of sensor attachment and sensor signal processing It can be applied to applications where the position or speed sensor cannot be attached to the synchronous motor. In addition, since it is not necessary to detect the induced voltage generated, there is no need to secure a non-energized section necessary for detecting the induced voltage, and the energization width is arbitrarily set between 120 degrees and 180 degrees by rectangular wave energization. In addition to the sine wave energization, a wide range of energization waveforms can be selected.

In sinusoidal energization, the output voltage vector command value Vo ^* is updated every control cycle Ts. Similarly, the output voltage vector command value Vo ^* may be updated every control cycle Ts even when the rectangular wave is energized. However, when the rectangular wave is energized, only six or twelve voltage vectors corresponding to the phase of the output voltage vector command value Vo ^* are output as described above. For this reason, in the stage-mode condition shown in FIG. 7, in the phase range in which the same vector is output, the phase angle of the output voltage vector is fixed to 6 or 12 voltage vectors, and every control cycle Ts. Only the magnitude of the voltage vector changes depending on the voltage vector command value Vo ^* . Here, in the rectangular wave energization, only the control electrical angle phase is updated, the calculation of the voltage vector command value at the control frequency Fs is stopped, and the calculation is performed once for six or twelve output voltage vectors. An output voltage vector command value may be obtained. At this time, the number of computations of the output voltage vector command value Vo ^{* in} one electrical angle cycle is 6 to 12 regardless of the control cycle Ts.

  The timing of the output voltage vector command value calculation may be performed when the voltage vector is switched when the rectangular wave is energized, that is, when the stage or mode is switched. However, at the time of rectangular wave energization, the current is in a state that changes transiently when the voltage vector is switched, so in a control method that detects current like a control method based on sine wave energization, There is a possibility that a deviation occurs in the control calculation and adversely affects the control. In such a case, current detection is performed after a predetermined time or a predetermined phase has elapsed since the switching of the output voltage vector, and the output voltage vector command value is calculated using the current signal at this time. You should do it. As an example of the predetermined phase, the stage shown in FIG. 7 and the half of the stage-mode phase range may be used.

  The synchronous motor 6 includes a permanent magnet type synchronous motor (PMSM), a brushless DC motor (BLDCM), a reluctance motor (RM), a synchronous motor, in addition to an embedded magnet type synchronous motor (IPMSM) and a surface magnet type synchronous motor (SPMSM). Any shape such as an eggplant reluctance motor (SyRM) or a switched reluctance motor (SRM) may be used.

  Here, the flow of the operation in the inverter control means 8 provided in the synchronous motor driving apparatus according to the present embodiment will be described with reference to FIG. FIG. 19 shows processing realized by microprocessor software inside the inverter control means 8. Of course, the same processing may be configured by hardware. The process shown in FIG. 19 is performed every control cycle Ts [s], and generates a PWM signal for performing on / off operations of the plurality of switching elements 3 provided in the inverter main circuit 5.

In FIG. 19, the phase current flowing into the synchronous motor 6 detected by the current detection means 7 in the phase current detection step ST1 is subjected to analog / digital conversion (hereinafter referred to as A / D conversion) to obtain a phase current signal. Here, in the phase current detection step ST1, another one-phase current Iw is obtained from the two-phase currents Iu and Iv as necessary. In the output voltage vector command computing step ST2, the and the phase current detection step ST1 obtained in the phase currents Iu, Iv, Iw, the rotation speed from the rotation speed command value omega ^* synchronous motor 6 satisfies the rotational speed command value omega ^* Thus, an output voltage vector command value Vo ^* for driving is obtained by calculation. In the DC voltage detection step ST3, the DC voltage Vdc detected by the DC voltage detection means 10 is A / D converted to obtain a DC voltage signal Vdc. In the modulation factor / phase calculation step ST4, modulation for obtaining a PWM signal from the output voltage vector command value Vo ^* obtained in the output voltage vector calculation step ST2 and the DC voltage signal Vdc obtained in the DC voltage detection step ST3. The rate Vk ^* and the output voltage vector phase θv are obtained.

In energization waveform selection step ST5, phase current signals Iu, Iv, Iw obtained in phase current detection step ST1, rotation speed command value ω ^* , modulation rate Vk ^* obtained in modulation rate / phase calculation step ST4, and synchronous motor drive conditions From Jk ^* , the energization method Md and the energization width θon of the energization waveform most suitable for the synchronous motor drive condition Jk ^* are obtained. If rectangular wave energization is selected as the energization method Md in the energization waveform selection step ST5, the process proceeds to the modulation rate correction step ST6. In the modulation factor correction step ST6, the modulation factor Vk ^* is corrected in accordance with the energization width θon so that the primary frequency component of the output voltage in the period of one electrical angle is the same as when the sine wave is energized, and the corrected modulation factor Vk ^* ′ Ask for. The rectangular wave PWM calculation step ST7 includes the energization width θon obtained in the energization waveform selection step ST5, the correction modulation rate Vk ^* ′ obtained in the modulation rate correction step ST6, and the output voltage obtained in the modulation rate / phase calculation step ST4. Based on the vector phase θv, a PWM signal for energizing the rectangular wave is obtained.

On the other hand, when the sine wave energization is selected as the energization method Md in the energization waveform selection step ST5, the process proceeds to the sine wave PWM calculation step ST8. In sine wave PWM calculation step ST8, a PWM signal for sine wave energization is obtained based on modulation factor Vk ^* and voltage vector phase θv obtained in modulation factor / phase calculation step ST4.

  Here, in the operation flow of the synchronous motor driving device shown in FIG. 19, all steps are performed at the control cycle Ts [s]. At this time, the rectangular wave PWM calculation step ST7 and the sine wave PWM calculation step are performed. The carrier frequency of the PWM signal obtained in ST8 is 1 / Ts [Hz]. The carrier generation cycle of the PWM signal and the control computation cycle are not necessarily synchronized. At this time, steps other than the rectangular wave PWM calculation step ST7 or the sine wave PWM calculation step ST8 are calculated at the control cycle Ts [s], and the processing of the rectangular wave PWM calculation step ST7 or the sine wave PWM calculation step ST8 is performed in a different cycle. You may carry out by Ts' [s]. In this way, it is not necessary to change the control cycle in accordance with the PWM carrier frequency.

  In the above-described configuration example of the synchronous motor driving device according to the first embodiment, the current detection unit 7 has been described with respect to the method of directly detecting and obtaining the two-phase current flowing into the synchronous motor 6. As another method, the same can be realized by using a method of detecting a direct current between the direct current power supply unit 1 and the inverter device 2 and converting it to a three-phase current from the relationship with the voltage vector output from the inverter device 2. It is.

  Since it comprised as mentioned above, it is the rectangular wave conduction | electrical_connection which can select the arbitrary conduction widths between 120 degree | times and 180 degree | times with one circuit structure, sine wave conduction | electrical_connection, and sine wave conduction | electrical_connection using an overmodulation area | region A plurality of energization waveforms can be realized. Further, it is possible to switch to the most suitable energization waveform in a short time according to the driving conditions (high efficiency driving condition, low noise driving condition) required for the synchronous motor to be driven. By switching the energization waveform, it is possible to set the total efficiency when the synchronous motor is driven to a predetermined value or more, or to reduce the noise when the synchronous motor is driven to a predetermined value or less. Furthermore, compared with a synchronous motor drive device having only one of sine wave energization or rectangular wave energization, overall efficiency can be improved or noise can be reduced.

  Since the rectangular wave PWM generating means 15 sets six voltage vectors to be output in one cycle in terms of electrical angle at 120 degrees energization and 180 degrees energization, the rectangular wave energization at 120 degrees and 180 degrees is output. It becomes possible to approximate the voltage vector to a circular locus of one electrical angle cycle when a sine wave is energized.

  Since the rectangular wave PWM generating means 15 is set to 12 voltage vectors to be output in one cycle in terms of electrical angle when energized with an energization width greater than 120 degrees and smaller than 180 degrees, 120 degrees and 180 degrees It is possible to approximate the output voltage vector of rectangular wave energization at 1 to a circular locus of one electrical angle cycle when energizing a sine wave.

  The rectangular wave PWM generation means 15 includes an output voltage correction means corresponding to the energization width, and includes a primary component included in the voltage output from the sine wave PWM generation means 16 and a voltage output from the rectangular wave PWM generation means 15. Since the pulse width of the output PWM is corrected according to the energization width so that the included primary component becomes equal, it is possible to suppress the output torque fluctuation at the time of switching the energization waveform. It is possible to switch the energization waveform instantaneously while maintaining the stability.

  Since the rectangular wave PWM generator 15 sets the PWM duty at the time of energization between the three phases to 1 / √3 of the energization between the two phases, the magnitude of the output voltage vector generated at the energization between the three phases and the energization between the two phases is set. It becomes possible to make it the same magnitude, and it is possible to suppress voltage fluctuation during one electrical angle cycle, approximate it to a circular locus at the time of sine wave energization, and maintain stable control in rectangular wave energization. In addition, since it is not necessary to detect the induced voltage generated, there is no need to secure a non-energized section necessary for detecting the induced voltage, and the energization width is arbitrarily set between 120 degrees and 180 degrees by rectangular wave energization. In addition to the sine wave energization, a wide range of energization waveforms can be selected.

  The energization waveform selection means 14 switches the energization waveform appropriately after the drive state is stabilized when performing high-efficiency operation including the inverter device 2 and the synchronous motor 6, and generates an energization waveform that minimizes the input current of the inverter device 2. Since the search is made, if the synchronous motor 6 and its load characteristics are not grasped or if preliminary verification by experiment cannot be performed, high-efficiency driving by switching the energized waveform can be realized. Further, even when there is a difference between the result obtained from the table or function and the actual driving situation due to variations in the motor characteristics and circuit component characteristics, it is possible to follow the maximum efficiency energization waveform.

  When switching the energization method from sine wave energization to rectangular wave energization, the energization waveform selection means 14 changes the energization width at a predetermined ratio so that a predetermined energization width is obtained after switching to 180-degree rectangular wave energization. As a result, it is possible to suppress the occurrence of sharp fluctuations in the current waveform, noise, and vibration when the energization waveform is switched instantaneously.

  When switching the energization method from rectangular wave energization to sine wave energization, the energization waveform selection means 14 changes the energization width to a 180-degree rectangular wave energization state at a predetermined rate and then switches to sine wave energization. When the energization waveform is switched instantaneously, it is possible to suppress the occurrence of sharp fluctuations in the current waveform, noise, and vibration.

  Since the energization waveform selection means 14 selects the sine wave energization when the output voltage approaches the saturation state and shifts to the overmodulation operation, the energization waveform selection means 14 is controlled when the rectangular wave energization is continued in the voltage saturation state. It becomes possible to suppress the state where the rotation speed becomes unstable or the rotation speed satisfying the rotation speed command value cannot be obtained.

  Since the energization waveform selection means 14 selects the sine wave energization when accelerating or decelerating the synchronous motor 6, the control is more stable when the control becomes unstable during acceleration / deceleration when energizing the rectangular wave. It is possible to shift to a new state.

  Since the energization waveform selection means 14 uses sine wave energization when the effective value of the current flowing into the synchronous motor 6 is smaller than a predetermined value, rectangular wave energization is not performed under the condition that the load is extremely small. When it becomes stable, it becomes possible to shift to a more stable state of control.

  The output voltage vector calculation means 12 includes a current detection means 7 for detecting a phase current flowing into the synchronous motor 6, and a three-phase 2 for converting the phase current obtained by the current detection means 7 into an excitation current component and a torque current component. An energization method selected by the energization waveform selection means, comprising: phase conversion means; and voltage command value calculation means for obtaining an output voltage command value based on the excitation current component and torque current component obtained by the three-phase two-phase conversion means Alternatively, the same output voltage vector command value calculation means is used regardless of the energization width, so that one piece of hardware is required for each of the rectangular wave energization and the sine wave energization, and further the output voltage vector command value calculation is performed. The software to be performed can be reduced to one, and the configuration of the inverter device can be simplified, downsized, and reduced in cost. Furthermore, the required amount of RAM (Random Access Memory) and ROM (Read Only Memory) of the microprocessor can be reduced, and the computational load is not increased significantly, so a lower cost microprocessor can be used. In addition, the cost of the inverter device can be reduced.

  Since the output voltage vector calculation means 12 is a position / speed sensorless control means that does not require a position sensor or a speed sensor for detecting the rotor position or speed of the synchronous motor 6, the sensor is attached and the sensor signal is processed. It is also possible to improve the reliability of the motor and to use in applications where a position or speed sensor cannot be attached to the synchronous motor. In addition, since it is not necessary to detect the induced voltage generated, there is no need to secure a non-energized section necessary for detecting the induced voltage, and the energization width is arbitrarily set between 120 degrees and 180 degrees by rectangular wave energization. In addition to the sine wave energization, a wide range of energization waveforms can be selected.

  The output voltage vector calculation means 12 detects the current after a lapse of a predetermined time from switching of six or twelve output voltage vectors outputted when the rectangular wave energization is selected by the energization waveform selection means. Since the output voltage vector calculation is performed, the control stability can be ensured in the rectangular wave energization, and the same control performance as in the sine wave energization can be obtained.

Embodiment 2. FIG.
FIG. 20 is a diagram showing the second embodiment and shows a synchronous motor driving device. In the present embodiment, when the output voltage vector command value Vo ^* is obtained, the three-phase current is converted into an excitation current component (d-axis current) and a torque current component (q-axis current), and the synchronous motor 6 A case where vector control for controlling to a predetermined value according to the driving state will be described.

  In the figure, a DC power supply unit 1 is connected to an inverter device 2 and supplies DC power to the inverter device 2. The synchronous motor 6 is connected to the inverter device 2, and electric power for driving the synchronous motor 6 is supplied from the inverter device 2.

  The inverter device 2 includes a U-phase upper switching element 3a, a V-phase upper switching element 3b, a W-phase upper switching element 3c, a U-phase lower switching element 3d, a V-phase lower switching element 3e, and a W-phase lower switching element 3f. An inverter main circuit 5 including a plurality of switching elements 3 and a free-wheeling diode 4 connected in parallel with the plurality of switching elements 3, a current detection means 7a for detecting a current for one phase flowing into the synchronous motor 6, and a current detection means Current detection means 7 comprising current detection means 7b for detecting a current of a phase different from 7a, and inverter control for generating PWM signals for on / off control of the plurality of switching elements 3 based on the current detected by the current detection means 7. A plurality of switching elements based on the PWM signal obtained from the means 8 and the inverter control means 8 The gate drive circuit 9 for turning on / off drive, a DC voltage detector 10.

  The inverter control means 8 outputs the output voltage vector command value based on the current signal obtained from the current detection means 7, the rotational speed command value given from the outside, and the excitation current command value obtained from the current phase control means 21. Based on the speed signal, voltage, and current signal used by the vector calculation means 20 and the output voltage vector calculation means, current phase control means 21 for obtaining an excitation current command value for realizing a current phase that maximizes the output torque of the synchronous motor. A modulation rate / phase calculation means 13 for obtaining a modulation rate / phase angle of the output voltage vector based on the DC voltage signal obtained from the DC voltage detection means 10 and the output voltage vector obtained from the output voltage vector calculation means 20, and an output voltage The current signal used in the vector calculation means 20, the output voltage vector calculation means 12 and the rotation speed given from the outside The energization waveform selection means 14 for selecting the energization waveform (energization method and energization width) based on the command value, and the energization width obtained from the output voltage vector modulation rate and energization waveform selection means 14 obtained from the modulation factor / phase calculation means 13 The rectangular wave PWM generation means 15 for obtaining a PWM signal for energizing the rectangular wave based on the command value, the output voltage vector modulation rate obtained from the modulation factor / phase calculation means 13 and the energization width command value obtained from the energization waveform selection means 14 Based on the sine wave PWM generation means 16 for obtaining a PWM signal for sine wave energization based on the energization method signal obtained from the energization waveform selection means 14, the PWM of the rectangular wave PWM generation means 15 and the sine wave PWM generation means 16 is selectively output. PWM output means 17 for performing the above operation.

  Next, the operation of the synchronous motor driving device will be described with reference to FIG. In the figure, an inverter control means 8 is different from the synchronous motor driving device in the first embodiment shown in FIG. Since the operations other than the inverter control means 8 are the same as those shown in FIG. 1, description of these operations is omitted.

The operation of the inverter control means 8 will be described. In the inverter control means 8, the current signals Iu and Iv for two phases detected by the current detection means 7a and 7b are given to the output voltage vector calculation means 20, and the output voltage vector calculation means 20 signals Iu, by calculation Iv and the rotation speed command value omega ^* and the excitation current command value i? ^* more synchronous output voltage vector command value for the rotational speed is driven to meet the rotational speed command value omega ^* of the electric motor 6 Vo ^* Ask.

Further, the current phase control means 21 uses the frequency signal ω1, voltage signals Vγ, Vδ, and current signals Iγ, Iδ, which are variables used in the calculation process in the output voltage vector calculation means 20, to calculate the output torque of the synchronous motor 6. An excitation current command value iγ ^* for realizing the maximum current phase is obtained by calculation and output.

The modulation factor / phase calculation means 13 is a ratio (modulation factor) of the output voltage vector to the DC voltage based on the DC voltage Vdc obtained from the DC voltage detection means 10 and the output voltage vector command value Vo ^* obtained from the output voltage vector calculation means 12. Vk ^* and the phase angle θv of the output voltage vector are obtained.

The energization waveform selection unit 14 includes a γ-axis current Iγ and a δ-axis current Iδ obtained from the output voltage vector calculation unit 12, an output voltage vector modulation rate Vk ^* obtained from the modulation rate / phase calculation unit 13, and a rotational speed command value ω ^*. Then, the most suitable energization method Md and energization width θon for realizing the synchronous motor drive requirement Jk ^* are obtained and output based on the synchronous motor drive requirement Jk ^* given from the outside. Here, the synchronous motor drive requirement condition Jk ^* is given from the host device of the inverter device 2 in the same manner as the rotational speed command value ω ^* . The condition required for driving the synchronous motor is, for example, a high efficiency drive request or a low noise drive request.

The rectangular wave PWM generation means 15 sets the energization width to θon from the output voltage vector modulation rate Vk ^* obtained from the modulation factor / phase calculation means 13, the output voltage vector phase angle θv, and the energization width θon obtained from the energization waveform selection means 14. A rectangular wave energizing PWM signal is generated.

The sine wave PWM generating means 16 generates a PWM signal for energizing a sine wave from the output voltage vector modulation rate Vk ^* obtained from the modulation rate / phase calculation means 13 and the output voltage vector phase angle θv.

  The PWM output means 17 selects and outputs one of the rectangular wave PWM generation means 15 and the sine wave PWM generation means 16 in accordance with the energization method signal Md output by the energization waveform selection means 14.

  An example of the configuration of the output voltage vector calculation means 20 in the second embodiment is shown in FIG. The configuration of the output voltage vector calculation means shown in FIG. 21 is used as a control method for a general-purpose inverter device as an example of the calculation means, so as to keep the magnetic flux on the control axis (γ-δ axis) constant. This shows an example of primary magnetic flux control to be controlled. In general vector control, a three-phase current is converted into an excitation current component (d-axis current) as a magnetic flux direction component of the rotor and a d-q of a torque current component (q-axis current) advanced by 90 degrees from the excitation current component. Convert to coordinates. However, in the primary magnetic flux control shown here, since the exact rotor position is not detected, the dq coordinates cannot be obtained accurately. For this reason, γ-δ coordinates are used as the coordinate system for control as the estimated dq coordinates.

  In FIG. 21, the primary angular frequency ω1 of the voltage vector output from the inverter device 2 is integrated by an integrator 22 to obtain an electrical angle phase θe. Using the electrical angle phase θe and the two-phase current out of the three-phase current flowing into the synchronous motor 6, for example, Iu and Iv, the γ-axis current Iγ and the δ-axis current Iδ on the γ-δ coordinates on the control coordinates are converted. .

The current filter means 24 removes specific frequency components from Iγ and Iδ to obtain respective filter values Iγf and Iδf. The speed compensator 25 receives the δ-axis current filter value Iδf and obtains a speed compensation amount Δω for compensating for the rotational speed command value ω ^* according to the fluctuation of Iδf. The speed error calculation means 26 subtracts the speed compensation amount Δω from the rotational speed command value ω ^* to obtain the primary angular frequency ω1 of the voltage vector.

The voltage command calculation means 27 uses the primary angular frequency ω1, the excitation current command value Iγ ^* , and the γ-δ axis current filter values Iγf and Iδf so that the rotation speed of the synchronous motor 6 satisfies the rotation speed command value ω ^*. The voltage command values Vγ ^* and Vδ ^* on the γ-δ axis for driving are obtained by a control arithmetic expression. The voltage vector calculation means 28 obtains an output voltage vector Vo ^* from the electrical angle phase θe and the voltage command values Vγ ^* and Vδ ^* on the γ-δ axis.

  FIG. 22 shows an example different from FIG. 21 of the configuration of the output voltage vector calculation means 20 in the second embodiment. The configuration of the output voltage vector calculation means 20 shown in FIG. 22 shows general vector control as an example of the calculation means. In general vector control, a three-phase current is converted into an excitation current component (d-axis current) as a magnetic flux direction component of the rotor and a d-q of a torque current component (q-axis current) advanced by 90 degrees from the excitation current component. Convert to coordinates. In FIG. 22 shown here, the dq coordinates are expressed as γ-δ coordinates, but either may be used.

In FIG. 22, the three-phase to two-phase conversion unit 23 converts the currents Iu and Iv for two phases obtained from the current detection unit 7 into currents Iγ and Iδ on the γ-δ coordinates. The current filter unit 24 removes specific frequency components from the currents Iγ and Iδ on the γ-δ coordinates obtained from the three-phase to two-phase conversion unit 23 to obtain γ-δ coordinate current filter values Iγf and Iδf. . The speed comparison means 30 obtains a speed error ωerr from the rotational speed command value ω ^* and the estimated speed value ω ^.

Based on the speed error ωerr obtained from the speed comparison means 30, the speed control means 31 obtains the δ-axis current command value Iδ ^* by proportional / integral control. The γ-axis current comparison means 32 compares the excitation current command value Iγ ^* with the γ-axis current filter value Iγf obtained from the current filter means to obtain a γ-axis current error Iγerr. The δ-axis current comparison means 33 compares the δ-axis current command value Iδ ^* obtained from the speed control means 31 with the δ-axis current filter value Iδf obtained from the current filter means 24 to obtain a δ-axis current error Iδerr. The current control means 34 uses the γ-axis current error Iγerr obtained from the γ-axis current comparison means 31 and the δ-axis current error Iδerr obtained from the δ-axis current comparison means 32 to perform a γ-axis voltage command value Vγ ^* and δ by proportional / integral control. Determine the shaft voltage command value Vδ ^* . The voltage vector calculation means 28 obtains an output voltage vector Vo ^* from the electrical angle phase estimated value θ ^ and the voltage command values Vγ ^* and Vδ ^* on the γ-δ axis.

The speed / position estimation means 35 is a motor of the synchronous motor 6 and the γ-δ-axis voltage command values Vγ ^* and Vδ ^* obtained from the current control means 34 and the γ-δ-axis current filter values Iγf and Iδf obtained from the current filter means 24. Using the constants, the rotor position θ ^ and speed ω ^ of the synchronous motor 6 are estimated. In the speed / position estimation means 35, the rotor speed / position estimation is synchronized with the γ-δ-axis voltage command values Vγ ^* and Vδ ^* and the γ-δ-axis current filter values Iγf and Iδf obtained from the current filter means 24. A method for obtaining a mathematical model of a synchronous motor using a motor constant of the motor 6, a method for using an induced voltage of the synchronous motor 6, a method for obtaining from a response of an applied voltage or current using a saliency of the synchronous motor 6, Although there is a method of obtaining from the neutral point voltage fluctuation of the three-phase winding of the synchronous motor 6, any method may be used.

Considering the case where an IPMSM is used as the synchronous motor 6, the output torque of the IPMSM can be classified into a magnet torque generated by a magnet used for the rotor and a reluctance torque generated by a difference between the d-axis inductance and the q-axis inductance. When the phase of the current vector viewed from the rotor magnetic flux axis (d-axis) is β and the magnitude of the current vector is constant, as shown in FIG. 23, the magnitude of the magnet torque and reluctance torque depends on the phase angle β. Changes, and the output combined torque of the IPMSM changes. The phase angle βm of the current vector that maximizes the combined torque shown in FIG. 23 varies depending on the rotational speed and load torque of the IPMSM. Therefore, the current phase control means 21 controls the current phase β so as to maximize the output torque according to the operating state (rotational speed and load torque) of the synchronous motor 6. At this time, since the current phase β is controlled using the excitation current Iγ, the current phase control means 21 outputs the excitation current command value Iγ ^* .

The current phase control means 21 uses a mathematical model of the synchronous motor 6 to output the primary rotation speed ω1, the voltage command values Vγ ^* and Vδ ^* , the current filter values Iδf and Iγf, which indicate the operating state of the synchronous motor 6. Accordingly, the excitation current command value Iγ ^* is output. Here, the mathematical model may be used as it is, or may be used as a table by simplifying the mathematical model and reducing input variables.

  The case where IPMSM is used as the synchronous motor 6 has been described above. However, when SPMSM is used, since SPMSM does not have reluctance torque, only magnet torque is obtained as output torque. For this reason, in SPMSM, the output torque becomes maximum when the current phase angle β = 0.

Regardless of IPMSM or SPMSM, when the voltage is saturated, there is a weakening excitation control in which the magnetic flux is generated in the direction of canceling the magnet magnetic flux in the d-axis direction and the magnet magnetic flux is weakened. When this control is used, it is necessary to use an excitation current command value different from the excitation current command value Iγ ^* obtained from the mathematical model or table that maximizes the output torque. In such a case, the modulation rate Vk ^* is monitored, and when a voltage saturation state (Vk ^* > 1) is reached, an excitation current command value calculating means for weak excitation control may be used.

  The configuration example of FIGS. 21 and 22 shown as an example of the output voltage vector calculation means 20 and the above-described current phase control means 21 are mainly used in a sine wave energization method. In the present invention, the same control method (output voltage vector calculation means) is used in the sine wave energization method and the rectangular wave energization method, and the control method is configured as follows for one method for sine wave energization.

  When the sine wave is energized, the phase current obtained by the current detection means 7 is converted into γ-δ coordinates by the three-phase to two-phase conversion means 23 if the load of the synchronous motor 6 is in an ideal state without torque ripple. In this case, the γ-δ axis currents Iγ and Iδ are substantially DC. However, when rectangular wave energization is performed, the phase current obtained by the current detection means 7 is converted to γ by the three-phase to two-phase conversion means 23 even in an ideal state where there is no torque ripple in the load of the synchronous motor 6. When converted to -δ coordinates, AC components are generated in the γ-δ axis currents Iγ and Iδ. This is because the voltage waveform output from the inverter device 2 becomes a rectangular waveform due to the rectangular wave energization, and the current waveform flowing into the synchronous motor 6 also becomes a rectangular waveform including a harmonic component.

When an alternating current component is included in Iγ and Iδ, the voltage command calculation means 27 or current control means 34 obtains an output voltage command value using Iγ and Iδ, and therefore is obtained by the voltage command calculation means 27 or current control means 34. The voltage command value also includes an AC component, and the output voltage vector command value Vo ^* obtained from the voltage vector calculation means 28 also includes an AC component. At this time, the control becomes oscillating, and the driving of the synchronous motor 6 may not be continued.

  Further, in the speed / position estimation means 35, the error of the estimated values of the speed and position increases due to the AC component included in Iγ and Iδ, the accuracy of the speed / position information decreases, and the controllability can be lowered. There is sex. Furthermore, the current phase control means 21 is also affected, and the efficiency of the excitation current command value that maximizes the output torque may be reduced, and the control may be vibrated.

In order to avoid the adverse effects on the control due to the AC components of Iγ and Iδ caused by the rectangular wave energization, the AC component that adversely affects the control is removed using the current filter means 24. Thus, the control is stabilized by removing the AC component included in the output voltage vector command value Vo ^* , ensuring the speed / position estimation accuracy, and ensuring and stabilizing the excitation current command value accuracy. When the sine wave energization and the rectangular wave energization of 120 to 180 degrees are compared experimentally, the AC component generated in Iγ and Iδ due to the rectangular wave energization is 6 f or 12 f of the electrical angular frequency (voltage primary angular frequency). Is a multiple of.

  Therefore, in the current filter means 24, a multiple component of 6 of the electrical angular frequency is removed using a band elimination filter in accordance with the electrical angular frequency, so that a current close to the current obtained during sine wave driving can be obtained. Here, when the band elimination filter is used for a plurality of components, the calculation load of the microprocessor is increased, so the components to be removed may be limited. In particular, only the 6f component having a large generation amount may be used. Further, as a simpler filter configuration, a harmonic component having an electrical angle of 6f or more may be removed as a low-pass filter.

In sinusoidal energization, the output voltage vector command value Vo ^* is updated every control cycle Ts. Similarly, the output voltage vector command value Vo ^* may be updated every control cycle Ts even when the rectangular wave is energized. However, when the rectangular wave is energized, only six or twelve voltage vectors corresponding to the phase of the output voltage vector command value Vo ^* are output as described above. For this reason, in the stage-mode condition shown in FIG. 7, in the phase range in which the same vector is output, the phase angle of the output voltage vector is fixed to 6 or 12 voltage vectors, and every control cycle Ts. Only the magnitude of the voltage vector changes depending on the voltage vector command value Vo ^* . Here, in the rectangular wave energization, only the control electrical angle phase is updated, the calculation of the voltage vector command value at the control frequency Fs is stopped, and the calculation is performed once for six or twelve output voltage vectors. An output voltage vector command value may be obtained. At this time, the number of computations of the output voltage vector command value Vo ^{* in} one electrical angle cycle is 6 to 12 regardless of the control cycle Ts.

  The timing of the output voltage vector command value calculation may be performed when the voltage vector is switched when the rectangular wave is energized, that is, when the stage or mode is switched. However, at the time of rectangular wave energization, the current is in a state that changes transiently when the voltage vector is switched, so in a control method that detects current like a control method based on sine wave energization, There is a possibility that a deviation occurs in the control calculation and adversely affects the control. In such a case, current detection is performed after a predetermined time or a predetermined phase has elapsed since the switching of the output voltage vector, and the output voltage vector command value is calculated using the current signal at this time. You should do it. As an example of the predetermined phase, the stage shown in FIG. 7 and the half of the stage-mode phase range may be used.

  In the above-described configuration example of the synchronous motor driving device according to the second embodiment, the current detection unit 7 has been described with respect to the method of directly detecting and obtaining the two-phase current flowing into the synchronous motor 6. As another method, the same can be realized by using a method of detecting a direct current between the direct current power supply unit 1 and the inverter device 2 and converting it to a three-phase current from the relationship with the voltage vector output from the inverter device 2. It is.

  Since it comprised as mentioned above, with one circuit structure, the rectangular wave which can select arbitrary energization widths between 120 degrees and 180 degrees, sine wave energization, and sine wave energization using an overmodulation field A plurality of energization waveforms can be realized. Further, it is possible to switch to the most suitable energization waveform in a short time according to the driving conditions (high efficiency driving condition, low noise driving condition) required for the synchronous motor to be electrically driven. By switching the energization waveform, it is possible to set the total efficiency when the synchronous motor is driven to a predetermined value or more, or to reduce the noise when the synchronous motor is driven to a predetermined value or less. Furthermore, compared with a synchronous motor drive device having only one of sine wave energization or rectangular wave energization, overall efficiency can be improved or noise can be reduced.

  Since the current filter means 24 described above is a method for current distortion that occurs during energization of a rectangular wave, it may be used only during energization of the rectangular wave. Further, there is no particular problem even if the current filter means 24 is used when a sine wave is energized.

  The output voltage vector calculation means 20 includes a current filter means 24 for removing a predetermined frequency component from the excitation current and torque current obtained from the three-phase to two-phase conversion means 23, and the energization waveform selection means 14 selects rectangular wave energization. Since the output voltage vector calculation uses the excitation current and torque current from which the predetermined frequency component obtained from the current filter means 24 is removed, the output voltage vector command value is obtained. It becomes difficult to be affected by the harmonic component of the current generated by the above, and rectangular wave energization can be realized using a control method used in sine wave energization. In addition, this makes it possible to use the same voltage command value calculation means as the control method for driving the synchronous motor regardless of the energization method or the energization width, and the reduction of hardware for detecting the rotor position; Switching processing at the time of switching the energization method and microprocessor processing can be reduced, and the synchronous motor driving device can be simplified, downsized, and reduced in cost.

  Here, as examples of the synchronous motor 6, examples of an embedded magnet type synchronous motor (IPMSM) and a surface magnet type synchronous motor (SPMSM) have been shown, but other than this, a permanent magnet type synchronous motor (PMSM), a brushless The same effect can be obtained with any shape such as a DC motor (BLDCM), a reluctance motor (RM), a synchronous reluctance motor (SyRM), and a switched reluctance motor (SRM).

Embodiment 3 FIG.
FIG. 24 is a diagram showing the third embodiment and shows a configuration example in a refrigeration / refrigerator using a synchronous motor driving device. 24, the refrigerator-freezer 40 includes an input / output device 41, a refrigerator-freezer control means 42, a damper 43, a fan and fan motor 44, a heater 45, a synchronous motor driving device 46, a hermetic compressor 47, and the like. The refrigerant circuit 48 and the temperature sensor 49 are provided. The power source 50 is a power source that supplies power to the refrigerator-freezer 40.

  In FIG. 24, an input / output device 41 is a user interface, and displays the operation state of the refrigerator / freezer such as the temperature display in the cabinet and various operation settings such as the temperature setting in the cabinet, the energy saving mode, and the low noise mode. . The refrigerator-freezer control means 42 controls the entire refrigerator-freezer 40, and the temperature and operating conditions are based on the internal temperature information obtained from the temperature sensor 49 in accordance with the operation setting conditions obtained from the input / output device 41. The operation of each component such as the damper 43, the fan and fan motor 44, the heater 45, and the hermetic compressor 47 is controlled so as to be a predetermined value. Also, various information for display is sent to the input / output device 41.

  The synchronous motor driving device 46 is a portion corresponding to the synchronous motor driving device of the present invention, and is connected to the refrigerant circuit 48 according to the rotational speed command value and the synchronous motor driving condition given from the refrigerator-freezer control means 46. The hermetic compressor 47 is driven. Further, the synchronous motor driving device 46 sends various information such as the operating state and abnormality of the hermetic compressor 47 to the refrigerator-freezer control means 42. Here, a synchronous motor is used for the hermetic compressor 47.

  The refrigerator-freezer control means 42 sends a rotational speed command value and a synchronous motor drive condition to the synchronous motor drive device 46 in accordance with an operation condition signal obtained from the input / output device 41, and causes the hermetic compressor 47 to It is made to drive with the energization waveform according to synchronous motor drive conditions. For example, when the energy saving operation mode is set in the input / output device 41, the refrigerator-freezer control means 42 performs an operation for causing the entire refrigerator-freezer 40 to perform energy-saving operation, and sends a high efficiency request signal to the synchronous motor control means 46. The synchronous motor driving device 46 receives this high efficiency request signal, selects an energization waveform (energization method and energization width) at which the synchronous motor provided in the hermetic compressor 47 becomes highly efficient, and performs high efficiency driving.

  Further, when the low noise operation mode is set in the input / output device 41, the refrigerator-freezer control means 42 performs an operation for operating the entire refrigerator-freezer 40 with low noise, and also sends a low noise request signal to the synchronous motor control means 46. send. The synchronous motor driving device 46 receives this low noise request signal, selects an energization waveform (energization method and energization width) that makes the synchronous motor provided in the hermetic compressor 47 low noise, and performs low noise driving. Under standard conditions, the energy saving operation mode may be selected, and the low noise operation mode may be selected as required.

  Further, the operating condition request signal may be automatically switched between daytime and nighttime by a timer device or a clock provided in the refrigerator-freezer control means 42. In order to reduce power consumption during the day, the energy-saving operation mode is selected. At night, the operation sound of the refrigerator / freezer is more conspicuous than during the day, so the low noise operation mode is selected.

  As described above, by applying the synchronous motor driving device according to the present invention to a refrigerator-freezer, compared to a synchronous motor driving device using either a sine wave energization method or a rectangular wave energization method, the motor efficiency and Overall efficiency including circuit efficiency can be improved, and energy saving of the refrigerator-freezer can be realized. In addition, it is possible to reduce noise compared to a synchronous motor driving device using either the sine wave energization method or the rectangular wave energization method.

  By applying the synchronous motor drive device according to the present invention to a refrigerator-freezer, it is possible to drive a hermetic compressor according to the operation requirement of the user. In addition, it is possible to realize the driving of the hermetic compressor in accordance with the daily use state of the refrigerator-freezer in terms of efficiency and sound.

Embodiment 4 FIG.
FIG. 25 is a diagram showing the fourth embodiment and shows a configuration example of an air conditioner using a synchronous motor driving device. In FIG. 25, a separate type air conditioner including an outdoor unit 51 and an indoor unit 60 is shown. The outdoor unit 51 includes an air conditioner control unit 52, a temperature sensor 53, an outdoor unit fan and fan motor 54, a synchronous motor driving device 55, a hermetic compressor 56, a four-way valve 57, and an outdoor heat exchange. And an expansion valve 59. The indoor unit 60 includes an indoor unit control means 61, a temperature sensor 62, an indoor unit fan and fan motor 63, and an indoor unit heat exchanger 64. The power source 50 is a power source that supplies power to the air conditioner.

  In the air conditioner shown in FIG. 25, the indoor unit control means 61 is connected to the temperature sensor 62 or the air conditioner in the outdoor unit 51 according to the operation state set in the indoor unit control means 61 via a user interface such as a remote controller. Based on the signal obtained from the control means 52, the fan motor and various components of the indoor unit are controlled so as to satisfy the set operation condition (cooling / heating / dehumidifying operation or set temperature). The indoor unit control means 61 sends an operating condition signal to the air conditioner control means 52 in the outdoor unit 51 and also displays the operating state.

  On the outdoor unit 51 side, the outdoor unit fan and fan motor 54 are arranged so as to satisfy the operation condition based on the operation condition signal obtained from the indoor unit control means 61 in the indoor unit 60 and the temperature signal obtained from the temperature sensor 53. The four-way valve 57, the expansion valve 59, and the hermetic compressor 56 are controlled. The synchronous motor driving device 55 is a portion corresponding to the synchronous motor driving device of the present invention, and drives the hermetic compressor 56 according to the rotational speed command value and the synchronous motor driving condition given from the air conditioner control means 52. A synchronous motor is used in the hermetic compressor 56.

  The air conditioner control means 52 sends a rotational speed command value and a synchronous motor drive condition to the synchronous motor drive device 55 in accordance with the drive condition signal obtained from the indoor unit control means 61, and sends the rotational speed command to the hermetic compressor 56. It is made to drive with the energization waveform according to a value and synchronous motor drive conditions.

  For example, when the energy saving operation mode is set in the indoor unit control means 61, the air conditioner control means 52 performs an operation for energy saving operation of the entire air conditioner and sends a high efficiency request signal to the synchronous motor control means 55. send. The synchronous motor driving device 55 receives this high efficiency request signal, selects an energization waveform (energization method and energization width) that can drive the synchronous motor provided in the hermetic compressor 56 with high efficiency, and performs high efficiency operation.

  Further, when the low noise operation mode is set in the indoor unit control means 61, the air conditioner control means 52 performs an operation for low noise operation of the entire air conditioner, and the synchronous motor control means 55 requires a low noise operation. Send a signal. The synchronous motor drive device 55 receives the low noise request signal, selects an energization waveform (energization method and energization width) that makes the synchronous motor provided in the hermetic compressor 56 low noise, and performs low noise driving. Under standard conditions, the energy saving operation mode may be selected, and the low noise operation mode may be selected as required.

  Furthermore, the operating condition request signal may be automatically switched between daytime and nighttime by a timer device or a clock provided in the air conditioner control means 52 or the indoor unit control means 61. In order to reduce power consumption during the day, the energy-saving operation mode is used. At night, the operation sound of the air conditioner is more conspicuous than during the day, so the low-noise operation mode is selected.

  As described above, by applying the synchronous motor driving device according to the present invention to the air conditioner, the motor efficiency is higher than that of the synchronous motor driving device using either the sine wave energization method or the rectangular wave energization method. It is possible to improve the overall efficiency including the circuit efficiency and energy saving of the air conditioner. In addition, it is possible to reduce noise compared to a synchronous motor driving device using either the sine wave energization method or the rectangular wave energization method.

  By applying the synchronous motor driving device according to the present invention to an air conditioner, it is possible to drive a hermetic compressor according to the operation requirement of the user. Further, it becomes possible to realize the driving of the hermetic compressor in accordance with the daily use state of the air conditioner in terms of efficiency and sound.

  In Embodiment 4 shown here, the motor for a compressor of an air conditioner is used as an application example of the synchronous motor drive device, but it can also be applied to a motor for a fan motor. Since both the objects driven by the synchronous motor driving device are synchronous motors, the configuration is the same. However, the driving load is different between the compressor and the fan motor, and the required specifications are also different. In both cases, it is desirable that both high efficiency and low noise can be achieved. However, it is difficult to achieve both, and there are cases in which both specifications are specialized. For compressor applications, efficiency is mainly important, and for fan motors, low noise is mainly important. What is necessary is just to comprise so that an energization waveform may be switched according to this required specification. As an example of switching the energization waveform, in a compressor application that requires efficiency, a rectangular wave energization at low speed and a sine wave energization at high speed may be used. In fan motor applications where low noise is required, a sine wave energization may be used on the low speed side and a rectangular wave energization may be used on the high speed side.

  Examples of utilization of the synchronous motor drive device of the present invention include a washing machine using a synchronous motor, a fan motor drive device using a synchronous motor, a disk drive drive device, a synchronous motor for power, an automobile and an electric machine. There is a drive device for a synchronous motor used in a railway.

FIG. 2 is a diagram illustrating the first embodiment and is a diagram illustrating a configuration of a synchronous motor driving device. It is a figure which shows the relationship between the phase relationship of the output voltage vector in a three-phase coordinate, and the stage for PWM generation. It is a figure which shows Embodiment 1, and is a figure which shows the relationship between the stage of sine wave PWM generation | occurrence | production, and various waveforms. It is a figure which shows Embodiment 1, and is a figure which shows the output voltage vector at the time of 120 degree | times energization. It is a figure which shows Embodiment 1, and is a figure which shows the output voltage vector in the conduction width larger than 120 degree | times and smaller than 180 degree | times. It is a figure which shows Embodiment 1, and is a figure which shows the output voltage vector at the time of 180 degree | times energization. It is a figure which shows Embodiment 1, and is a figure which shows the relationship between the stage in rectangular wave PWM, and various waveforms. It is a figure which shows Embodiment 1, and is a figure which shows the relationship of the output voltage vector at the time of the electricity supply between 2 phases, and the electricity supply between 3 phases. It is a figure which shows ratio of the primary component of the output phase voltage of rectangular wave energization with respect to sine wave energization. It is sectional drawing which shows an example of the structure of IPMSM. It is a figure which shows the electricity supply waveform at the time of IPMSM drive, and the circuit efficiency of an inverter apparatus. It is a figure which shows the relationship between the electricity supply waveform at the time of IPMSM drive, and motor efficiency. It is a figure which shows the relationship between the electricity supply waveform at the time of IPMSM drive, and total efficiency. It is a figure which shows Embodiment 1, and is a figure which shows an example of the energization width selection table of the energization waveform selection means at the time of high efficiency drive. It is a figure which shows Embodiment 1, and is a figure which shows another example of a structure of the electricity supply width selection table of the electricity supply waveform selection means at the time of a highly efficient drive. It is a figure which shows Embodiment 1, and is a figure which shows another example of a structure of the energization width selection table of the energization waveform selection means at the time of a highly efficient drive. It is a figure which shows Embodiment 1, and is a flowchart figure which shows the flow of the energization width selection method of the energization waveform selection means at the time of a highly efficient drive. It is a figure which shows Embodiment 1, and is a figure which shows an example of the energization width selection table of the energization waveform selection means at the time of low noise drive. FIG. 5 is a diagram showing the first embodiment and a flowchart showing an operation flow. It is a figure which shows Embodiment 2, and is a figure which shows the structure of a synchronous motor drive device. It is a figure which shows Embodiment 2, and is a figure which shows an example of the control block of the output voltage vector calculating means of a synchronous motor drive device. It is a figure which shows Embodiment 2, and is a figure which shows another example of the control block of the output voltage vector calculating means of a synchronous motor drive device. It is a figure which shows the relationship between the electric current phase in IPMSM, and an output torque. It is a figure which shows Embodiment 3, and is a figure which shows the structural example of a refrigerator-freezer. It is a figure which shows Embodiment 4, and is a figure which shows the structural example of an air conditioner.

Explanation of symbols

  1 DC power supply unit, 2 inverter device, 3a U-phase upper switching element, 3b V-phase upper switching element, 3c W-phase upper switching element, 3d U-phase lower switching element, 3e V-phase lower switching element, 3f W-phase lower Side switching element, 4 freewheeling diode, 5 inverter main circuit, 6 synchronous motor, 7 current detection means, 8 inverter control means, 9 gate drive circuit, 10 DC voltage detection means, 11 phase current calculation means, 12 output voltage vector calculation means , 13 Modulation rate / phase calculation means, 14 Energized waveform selection means, 15 Rectangular wave PWM generation means, 16 Sine wave PWM generation means, 17 PWM output means, 20 Output voltage vector calculation means, 21 Current phase control means, 22 Integrator , 23 Three-phase to two-phase conversion means 23, 24 Current filter means, 5 Speed compensator, 26 Speed error calculating means, 27 Voltage command calculating means, 28 Voltage vector calculating means, 30 Speed comparing means, 31 Speed control means, 32 γ-axis current comparing means, 33 δ-axis current comparing means, 34 Current control Means, 35 speed / position estimation means, 40 refrigerator, 41 input / output device, 42 refrigerator-freezer control means, 43 damper, 44 fan and fan motor, 45 heater, 46 synchronous motor drive device, 47 hermetic compressor, 48 refrigerant circuit , 49 Temperature sensor, 50 Power supply, 51 Outdoor unit, 52 Air conditioner control means, 53 Temperature sensor, 54 Fan and fan motor for outdoor unit, 55 Synchronous motor drive device, 56 Hermetic compressor, 57 Four-way valve, 58 Outdoor Machine heat exchanger, 59 expansion valve, 60 indoor unit, 61 indoor unit control means, 62 temperature sensor, 63 room Fan for indoor unit and fan motor, 64 indoor unit heat exchanger.

Claims (21)

In the synchronous motor drive device that drives the synchronous motor by the inverter device,
DC voltage detection means for detecting the DC voltage of the DC power supply unit;
An output voltage vector computing means for obtaining an output voltage vector command value based on a current signal of the synchronous motor and a rotational speed command value given from the outside ;
Energization waveform selection means for selecting an energization method and an energization width based on a current signal of the synchronous motor, the output voltage vector calculation means, and a rotational speed command value given from the outside ;
Rectangular wave energization generating means for obtaining a PWM signal for energizing rectangular waves based on an energization width command value obtained from the energization waveform selecting means ;
Sine wave energization means for obtaining a PWM signal for sine wave energization based on an energization width command value obtained from the energization waveform selection means ;
A synchronous motor drive device comprising: a rectangular wave energizing unit and a PWM output unit for selectively outputting a PWM of the sine wave energizing unit based on an energization method signal obtained from the energized waveform selecting unit .   2. The synchronous motor drive device according to claim 1, wherein the sine wave energizing unit energizes in an overmodulated state when the voltage is saturated.   2. The synchronous motor drive apparatus according to claim 1, wherein the condition required for driving the synchronous motor is that the total drive efficiency including the inverter efficiency and the motor efficiency is a predetermined value or more.   2. The synchronous motor drive device according to claim 1, wherein the condition required for driving the synchronous motor is that the noise of the synchronous motor is not more than a predetermined value.   2. The synchronous motor drive device according to claim 1, wherein the rectangular wave energizing unit sets six voltage vectors to be output in one cycle in terms of electrical angle when energizing 120 degrees and energizing 180 degrees.   2. The rectangular wave energizing means, when energizing with an energizing width larger than 120 degrees and smaller than 180 degrees, sets 12 voltage vectors to be output in one cycle in terms of electrical angle. Synchronous motor drive device.   The rectangular wave energization means includes output voltage correction means corresponding to the energization width, and includes a primary component included in the voltage output from the sine wave energization means and a primary component included in the voltage output from the rectangular wave energization means. The synchronous motor drive device according to claim 1, wherein the pulse width of the output PWM is corrected in accordance with the energization width so that.   2. The synchronous motor driving device according to claim 1, wherein the rectangular wave energizing unit sets a PWM on-pulse width at the time of energization between three phases to 1 / √3 at the time of energization between two phases.   The energization waveform selecting means selects an energization waveform using a preset table or function when selecting an energization waveform according to a condition required for driving the synchronous motor. 2. The synchronous motor drive device according to 1.   2. The synchronous motor drive device according to claim 1, wherein the energization waveform selection unit appropriately switches the energization waveform and searches for an energization waveform that minimizes an input current of the inverter device when performing high-efficiency operation. .   When switching the energization method from sine wave energization to rectangular wave energization, the energization waveform selection means changes the energization width at a predetermined ratio so as to become a predetermined energization width after switching from 180-degree rectangular wave energization. The synchronous motor drive device according to claim 1, wherein   The energization waveform selection means is characterized in that when the energization method is switched from rectangular wave energization to sine wave energization, the energization width is changed to a 180-degree rectangular wave energization state at a predetermined rate and then switched to sine wave energization. The synchronous motor drive device according to claim 1.   2. The synchronous motor drive device according to claim 1, wherein when the output voltage approaches a saturation state, the energization waveform selection unit selects a sine wave energization and shifts to an overmodulation operation.   The synchronous motor drive device according to claim 1, wherein the energization waveform selection means selects sinusoidal energization when accelerating or decelerating the synchronous motor.   The synchronous motor drive device according to claim 1, wherein the energization waveform selection means uses sine wave energization when the effective value of the current flowing into the synchronous motor is smaller than a predetermined value.   The output voltage vector calculation means includes a current detection means for detecting a current phase current flowing into the synchronous motor, and a three-phase 2 for converting the phase current obtained by the current detection means into an excitation current component and a torque current component. An energization method selected by the energization waveform selection means, comprising: phase conversion means; and voltage command value calculation means for obtaining an output voltage command value based on the excitation current component and torque current component obtained by the three-phase two-phase conversion means Alternatively, the synchronous motor driving device according to claim 1, wherein the same voltage command value calculation means is used regardless of the energization width.   2. The output voltage vector calculation means is position / speed sensorless control means that does not require a position sensor or a speed sensor for detecting a rotor position or a rotor speed of the synchronous motor. The synchronous motor drive device described in 1.   When the energization waveform selection unit selects rectangular wave energization, the output voltage vector calculation unit is configured to detect current after a predetermined time has elapsed from switching of 6 or 12 output voltage vectors output when energizing the rectangular wave. The synchronous motor drive device according to claim 1, wherein an output voltage vector calculation is performed.   The output voltage vector calculation means includes a current filter means for removing a predetermined frequency component from the excitation current and torque current obtained from the three-phase two-phase conversion means for converting the phase current into an excitation current component and a torque current component, When the energization waveform selection means selects rectangular wave energization, the output voltage calculation means uses the excitation current and torque current obtained by removing the predetermined frequency component obtained from the current filter means to output the output voltage vector command value. The synchronous motor drive device according to claim 1, wherein the synchronous motor drive device is obtained.   The refrigerator-freezer characterized by using the synchronous motor drive device of Claim 1 for the drive means of the synchronous motor with which the hermetic compressor contained in a refrigerant circuit was equipped in the refrigerator refrigerator.   An air conditioner using the synchronous motor drive device according to claim 1 as a drive means of a synchronous motor provided in a hermetic compressor included in a refrigerant circuit.

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