JP3656944B2 - Control method of synchronous motor - Google Patents
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Description
【０００１】
【発明の属する技術分野】
この発明は電圧形インバータにより給電される同期電動機を、磁極位置センサおよび回転速度センサを使用しない、所謂、磁極，速度センサレスベクトル制御に基づいて可変速制御する同期電動機の制御方法に関する。
【０００２】
【従来の技術】
電圧形インバータにより給電される同期電動機の可変速制御方法の第１の従来例としては、該電動機に磁極位置センサおよび回転速度センサを装着し、これらのセンサからの検出信号などに基づくベクトル制御により、該電動機を可変速制御することが行われている。
【０００３】
また上述の制御方法とは異なった第２の従来例としては、前記磁極，速度センサレスベクトル制御を行う際に、前記電動機の回転子位置の推定演算を行い、この推定演算値を単に微分演算し、この演算値を回転速度推定値にしていた。
【０００４】
【発明が解決しようとする課題】
前述の第１の従来例の如く、同期電動機に磁極位置センサおよび回転速度センサを装着すると該電動機の双方の出力軸を負荷に連結する用途に不向きであり、また、磁極位置センサおよび回転速度センサは高価であり、さらに、該電動機の設置場所と前記電圧形インバータなどの駆動装置の設置場所との距離が離れている場合には、それぞれのセンサからの検出信号にノイズが混入するなど、該駆動装置の動作信頼性を阻害する要因となっていた。
【０００５】
また前述の第２の従来例では、先ず最初に同期電動機の回転子位置を推定するが、この推定値は検出信号に混入するノイズによって適正でないことがよくあるので、そのとき、該電動機は脱調現象を起こしてしまう。さらに、この推定値の微分値である回転速度推定値は、検出信号に混入するノイズに対してとても感度が高くなるので、回転速度推定値の誤差は大きくなり、速度制御系はその誤差により、振動的・不安定な応答結果になる。
【０００６】
この発明の目的は上記問題点を解決し、電圧形インバータにより給電される同期電動機を磁極，速度センサレスで好適に可変速制御できる該電動機の制御方法を提供することにある。
【０００７】
【課題を解決するための手段】
この第１の発明は、電圧インバータにより給電される同期電動機であって、該電動機の磁極位置推定値と回転速度推定値とに基づくベクトル制御によって該電動機を可変速制御する同期電動機の制御方法において、
前記ベクトル制御による前記電動機のｄ軸電流指令値およびｑ軸電流指令値に微小振幅の高周波信号を加算した値を該電動機の新たなｄ軸電流指令値およびｑ軸電流指令値とし、この新たなｄ軸電流指令値と前記電動機の電流を座標変換して得られるｄ軸電流検出値との偏差を零にする調節演算を行い、この演算結果を該電動機のｄ軸電圧指令値とし、前記新たなｑ軸電流指令値と前記電動機の電流を座標変換して得られるｑ軸電流検出値との偏差を零にする調節演算を行い、この演算結果を該電動機のｑ軸電圧指令値とし、前記ｄ，ｑ軸それぞれの電流検出値と電圧指令値とに基づく前記高周波信号成分の瞬時無効電力から前記電動機の磁極位置補正値を導出し、前記回転速度推定値を積分演算した値に前記磁極位置補正値を加算した値を前記磁極位置推定値としたことを特徴とする。
【０００８】
第２の発明は前記第１の発明において、前記高周波信号のｄ軸成分とｑ軸成分とは、その振幅が等しく、且つ、互いに９０°の位相差を有する正弦波にしたことを特徴とする。
【０００９】
第３の発明は前記第１又は第２の発明において、前記同期電動機の始動時に、前記磁極位置補正値の導出演算を行わせることを特徴とする。
【００１０】
第４の発明は前記第１又は第２の発明において、前記同期電動機は突極性を有する永久磁石同期電動機とし、該電動機の運転中は、前記磁極位置補正値の導出演算を常時行わせることを特徴とする。
【００１１】
この発明によれば、磁極，速度センサレスのベクトル制御系に注入した微小振幅の高周波信号により、同期電動機の磁極位置推定値を該電動機の始動時を含めて、後述の如く、より真値に近づけることができるので、磁極，速度センサレスで該電動機を好適に可変速制御することができる。
【００１２】
【発明の実施の形態】
図１は、この発明の実施の形態を示す同期電動機の制御装置のブロック構成図である。
【００１３】
図１において、１は三相の電圧指令値（ｖu * ，ｖv * ，ｖw * ）に基づいた所望の周波数，振幅の三相交流電圧（ｖu ，ｖv ，ｖw ）を出力する電圧形インバータ、２は電圧形インバータ２により給電される同期電動機、２ａは同期電動機２の出力軸に連結された負荷、３は電圧形インバータ２を介して同期電動機２を可変速制御する制御装置である。
【００１４】
この制御装置３において、速度調節器３１とｑ軸電流調節器３２とｄ軸電流調節器３３と座標変換器３４と電圧指令発生器３５と電流検出器３６と座標変換器３７と速度オブザーバ３８と積分器３９とは磁極，速度センサレスのベクトル制御に基づく同期電動機２の可変速制御を電圧形インバータ１を介して行う基本的な制御要素であり、さらに、信号発生器４１と磁極位置補正器４２と加算器４３とはこの発明に基づいて付加された制御要素である。
【００１５】
図３に示した制御装置３の動作を以下に説明する。
【００１６】
先ず、速度オブザーバ３８では、同期電動機２の三相電流を電流検出器３６で検出した電流（ｉu ，ｉv ，ｉw ）に座標変換器３７を介してｄ軸電流検出値（ｉd ）とｑ軸電流値（ｉq ）とをそれぞれ３相−２相座標変換した値と、ｑ軸電流調節器３３の出力であるｑ軸電圧指令値（ｖq * ）とを入力して、下記式（１）に示す演算を行い、同期電動機２の回転速度推定値（ωmE）を得ている。
【００１７】
【数１】
ここで、Δｖq ：電圧形インバータ２を構成する半導体素子間のデッドタイムおよび電圧降下に基づく出力電圧誤差のｑ軸変換成分（計算値）
Ｒq ：同期電動機２の抵抗のｑ軸成分
Ｌq ：同期電動機２のリアクタンスのｑ軸成分
Ｌd ：同期電動機２のリアクタンスのｄ軸成分
Φ ：同期電動機２の鎖交磁束数
Ｐ ：同期電動機２の磁極対数
α ：速度オブザーバ３８の時定数の逆数値
ｓ ：ラプラス演算子
である。
【００１９】
速度調節器３１では指令される回転速度指令値（ωm * ）と、上記式（１）で得られた回転速度推定値（ωmE）との偏差を零にする調節演算を行い、この演算値をｑ軸電流指令値（ｉq * ）として出力する。
【００２０】
ｑ軸電流調節器３２では前記ｑ軸電流指令値（ｉq * ）と後述の信号発生器４１からのｑ軸高周波信号（ｉqh）とを加算した値と、前記ｑ軸電流値（ｉq ）との偏差を零にする調節演算を行い、この演算値をｑ軸電圧指令値（ｖq * ）として出力する。
【００２１】
同様に、ｄ軸電流調節器３３では指令されるｄ軸電流指令値（ｉd * ）と後述の信号発生器４１からのｄ軸高周波信号（ｉdh）とを加算した値と、前記ｄ軸電流値（ｉd ）との偏差を零にする調節演算を行い、この演算値をｄ軸電圧指令値（ｖd * ）として出力する。
【００２２】
また、座標変換器３４では前記ｑ軸電圧指令値（ｖq * ）とｄ軸電圧指令値（ｖd * ）とを２相−３相座標変換した三相の電圧設定値（ｖu ref ，ｖv ref ，ｖw ref ）とを出力している。
【００２３】
さらに、電圧指令発生器３５では前記三相の電圧設定値（ｖu ref ，ｖv ref ，ｖw ref ）からＰＷＭ制御された三相の電圧指令値（ｖu * ，ｖv * ，ｖw * を生成し、電圧形インバータ１へ出力している。
【００２４】
次に、積分器３９では下記式（２）の積分演算を行い磁極位置推定値（θE1）を出力している。
【００２５】
【数２】
θE1＝（１／ｓ）ωmE …（２）
なお、マクイロコンピュータによるデジタル制御の場合には、速度オブザーバ３８と積分器３９とにおける演算周期は、ｑ軸電流調節器３２，ｄ軸電流調節器３３の演算周期に等しくすることが好適である。
【００２６】
しかしながら従来の制御方法の如く、上記式（２）で得られた磁極位置推定値（θE1）により座標変換器３４および座標変換器３７の演算を行わせると、同期電動機２の始動時には磁極位置推定値の初期値を必要とし、この初期値が適正でないときにはベクトル制御における座標変換値に誤差を生ずる。
【００２７】
そこで、信号発生器４１から出力するｄ軸高周波信号（ｉdh）とｑ軸高周波信号（ｉqh）とに下記式（３），式（４）の関係を持たせる。
【００２８】
【数３】
ｉdh＝Ｉh ｃｏｓ（ωh ｔ） …（３）
【００２９】
【数４】
ｉqh＝Ｉh ｓｉｎ（ωh ｔ） …（４）
磁極位置補正器４２では、上記式（３），式（４）の関係にあるそれぞれの高周波信号をｑ軸電流調節器３２とｄ軸電流調節器３３とに注入し、その結果、この電流制御系に現れるｑ軸電圧指令値（ｖqh * ）とｄ軸電圧指令値（ｖdh * ）とｑ軸電流検出値（ｉqh）とｄ軸電流検出値（ｖdh * ）とから、下記式（５）に示す高周波信号成分の瞬時無効電力（Ｑh ）を求めている。
【００３０】
【数５】
ここで、ωmE ：回転速度の推定値
ωm * ：回転速度指令値
θ0 ：磁極位置の真値
θE ：磁極位置の推定値
である。
【００３２】
上記式（５）から明らかなように、式（５）の右辺第２項の位相成分（θ0 −θE ）は磁極位置の真値（θ0 ）と推定値（θE ）との誤差（θerr ）に関係した値である。そこで、磁極位置補正器４２に内蔵するバンドパスフィルタなどにより、右辺第２項のみを抽出し、この抽出値と基準になるＩh 2 ｃｏｓ（ωh ｔ）とを比較することにより２（θ0 −θE ）を求めることができ、従って、磁極位置補正値θerr （＝θ0 −θE ）が演算できる。
【００３３】
すなわち、加算器４３の加算値（θE2）は、上記式（５）の推定値（θE ）を積分器３９の演算値（θE1）とすることにより、磁極位置の真値（θ0 ）により近い値となる。
【００３４】
なお、信号発生器４１からの高周波信号の周波数は、電圧指令発生器３５におけるＰＷＭ演算のキャリア周波数の１／５〜１／１０程度が望ましい。
【００３５】
さらに、マクイロコンピュータによるデジタル制御の場合には、磁極位置補正器４２と加算器４３とで得られる今回の磁極位置推定値（θE2）を、座標変換器３４および座標変換器３７での次回の座標変換演算の際に使用するとよい。
【００３６】
【発明の効果】
この発明によれば、磁極，速度センサレスのベクトル制御系に注入した微小振幅の高周波信号により、同期電動機の磁極位置推定値を該電動機の始動時を含めて、より真値に近づけることができるので、磁極，速度センサレスで該電動機を零速から定格速度域まで好適に可変速制御することができる。
【００３７】
特に、突極性を有する永久磁石同期電動機の如く、その電気的パラメータが変動する同期電動機に対して、この発明の制御方法による磁極位置推定値は前記電気的パラメータの変動に不感であるので、ロバストに該電動機を可変速制御することができる。
【図面の簡単な説明】
【図１】この発明の実施の形態を示す同期電動機の制御装置のブロック構成図
【符号の説明】
１…電圧形インバータ、２…同期電動機、２ａ…負荷、３…制御装置、３１…速度調節器、３２…ｑ軸電流調節器、３３…ｄ軸電流調節器、３４…座標変換器、３５…電圧指令発生器、３６…電流検出器、３７…座標変換器、３８…速度オブザーバ、３９…積分器、４１…信号発生器、４２…磁極位置補正器、４３…加算器。[0001]
BACKGROUND OF THE INVENTION
The present invention relates to a control method for a synchronous motor in which a synchronous motor fed by a voltage source inverter is controlled at a variable speed based on so-called magnetic pole and speed sensorless vector control without using a magnetic pole position sensor and a rotational speed sensor.
[0002]
[Prior art]
As a first conventional example of a variable speed control method for a synchronous motor fed by a voltage source inverter, a magnetic pole position sensor and a rotational speed sensor are mounted on the motor, and vector control based on detection signals from these sensors is used. The electric motor is subjected to variable speed control.
[0003]
Further, as a second conventional example different from the above-described control method, when performing the magnetic pole and speed sensorless vector control, an estimation calculation of the rotor position of the electric motor is performed, and this estimated calculation value is simply differentiated. The calculated value was used as the estimated rotational speed.
[0004]
[Problems to be solved by the invention]
If the magnetic pole position sensor and the rotational speed sensor are mounted on the synchronous motor as in the first conventional example described above, it is not suitable for use in connecting both output shafts of the motor to a load, and the magnetic pole position sensor and the rotational speed sensor. Is expensive, and further, when the installation location of the electric motor and the installation location of the drive device such as the voltage source inverter are separated, noise is mixed in the detection signal from each sensor, etc. This is a factor that hinders the operational reliability of the driving device.
[0005]
In the second conventional example, the rotor position of the synchronous motor is first estimated, but this estimated value is often not appropriate due to noise mixed in the detection signal. Cause a tonal phenomenon. Furthermore, since the rotational speed estimated value, which is a differential value of this estimated value, is very sensitive to noise mixed in the detection signal, the error of the rotational speed estimated value becomes large, and the speed control system has the error, Response results are oscillating and unstable.
[0006]
SUMMARY OF THE INVENTION An object of the present invention is to solve the above-mentioned problems and to provide a method for controlling a motor that can suitably control a synchronous motor fed by a voltage source inverter without a magnetic pole and a speed sensor.
[0007]
[Means for Solving the Problems]
The first aspect of the present invention is a synchronous motor fed by a voltage inverter, wherein the motor is variable-speed controlled by vector control based on the magnetic pole position estimated value and the rotational speed estimated value of the motor. ,
A value obtained by adding a high-frequency signal having a small amplitude to the d-axis current command value and q-axis current command value of the motor by the vector control is set as a new d-axis current command value and q-axis current command value of the motor. An adjustment calculation is performed so that the deviation between the d-axis current command value and the d-axis current detection value obtained by coordinate conversion of the current of the motor is zero, and the calculation result is used as the d-axis voltage command value of the motor. Adjustment calculation is performed so that the deviation between the q-axis current command value and the q-axis current detection value obtained by coordinate conversion of the current of the motor is zero, and the calculation result is used as the q-axis voltage command value of the motor. The magnetic pole position correction value of the motor is derived from the instantaneous reactive power of the high-frequency signal component based on the current detection value and the voltage command value for each of the d and q axes, and the magnetic pole position is obtained by integrating the rotational speed estimated value. Add the correction value to And characterized in that a serial magnetic pole position estimation value.
[0008]
According to a second invention, in the first invention, the d-axis component and the q-axis component of the high-frequency signal are sine waves having the same amplitude and a phase difference of 90 ° from each other. .
[0009]
According to a third aspect, in the first or second aspect, the magnetic pole position correction value is derived when the synchronous motor is started.
[0010]
According to a fourth aspect of the present invention, in the first or second aspect of the invention, the synchronous motor is a permanent magnet synchronous motor having a saliency, and the derivation calculation of the magnetic pole position correction value is always performed during operation of the motor. Features.
[0011]
According to the present invention, the estimated magnetic pole position of the synchronous motor is brought closer to the true value as will be described later, including when the motor is started, by the high-frequency signal having a small amplitude injected into the magnetic pole and speed sensorless vector control system. Therefore, the motor can be suitably controlled at a variable speed without a magnetic pole and a speed sensor.
[0012]
DETAILED DESCRIPTION OF THE INVENTION
FIG. 1 is a block diagram of a control apparatus for a synchronous motor showing an embodiment of the present invention.
[0013]
In FIG. 1, 1 outputs a three-phase AC voltage (v u , v v , v w ) having a desired frequency and amplitude based on a three-phase voltage command value (v u * , v v * , v w * ). 2 is a synchronous motor fed by the
[0014]
In this
[0015]
The operation of the
[0016]
First, the speed observer 38 detects a three-phase current of the
[0017]
[Expression 1]
Here, Δv q : q-axis conversion component (calculated value) of output voltage error based on dead time and voltage drop between semiconductor elements constituting
R q : q-axis component L q of the resistance of the
[0019]
The speed adjuster 31 performs an adjustment operation to make the deviation between the commanded rotational speed command value (ω m * ) and the estimated rotational speed value (ω mE ) obtained by the above equation (1) zero. The value is output as the q-axis current command value (i q * ).
[0020]
The q-axis current regulator 32 adds a value obtained by adding the q-axis current command value (i q * ) and a q-axis high-frequency signal (i qh ) from the signal generator 41 described later, and the q-axis current value (i q ) Is adjusted to zero, and the calculated value is output as a q-axis voltage command value (v q * ).
[0021]
Similarly, the d-axis current regulator 33 adds a commanded d-axis current command value (i d * ) and a d-axis high-frequency signal (i dh ) from the signal generator 41 described later, and the d-axis. performs an adjustment operation for the deviation between the current value (i d) to zero, and outputs the calculated value d-axis voltage command value as (v d *).
[0022]
In the coordinate converter 34, a three-phase voltage setting value (v u ref ,) obtained by two-phase to three-phase coordinate conversion of the q-axis voltage command value (v q * ) and the d-axis voltage command value (v d * ). v v ref , v w ref ) are output.
[0023]
Further, in the voltage command generator 35, three-phase voltage command values (v u * , v v * , v w ) PWM-controlled from the three-phase voltage setting values (v u ref , v v ref , v w ref ). * Is generated and output to the
[0024]
Next, the
[0025]
[Expression 2]
θ E1 = (1 / s) ω mE (2)
In the case of digital control by a Maciro computer, it is preferable that the calculation cycle of the speed observer 38 and the
[0026]
However, if the calculation of the coordinate converter 34 and the coordinate
[0027]
In view of this, the d-axis high-frequency signal (i dh ) and the q-axis high-frequency signal (i qh ) output from the signal generator 41 have a relationship of the following expressions (3) and (4).
[0028]
[Equation 3]
i dh = I h cos (ω h t) (3)
[0029]
[Expression 4]
i qh = I h sin (ω h t) (4)
In the magnetic pole position corrector 42, the respective high frequency signals having the relations of the above formulas (3) and (4) are injected into the q-axis current regulator 32 and the d-axis current regulator 33. As a result, this current control is performed. From the q-axis voltage command value (v qh * ), d-axis voltage command value (v dh * ), q-axis current detection value (i qh ), and d-axis current detection value (v dh * ) appearing in the system, The instantaneous reactive power (Q h ) of the high frequency signal component shown in (5) is obtained.
[0030]
[Equation 5]
Here, ω mE : Estimated value of rotational speed ω m * : Rotational speed command value θ 0 : True value of magnetic pole position θ E : Estimated value of magnetic pole position.
[0032]
As is clear from the above equation (5), the phase component (θ 0 −θ E ) of the second term on the right side of equation (5) is the error between the true value (θ 0 ) and the estimated value (θ E ) of the magnetic pole position. It is a value related to (θ err ). Therefore, only the second term on the right side is extracted by a bandpass filter or the like built in the magnetic pole position corrector 42, and this extracted value is compared with the reference I h 2 cos (ω h t) to obtain 2 (θ 0− θ E ) can be obtained, and therefore the magnetic pole position correction value θ err (= θ 0 −θ E ) can be calculated.
[0033]
That is, the added value (θ E2 ) of the adder 43 is obtained by setting the estimated value (θ E ) of the above equation (5) as the calculated value (θ E1 ) of the
[0034]
The frequency of the high frequency signal from the signal generator 41 is preferably about 1/5 to 1/10 of the carrier frequency of the PWM calculation in the voltage command generator 35.
[0035]
Furthermore, in the case of digital control by a Maciro computer, the current magnetic pole position estimated value (θ E2 ) obtained by the magnetic pole position corrector 42 and the adder 43 is used for the next time in the coordinate converter 34 and the coordinate
[0036]
【The invention's effect】
According to the present invention, the estimated magnetic pole position value of the synchronous motor can be made closer to the true value including when the motor is started, by using a high-frequency signal with a small amplitude injected into the magnetic pole and speed sensorless vector control system. The motor can be suitably controlled at a variable speed from zero speed to a rated speed range without a magnetic pole and a speed sensor.
[0037]
In particular, for a synchronous motor whose electrical parameters fluctuate, such as a permanent magnet synchronous motor having a saliency, the magnetic pole position estimation value according to the control method of the present invention is insensitive to the fluctuations of the electric parameters. In addition, the electric motor can be controlled at a variable speed.
[Brief description of the drawings]
FIG. 1 is a block diagram of a synchronous motor control apparatus showing an embodiment of the present invention.
DESCRIPTION OF
Claims (4)
前記ベクトル制御による前記電動機のｄ軸電流指令値およびｑ軸電流指令値に微小振幅の高周波信号を加算した値を該電動機の新たなｄ軸電流指令値およびｑ軸電流指令値とし、
この新たなｄ軸電流指令値と前記電動機の電流を座標変換して得られるｄ軸電流検出値との偏差を零にする調節演算を行い、この演算結果を該電動機のｄ軸電圧指令値とし、
前記新たなｑ軸電流指令値と前記電動機の電流を座標変換して得られるｑ軸電流検出値との偏差を零にする調節演算を行い、この演算結果を該電動機のｑ軸電圧指令値とし、
前記ｄ，ｑ軸それぞれの電流検出値と電圧指令値とに基づく前記高周波信号成分の瞬時無効電力から前記電動機の磁極位置補正値を導出し、
前記回転速度推定値を積分演算した値に前記磁極位置補正値を加算した値を前記磁極位置推定値としたことを特徴とする同期電動機の制御方法。A synchronous motor fed by a voltage inverter, wherein the motor is variable-speed controlled by vector control based on a magnetic pole position estimated value and a rotational speed estimated value of the motor.
A value obtained by adding a high-frequency signal having a minute amplitude to the d-axis current command value and q-axis current command value of the motor by the vector control is set as a new d-axis current command value and q-axis current command value of the motor,
An adjustment calculation is performed so that the deviation between the new d-axis current command value and the d-axis current detection value obtained by coordinate conversion of the current of the motor is zero, and this calculation result is used as the d-axis voltage command value of the motor. ,
An adjustment calculation is performed so that the deviation between the new q-axis current command value and the q-axis current detection value obtained by coordinate conversion of the current of the motor is zero, and this calculation result is used as the q-axis voltage command value of the motor. ,
Deriving the magnetic pole position correction value of the motor from the instantaneous reactive power of the high-frequency signal component based on the current detection value and the voltage command value for each of the d and q axes,
A method for controlling a synchronous motor, wherein a value obtained by adding the magnetic pole position correction value to a value obtained by integrating the rotational speed estimated value is used as the magnetic pole position estimated value.
前記高周波信号のｄ軸成分とｑ軸成分とは、その振幅が等しく、且つ、互いに９０°の位相差を有する正弦波にしたことを特徴とする同期電動機の制御方法。In the control method of the synchronous motor according to claim 1,
A control method for a synchronous motor, wherein the d-axis component and the q-axis component of the high-frequency signal are sine waves having the same amplitude and a phase difference of 90 ° from each other.
前記同期電動機の始動時に、前記磁極位置補正値の導出演算を行わせることを特徴とする同期電動機の制御方法。In the synchronous motor control method according to claim 1 or 2,
A control method for a synchronous motor, wherein a calculation for deriving the magnetic pole position correction value is performed when the synchronous motor is started.
前記同期電動機は突極性を有する永久磁石同期電動機とし、
該電動機の運転中は、前記磁極位置補正値の導出演算を常時行わせることを特徴とする同期電動機の制御方法。In the synchronous motor control method according to claim 1 or 2,
The synchronous motor is a permanent magnet synchronous motor having saliency,
A method of controlling a synchronous motor, wherein the derivation calculation of the magnetic pole position correction value is always performed during operation of the motor.
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| JP26772299A JP3656944B2 (en) | 1999-09-21 | 1999-09-21 | Control method of synchronous motor |
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| JP5148789B2 (en) * | 2001-07-25 | 2013-02-20 | オリエンタルモーター株式会社 | Identification method of flux linkage and electrical time constant of permanent magnet synchronous motor |
| US7161324B1 (en) | 2003-07-16 | 2007-01-09 | Mitsubishi Denki Kabushiki Kaisha | Device for estimating pole position of synchronous motor |
| JP4592385B2 (en) * | 2004-10-27 | 2010-12-01 | 株式会社東芝 | Control device for synchronous machine |
| KR100645807B1 (en) * | 2004-12-06 | 2007-02-28 | 엘지전자 주식회사 | Apparatus and method for startup synchronous reluctance motor |
| JP5176420B2 (en) | 2007-08-02 | 2013-04-03 | 株式会社ジェイテクト | Sensorless control device for brushless motor |
| JP5194886B2 (en) * | 2008-03-03 | 2013-05-08 | 株式会社明電舎 | Variable speed drive for synchronous motor |
| KR20140116728A (en) * | 2013-03-25 | 2014-10-06 | 엘지전자 주식회사 | Apparatus and method for initially driving a sensorless bldc motor |
| WO2016038992A1 (en) * | 2014-09-12 | 2016-03-17 | 三菱電機株式会社 | Ac rotating machine control device and method for calculating magnetic pole position correction amount |
| JP6390649B2 (en) * | 2016-03-18 | 2018-09-19 | 株式会社安川電機 | Power converter, motor power estimation method and motor control method |
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