JP2672797B2 - Ultrasonic transducer drive circuit - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to an ultrasonic transducer driving circuit.

[0002]

2. Description of the Related Art Conventionally, various devices using an ultrasonic transducer have been proposed, and for example, ultrasonic scalpels for surgical use, ultrasonic cleaners, ultrasonic processing devices, etc. are known. The ultrasonic transducer used in such an ultrasonic device is preferably driven at its resonance frequency in order to increase efficiency,
The resonance frequency varies depending on load conditions and temperature.

From the above, conventionally, for example, the phase difference between the drive voltage and the drive current of the ultrasonic transducer or the vibration velocity detection signal is detected, and the drive frequency of the ultrasonic transducer is controlled based on the output. A so-called phase-locked loop (PLL) type resonance point tracking circuit has been proposed.

Here, as shown in the equivalent circuit of the piezoelectric vibrator 1 in FIG. 9, the ultrasonic transducer is connected in series with a resistor R, an inductor L and a capacitor C which are connected in series. In addition, in actual use, in order to cancel the braking capacity Cd,
The correction inductor Ld is connected in parallel.

In this case, the frequency characteristic of the phase difference θ between the driving voltage and the driving current of the vibrator 1 is as shown in FIG. 10A, and the frequency characteristic of the impedance | Z |
As shown in B. That is, the phase difference θ is equal to the resonance point f.
_r and the antiresonance point f ₁ before and after, f ₂ at zero, and the impedance | Z | is minimum at the resonance point f _r, antiresonance point f
It becomes maximum at ₁ and f ₂ . Therefore, by using the above-described PLL type resonance point tracking circuit, as is clear from FIGS. 10A and 10B, the resonance point can be effectively tracked between the anti-resonance points f ₁ and f ₂ .

[0006]

However, this is not the case.
When driving such an ultrasonic transducer, for example, start oscillation
Driving frequency is f₁Less than or f_TwoIf the above
Feedback control of the loop is not effective and tracking becomes impossible, and the drive frequency is
f₁Less than or f_TwoIt becomes the above value. This phenomenon
Is the resonance frequency of the ultrasonic transducer, not only at the start of oscillation.
However, for example, it may fluctuate greatly depending on load conditions and temperature,
If an abnormality occurs in the ultrasonic transducer, f ₁To f_TwoCharacteristics
When an abnormality occurs in, for example, the resonance point f_rNo longer exists
It also occurs when there is. Furthermore, under the above circumstances,
Even if there is a resonance point, its frequency value is outside the specified range.
May be. In this case, the correct ultrasonic transducer
If the driving frequency is outside the specified range as if it were a swing point,
The wave converter continues to be supplied.

As a device capable of preventing driving at a point other than the correct resonance point, for example, Japanese Patent Laid-Open No. 60-34776.
In Japanese Patent Laid-Open No. 62-140686 and Japanese Patent Laid-Open No. 62-140686, the output frequency of a voltage controlled oscillator for driving an ultrasonic transducer is swept by an external signal from a D / A converter before entering lock-in. , The transducer current data for each frequency step is obtained and stored, then the stored transducer current data is searched to determine the basic resonance point of the ultrasonic transducer, and then the output frequency of the voltage controlled oscillator is Locking in to the frequency of the determined basic resonance point and switching to PLL control, and in the PLL control, the PLL
It is disclosed that the output of a phase comparator of a control loop is monitored by a window comparator, and when the output exceeds a set range, driving of an ultrasonic transducer is stopped. However, in these conventional drive circuits, after the output frequency of the voltage controlled oscillator is locked in to the frequency of the basic resonance point and the tracking of the resonance point by the PLL is started,
Since the drive stop of the ultrasonic transducer is controlled based on the output of the phase comparator of the L control loop, before entering the lock-in, it is determined whether or not there is an abnormality in the ultrasonic transducer. Cannot be detected. Thus, for example,
When there is an abnormality in the mechanical vibration system due to cracks in the ultrasonic transducer, the resonance frequency other than the correct resonance point is searched by sweeping the drive frequency by the external signal from the D / A converter, and the frequency is searched. The ultrasonic transducer is driven by being locked in, and this not only significantly reduces the driving efficiency of the ultrasonic transducer, but if driving in that state continues, the drive circuit is configured by mismatch. There is a problem that the electronic parts etc. to be burned out will be burned out.

The present invention has been made by paying attention to such a conventional problem, and detects an abnormality of an ultrasonic transducer in a resonance point tracking state before lock-in and after lock-in. It is an object of the present invention to provide an ultrasonic transducer drive circuit that is appropriately configured so that its drive can be automatically stopped, and therefore burnout of components due to misalignment can be effectively prevented.

[0009]

In order to achieve the above object, an ultrasonic transducer driving circuit according to the present invention includes a feedback signal generating means for generating two feedback signals based on a driving signal of the ultrasonic transducer. A reference signal generating means for generating a reference signal having a variable frequency, a signal switching means for receiving one of the two feedback signals and the reference signal and selectively outputting one of them, and the signal switching means. And a phase comparator that receives the other of the two feedback signals and compares their phases, and forms a phase-locked loop with this phase comparator, and based on the output of the phase comparator, An oscillator that generates a signal for driving the ultrasonic transducer, and a frequency of the drive signal that is supplied to the ultrasonic transducer based on the signal generated from the oscillator are the ultrasonic waves. Resonance point detecting means for detecting whether or not the resonance frequency of the transducer is substantially equal, switching control means for controlling the signal switching means based on the output of the resonance point detecting means, and driving start of the ultrasonic transducer In the ultrasonic transducer drive circuit having switch means for instructing, the reference signal from the reference signal generating means is supplied to the phase comparator through the signal switching means based on a start instruction from the switch means. Then, the frequency of the drive signal supplied to the ultrasonic transducer is changed according to the change of the frequency of the reference signal by a phase-locked loop, and the frequency of the drive signal is substantially equal to the resonance frequency. On the basis of the fact that the resonance point detection means has detected equality, the switching control means controls the signal switching means, One of the feedback signals is supplied to the phase comparator, and when the frequency of the drive signal is not detected by the resonance point detecting means, the frequency of the drive signal is changed by the phase lock loop to the frequency of the reference signal. If the frequency of the drive signal is not detected by the resonance point detecting means even if the process of changing the frequency is restarted and the process is repeated a predetermined number of times, the drive of the ultrasonic transducer is controlled to be stopped. It is characterized by being configured.

[0010]

According to the present invention, one of the two feedback signals based on the drive signal of the ultrasonic transducer and the reference signal from the reference signal generating means, the frequency of which changes, form a phase-locked loop comparator. Is supplied to the ultrasonic transducer, and a process of changing the frequency of the drive signal of the ultrasonic transducer according to the change of the frequency of the reference signal is performed by the phase lock loop. Here, when the frequency of the drive signal is detected by the resonance point detecting means, another one of the two feedback signals is supplied to the phase comparator instead of the reference signal, and the phase comparison of these two feedback signals is performed. The control shifts to resonance point tracking control based on. When the frequency of the drive signal is not detected by the resonance point detecting means, the process of changing the frequency of the drive signal according to the change of the frequency of the reference signal by the phase lock loop is restarted, and this process is repeated a predetermined number of times. Even if the frequency of the driving signal is not detected by the resonance point detecting means, the driving of the ultrasonic transducer is stopped.

Therefore, the abnormality of the ultrasonic transducer can be detected and the driving thereof can be automatically stopped both before and after the lock-in and the resonance point tracking state after the lock-in. It is possible to effectively prevent the burnout and the like. In addition, it is preferable to provide an alarm generating means for generating an alarm in synchronization with the driving of the ultrasonic transducer in order to notify the abnormality of the ultrasonic transducer.

[0012]

FIG. 1 shows an embodiment of the present invention. This example is applied to an ultrasonic scalpel device,
Langevin type vibrator 3 provided on the handpiece 35
6 is driven via the matching transformer 38 based on the output of the PLL 37. The vibrator 36 is connected to the secondary side of the matching transformer 38, and the correction inductor 39 for canceling the braking capacitance of the vibrator 36 is connected in parallel to the secondary side of the matching transformer 38.

The PLL 37 includes a phase comparator (PC) 40.
And a charge pump 41 for converting its digital output into an analog signal, a loop filter 42, and a voltage controlled oscillator (VCO) 43.
The output of 1 is supplied as a control voltage to the VCO 43 via the loop filter 42.

The output of the VCO 43 is supplied to the filter 44, divided by the frequency dividing circuit 45, and supplied to the filter 44. As a result, the rectangular-wave drive signal output from the VCO 43 is supplied to the resonance frequency component of the oscillator 36. It is converted into a drive signal of only a sine wave so as to prevent unnecessary heat generation in the vibrator 36. In this example, a switched capacitor filter (SCF) whose cutoff frequency can be changed by an external clock input is used as the filter 44. As described above, when the SCF is used as the filter 44, even if the oscillation frequency of the VCO 43 changes, the size of the filter output waveform and the fluctuation of the phase rotation are eliminated, and as a result, the influence on the constant current control and the PLL operation described later is exerted. It is possible to reduce an ideal rectangular wave-sinusoidal wave conversion.

The output of the filter 44 is a voltage control amplifier circuit (VCA) 46 and a buffer amplifier 4 whose gain can be changed.
7, through the switch circuit 48 and the power amplifier 49, and supplied to the primary side of the matching transformer 38. In this way, the matching transformer 38 separates the circuit system of the vibrator 36 from the drive circuit system for electrical insulation, and at the same time, matches the power amplifier 49 with the vibrator 36 serving as a load. To

The voltage applied to the vibrator 36 via the power amplifier 49 and the current flowing in the vibrator 36 are detected by the voltage / current detection circuit 50 provided on the primary side of the matching transformer 38, and the voltage detection signal and the current are detected. The signals are supplied to the differential amplifiers 51-1 and 51-2, respectively, so that common mode noise is removed.

FIG. 2 shows an example of the configuration of the voltage / current detection circuit 50 and the differential amplifiers 51-1 and 51-2. The voltage applied to the oscillator 36 is detected by voltage division by the resistor 52, and the output is supplied to the differential amplifier 51-1 to obtain the voltage detection signal V. Also, the oscillator 3
The current flowing through 6 is detected by the current sensor 53, and its output is supplied to the differential amplifier 51-2 to obtain the current detection signal I.

As described above, the voltage / current detected by the voltage / current detection circuit 50 is supplied to the differential amplifiers 51-1, 5 respectively.
1-2 to supply voltage detection signal V and current detection signal I
Thus, it is possible to effectively solve the problem of common-mode noise in the case of detecting a high voltage and a large current at a low voltage, and even if the positive and negative connections of the output of the power amplifier 49 are reversed,
Further, even if one of the outputs is not of the grounded output type, the voltage and current signals V and I can be stably detected.

In FIG. 1, the voltage detection signal obtained from the differential amplifier 51-1 is supplied to a comparator 54 and an absolute value detection circuit 55, respectively, and the voltage phase signal θ _V is supplied to the absolute value detection circuit in the comparator 54. At 55, the absolute value | V | of the voltage detection signal is detected. Similarly, the current detection signal obtained from the differential amplifier 51-2 is the comparator 56.
And the absolute value detection circuit 57, and the comparator 5
6, the current phase signal θ _I is detected, and the absolute value detection circuit 57 detects the absolute value | I | of the current detection signal.

The voltage phase signal θ _V obtained from the comparator 54
Is supplied to the phase comparator 58 and the variable input terminal V of the PC 40 constituting the PLL 37. The current phase signal θ _I obtained from the comparator 56 is supplied to the phase comparator 5
8 and the PC4 via the switch circuit 59
The reference input terminal R of 0 is supplied. Further, the absolute value | V | of the voltage detection signal obtained from the absolute value detection circuit 55.
Is supplied to the voltage comparator 60 and compared with a predetermined set value,
The output is supplied to the phase comparator 58.

In this way, the phase comparator 58 applies the output to the oscillator 36 based on the output of the voltage comparator 60, the voltage phase signal θ _V from the comparator 54 and the current phase signal θ _I from the comparator 56. Detecting whether the frequency of the driving signal being applied is substantially equal to the resonance frequency of the vibrator 36,
Based on the output, the switching operation of the current phase signal θ _I in the switch circuit 59 and the reference signal θ _ref from the oscillator to be described later is controlled, and at the same time, the lighting of the light emitting diode 61 is controlled so that the PLL 37 performs the resonance point tracking operation. Display whether or not the transition has been made.

FIG. 3 shows an example of the configuration of the phase comparator 58 and the voltage comparator 60. Phase comparator 58
Are three D-flip-flops (D-FF) 62, 6
3 and 64 and an OR gate 65.
The voltage phase signal θ _V obtained from the comparator 54 is D-FF
The current phase signal θ _I obtained from the comparator 56 is supplied to the D input terminal of 62, and is supplied to the clock input terminal of the D-FF 62. The Q output and inverted Q output of the D-FF 62 are the clock input terminal of the D-FF 63 and the D-FF 6 respectively.
4 to the clock input terminals, and these DF
The Q outputs of F63 and 64 are supplied to the OR gate 65 so that the control signals of the switch circuit 59 and the light emitting diode 61 are obtained. In addition, D-FF 63 and 64
V _CC is applied to the D input terminal of.

Further, the voltage comparator 60 is composed of an OP amplifier, the inverting input terminal thereof is supplied with the absolute value | V | of the voltage detection signal obtained from the absolute value detection circuit 55, and the non-inverting input terminal thereof is supplied with the set voltage V. _{Apply SET} and set its output to D-
It is supplied to the clear terminals of the FFs 63 and 64. In addition, O
The output of the R gate 65 is supplied to the control circuit 66 (see FIG. 1), and the control circuit 66 outputs D
-A reset signal is supplied to the clear terminal of FF62.

In FIG. 1, the absolute value | I | of the current detection signal obtained from the absolute value detection circuit 57 is the differential amplifier circuit 6
7 is supplied to the inverting input terminal. The setting signal from the current value setting circuit 68 is supplied to the non-inverting input terminal of the differential amplifier circuit 67, and the oscillator 36 is always driven by a constant current corresponding to the setting signal based on the output thereof. , The amplification factor of the VCA 46 is controlled via the limiter circuit 69. The current value setting circuit 68 is provided with an output amplitude setting variable resistor 70 and a low constant current drive setting variable resistor 71, which are selected on the basis of a signal from the control circuit 66, and at the time of startup. The output of the low constant current drive setting variable resistor 71 and the output of the output amplitude setting variable resistor 70 are supplied to the differential amplifier circuit 67 in the resonance point tracking operation.

Thus, the absolute value | I | of the current detection signal obtained from the absolute value detection circuit 57 and the current value setting circuit 6
8 is compared with the setting signal from 8 by the differential amplifier circuit 67, and the amplification factor of the VCA 46 is controlled based on the output to control the signal voltage input to the buffer amplifier 47 and the power amplifier 49. The vibrator 36 can always be driven with a constant current corresponding to the setting signal from the current value setting circuit 68 even if the impedance changes due to a load change or the like, and the amplitude of the handpiece 35 can be kept constant.

A trigger output circuit 72 is connected to the control circuit 66 so that a trigger signal is generated under the control of the control circuit 66. This trigger signal is supplied to the generator 73 to generate a sawtooth wave, and this is supplied to the oscillator 74 so that the oscillator 74 generates the reference signal θ _ref whose frequency changes.
This reference signal θ _ref is _supplied to the phase comparator 58 as described above.
Under the control of the above, it is supplied to the R input terminal of the PC 40 which constitutes the PLL 37 via the switch circuit 59.

4 to 7 show the switch circuit 5 shown in FIG.
9 shows four examples. The switch circuit 59 shown in FIG. 4 supplies the current phase signal θ _I and the reference signal θ _ref to the 3-state buffers 25-1 and 25-2, respectively, and the phase comparator 58 to the control terminal of the 3-state buffer 25-1. Is directly supplied to the control terminal of the 3-state buffer 25-2 and the output of the phase comparator 58 is supplied via the inverter 26. When the output of the phase comparator 58 is at the high (H) level, For example, current phase signal θ
_{When I} is at the low (L) level, the reference signal θ
_{The ref} is supplied to the R input terminal of the PC 40 in the PLL 37.

The switch circuit 59 shown in FIG. 5 supplies the current phase signal θ _I and the reference signal θ _ref to one input terminals of the AND gates 27-1 and 27-2, respectively, and at the same time, the AND gate 27-1. The output of the phase comparator 58 is directly supplied to the other input terminal of the AND gate 27-2.
When the output of the phase comparator 58 is at H level, for example, when the current phase signal θ _I is at L level, the output of the phase comparator 58 is supplied to the other input terminal of The reference signal θ _ref is _supplied to the OR gate 2 respectively.
The signal is supplied to the R input terminal of the PC 40 in the PLL 37 via the line 9.

The switch circuit 59 shown in FIG. 6 supplies the current phase signal θ _I and the reference signal θ _ref to the analog switches 30-1 and 30-2, respectively, and a phase comparator is connected to the control terminal of the analog switch 30-1. The output of the phase comparator 58 is directly supplied to the control terminal of the analog switch 30-2 through the inverter 28, and when the output of the phase comparator 58 is at the H level, for example, the current phase The reference signal θ _ref is supplied to the R input terminals of the PC 40 in the PLL 37 when the signal θ _I is at the L level.

The switch circuit 59 shown in FIG. 7 outputs the current phase signal θ _I and the reference signal θ _ref to the relay contacts 3 respectively.
When the output of the phase comparator 58 is supplied to the relay 32 and the output of the phase comparator 58 is at the H level, for example, the relay 32 is energized to supply the current phase. Relay 3 when signal θ _I is at L level
2 is turned off and the reference signal θ _ref is supplied to the R input terminals of the PC 40 in the PLL 37.

In FIG. 1, the trigger signal from the trigger output circuit 72 is also supplied to a counter 75 for counting, and when the count value exceeds a set value, the switch circuit 48 is turned off and the handpiece 35 is turned off. As an abnormal condition, the probe check light emitting diode 76 is turned on. The counter 75 is reset by a reset signal from the control circuit 66.

On the other hand, the output of the loop filter 42 constituting the PLL 37 is also supplied to the low pass filter (LPF) 77 so that spike-like noise contained in the control voltage of the VCO 43 is removed. This LPF77
Is supplied to the window comparator 78, where the output frequency range of the VCO 43 is monitored, and when it is out of a predetermined range, a reset signal is output to the control circuit 66.

That is, the VCO 43 is the loop filter 4
The oscillation frequency changes depending on the control voltage from 2.
When the LL 37 deviates from the resonance point tracking control of the vibrator 36,
The oscillation frequency of the VCO 43 saturates at its maximum or minimum oscillation frequency. Therefore, in order to detect this unlocked state, the control voltage of the VCO 43 is monitored by the window comparator 78.

Here, the control voltage supplied to the VCO 43 becomes a signal smoothed to some extent by the loop filter 42. For example, the PC 40 of the PLL 37 is of the edge trigger type and the loop characteristics are set to the loop filter. If the design of 42 makes the speed to some extent, V
Spike-like noise leaks into the control voltage supplied to the CO 43 at the comparison part of the edges of the two input signals of the PC 40. Since this spike-like noise adversely affects the operation of the window comparator 78, the LPF 77 is inserted in this example as described above to remove the spike-like noise.

A foot switch 79 for controlling ON / OFF of the vibrator 36 is connected to the control circuit 66, and the signal from the foot switch 79, the signal from the phase comparator 58 described above, and the window comparator 7 are connected.
Based on the reset signal from 8, the operation of each of the above parts is controlled, the suction unit 80 for removing the tissue excised by the ultrasonic scalpel and the probe of the handpiece 35 are cooled, and the excision site is The operation of the water supply unit 81 for flushing is controlled.

The operation of this ultrasonic scalpel device will be described below with reference to FIG.
This will be described with reference to the flowchart shown in FIG. When the foot switch 79 is OFF, the switch circuit 48 is O
The FF and switch circuit 59 outputs the output of the oscillator 74 to the PLL.
It is connected to the R input terminal of the PC 40 of 37. When the foot switch 79 is turned on in this state, the control circuit 66 receives a start signal,
The phase comparator 58 and the counter 75 are reset, and the low constant current drive setting variable resistor 71 of the current value setting circuit 68 is selected and its output is supplied to the differential amplifier circuit 67. Further, the switch circuit 48 is turned on and the trigger output circuit 72 is operated to generate a trigger signal from the trigger output circuit 72. Thereby, PL
The reference signal θ _ref whose frequency changes according to the output of the generator 73 is supplied to the L37 from the oscillator 74 via the switch circuit 59, and as a result, the PLL 37 sets the drive signal frequency of the oscillator 36 to the reference signal θ _ref. Scan to lock to.

Here, the vibrator 36 is the current value setting circuit 6
Since the driving is controlled by the low constant current set by the variable resistor 71 of No. 8, the voltage applied to the vibrator 36 is proportional to the magnitude of its impedance, and therefore the voltage applied to the vibrator 36 is equal to the magnitude of its impedance. It changes by scanning in a manner similar to the frequency characteristic. This change in drive voltage is caused by the voltage / current detection circuit 50 and the differential amplifier 51-
1. Monitored by the voltage comparator 60 via the absolute value detection circuit 55, which is equal to or lower than the set voltage V _SET (see FIG. 3), that is, the frequency of the reference signal θ _ref is close to the resonance frequency of the handpiece 35 and the impedance. When is less than or equal to the set value, the enable signal is output to the phase comparator 58.

In the phase comparator 58, the voltage comparator 6
The enable signal is output from 0, and the comparator 54 outputs
Voltage phase signal θ_VAnd the current phase signal from the comparator 56
Number θ _IWhen the phase difference of becomes 0, its output becomes H level.
And the switch circuit 59 is held.
The current phase signal θ from the comparator 56._IIs P
It is supplied to the R input terminal of C40, and the light emitting diode
LED 61 lights up to indicate that resonance point tracking operation has started
Is done.

At the same time, based on the output of the phase comparator 58, the current value setting circuit 68 is passed through the control circuit 66.
Output amplitude setting variable resistor 70 is selected, and its output is supplied to the differential amplifier circuit 67. Therefore, thereafter, the handpiece 35 is drive-controlled with a predetermined current set by the variable resistor 70 so that the voltage phase signal θ _V and the current phase signal θ _I always have the same phase. Further, by shifting to the resonance point tracking operation, a drive signal is supplied from the control circuit 66 to the suction unit 80 and the water supply unit 81, and each operation is performed. The above operation is canceled by turning off the foot switch 79.

On the other hand, when the resonance point is not detected by one scan, the oscillation frequency of the VCO 43 rises or falls while being locked by the reference signal θ _ref from the oscillator 74, and the VCO 43 rises or falls from the predetermined frequency range in the window comparator 78. When the disconnection is detected, the control circuit 6
A reset signal is output to 6. As a result, a signal is sent from the control circuit 66 to the trigger output circuit 72 so as to output a retrigger, and the above operation is repeated.

The number of times the trigger signal is output from the trigger output circuit 72, that is, the number of scans of the drive signal frequency is counted by the counter 75, and when it reaches a predetermined value, one foot switch 79 is used in this example. ON
If the drive signal frequency cannot be locked in to the resonance frequency even if the drive signal frequency is scanned 10 times, it is assumed that the resonance point of the handpiece 35 does not exist within the predetermined frequency range, and at that time, the output of the counter 75 causes the switch circuit 48 to switch. Is turned off, the driving of the vibrator 36 is stopped, and the light emitting diode 76 is turned on to display the abnormality of the handpiece 35. As a result, it is possible to effectively prevent the risk of the handpiece 35 continuing to be driven in an abnormal state.

Further, even after shifting to the resonance point tracking operation, V
The control voltage of the CO 43 is monitored by the window comparator 78, and when it is detected that it is out of the predetermined range, a reset signal is output to the control circuit 66, whereby the lock-in operation described above is repeated. That is, when the reset signal is output from the window comparator 78 after the resonance point tracking operation, the phase comparator 58 and the counter 75 are reset, and the low constant current drive setting variable resistor 71 of the current value setting circuit 68 is reset.
Is selected, the trigger output circuit 72 is activated, and the reference signal θ _ref is output from the oscillator 74 via the generator 73.
Is output, the reference signal θ _ref is supplied to the PLL 37 via the switch circuit 59, and the lock-in operation is repeated in the same manner as described above.

According to the above ultrasonic scalpel device, when the resonance lock-in cannot be performed, or when the resonance point tracking becomes impossible after the resonance driving, that is, when the driving frequency is out of the resonance point tracking range, Since the driving of the vibrator 36 is stopped and the light emitting diode 76 is turned on, it is possible to effectively prevent burning of electronic components and the like due to mismatching, and it is also possible to detect an abnormality of the handpiece 35.

By combining a general constant current drive circuit, a constant amplitude operation can be performed, and the frequency characteristic of impedance can be detected by a simple method, so that the resonance point can be accurately and surely detected. Can be detected. Further, since the voltage applied to the vibrator 36 and the current flowing through the vibrator 36 are detected by the differential method, common-mode noise can be effectively removed, and a desired voltage can be obtained regardless of the output format of the power amplifier 49. And the current can be detected effectively, which makes it possible to float the circuit part around the power amplifier that is generating a high voltage from the ground, greatly reducing the leakage current to ground of the oscillator circuit, that is, the patient circuit. it can.

In the above ultrasonic scalpel device,
The voltage applied to the vibrator 36 and the current flowing through the vibrator 36 are detected on the primary side of the matching transformer 38.
It is also possible to detect on the secondary side of 8. Also,
The frequency range of the reference signal θ _ref generated from the oscillator 74 includes the frequency range corresponding to the setting range of the control voltage in the window comparator 78, and is slightly wider than the frequency range, and the anti-resonance point of the vibrator 36 is further increased. As the range not included, undesired lock-in at the anti-resonance point can be more surely prevented.

The present invention is not limited to the ultrasonic scalpel device described above, but is effectively applied to a driving circuit of an ultrasonic transducer used in other ultrasonic devices such as an ultrasonic processing device and an ultrasonic cleaning device. can do.

Appendix 1. An ultrasonic scalpel device having a handpiece including an ultrasonic transducer and a probe that is fastened to this handpiece and transmits ultrasonic vibration to a surgical site, the ultrasonic transducer, tracking its resonance point. Driving means for driving at the resonance frequency, monitoring means for monitoring whether or not the frequency for driving the ultrasonic transducer by the driving means is within a predetermined range, and the driving frequency is controlled by the monitoring means. When it is detected that it is out of the range, the driving means restarts the driving of the ultrasonic transducer at a driving frequency in a range including the frequency in the predetermined range, and even if this operation is repeated a predetermined number of times, 1. An ultrasonic scalpel device comprising: a control unit that controls the driving of the ultrasonic transducer to be stopped when the driving frequency deviates from the predetermined range. An ultrasonic scalpel device having a handpiece including an ultrasonic transducer and a probe that is fastened to this handpiece and transmits ultrasonic vibration to a surgical site, the ultrasonic transducer, tracking its resonance point. It has a drive means for driving at a resonance frequency and a phase comparison means for comparing the phase of the drive voltage and the current of the ultrasonic converter by this drive means, and based on the output of this phase comparison means, the ultrasonic wave An ultrasonic scalpel device configured to stop driving of the ultrasonic transducer when the transducer is not in a resonance state. 3. 3. The ultrasonic scalpel device according to appendix 1 or 2, further comprising alarm generating means for generating an alarm in synchronization with the driving of the ultrasonic transducer being stopped.

[0048]

As described above, according to the present invention, one of the two feedback signals based on the drive signal of the ultrasonic transducer and the reference signal whose frequency changes from the reference signal generating means are It is supplied to the phase comparator that composes the phase-locked loop, and the frequency of the drive signal of the ultrasonic transducer is changed according to the change of the frequency of the reference signal by the phase-locked loop, and the frequency of the drive signal resonates. When the resonance point detecting means detects that the frequency is substantially equal to the frequency, another one of the two feedback signals is supplied to the phase comparator in place of the reference signal to shift to resonance point tracking control.
If the frequency of the drive signal is not detected by the resonance point detecting means, the above process is restarted, and if the frequency of the drive signal is not detected by the resonance point detecting means even after repeating this process a predetermined number of times, the ultrasonic transducer Since the drive is stopped, it is possible to detect the abnormality and automatically stop the drive both before and after entering lock-in and at the resonance point tracking state after lock-in. It is possible to effectively prevent the burnout and the like.

[Brief description of the drawings]

FIG. 1 is a block diagram showing one embodiment of the present invention.

FIG. 2 is a diagram showing a configuration of an example of a voltage / current detection circuit portion shown in FIG.

FIG. 3 is a diagram showing a configuration of an example of a phase comparator and a voltage comparator shown in FIG.

4 is a diagram showing a configuration of an example of a switch circuit shown in FIG.

FIG. 5 is a diagram similarly showing the configuration of another example of the switch circuit.

FIG. 6 is a diagram similarly showing the configuration of another example of the switch circuit.

FIG. 7 is a diagram similarly showing the configuration of still another example of the switch circuit.

FIG. 8 is a flowchart for explaining the operation of FIG.

FIG. 9 is a diagram showing an equivalent circuit of a piezoelectric vibrator.

10 is a diagram showing frequency characteristics of phase difference between driving voltage and current and frequency characteristics of impedance of the vibrator shown in FIG.

[Explanation of symbols]

 35 Handpiece 36 Transducer 37 PLL 38 Matching Transformer 39 Correction Inductor 40 Phase Comparator (PC) 41 Charge Pump 42 Loop Filter 43 Voltage Controlled Oscillator (VCO) 44 Filter 45 Frequency Divider 46 Voltage Controlled Amplifier (VCA) 47 Buffer amplifier 48 Switch circuit 49 Power amplifier 50 Voltage / current detection circuit 51-1, 51-2 Differential amplifier 54,56 Comparator 55,57 Absolute value detection circuit 58 Phase comparator 59 Switch circuit 60 Voltage comparator 61,76 Light emitting diode 66 Control circuit 67 Differential amplifier circuit 68 Current value setting circuit 69 Limiter circuit 70 Variable resistor for output amplitude setting 71 Variable resistor for low constant current drive setting 72 Trigger output circuit 73 Generator 74 Oscillator 75 Counter 77 Low pass Filter (LPF) 78 Wind comparator 79 Foot switch 80 Suction unit 81 Water supply unit

Claims (1)

(57) [Claims] 1. A feedback signal generation means for generating two feedback signals based on a drive signal of an ultrasonic transducer, a reference signal generation means for generating a reference signal whose frequency changes, and one of the two feedback signals. And the reference signal,
A signal switching means for selectively outputting one of them, a phase comparator for receiving the signal output from the signal switching means and the other of the two feedback signals and comparing their phases, and this phase comparison And an oscillator for generating a signal for driving the ultrasonic transducer based on the output of the phase comparator, and the ultrasonic transducer based on the signal generated from the oscillator. Resonance point detecting means for detecting whether or not the frequency of the drive signal supplied to the ultrasonic transducer is substantially equal to the resonance frequency of the ultrasonic transducer, and the signal switching means is controlled based on the output of the resonance point detecting means. In the ultrasonic transducer drive circuit having switching control means for controlling the ultrasonic transducer and switch means for instructing to start driving the ultrasonic transducer, a start finger from the switch means is provided. On the basis of, the reference signal from the reference signal generating means, is supplied to the phase comparator through the signal switching means, the frequency of the drive signal supplied to the ultrasonic transducer, by the phase lock loop A process of changing the frequency of the reference signal is performed, and based on the fact that the resonance point detecting means detects that the frequency of the drive signal is substantially equal to the resonance frequency, the switching control means performs the operation. The signal switching means is controlled to supply one of the two feedback signals to the phase comparator, and when the frequency of the drive signal is not detected by the resonance point detecting means, the frequency of the drive signal is phase locked. The process of changing the frequency of the reference signal by a loop is restarted, and even if this process is repeated a predetermined number of times, the frequency of the drive signal is changed. Wherein when not detected by the resonance point detector, ultrasonic transducer drive circuit which is characterized by being configured such that the controls to stop driving the ultrasonic transducer.