JP2007208748A - Ofdm demodulator, ofdm demodulation method, program and computer readable recording medium - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide an OFDM demodulator with which a state of a received signal is recognized without time lag by high responsiveness. <P>SOLUTION: The OFDM demodulator 1 has: a digital quadrature demodulation circuit 3 which performs quadrature demodulation of an intermediate frequency signal received by an antenna 13 and frequency-converted from a high frequency signal by a tuner 14 to generate a baseband signal; a narrow band carrier frequency error correction circuit 4 which corrects a narrow band carrier frequency error of the baseband signal; an FFT operation circuit 5 which applies an FFT operation to the baseband signal whose narrow band carrier frequency error has been corrected; and an SNR calculation circuit 6 which calculates SNR based on the baseband signal whose narrow carrier frequency error has been corrected by the narrow band carrier frequency error correction circuit 4. <P>COPYRIGHT: (C)2007,JPO&INPIT

Description

  The present invention relates to an orthogonal frequency division multiplexing (OFDM) demodulation apparatus, an OFDM demodulation method, a program, and a computer readable medium that can efficiently transmit a video signal and an audio signal by a digital transmission method. The present invention relates to various recording media.

(OFDM broadcast)
In terrestrial digital broadcasting, a multicarrier OFDM modulation / demodulation method is known as a modulation method suitable for overcoming ghost interference (fading, multipath) by buildings. The OFDM modulation / demodulation method is a digital modulation / demodulation method in which a large number (about 256 to 1024) of sub-carriers are provided in one channel band and video signals and audio signals can be efficiently transmitted. A baseband (BB) signal in which all the carriers are OFDM-modulated by an inverse fast Fourier transform (IFFT) is generated.

  FIG. 10 is a diagram illustrating an example of a transmission symbol of an OFDM modulated wave. The period of the IFFT conversion processing window is an effective symbol period ts. The effective symbol period corresponds to Fs clock N cycles. The sum of all digitally modulated carriers with the effective symbol period ts as a basic unit is called an OFDM transmission symbol.

  As shown in FIG. 10, an actual transmission symbol is usually configured by adding a period tg called a guard interval (GI) 202a to an effective symbol period 201. The waveform of the GI period tg (202a) is obtained by repeating the signal waveform of the rear part 202b of the effective symbol period ts. The symbol period 203 of the transmission symbol is the sum of the effective symbol period 201 and the GI period 202a. For example, according to the broadcasting standard of Non-Patent Document 1, the effective symbol period length is defined as shown in the following table 1 by a parameter called MODE.

  Further, the GI period (unit: μs) is defined as shown in the following Table 2 by a parameter called GI period length (GI ratio) which is a ratio to each effective symbol period length.

  A collection of several transmission symbols is called a transmission frame. This is a collection of about 100 information transmission symbols plus frame synchronization symbols and service identification symbols. For example, in Non-Patent Document 1, one frame is defined as 204 symbols.

  Further, according to Non-Patent Document 1, a carrier shown in the following table 3 is arranged for one segment in one transmission symbol subjected to QPSK, 16QAM, or 64QAM modulation.

  In this table, SP means an SP (Scattered Pilot) signal. This SP signal is a pilot signal periodically inserted. For example, the SP signal is inserted once in 12 carriers in the carrier direction and once in 4 symbols in the symbol direction. TMCC means TMCC (Transmission and Multiplexing Configuration Control) signal. This TMCC signal is a signal for transmitting a frame synchronization signal and a transmission parameter. AC1 means an AC1 (Auxiliary Channel) signal. This AC1 signal is a signal for transmitting additional information. Unlike SP, TMCC and AC1 are arranged aperiodically in each carrier.

(Basic configuration of conventional OFDM demodulator)
A configuration example of a conventional OFDM demodulator is shown in Non-Patent Document 2, for example. Therefore, the OFDM demodulator disclosed in Non-Patent Document 2 will be described below.

  FIG. 11 is a block diagram showing a configuration of a conventional OFDM demodulator 900. As shown in FIG. 11, the OFDM demodulator 900 includes an antenna 83, a tuner 84, a bandpass filter (BPF) 903, an A / D conversion circuit 904, a DC cancellation circuit 905, a baseband signal processing unit 92, and an error correction processing unit. 85, a transport stream generation circuit 919, and an RS decoding circuit 920.

  The baseband signal processing unit 92 includes a digital orthogonal demodulation circuit 93, a narrowband carrier frequency error correction circuit 94, a narrowband carrier frequency error detection circuit 97, an FFT operation circuit 95, a waveform equalization circuit 910, and a MER calculation circuit 96. Yes.

  The error correction processing unit 85 includes a frequency deinterleave circuit 911, a time deinterleave circuit 912, a demapping circuit 913, a bit deinterleave circuit 914, a depuncture circuit 915, a Viterbi circuit 916, a byte deinterleave circuit 917, and a spread signal removal circuit 918. Contains.

  A broadcast wave of a digital broadcast broadcast from a broadcast station is received by the antenna 83 of the OFDM demodulator 900 and supplied to the tuner 84 as an RF signal. The tuner 84 includes a multiplier 902a and a local oscillator 902b, and converts the frequency of the RF signal received through the antenna 83 into an IF signal. The tuner 84 supplies the frequency-converted IF signal to the BPF 903.

  The oscillation frequency of the reception carrier signal oscillated from the local oscillator 902b is switched according to a channel selection signal supplied from a channel selection circuit (not shown). The IF signal output from the tuner 84 is filtered by the BPF 903 and then digitized by the A / D conversion circuit 904. The digitized IF signal has its DC component removed by the DC cancellation circuit 905 and is supplied to the digital quadrature demodulation circuit 93.

  The digital orthogonal demodulation circuit 93 performs orthogonal demodulation on the digitized IF signal using a carrier signal having a predetermined frequency (carrier frequency), and outputs a baseband OFDM signal. As a result of orthogonal demodulation, the baseband OFDM signal becomes a complex signal composed of a real axis component (I channel signal) and an imaginary axis component (Q channel signal). The baseband OFDM signal output from the digital quadrature demodulation circuit 93 is supplied to the narrowband carrier frequency error correction circuit 94.

  The narrowband carrier frequency error correction circuit 94 corrects the narrowband carrier frequency error of the baseband OFDM signal supplied from the digital quadrature demodulation circuit 93 and supplies it to the narrowband carrier frequency error detection circuit 97 and the FFT operation circuit 95. To do.

  The narrowband carrier frequency error detection circuit 97 detects a narrowband carrier frequency error of the baseband OFDM signal and supplies a control signal for correcting this to the narrowband carrier frequency error correction circuit 94.

  The FFT operation circuit 95 performs an FFT operation on the baseband OFDM signal with the center frequency error corrected, and extracts and outputs a signal that is orthogonally modulated on each subcarrier. The FFT operation circuit 95 extracts an effective symbol length signal from one OFDM symbol, and performs an FFT operation on the extracted signal. That is, the FFT operation circuit 95 removes a signal corresponding to the guard interval length from one OFDM symbol, and performs an FFT operation on the remaining signal. The range of the signal extracted for performing the FFT operation may be an arbitrary position of one OFDM transmission symbol as long as the extracted signal points are continuous. That is, the start position of the extracted signal range is any position during the GI period. The signal modulated by each subcarrier extracted by the FFT operation circuit 95 is a complex signal composed of a real axis component (I channel signal) and an imaginary axis component (Q channel signal). The signal extracted by the FFT operation circuit 95 is supplied to the waveform equivalent circuit 910.

  The waveform equivalent circuit 910 is supplied with a signal demodulated from each subcarrier output from the FFT operation circuit 95. The waveform equivalent circuit 910 detects a pilot signal, performs waveform equalization based on the pilot signal, and performs carrier demodulation on the signal. In the case of demodulating an ISDB-T standard OFDM signal, the waveform equivalent circuit 910 performs, for example, DQPSK differential demodulation or synchronous demodulation such as QPSK, 16QAM, and 64QAM. The signal demodulated by the carrier by the waveform equivalent circuit 910 is supplied to the MER calculation circuit 96 and the frequency deinterleave circuit 911.

  Here, there is a MER (Modulation Error Ratio) as an index representing a noise component included in a signal. The MER is a value defined as a ratio between the power conversion value of the vector error from the ideal constellation point and the power of the ideal constellation point in the demodulated constellation. Therefore, since the MER needs to be calculated based on the output after waveform equalization, the MER calculation circuit 96 is provided at the subsequent stage of the waveform equalization circuit 910.

  The signal demodulated by the carrier by the waveform equivalent circuit 910 is deinterleaved in the frequency direction by the frequency deinterleave circuit 911. Subsequently, the time deinterleave circuit 912 performs deinterleave processing in the time direction and then supplies the demapper circuit 913.

  The demapping circuit 913 performs data reassignment processing (demapping processing) on the carrier demodulated signal (complex signal). Thereby, the transmission data series is restored. For example, in the case of demodulating an ISDB-T standard OFDM signal, the demapping circuit 913 performs demapping processing corresponding to QPSK, 16QAM, or 64QAM.

  The transmission data series output from the demapping circuit 913 passes through the bit deinterleave circuit 914, the depuncture circuit 915, the Viterbi circuit 916, the byte deinterleave circuit 917, and the spread signal removal circuit 918. This enables deinterleaving processing corresponding to bit interleaving for error dispersion of multilevel symbols, depuncturing processing corresponding to puncturing processing for reducing transmission bits, and decoding of convolutionally encoded bit strings. Energy despreading processing corresponding to Viterbi decoding processing, deinterleaving processing in byte units, and energy diffusion processing is performed. Thereafter, the transmission data sequence is input to the transport stream generation circuit 919.

  For example, the transport stream generation circuit 919 inserts data defined by each broadcasting system such as a null packet at a predetermined position in the stream. In addition, the transport stream generation circuit 919 performs a so-called smoothing process in which the bit interval of the intermittently supplied stream is smoothed to obtain a temporally continuous stream. The smoothed transmission data sequence is supplied to the RS decoding circuit 920.

The RS decoding circuit 920 performs a Reed-Solomon decoding process on the input transmission data sequence. Thereby, it outputs as a transport stream prescribed | regulated by MPEG-2 systems.
JP 2004-266747 A (published on September 24, 2004 (2004.24.24)) Published Japanese Patent Laying-Open No. 2005-229207 (August 25, 2005 (2005. 8.25)) "Transmission method for digital terrestrial television broadcasting ARIB STD-B31 1.5 edition", the radio industry, the first edition of May 31, 2001, revised version 1.5 on July 29, 2003 "Digital Terrestrial Audio Broadcasting Demodulator Standard (desired specification) ARIB STD-B30 version 1.2", Radio Industry, first edition of May 31, 2001, July 29, 2003 version 1.2 Revision VLSI algorithm for arithmetic operations, Naofumi Takagi, Corona, 2005

  However, in the above-described conventional configuration shown in FIG. 11, the MER is calculated by storing an OFDM baseband signal for one symbol and then performing an FFT operation so that a delay of one symbol period or more is generated. After the waveform equalization by 95, it takes a delay time of one symbol or more to calculate the MER, and there is a problem that the responsiveness for calculating the noise component included in the received signal is poor. .

  The present invention has been made in view of the above problems, and an object of the present invention is to realize an OFDM demodulator that can grasp the state of a received signal without time lag due to high responsiveness.

  In order to solve the above problem, an OFDM demodulator according to the present invention includes a digital quadrature demodulator that generates a baseband signal by orthogonally demodulating an intermediate frequency signal received from an antenna and frequency-converted from a high frequency signal by a tuner. A narrowband carrier frequency error correction circuit that corrects a narrowband carrier frequency error of the baseband signal, an FFT operation circuit that performs an FFT operation on the baseband signal that has been corrected for the narrowband carrier frequency error, and the narrowband And an SNR calculation circuit that calculates an SNR based on a baseband signal whose narrowband carrier frequency error is corrected by the band carrier frequency error correction circuit.

  According to the above feature, since the SNR (Signal To Noise Ratio) is calculated based on the baseband signal before the FFT operation is performed by the FFT operation circuit, the state of the received signal can be grasped without delay due to the FFT operation. As a result, it is possible to provide an OFDM demodulator capable of grasping the state of a received signal without time lag due to high responsiveness.

  The OFDM demodulator according to the present invention further comprises a narrowband carrier frequency error detection circuit for detecting a narrowband carrier frequency error from the baseband signal with the narrowband carrier frequency error corrected, and the narrowband carrier frequency error detection circuit Includes a complex correlation circuit that calculates a complex correlation of the baseband signal to generate a complex correlation signal, and the SNR calculation circuit calculates the SNR based on the complex correlation signal generated by the complex correlation circuit It is preferable.

  According to the above configuration, the SNR is calculated based on the complex correlation signal generated by the complex correlation circuit provided in the narrowband carrier frequency error detection circuit. Therefore, the variance of the narrowband carrier frequency error can be obtained based on the complex correlation signal, and the SNR can be estimated based on the variance of the narrowband carrier frequency error.

  In the OFDM demodulator according to the present invention, the narrowband carrier frequency error detection circuit includes a complex correlation circuit that calculates a complex correlation of the baseband signal and generates a complex correlation signal. A section average circuit that generates a section average signal based on the complex correlation signal, and the SNR calculation circuit calculates the SNR based on the section average signal generated by the section average circuit.

  According to the above configuration, the SNR is calculated based on the section average signal generated by the section average circuit provided in the narrowband carrier frequency error detection circuit. Therefore, the variance of the narrowband carrier frequency error can be obtained based on the section average signal, and the SNR can be estimated based on the variance of the narrowband carrier frequency error.

  In the OFDM demodulator according to the present invention, the narrowband carrier frequency error detection circuit calculates a complex correlation of the baseband signal to generate a complex correlation signal, and an interval average based on the complex correlation signal A section average circuit for generating a signal and a phase shift amount circuit for generating a phase shift amount based on the section average signal, wherein the SNR calculation circuit is based on the phase shift amount generated by the phase shift amount circuit. It is preferable to calculate the SNR.

  According to the above configuration, the SNR is calculated based on the phase shift amount generated by the phase shift amount circuit provided in the narrowband carrier frequency error detection circuit. Therefore, the variance of the narrow band carrier frequency error can be obtained based on the phase shift amount, and the SNR can be estimated based on the variance of the narrow band carrier frequency error.

  In the OFDM demodulator according to the present invention, the narrowband carrier frequency error detection circuit calculates a complex correlation of the baseband signal to generate a complex correlation signal, and an interval average based on the complex correlation signal A section average circuit that generates a signal, a phase shift amount circuit that generates a phase shift amount based on the section average signal, and an integration circuit that integrates the phase shift amount in symbol units, and the SNR calculation circuit includes: It is preferable to calculate the SNR based on an integration value integrated by the integration circuit.

  According to the above configuration, the SNR is calculated based on the integration value integrated by the integration circuit provided in the narrowband carrier frequency error detection circuit. Therefore, the variance of the narrowband carrier frequency error can be obtained based on the integral value, and the SNR can be estimated based on the variance of the narrowband carrier frequency error.

  In the OFDM demodulator according to the present invention, the narrowband carrier frequency error detection circuit calculates a complex correlation of the baseband signal to generate a complex correlation signal, and an interval average based on the complex correlation signal An interval average circuit that generates a signal, a phase shift amount circuit that generates a phase shift amount based on the interval average signal, an integration circuit that integrates the phase shift amount in symbol units, and an integration value by the integration circuit is sampled It is preferable that the SNR calculation circuit calculates the SNR based on the integration value integrated by the sampling unit integration circuit.

  According to the above configuration, the SNR is calculated based on the integration value integrated by the sampling unit integration circuit provided in the narrowband carrier frequency error detection circuit. Therefore, the variance of the narrowband carrier frequency error can be obtained based on the integration value integrated by the sampling unit integration circuit, and the SNR can be estimated based on the variance of the narrowband carrier frequency error.

  In order to solve the above problem, an OFDM demodulation method according to the present invention generates a baseband signal by orthogonally demodulating an intermediate frequency signal received by an antenna and frequency-converted from a high frequency signal by a tuner, and generating the baseband signal. The SNR is calculated based on the baseband signal with the narrowband carrier frequency error corrected before performing the FFT operation on the baseband signal with the narrowband carrier frequency error corrected. It is characterized by doing.

  According to the above feature, since the SNR is calculated based on the baseband signal before performing the FFT operation, the state of the received signal can be grasped without delay due to the FFT operation. As a result, it is possible to provide an OFDM demodulation method capable of grasping the state of a received signal without time lag due to high responsiveness.

  In order to solve the above-described problem, the program according to the present invention is configured to generate a baseband signal by orthogonally demodulating an intermediate frequency signal received by an antenna and frequency-converted from a high-frequency signal by a tuner. The procedure for correcting the narrowband carrier frequency error of the baseband signal and the baseband signal corrected for the narrowband carrier frequency error before performing the FFT operation on the baseband signal corrected for the narrowband carrier frequency error. And a step of calculating an SNR based on the procedure.

  In order to solve the above problems, a computer-readable recording medium according to the present invention generates a baseband signal by orthogonally demodulating an intermediate frequency signal received by an antenna and frequency-converted from a high frequency signal by a tuner. Correcting the narrowband carrier frequency error of the baseband signal, and correcting the narrowband carrier frequency error before performing an FFT operation on the baseband signal corrected for the narrowband carrier frequency error. A program for executing a procedure for calculating an SNR based on the baseband signal thus recorded is recorded.

  As described above, the OFDM demodulator according to the present invention includes the SNR calculation circuit that calculates the SNR based on the baseband signal whose narrowband carrier frequency error is corrected by the narrowband carrier frequency error correction circuit. In addition, there is an effect that it is possible to provide an OFDM demodulator that can grasp the state of a received signal without time lag due to high responsiveness.

  As described above, the OFDM demodulation method according to the present invention is based on the baseband signal corrected for the narrowband carrier frequency error before performing the FFT operation on the baseband signal corrected for the narrowband carrier frequency error. Thus, the SNR is calculated, so that it is possible to provide an OFDM demodulation method capable of grasping the state of the received signal without time lag due to high responsiveness.

  An embodiment of the present invention will be described with reference to FIGS. 1 to 9 as follows.

  FIG. 1 is a block diagram illustrating a configuration of an OFDM demodulator 1 according to an embodiment. As shown in FIG. 1, the OFDM demodulator 1 includes an antenna 13, a tuner 14, a bandpass filter (BPF) 103, an A / D conversion circuit 104, a DC cancellation circuit 105, a baseband signal processing unit 2, and an error correction processing unit. 15, a transport stream generation circuit 119 and an RS decoding circuit 120 are provided.

  The baseband signal processing unit 2 includes a digital orthogonal demodulation circuit 3, a narrowband carrier frequency error correction circuit 4, a narrowband carrier frequency error detection circuit 7, an FFT operation circuit 5, a waveform equalization circuit 110 and an SNR calculation circuit 6. Yes.

  The error correction processing unit 15 includes a frequency deinterleave circuit 111, a time deinterleave circuit 112, a demapping circuit 113, a bit deinterleave circuit 114, a depuncture circuit 115, a Viterbi circuit 116, a byte deinterleave circuit 117, and a spread signal removal circuit 118. Contains.

  A broadcast wave of a digital broadcast broadcast from a broadcast station is received by the antenna 13 of the OFDM demodulator 1 and supplied to the tuner 14 as an RF signal. The tuner 14 includes a multiplier 102a and a local oscillator 102b, and converts the frequency of the RF signal received through the antenna 13 into an IF signal. The tuner 14 supplies the IF signal subjected to frequency conversion to the BPF 103.

  The oscillation frequency of the reception carrier signal oscillated from the local oscillator 102b is switched according to a channel selection signal supplied from a channel selection circuit (not shown). The IF signal output from the tuner 14 is filtered by the BPF 103 and then digitized by the A / D conversion circuit 104. The digitized IF signal has its DC component removed by the DC cancellation circuit 105 and is supplied to the digital quadrature demodulation circuit 3.

The digital quadrature demodulation circuit 3 performs quadrature demodulation on the digitized IF signal using a carrier signal having a predetermined frequency (carrier frequency), and outputs a baseband OFDM signal. As a result of orthogonal demodulation, the baseband OFDM signal becomes a complex signal composed of a real axis component (I channel signal) and an imaginary axis component (Q channel signal). The IQDMD output (AFC1 output, before frequency correction) baseband signal Z ₀ (m, n) from the digital quadrature demodulation circuit 3 is supplied to the narrowband carrier frequency error correction circuit 4.

This baseband signal Z ₀ (m, n)
Z ₀ (m, n) = I ₀ (m, n) + jQ ₀ (m, n),
Represented by

The narrowband carrier frequency error correction circuit 4 is an AFC1 output (after frequency correction) baseband signal Z ₁ obtained by correcting the narrowband carrier frequency error of the baseband signal Z ₀ (m, n) supplied from the digital quadrature demodulation circuit 3. (M, n) is generated and supplied to the narrowband carrier frequency error detection circuit 7, the SNR calculation circuit 6, and the FFT operation circuit 5.

This baseband signal Z ₁ (m, n) is
Z ₁ (m, n) = I ₁ (m, n) + jQ ₁ (m, n),
Represented by

The SNR calculation circuit 6 calculates the SNR based on the baseband signal Z ₁ (m, n) generated by the narrowband carrier frequency error correction circuit 4.

  FIG. 2 is a diagram for explaining the difference between the guard interval signal of the OFDM symbol and the copy source signal. Here, the OFDM symbol will be described before the description of the SNR measurement method. The OFDM symbol is composed of an effective symbol corresponding to a signal period in which IFFT is performed at the time of transmission, and a guard interval in which a waveform of the latter half of the effective symbol is copied as it is. The guard interval is provided in the first half of the OFDM symbol. Under ideal conditions without noise and fading, the copy source signal and the guard interval signal completely match. However, when noise is superimposed, the copy source signal and the guard interval signal do not match due to the influence of noise.

Therefore, the amount of noise is calculated by obtaining δZ (m, n), which is the difference between the guard interval signal Z ₁ (m, n−N) and the copy source signal Z ₁ (m, n).

Here, as shown in FIG.
δZ (m, n) = Z ₁ (m, n−N) −Z ₁ (m, n),
here,
N: number of sampling points in the effective symbol period length,
For example, 512 (mode 2), 1024 (mode 3),
It is.

  Further, the reference sampling clock Fs = 64/63 MHz, and the reference sampling clock period Ts = 1 / Fs.

If the SNR is calculated using the baseband signal Z ₁ (m, n) whose narrowband carrier frequency error is corrected by the narrowband carrier frequency error correction circuit 4, the narrowband carrier frequency error is corrected at this time. Therefore, δZ (m, n) obtained by taking the difference between the copy source signal and the guard interval signal is the sum of the noise component of the copy source signal and the noise component of the guard interval signal, and δZ (m, n) / 2. Is calculated, the average value of the noise superimposed at the n points of m symbols and the noise superimposed at the nN points of m symbols can be obtained.

here,
m: symbol count,
n: Count of the number of sampling points, n = 0 to N + N _GI −1,
N _GI : Number of sampling points in the guard interval (GI) period length,
N _GI = N · g,
g: Guard interval (GI) period length ratio, 1/4, 1/8, 1/16,
It is.

  Next, the SNR calculation method will be described in detail. SNR is the result of decibel conversion of the ratio between the average value Ps of signal power and the average value Pn of noise power, as in (Equation 1).

Therefore, if the average value Ps of the signal power and the average value Pn of the noise power are obtained, the SNR can also be obtained. Here, in order to calculate the average value Pn of the noise power, first, v ₁ defined by (Equation 2) is calculated.

Next, in order to calculate the average value Ps of the signal power, v ₀ defined by (Equation 3) is calculated.

  Using the above (Formula 2) and (Formula 3), the SNR can be calculated as in the following (Formula 4).

  FIG. 3 is a diagram for explaining noise power of delayed waves in a multipath environment. The preceding wave has guard interval data 24a copied from the guard interval copy source data 25a, and the delayed wave has guard interval data 24b copied from the guard interval copy source data 25b. The preceding wave guard interval data 24a starts from time t1, the delayed wave guard interval data 24b starts from time t2, and the delayed wave is delayed by time t5 (= t2-t1) from the preceding wave. Yes. The preceding wave guard interval copy source data 25a is started from time t3, and the delayed wave guard interval copy source data 25b is started from time t4 which is delayed by time t5 from time t3.

  In the description of the calculation of SNR based on the above-described (Expression 1) to (Expression 3), assuming that there is a noiseless environment, it is assumed that the guard interval data 24a and the guard interval copy source data 25a completely match. Yes.

However, in the case of a two-wave multipath as shown in FIG. 3, if a delayed wave is mixed into the preceding wave, the guard interval data 24a (Z ₁ (m, n−N)) and the guard are generated even in a noiseless environment. The interval copy source data 25a (Z ₁ (m, n)) does not completely match. This is because the data 26 from the time t1 to the time t2 of the delayed wave mixed in the preceding wave is not the guard interval data 24a but the data having no correlation, and from the time t3 to the time t4 of the mixed delayed wave. This is because the data 27 is not the guard interval copy source data 25b but data that has no correlation. Therefore, when the SNR is calculated based on the above-described (Expression 1) to (Expression 3), noise power based on the data 26 and 27 is detected even in a noiseless environment. Therefore, the SNR having a value larger than the actually superimposed noise is calculated.

  In a multipath environment, the SNR is calculated in consideration of the above circumstances. Here, methods for detecting a multipath state are introduced in Japanese Patent Application Laid-Open Nos. 2000-165338, 2001-251272, and 2003-92560. If these methods are used, the delay profile can be calculated. Here, it is assumed that the delay profile is already known.

As an example, SNR calculation in a two-wave multipath environment as shown in FIG. 3 will be described. At this time, the delay time of the delay wave tau, DU ratio (intensity of the intensity ÷ delayed wave of the preceding wave) and [rho ^2.

In this case, the noise power vm ₁ can be expressed by (Equation 5), and the signal power vm ₂ can be expressed by (Equation 6).

  When the average value Ps of the signal power and the average value Pn of the noise power are obtained from (Expression 5) and (Expression 6), they are as shown in (Expression 7) and (Expression 8).

Substituting (Expression 7) and (Expression 8) into (Expression 1), the SNR is calculated. Thus, if there is a delay profile (delay time τ, DU ratio ρ ² ), the SNR can be calculated correctly even in a multipath environment.

FIG. 4 is a block diagram for explaining the configuration of the narrowband carrier frequency error detection circuit 7. The narrowband carrier frequency error detection circuit 7 calculates a complex correlation of the symbol m based on the baseband signal Z ₁ (m, n) output from the AFC1, and generates a complex correlation signal c (m, n). 8 is included. FIG. 5 is a block diagram showing a configuration of the complex correlation circuit 8. FIG. 6 is a diagram for explaining the complex correlation calculated by the complex correlation circuit 8. The complex correlation signal c (m, n) is expressed by the following (Equation 9).

  The complex correlation circuit 8 has a pair of delay FIFOs 16a and 16b arranged in parallel with each other. The complex correlation circuit 8 is provided with a complex multiplier 17. The complex multiplier 17 includes four multipliers 18a, 18b, 18c, and 18d arranged in parallel to each other, and two adders 19a and 19b arranged in parallel to each other.

The I component of the baseband signal Z ₁ (m, n) generated by the narrowband carrier frequency error correction circuit 4 is supplied to the delay FIFO 16a and the multipliers 18a and 18c, and the baseband signal Z ₁ (m, n) ) Q component is provided to the delay FIFO 16b and the multipliers 18b and 18d. The delay FIFO 16a delays the I component of the baseband signal Z ₁ (m, n) and supplies it to the multipliers 18a and 18b. The delay FIFO 16b delays the Q component of the baseband signal Z ₁ (m, n) and supplies it to the multipliers 18c and 18d. The adder 19a outputs an autocorrelation signal obtained by adding the output from the multiplier 18a and the output from the multiplier 18d. The adder 19b outputs a cross-correlation signal obtained by adding the output from the multiplier 18b and the output from the multiplier 18c.

  The narrow band carrier frequency error detection circuit 7 is provided with a section average circuit 9. FIG. 7 is a block diagram showing the configuration of the interval averaging circuit 9. The section average circuit 9 calculates a section average in the symbol m based on the complex correlation signal c (m, n), and generates a section average signal d (m). The section average signal d (m) is a section average of the complex correlation c (m, n) and is expressed by the following (Equation 10).

  The section average circuit 9 includes a FIFO 20, an adder / subtractor 21, a calculator 22, and a division bit shifter 23. The complex correlation signal c (m, n) is supplied to the FIFO 20 and the adder / subtractor 21. The adder / subtractor 21 delays the complex correlation signal c (m, n) from the complex correlation signal c (m, n) by the FIFO 20. Is supplied to the calculator 22. The output from the arithmetic unit 22 is added to the adder / subtracter 21 and is also supplied to the division bit shifter 23 to be subjected to division bit processing and output.

The narrow band carrier frequency error detection circuit 7 includes a phase shift amount circuit 10. The phase shift amount circuit 10 generates a phase shift amount δθ (m) per effective symbol period length N based on the section average signal d (m). As shown in FIG. 6, the phase shift amount δθ (m) is the phase rotation amount of the AFC1 output Z ₁ per effective symbol period length N in the symbol m. Since this is the amount of phase rotation per N samples (= effective symbol period length), it may be considered that it corresponds to a “normalized angular frequency” in which the angular frequency is normalized by the sampling angular frequency. This phase shift amount δθ (m) is expressed by the following (formula 11).

  An example of a specific mounting method of the phase shift amount circuit 10 is a CORDIC method described in Non-Patent Document 3. The CORDIC is a circuit that performs processing related to trigonometric functions.

The narrow band carrier frequency error detection circuit 7 is provided with a symbol unit integration circuit 11. The symbol unit integration circuit 11 integrates the phase shift amount δθ (m) from 0 to m to generate a symbol unit integration value γ (m) of the symbol m + 1. The symbol unit integral value γ (m) is a phase rotation amount (feedback value) per effective symbol period length N to be corrected by the symbol m. The amount of phase rotation detected from the AFC1 output Z ₁ (m, n) obtained by correcting γ (m) with AFC1 is δθ (m). That is, δθ (m) is a further phase rotation detected by the symbol m when γ (m) is used as a reference. Therefore, γ (m) to be corrected with the symbol m is expressed by the following (formula 12).

The narrow band carrier frequency error detection circuit 7 includes a sampling unit integration circuit 12. FIG. 8 is a diagram for explaining the phase correction amount generated by the sampling unit integration circuit 12. The sampling unit integration circuit 12 calculates a sampling unit integration value φ (m, n) which is an AFC1 phase correction amount for correcting the IQDMD output Z ₀ (m + 1, n) based on the symbol unit integration value γ (m). calculate. The sampling unit integral value φ (m + 1, n) is expressed by the following (formula 13).

The sampling unit integral value φ (m, n) is an AFC1 phase correction amount that is corrected by the AFC1 phase rotation circuit at the sampling point n of the symbol m. In AFC1, the IQDMD output Z ₀ (m, n) is corrected for each sampling point, so this variable is defined.

φ (m + 1, 0) = φ (m, 0) + γ (m) (N + N _GI ) / N = φ (m, N + N _GI ),
φ (m + 1, n) = φ (m + 1, 0) + γ (m + 1) × n / N (n = 1 to N + N _GI ),
It is.

The narrow-band carrier frequency error correction circuit 4 is configured to phase the IQDMD output signal Z ₀ (m, n) by an AFC1 phase correction amount −φ (m−1, n) based on the sampling unit integral value φ (m, n). Rotate to AFC1 output Z ₁ (m, n). The AFC1 output Z ₁ (m, n) is expressed by the following (formula 14).

  FIG. 9 is a block diagram for explaining another configuration of the narrowband carrier frequency error detection circuit 7. The same reference numerals are given to the same components as those described above. Therefore, detailed description of these components is omitted. The SNR calculation circuit 6 estimates the SNR from one of the intermediate signals in the narrowband carrier frequency error detection circuit 7. That is, the SNR calculation circuit 6 includes a complex correlation signal c (m, n) generated by the complex correlation circuit 8, a section average signal d (m) generated by the section average circuit 9, and a phase shift amount circuit 10. The generated phase shift amount δθ (m), the symbol unit integration value γ (m) integrated by the symbol unit integration circuit 11, and the sampling unit integration value φ (m, n) integrated by the sampling unit integration circuit 12. SNR is calculated based on at least one of the following.

  As described above, the guard interval signal and the copy source signal are completely coincident under the conditions without noise and fading. However, even under such conditions, since the input baseband signal includes a narrowband carrier frequency error, the narrowband carrier frequency error detection circuit 7 always detects a certain narrowband carrier frequency error.

  As described above, when noise is not superimposed on the input baseband signal, the narrowband carrier frequency error circuit 7 continues to detect a constant narrowband carrier frequency error without fluctuation. However, when noise is superimposed on the input baseband signal, the detected narrowband carrier frequency error fluctuates.

  The fluctuation of the narrow band carrier frequency error increases as the SNR decreases. Therefore, it is possible to obtain the variance (fluctuation) of the narrow band carrier frequency error and estimate the SNR therefrom. The data used for the estimation may be data based on an intermediate signal generated in the middle of the narrowband carrier frequency error circuit 7 as well as the narrowband carrier frequency error. This is because the intermediate signal generated in the middle also changes in dispersion due to the influence of the SNR, similarly to the narrow-band carrier frequency error. There is no clear relationship between dispersion and SNR. Therefore, the relational expression between the variance and the SNR is calculated back from the result of evaluation using a model simulation or a circuit such as an ASIC. That is, the dispersion value is evaluated in a state where the SNR of the propagation path is known, and a relational expression for calculating the SNR of the propagation path from the dispersion value is obtained based on the result.

  As described above, by obtaining the SNR based on the signal before being processed by the FFT operation circuit 5, the responsiveness can be improved and the state of the signal can be grasped without a time lag. The SNR does not include the amount of error generated in the demodulation process, but represents the amount of noise superimposed on the transmission path as a ratio. On the other hand, MER represents the sum of the noise in the transmission path and the noise during demodulation as a ratio. Since the transmission path status can be read not only from the MER but also from the SNR, it is effective to calculate the SNR.

  In the present embodiment, an example of an OFDM demodulator for receiving terrestrial digital broadcasting has been described, but the present invention is not limited to this. Any device that receives signals in accordance with the OFDM system may be used. For example, the present invention is also applied to a demodulator for wireless LAN, a demodulator for receiving BS digital broadcast and CS digital broadcast, and a demodulator for cable television. Can be applied.

  The present invention is not limited to the above-described embodiments, and various modifications can be made within the scope of the claims. That is, embodiments obtained by combining technical means appropriately changed within the scope of the claims are also included in the technical scope of the present invention.

  Note that in each part and each processing step of the OFDM demodulator according to the above-described embodiment, a calculation unit such as a CPU executes a program stored in a storage unit such as a ROM (Read Only Memory) or a RAM, and a communication such as an interface circuit. This can be realized by controlling the means. Therefore, various functions and various processes of the OFDM demodulator according to the present embodiment can be realized simply by a computer having these means reading the recording medium storing the program and executing the program. In addition, by recording the program on a removable recording medium, the various functions and various processes described above can be realized on an arbitrary computer.

  As this recording medium, a program medium such as a memory (not shown) such as a ROM may be used for processing by the microcomputer, or a program reader is provided as an external storage device (not shown). It may be a program medium that can be read by inserting a recording medium therein.

  In any case, the stored program is preferably configured to be accessed and executed by the microprocessor. Furthermore, it is preferable that the program is read out, and the read program is downloaded to a program storage area of the microcomputer and the program is executed. It is assumed that this download program is stored in advance in the main unit.

  The program medium is a recording medium configured to be separable from the main body, such as a tape system such as a magnetic tape or a cassette tape, a magnetic disk such as a flexible disk or a hard disk, or a disk such as a CD / MO / MD / DVD. Fixed disk, IC card (including memory card), etc., or semiconductor ROM such as mask ROM, EPROM (Erasable Programmable Read Only Memory), EEPROM (Electrically Erasable Programmable Read Only Memory), flash ROM, etc. In particular, there are recording media that carry programs.

  In addition, if the system configuration is capable of connecting to a communication network including the Internet, the recording medium is preferably a recording medium that fluidly carries the program so as to download the program from the communication network.

  Further, when the program is downloaded from the communication network as described above, it is preferable that the download program is stored in the main device in advance or installed from another recording medium.

  The present invention can be applied to an OFDM demodulation device, an OFDM demodulation method, a program, and a computer-readable recording medium that can efficiently transmit a video signal and an audio signal by a digital transmission method. The present invention is also applied to a device that receives a signal in accordance with the OFDM system, for example, a demodulator for a wireless LAN, a demodulator for receiving BS digital broadcast and CS digital broadcast, and a demodulator for cable television. be able to.

1, showing an embodiment of the present invention, is a block diagram showing a configuration of an OFDM demodulator. FIG. It is a figure for demonstrating the difference of the guard interval signal of an OFDM symbol, and a copy source signal. It is a figure for demonstrating the noise power of the delay wave in a multipath environment. It is a block diagram for demonstrating the structure of the narrow-band carrier frequency error detection circuit provided in the said OFDM demodulator. It is a block diagram which shows the structure of the complex correlation circuit provided in the said narrow-band carrier frequency error detection circuit. It is a figure for demonstrating the complex correlation which the said complex correlation circuit handles. It is a block diagram which shows the structure of the area average circuit provided in the said narrow-band carrier frequency error detection circuit. It is a figure for demonstrating the phase correction amount produced | generated by the sampling unit integration circuit provided in the said narrow-band carrier frequency error detection circuit. It is a block diagram for demonstrating the other structure of the narrow-band carrier frequency error detection circuit provided in the said OFDM demodulator. It is a figure which shows an example of the transmission symbol of an OFDM modulation wave. It is a block diagram which shows a prior art and shows the structure of an OFDM demodulation apparatus.

Explanation of symbols

DESCRIPTION OF SYMBOLS 1 OFDM demodulator 2 Baseband signal processing part 3 Digital orthogonal demodulation circuit 4 Narrow-band carrier frequency error correction circuit 5 FFT operation circuit 6 SNR calculation circuit 7 Narrow-band carrier frequency error detection circuit 8 Complex correlation circuit 9 Section average circuit 10 Phase shift Quantity circuit 11 Symbol unit integration circuit (integration circuit)
12 sampling unit integration circuit 13 antenna 14 tuner

Claims (9)

A digital quadrature demodulation circuit that generates a baseband signal by orthogonally demodulating an intermediate frequency signal received by an antenna and frequency-converted from a high-frequency signal by a tuner;
A narrowband carrier frequency error correction circuit for correcting a narrowband carrier frequency error of the baseband signal;
An FFT operation circuit that performs an FFT operation on the baseband signal with the narrowband carrier frequency error corrected;
An OFDM demodulator comprising: an SNR calculation circuit that calculates an SNR based on a baseband signal whose narrowband carrier frequency error has been corrected by the narrowband carrier frequency error correction circuit. A narrowband carrier frequency error detection circuit for detecting a narrowband carrier frequency error from the baseband signal corrected for the narrowband carrier frequency error;
The narrowband carrier frequency error detection circuit includes a complex correlation circuit that calculates a complex correlation of the baseband signal to generate a complex correlation signal,
The OFDM demodulator according to claim 1, wherein the SNR calculation circuit calculates the SNR based on a complex correlation signal generated by the complex correlation circuit. The narrowband carrier frequency error detection circuit calculates a complex correlation of the baseband signal and generates a complex correlation signal; and
An interval average circuit that generates an interval average signal based on the complex correlation signal,
The OFDM demodulator according to claim 1, wherein the SNR calculation circuit calculates the SNR based on a section average signal generated by the section average circuit. The narrowband carrier frequency error detection circuit calculates a complex correlation of the baseband signal and generates a complex correlation signal; and
An interval averaging circuit for generating an interval average signal based on the complex correlation signal;
A phase shift amount circuit that generates a phase shift amount based on the section average signal,
2. The OFDM demodulator according to claim 1, wherein the SNR calculation circuit calculates the SNR based on a phase shift amount generated by the phase shift amount circuit. The narrowband carrier frequency error detection circuit calculates a complex correlation of the baseband signal and generates a complex correlation signal; and
An interval averaging circuit for generating an interval average signal based on the complex correlation signal;
A phase shift amount circuit for generating a phase shift amount based on the section average signal;
An integration circuit for integrating the phase shift amount in symbol units,
The OFDM demodulator according to claim 1, wherein the SNR calculation circuit calculates the SNR based on an integration value integrated by the integration circuit. The narrowband carrier frequency error detection circuit calculates a complex correlation of the baseband signal and generates a complex correlation signal; and
An interval averaging circuit for generating an interval average signal based on the complex correlation signal;
A phase shift amount circuit for generating a phase shift amount based on the section average signal;
An integration circuit for integrating the phase shift amount in symbol units;
A sampling unit integration circuit that integrates the integration value by the integration circuit in sampling units,
The OFDM demodulator according to claim 1, wherein the SNR calculation circuit calculates the SNR based on an integration value integrated by the sampling unit integration circuit. A baseband signal is generated by orthogonally demodulating an intermediate frequency signal received by an antenna and frequency-converted from a high frequency signal by a tuner,
Correct the narrowband carrier frequency error of the baseband signal,
An OFDM demodulation method, comprising: calculating an SNR based on a baseband signal corrected for the narrowband carrier frequency error before performing an FFT operation on the baseband signal corrected for the narrowband carrier frequency error. A procedure for generating a baseband signal by orthogonally demodulating an intermediate frequency signal received by an antenna and frequency-converted from a high frequency signal by a tuner to a computer;
Correcting the narrowband carrier frequency error of the baseband signal;
A step of calculating an SNR based on the baseband signal corrected for the narrowband carrier frequency error before performing an FFT operation on the baseband signal corrected for the narrowband carrier frequency error. Program to do. A procedure for generating a baseband signal by orthogonally demodulating an intermediate frequency signal received by an antenna and frequency-converted from a high frequency signal by a tuner to a computer;
Correcting the narrowband carrier frequency error of the baseband signal;
A program for executing an SNR calculation procedure based on the baseband signal corrected for the narrowband carrier frequency error before performing an FFT operation on the baseband signal corrected for the narrowband carrier frequency error A computer-readable recording medium characterized by being recorded.

Images