Antenna Designs For Recent Millimeter and THZ

AIN SHAMS UNIVERSITY
FACULTY OF ENGINEERING
Electronics and Communication Engineering Department
Antenna Designs for Recent Millimeter and THz

Applications
A Thesis
Submitted in Partial Fulfillment of the Requirements
For the Degree of Doctor of Philosophy in Electrical Engineering
(Electronics and Communication Engineering)
Submitted By
Eng. Kamel Salah Kamel Sultan
Supervised By
Prof. Dr. Esmat Abdel-Fattah Abdallah

Prof. Dr. Hadia Mohamed Saied El Hennawy
Prof. Haythem Hussien Abdullah
Assoc. Prof. Mohamed Ali Basha
Egypt
2022
FACULTY OF ENGINEERING
Electronics and Communication Engineering Department
Antenna Designs for Recent Millimeter and THz Applications

By
Kamel Salah Kamel Sultan
Doctor of Philosophy in Electrical Engineering

(Electronics and Communications Engineering)
Faculty of Engineering, Ain Shams University, 2021
Examiners’ Committee
Approved by:
Name and Affiliation Signature
1- Prof. Mostafa Elsaid Mostafa ………………..
Electronics and Electrical Communications Department,
Faculty of Engineering, Cairo University.
(Examiner)
2- Prof. Amr Mohamed Ezt Safwat ………………..
Electronics and Communications Department,
(Examiner)
Faculty of Engineering, Ain Shams University.
3- Prof. Esmat Abdel-Fattah Abdallah ………………
Microstrip Department,
(Supervisor)
Electronics Research Institute.
1- Prof. Hadia Mohamed Saied El Hennawy …………………
Electronics and Communications Department,
(Supervisor)
Faculty of Engineering, Ain Shams University.
Date: 13/12/2021
STATEMENT
This dissertation is submitted as a partial fulfillment of the degree of Doctor of Philosophy in

Electrical Engineering (Electronics and Communications Engineering) Faculty of Engineering, Ain
Shams University.
The work included in this thesis was carried out by the author at the Electronics and
Communications Department, Faculty of Engineering, Ain Shams University, Cairo, Egypt.
No part of this thesis was submitted for a degree or a qualification at any other university or
institution.
Student Name: Kamel Salah Kamel Sultan

Signature:…………………………………….
Date:
ii
ABSTRACT
Antenna Designs for Recent Millimeter and THz
Applications
by
KAMEL SALAH KAMEL SULTAN
DOCTOR OF PHILOSOPHY IN ELECTRICAL ENGINEERING THESIS
Over the previous few years, the millimetre wave frequency range and sub-THz range have been
received a lot of attention as they include unused frequency spectrum resources that are appropriate for
providing a lot of applications such as automotive radars, short communication, medical imaging,
security, and 5G communications to provide end-user access to Multi-Gbit/s services. On the other
hand and because of the restrictions on the communication systems in these ranges such as power,
path losses, and attenuation, the present RF antennas can't be used for millimeter and THz
applications. The needed antennas should have low-profile, high-gain, acceptable technology, high
efficiency and low cost to compensate these ranges restrictions.
Although there are some antenna designs for millimeter and sub THz ranges through the last decay,
the antenna designs in these ranges still under investigation and studies to complete the vision of the
proposed designs for the aforementioned applications. Thus, they cannot be carried by the optimum
response and efficiency for these applications. This thesis aims to introduce a comprehensive study
for the main applications that occupy these ranges of frequencies. In addition to introduce efficient
antennas for each application with acceptable technology for it.
As the automotive radar sensors play an important role in driver safety and assist him in the
actions, the antennas that are used in this application is a key component in the sensor because they
need to provide high gain to increase the radar range and to provide wideband to increase the radar
resolution. The virtual antenna array (VAA) concept is introduced to provide low profile radar antenna
array with high gain and serve the long range radar (LRR) and medium range radar (MRR). The
analysis of VAA are introduced and verified. The antenna is fabricated and measured. Furthermore,
hybrid linear antenna arrays with two different configurations are introduced to achieve a suitable
iii
range and HPBW for LRR. The frequencies that can be used for automotive radar sensors are 24 GHz
and 76 GHz; those two bands are covered in this thesis.
The second contribution in this thesis is introducing an antenna for one of the future
communication systems (5G). The proposed antenna has a dual-polarization to overcome the high
losses at 28 GHz (the best-recommended band for 5G). Furthermore, the multiple-input multiple-
output (MIMO) antenna for 5G is introduced with a complete study of the MIMO parameters. This
antenna is based on characteristic mode analysis to study the antenna's performance. The metasurface
is combined with the slot antenna to enhance its performance and increases the gain. The detailed
illustrations of the dual-polarized antenna for handheld 5G systems with the comprehensive study of
the interaction of an antenna with the human body and vice versa are considered. The antenna is
fabricated and measured.
The third contribution in this dissertation mainly focused on the antennas designed for short
communications and multi-Giga-bit data rate applications. Two different endfire on-chip antennas
(OCA) using CMOS technology are introduced. These antennas are the Yagi-uda antenna and tapered
slot Vivaldi antenna. These antennas succeeded in achieving high performance compared to the
previous published OCAs because of the integration of different techniques to increase the radiation
characteristics of these antennas. Furthermore, a MIMO on-chip antenna is introduced to overcome
the high losses and high attenuation at 60 GHz. Three different configurations from two elements of
MIMO are presented in addition to one configuration from four elements of MIMO based on the
diversity technique to increase the isolation between the elements is also introduced. In terms of the
on-chip antenna, it is observed that the introduced MIMO antenna overcomes high CMOS losses.
The last contribution in this thesis is an antenna array that is based on the dielectric waveguide
and silicon on glass technology. The mode analysis, dielectric rod design, a transition between the
metallic waveguide and dielectric waveguide, and disc dielectric antenna design are introduced to the
antenna array design. The antenna meets the high gain and low profile structure requirements for the
sub-THz applications. In addition, the CPW feeding network is compatible with the other components
in the THz devices.
The introduced antennas with different techniques and different technologies in this thesis
positively contribute to the millimeter and the sub-THz applications. They are expected to enhance the
performance of the antennas for automotive radars, 5G handheld devices, multi-giga-bit
communications devices, short-range networks, and biomedical imaging.
iv
Key Words: Millimeter antennas, sub-THz, automotive radar sensors, virtual antenna arrays,
dielectric resonator antenna, on-chip antennas, hybrid antenna array, Yagi-Uda
antenna, Vivaldi antenna, dielectric waveguide, dual polarized antenna, metasurface,
characteristic mode analysis, Multiple Input Multiple Output (MIMO).
Thesis supervisors:
• Prof. Dr. Esmat Abdel-Fattah Abdallah
Electronics Research Institute,
Giza, EGYPT.
• Prof. Dr. Hadia Mohammed Said El-Hennawy
Ain Shams University,
Cairo, EGYPT.
• Prof. Dr. Haythm Hussien Abdullah
Electronics Research Institute,
Giza, EGYPT.
• Assoc. Prof. Mohamed Basha

Waterloo University,
Canada.
v
PUBLICATIONS
A. Journals
1. K. S. Sultan, H. H. Abdullah, E. A. Abdallah, and H. El-Hennawy “MOM/GA-based virtual

array for radar systems” MDPI, Sensors (Basel, Switzerland), vol. 20, no. 3, pp.1-16, 2020.
2. K. S. Sultan, H. H. Abdullah, E. A. Abdallah, and H. S. El-Hennawy “Metasurface based dual
polarized MIMO antenna for 5G smartphones using CMA ” IEEE Access, vol. 8, pp. 37250-
37264, 2020.
3. K. S. Sultan, E. A. Abdallah, and H. El-Hennawy “A MIMO on-chip Quasi-Yagi-Uda antenna
for Multi-gigabits Communications” Wiley: Engineering Reports, pp. 1-14, March 2020.
doi.org/10.1002/eng2.12133.
B. Conferences
1. K. S. Sultan, H. H. Abdullah, E. A. Abdallah, M.A. Basha and H. El-Hennawy “A 60-GHz

CMOS quasi-Yagi antenna with enhancement of radiation properties” 12th European
Conference on Antennas and Propagation (EuCAP 2018), pp. 1-3, April 2018.
2. K. S. Sultan, H. H. Abdullah, E. A. Abdallah, M.A. Basha and H. El-Hennawy “A 60-GHz
gain enhanced Vivaldi antenna on-chip” 2018 IEEE International Symposium on Antennas
and Propagation, pp. 1821-1822, 8-13 July 2018.
3. K. S. Sultan, H. H. Abdullah, E. A. Abdallah, M.A. Basha and H. El-Hennawy “Dielectric
resonator antenna with AMC for long range automotive radar applications at 77 GHz” 2018
IEEE International Symposium on Antennas and Propagation, 8-13 July 2018..
4. K.S. Sultan and M. A. Basha “High gain disc resonator antenna array with CPW coupled for
THz applications” 2018 IEEE International Symposium on Antennas and Propagation, pp.
1821-1822, 8-13 July 2018.
5. K. S. Sultan and M. A. Basha “High gain CPW coupled disc resonator antenna for THz
applications”, IEEE Antennas and Propagation Symposium (APS), Fajardo, Puerto Rico, pp.
263-264, June 26 - July 1, 2016.
vi
TABLE OF CONTENTS
STATEMENT .............................................................................................................................................. II
RESEARCHER DATA ........................................................... ERROR! BOOKMARK NOT DEFINED.
ABSTRACT ................................................................................................................................................ III
PUBLICATIONS ...................................................................................................................................... VI
ACKNOWLEDGEMNT .......................................................... ERROR! BOOKMARK NOT DEFINED.
TABLE OF CONTENTS ..................................................................................................................... VII
LIST OF FIGURES................................................................................................................................. XI
LIST OF TABLES ................................................................................................................................ XIV
LIST OF ABBREVIATIONS .............................................................................................................XV
1 CHAPTER ONE: INTRODUCTION .............................................................................................. 1
1.1 Introduction ........................................................................................................................................................ 1
1.2 Objectives ........................................................................................................................................................... 2
1.3 Features of mmW and S-THz ............................................................................................................................... 2
1.4 Applications of mmW and S-THz.......................................................................................................................... 3

1.4.1 5G Application................................................................................................................................................4
1.4.2 Automotive Radar Applications .................................................................................................................4
1.4.3 Short Range Communications ....................................................................................................................5
1.4.4 S-THz Applications .......................................................................................................................................5
1.5 Original Contribution ........................................................................................................................................... 6
1.6 Software Packages Used ...................................................................................................................................... 7
1.7 Thesis Organization ............................................................................................................................................. 7
2 CHAPTER TWO: ANTENNAS FOR MILLIMETER AND SUB-THZ

APPLICATIONS ..................................................................................................................................... 10
2.1 Introduction ............................................................................................................................................ 10
2.2 Automotive Radar Sensors (ARS) .................................................................................................... 10

2.2.1 ARS Bands ....................................................................................................................................................10
vii
2.2.2 Commercial Sensors ...................................................................................................................................14
2.2.3 Types of Radars............................................................................................................................................16
2.2.4 Antenna Design for ARS ............................................................................................................................21
2.3 Antennas for 5G Mobile Handset...................................................................................................... 24

2.3.1 Beam Steerable Antenna ...........................................................................................................................25
2.3.2 Switchable Phased Array...........................................................................................................................27
2.3.3 Dual Polarized Antenna .............................................................................................................................28
2.4 Short Range Communications (SRC) .............................................................................................. 29

2.4.1 CMOS Technology .......................................................................................................................................30
2.4.2 AoC Designs ..................................................................................................................................................30
2.5 Sub-THz Applications........................................................................................................................... 31

2.5.1 THz Applications .........................................................................................................................................32
2.6 Conclusion ............................................................................................................................................... 35
3 CHAPTER THREE: ANTENNA DESIGN FOR AUTOMOTIVE RADARS................ 36
3.1 Introduction ............................................................................................................................................ 36
3.2 MIMO/Phased Array ............................................................................................................................. 36
3.3 Linear Virtual Antenna Array ........................................................................................................... 38

3.3.1 Equal Distance .............................................................................................................................................39
3.3.2 Unequal Distance ........................................................................................................................................42
3.3.3 ULA Vs. VAA ................................................................................................................................................44
3.4 Virtual Antenna Array Design .......................................................................................................... 45
3.5 Design Antenna Array for LRR and MRR ..................................................................................... 54

3.5.1 Proposed Radiation Pattern of LRR and MRR .....................................................................................54
3.5.2 Excitation Coefficients ...........................................................................................................................55
3.5.3 Power Divider Design .................................................................................................................................56
3.5.4 Antenna Array Design ......................................................................................................................................58
3.6 Hybrid Antenna for 77 GHz Automotive Radar........................................................................... 63

3.6.1 Antenna Design ...........................................................................................................................................64
3.6.2 Results and Discussion for Single Element ...........................................................................................67
3.6.3 Antenna Array .............................................................................................................................................68
3.7 Conclusion ............................................................................................................................................... 75
4 CHAPTER FOUR: ANTENNA DESIGN FOR 5G .................................................................. 76
4.1 Introduction ............................................................................................................................................ 76
4.2 Characteristics Mode Theory ............................................................................................................ 76

4.2.1 Analysis of CMs ...........................................................................................................................................77
4.3 Dual Polarized Antenna for 5G ......................................................................................................... 79

4.3.1 Antenna Design ...........................................................................................................................................80
4.3.2 CM Analysis of Antennas ..........................................................................................................................82
viii
4.3.3 Slot Antenna Designs .................................................................................................................................85
4.4 Metasurface Design .............................................................................................................................. 93

4.4.1 Analyze of Metasurfce using CMA ..........................................................................................................93
4.5 Proposed Antenna with MTS ........................................................................................................... 100
4.6 MIMO Antenna Design ....................................................................................................................... 104

4.6.1 Fabrication and Measurements .............................................................................................................104
4.6.2 Envelope Correlation Coefficient ...........................................................................................................107
4.6.3 Smartphone Modeling ..............................................................................................................................109
4.7 Conclusion ............................................................................................................................................. 115
5 CHAPTER FIVE: ANTENNA DESIGN FOR SHORT RANGE

COMMUNICATIONS ........................................................................................................................ 117
5.1 Introduction .......................................................................................................................................... 117
5.2 Yagi-Uda Antenna................................................................................................................................ 119
5.3 Tapered Slot Antenna ........................................................................................................................ 123
5.4 Design of MIMO on-chip Antenna .................................................................................................. 129

5.4.1 Design of Two Elements...........................................................................................................................129
5.4.2 Four Elements MIMO ..............................................................................................................................132
5.4.3 MIMO Parameters ....................................................................................................................................133
5.5 Conclusion ............................................................................................................................................. 136
6 CHAPTER SIX: ANTENNA DESIGNS FOR THZ APPLICATIONS ....................... 138
6.1 Introduction .......................................................................................................................................... 138
6.2 THz ........................................................................................................................................................... 138
6.3 Transition from MWG to DRW ........................................................................................................ 138
6.4 Design of DRW ...................................................................................................................................... 140
6.5 Transition from CPW to DRW ......................................................................................................... 142
6.6 Design of One Element....................................................................................................................... 144
6.7 Design of Two Elements .................................................................................................................... 147
6.8 Conclusion ............................................................................................................................................. 149
7 CHAPTER SEVEN: CONCLUSION AND FUTURE WORK .......................................... 150
7.1 Thesis Conclusions.............................................................................................................................. 150
7.2 Suggestions for Future Works ......................................................................................................... 151
ix
REFERENCES...................................................................................................................................... 153
x
LIST OF FIGURES
PAGE
FIGURE 1.1 ATMOSPHERIC ATTENUATION VERSUS FREQUENCY. ..........................................................................................................3

FIGURE 1.2 PATH LOSS FOR DIFFERENT RANGES (R=100 M, R=1 KM, R=10 KM) ..................................................................................3
FIGURE 1.3 PROPOSED APPLICATIONS IN THE THESIS (SPECTRUM OF MILLIMETER AND SUB-THZ APPLICATIONS)...........................................4
FIGURE 2.1 AUTOMOTIVE RADAR FREQUENCY BANDS. ....................................................................................................................10

FIGURE 2.2 (A) THE BEAM COVERAGE OF THE RADAR MODES AND (B) THE DISTANCE COVERED IN METER [5]. ............................................12
FIGURE 2.3 ANGULAR RESOLUTION AND ANGULAR ACCURACY VERSUS FREQUENCY, ..............................................................................13
FIGURE 2.4 DESCRIPTION OF AUTOMOTIVE RADAR [43]. .................................................................................................................14
FIGURE 2.5 BOSCH LRR3 [5] AND LRR4 STRUCTURE [24]. ..........................................................................................................15
FIGURE 2.6 ARS 300 SENSOR WITH OPENED COVER [24] ...............................................................................................................15
FIGURE 2.7 RADAR CONFIGURATION ...........................................................................................................................................16
FIGURE 2.8 PULSED RADAR SIGNAL (A) TRANSMITTED SIGNALS, (B) RECEIVED SIGNALS..........................................................................19
FIGURE 2.9 RANGING WITH FMCW RADAR [47]. .........................................................................................................................20
FIGURE 2.10 EARLY AUTOMOTIVE RADAR SYSTEM (A) PARABOLIC ANTENNA[50], (B) HORN ANTENNA[47]. ..............................................21
FIGURE 2.11 DESIGN OF CYLINDER LENS ANTENNA IN THE ELEVATION PLANE (LEFT) AND AZIMUTH PLANE (RIGHT)[53]. ...............................22
FIGURE 2.12 SERIES ANTENNA ARRAY[55, 56]..............................................................................................................................23
FIGURE 2.13 PHOTOGRAPH OF DEVELOPED SERIES ANTENNA[57]. ....................................................................................................24
FIGURE 2.14 SERIES MICROSTRIP ANTENNA ARRAY INTEGRATED WITH AMC[58]. ................................................................................24
FIGURE 2.15 BEAM STEERING IDEA FOR A TALK MODE AND BROWSING MODE [9] ................................................................................25
FIGURE 2.16 STEERABLE ANTENNA ARRAY WITH FULL BOARD (B) FEEDING DETAILS [9] ..........................................................................25
FIGURE 2.17 GEOMETRY OF ANTENNA ARRAY (A) 3-D VIEW, (B) DETAILED VIEW, (C) EXPLODED VIEW, (D) BACK VIEW WITH SURFACE COPPER (E)
BACK VIEW WITHOUT SURFACE COPPER (UNITS (MM)) [11] .................................................................................................... 27
FIGURE 2.18 PROPOSED RADIATION PATTERN OF MOBILE PHONE (A)4G AND (B)5G [64] .....................................................................28
FIGURE 2.19 SWITCHABLE PHASED ARRAYS (A) SIDE VIEW WITH FULL PCB, (B) 3 ANTENNA ARRAY, (C) TOP LAYER VIEW OF ONE ARRAY, (D)
BOTTOM LAYER VIEW OF ONE ARRAY [64]. WSUB=55, LSUB=110, HSUB=0.787, WP=4.32, LP=2, D=6.5, D1=4.5, W=4.574, L=3.787,
W1=1.72, L1=3, W2=0.5 (UNITS (MM)). ..........................................................................................................................28
FIGURE 2.20 DUAL-POLARIZED SQUARE RING SLOT ANTENNA [65] ....................................................................................................29
FIGURE 2.21 STRUCTURE OF 0.18 µM CMOS ..............................................................................................................................30
FIGURE 2.22 DIFFERENT CONFIGURATIONS OF ON-CHIP ANTENNA DESIGNS.........................................................................................31
FIGURE 2.23 DIFFERENT CONFIGURATIONS OF ANTENNAS FOR THZ SHORT COMMUNICATIONS. .........................................33
FIGURE 2.24 REAL TIME FOCAL PLANE ANTENNA ARRAY CAMERA [134]. ...........................................................................34
FIGURE 2.25 IMAGING SYSTEM BASED ON HORN ANTENNA AT 200 GHZ [123]. ...................................................................34
FIGURE 3.1 MODES OF MIMO RADAR ANTENNA. .........................................................................................................................37

FIGURE 3.2 MIMO RADAR MODEL [139]. ...................................................................................................................................37
FIGURE 3.3 EXAMPLE OF MIMO SYSTEM [138] ...........................................................................................................................38
FIGURE 3.4 VIRTUAL PHASE CENTER OF UNIFORM ARRAY (M=4, N=4) ..............................................................................................41
FIGURE 3.5 VIRTUAL PHASE CENTER OF NON-UNIFORM ELEMENTS (M=N) ..........................................................................................42
FIGURE 3.6 EVPC OF M=3, N=4, 𝜷𝜷 = 𝟒𝟒 OVERLAPPED ................................................................................................................43
FIGURE 3.7 EVPC OF M=3, N=4, 𝜷𝜷 = 𝟒𝟒 NO-OVERLAPPED ..........................................................................................................44
FIGURE 3.8 ARRAY FACTOR OF LINEAR UNIFORM ARRAY AND VMIMO ..............................................................................................45
FIGURE 3.9 VIRTUAL ANTENNA ARRAY CONCEPT............................................................................................................................46
FIGURE 3.10 REFLECTION COEFFICIENT OF THE RECTANGULAR PATCH ANTENNA, 𝜀𝜀𝜀𝜀 = 3.38, ℎ = 0.2 𝑚𝑚𝑚𝑚............................................48
FIGURE 3.11 COMPARISON BETWEEN VAA AND PAA, CASE I. .........................................................................................................49
FIGURE 3.12 COMPARISON BETWEEN VAA AND PAA, CASE II. ........................................................................................................51
FIGURE 3.13 COMPARISON BETWEEN VAA AND PAA, CASE III. .......................................................................................................53
xi
FIGURE 3.14 COMPARISION BETWEEN RADIATION PATTERN OF PAA AND VAA ..................................................................54
FIGURE 3.15 THE SUGGESTED RADIATION PATTERN OF THE ANTENNA TO SUPPORT MRR AND LRR (LMRR) .............................................55
FIGURE 3.16 THE CONFIGURATION OF WILKINSON POWER DIVIDER WITH IMPEDANCE DISTRIBUTIONS......................................................57
FIGURE 3.17 S-PARAMETERS OF POWER DIVIDER ...........................................................................................................................58
FIGURE 3.18 LINEAR ANTENNA ARRAY (LP=3.2MM, WP=3.45 MM, S1=24 MM, S2= 12MM, AND S3=6 MM) .........................................59
FIGURE 3.19 GEOMETRY AND PHOTO OF FABRICATED VAA .............................................................................................................60
FIGURE 3.20 SIMULATED AND MEASURED S-PARAMETERS OF VAA ..................................................................................................61
FIGURE 3.21 SETUP SYSTEM FOR MEASUREMENTS AND RADIATION PATTERNS. ....................................................................62
FIGURE 3.22 ANTENNA GEOMETRY .......................................................................................................................................64
FIGURE 3.23 OUTER RADIUS OF DIELECTRIC RESENATOR VERSUS FREQUENCY FOR THE FIRST THREE MODES................................................66
FIGURE 3.24 ARTIFICIAL MAGNETIC CONDUCTOR RESULT ................................................................................................................66
FIGURE 3.25 SIMULATED REFLECTION COEFFICIENT OF THE PROPOSED ANTENNA..................................................................................67
FIGURE 3.26 RADIATION PATTERN AT 77 GHZ ..............................................................................................................................68
FIGURE 3.27 GAIN AND RADIATION EFFICIENCY OF THE PROPOSED ANTENNA .......................................................................................68
FIGURE 3.28 GEOMETRY OF THE PROPOSED ANTENNA ARRAYS WITH 8 ELEMENTS ................................................................69
FIGURE 3.29 GEOMETRY OF THE PROPOSED ANTENNA ARRAYS WITH 16 ELEMENTS ..............................................................70
FIGURE 3.30 S-PARAMETERS OF PROPOSED ANTENNA ARRAY CONFIGURATIONS. .................................................................72
FIGURE 3.31 3-D RADIATION PATTERN OF PROPOSED ANTENNA ARRAY CONFIGURATIONS. ..................................................73
FIGURE 3.32 2-D RADIATION PATTERN OF PROPOSED ANTENNA ARRAY CONFIGURATIONS. ..................................................74
FIGURE 3.33 GAIN AND EFFICIENCY OF PROPOSED DIFFERENT ANTENNA ARRAY CONFIGURATIONS. .....................................74
FIGURE 4.1 GEOMETRY OF PROPOSED ANTENNA............................................................................................................................81

FIGURE 4.2 CHARACTERISTIC MODE ANALYSIS OF ANT. I ..................................................................................................................83
FIGURE 4.3 CHARACTERISTIC MODE ANALYSIS OF ANT. II .................................................................................................................84
FIGURE 4.4 CHARACTERISTIC MODE ANALYSIS OF ANT. III ................................................................................................................84
FIGURE 4.5 EXPECTED AND SIMULATED ELECTRIC FIELD AND CURRENT DISTRIBUTIONS FOR ANT. I AND ANT. II. ..........................................86
FIGURE 4.6 REFLECTION COEFFICIENT OF ANT. I AT DIFFERENT VALUES OF SLOT WIDTH AND SLOT LENGTH.................................................87
FIGURE 4.7 REFLECTION COEFFICIENT OF ANT. II AT DIFFERENT VALUES OF SLOT WIDTH AND SLOT LENGTH................................................88
FIGURE 4.8 PROPOSED ANTENNA DESIGN WITH CONNECTORS. .........................................................................................................89
FIGURE 4.9 SIMULATED S-PARAMETERS OF TWO PORTS ANTENNA. ...................................................................................................89
FIGURE 4.10 SIMULATED S-PARAMETERS OF THE TWO PORTS ANTENNA (CPW WITHOUT GROUND IN THE BOTTOM). .................................90
FIGURE 4.11 CURRENT DISTRIBUTION OF PROPOSED ANTENNA AT 28 GHZ FROM TWO PORTS ................................................................91
FIGURE 4.12 ELECTRIC FIELD DISTRIBUTIONS FROM TWO PORTS. .......................................................................................................91
FIGURE 4.13 CO-POLARIZED AND CROSS POLARIZED FOR PORT1 AND PORT 2. ....................................................................................92
FIGURE 4.14 GAIN OF PROPOSED ANTENNA..................................................................................................................................93
FIGURE 4.15 METASURFACE STRUCTURE (W1=1.7 MM, G=0.2 MM, WM=13.3 MM). .......................................................................94
FIGURE 4.16 CMA PARAMETERS OF MTS. ..................................................................................................................................96
FIGURE 4.17 MODAL SURFACE CURRENT OF MTS. ........................................................................................................................98
FIGURE 4.18 MODAL RADIATION PATTERNS OF MTS. ....................................................................................................................99
FIGURE 4.19 CONFIGURATION OF PROPOSED ANTENNA WITH MTS.................................................................................................101
FIGURE 4.20 PROTOTYPE OF THE PROPOSED ANTENNA WITH MTS..................................................................................................101
FIGURE 4.21 SIMULATED AND MEASURED S-PARAMETERS OF SINGLE ANTENNA (PORTS 1 AND 2). ........................................................102
FIGURE 4.22 CO-POLARIZED AND CROSS-POLARIZED FOR PORT1 AND PORT 2 OF MTS ANTENNA AT 28GHZ. ........................................103
FIGURE 4.23 EFFICIENCY OF PROPOSED ANTENNA. .......................................................................................................................104
FIGURE 4.24 GAIN OF PROPOSED MTS ANTENNA. .......................................................................................................................104
FIGURE 4.25 MIMO CONFIGURATION AND PROTOTYPE OF PROPOSED ANTENNA WITH MTS. ..............................................................105
FIGURE 4.26 S-PARAMETERS OF PROPOSED ANTENNA WITH MTS...................................................................................................106
FIGURE 4.27 THE CONFIGURATION OF MIMO ANTENNA WITH COMMON GROUND (A)FRONT VIEW, (B)BACK VIEW AND (C)3-D VIEW. ......107
FIGURE 4.28 S-PARAMETERS OF PROPOSED MIMO ANTENNA WITH COMMON GROUND PLANE. ..........................................................107
FIGURE 4.29 ECC PARAMETER OF PROPOSED ANTENNA WITH MTS. ...............................................................................................108
FIGURE 4.30 ECC PARAMETER OF THE PROPOSED ANTENNA WITH MTS. ...........................................................................109
FIGURE 4.31 MOBILE MODELLING WITH THE COMPONENTS. ..........................................................................................................111
FIGURE 4.32 S-PARAMETERS OF THE PROPOSED MIMO ANTENNA INSIDE HOUSING. .........................................................................111
FIGURE 4.33 3-D RADIATION PATTERN OF ALL PORTS AT 28 GHZ (WITH HOUSING). ...........................................................................112
FIGURE 4.34 SAR DISTRIBUTION FROM MIMO ELEMENTS. ...........................................................................................................113
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FIGURE 5.1 BUILD 3-D SYSTEM PACKAGING ...............................................................................................................................119
FIGURE 5.2 YAGI ANTENNA GEOMETRY ......................................................................................................................................120
FIGURE 5.3 RETURN LOSS OF THE YAGI ANTENNA.........................................................................................................................121
FIGURE 5.4 GAIN OF THE YAGI-UDA ANTENNA. ...........................................................................................................................122
FIGURE 5.5 RADIATION EFFICIENCY OF THE YAGI-UDA ANTENNA. ....................................................................................................122
FIGURE 5.6 YAGI ANTENNA POSITIONS. ...............................................................................................................................123
FIGURE 5.7 THE RADIATION PATTERN OF THE ANTENNA IN (A)XY PLANE AND (B)YZ PLANE. ..............................................123
FIGURE 5.8 VIVALDI ANTENNA GEOMETRY .........................................................................................................................125
FIGURE 5.9 DESIGN STEPS OF VIVALDI ANTENNA ................................................................................................................126
FIGURE 5.10 S-PARAMETERS OF TRANSITION FROM CPW TO CPS .......................................................................................126
FIGURE 5.11 REFLECTION COEFFICIENT OF THE PROPOSED VIVALDI ANTENNA FOR FOUR CASES USING CST AND HFSS ....127
FIGURE 5.12 GAIN OF VIVALDI ANTENNAS ..........................................................................................................................127
FIGURE 5.13 RADIATION EFFICIENCY OF VIVALDI ANTENNAS .............................................................................................128
FIGURE 5.14 THE RADIATION PATTERN OF THE VIVALDI ANTENNA IN (A)XY PLANE AND (B)YZ PLANE ..............................128
FIGURE 5.15 DIFFERENT CONFIGURATIONS OF TWO ELEMENTS MIMO YAGI-UDA ANTENNA ...............................................................130
FIGURE 5.16 S-PARAMETERS OF DIFFERENT CONFIGURATION OF TWO ELEMENTS MIMO YAGI-UDA ANTENNA. .......................................131
FIGURE 5.17 CONFIGURATION OF FOUR ELEMENTS MIMO ANTENNA..............................................................................................132
FIGURE 5.18 S-PARAMETERS OF PROPOSED FOUR ELEMENTS MIMO ANTENNA.................................................................................133
FIGURE 5.19 RADIATION PATTERN OF PROPOSED MIMO ANTENNA AT DIFFERENT PORTS ....................................................................133
FIGURE 5.20 ECC OF FOUR ELEMENTS PROPOSED MIMO ANTENNA ...............................................................................................134
FIGURE 5.21 DG OF FOUR ELEMENTS PROPOSED MIMO ANTENNA ................................................................................................135
FIGURE 5.22 TARC OF PROPOSED FOUR ELEMENTS MIMO ANTENNA .............................................................................................136
FIGURE 5.23 CCL OF FOUR ELEMENTS PROPOSED MIMO ANTENNA ...............................................................................................136
FIGURE 6.1 GEOMETRY OF THE DRW FED BY MWG ....................................................................................................................139

FIGURE 6.2 GEOMETRY OF THE PROPOSED DRW.........................................................................................................................141
FIGURE 6.3 TRANSMISSION COEFFICIENT OF THE DRW .......................................................................................................141
FIGURE 6.4 FIRST MODE IN DRW ........................................................................................................................................142
FIGURE 6.5 TRANSITION FROM CPW TO DRW ....................................................................................................................143
FIGURE 6.6 ELECTRIC FIELD DISTRIBUTION IN TRANSITION FROM CPW TO DRW ...............................................................143
FIGURE 6.7 S-PARAMETERS OF TRANSITION FROM CPW TO DRW ..................................................................................................144
FIGURE 6.8 ANTENNA GEOMETRY .......................................................................................................................................144
FIGURE 6.9 ELECTRIC FIELD DISTRIBUTION. ........................................................................................................................145
FIGURE 6.10 RETURN LOSSES OF THE ONE ELEMENT ANTENNA ...........................................................................................146
FIGURE 6.11 3-D RADIATION PATTERN OF THE ANTENNA AT 490 GHZ. ...............................................................................146
FIGURE 6.12 HYBRID MODE DISTRIBUTION FOR TWO ANTENNAS AT 490 GHZ ...................................................................................146
FIGURE 6.13 GAIN OF THE PROPOSED ANTENNA ..........................................................................................................................147
FIGURE 6.14 GEOMETRY OF THE ANTENNA ARRAY ..............................................................................................................147
FIGURE 6.15 RETURN LOSS OF THE ANTENNA ......................................................................................................................148
FIGURE 6.16 3-D RADIATION PATTERN OF ANTENNA ARRAY AT 490 GHZ ............................................................................148
FIGURE 6.17 RADIATION PROPERTIES OF ANTENNA ARRAY. ................................................................................................149
xiii
LIST OF TABLES
PAGE
TABLE 2. 1 COMPARISON BETWEEN TWO BANDS OF SRR ................................................................................................................11
TABLE 2. 2 AUTOMOTIVE RADAR CLASSIFICATION [5] .....................................................................................................................11
TABLE 2. 3 COMMERCIAL RADAR SENSORS....................................................................................................................................16
TABLE 2. 4 COMPARISON BETWEEN CW AND PULSED RADAR ...........................................................................................................19
TABLE 3. 1 AMPLITUDE OF POWER DIVIDER OUTPUTS .....................................................................................................................56

TABLE 3. 2 IMPEDANCE OF POWER DIVIDERS (OPTIMIZATION VALUES), (ALL VALUES IN Ω) .....................................................................57
TABLE 3. 3 COMPARISON BETWEEN PROPOSED ANTENNA AND ANTENNAS IN LITERATURE ......................................................................63
TABLE 3. 4 ANTENNA PARAMETERS (ALL DIMENSIONS IN MM) ..........................................................................................................65
TABLE 3. 5 COMPARISON BETWEEN ANTENNA ARRAYS PERFORMANCE ...............................................................................................75
TABLE 4. 1 GEOMETRIC PARAMETERS OF THE PROPOSED ANTENNAS (MM) .........................................................................................82

TABLE 4. 2 COMPARISON BETWEEN RESONANT FREQUENCIES FOR CM OF THREE ANTENNAS ..................................................................85
TABLE 4. 3 DIMENSIONS OF PROPOSED MTS ANTENNA (MM) ........................................................................................................101
TABLE 4. 4 MATERIALS OF SMARTPHONE ...................................................................................................................................110
TABLE 4. 5 SAR VALUES (W/KG) ..............................................................................................................................................113
TABLE 4. 6 POWER DENSITY LIMITS FROM DIFFERENT STANDARDS ...................................................................................................114
TABLE 4. 7 POWER DENSITY VALUES AT 28 GHZ FROM DIFFERENT PORTS ACCORDING TO DIFFERENT STANDARDS. ....................................114
TABLE 4. 8 COMPARISON BETWEEN REFERENCED ANTENNAS AND THE PROPOSED ANTENNA .................................................................115
TABLE 5.1 ANTENNA DIMENSIONS (µM) .....................................................................................................................................121

TABLE 5.2 COMPARISON BETWEEN DIFFERENT POSITIONS OF THE ANTENNA ......................................................................................123
TABLE 5.3 ANTENNA DIMENSIONS (µM).....................................................................................................................................125
TABLE 5.4 COMPARISON WITH PREVIOUS CMOS ANTENNAS .........................................................................................................129
TABLE 5.5 COMPARISON BETWEEN THREE CONFIGURATIONS OF MIMO ANTENNAS ...........................................................................131
xiv
LIST OF ABBREVIATIONS
4G Fourth Generation
5G Fifth Generation
AA Atmospheric Absorption
Ae Effective Area
AMC Artificial Magnetic Conductor
AOC Antenna-On-Chip
AR Automotive Radar
ARS Automotive Radar Sensor
BW Bandwidth
CCL Channel Capacity Loss
CMA Characteristics Mode Analysis
CMOS Complementary Metal-Oxide-Semiconductor
CPS Coplanar Slot
CPW Coplanar Waveguide
CST Computer Simulation Technology
CST-MS CST Microwave Studio
CW Continuous-Wave
DG Diversity Gain
DR Dielectric Resonator
DRA Dielectric Resonator Antenna
DRW Dielectric Rode Waveguide
DWG Dielectric Waveguide
EBG Electromagnetic Band Gap
ECC Envelope Correlation Coefficient
ETSI European-Telecommunications-Standards-Institute
FCC Federal-Communications-Commission
FMCW Frequency-Modulated Continuous-Wave
FSL Free Space Loss
GA Genetic Algorithm
HDMI High-Definition-Multimedia-Interface
HP Horizontal Polarization
HPBW Half Power Beam Width
IR Infrared
ITU International Telecommunication Union
LAA Linear Antenna Arrays
LRR Long-Range Radar
MGps Multi-Gigabit-Per-Second
MIMO Multiple Input Multiple Output
MLS Multilayer Solver
mm-Wave Millimeter Wave
Mom Method of Moment
MRR Medium-Range Radar
MS Modal Significance
MTS Metasurface
MWG Metallic Wave Guide
NB Narrowband
xv
OCA On-Chip Antenna
ODS Outdoor Systems.
PAA Planar Antenna Array
PET Positron Emission Tomography
PIFA Printed Inverted F Antenna
PR Pulsed Radar
QYA Quasi Yagi Antenna
RA Resonator Antenna
RCS Radar Cross Section
SAR Specific Absorption Rate
SIMO Single-Input-Multiple-Output
SNR Signal to Noise Ratio
SOC System on Chip
SRC Short Range Communications
SRR Short-Range Radar
S-THz Sub-Terahertz
TCM Theory of Characteristic Mode
TSVA Tapered Slot Vivaldi Antenna
ULA Uniform Linear Array
UWB Ultra-Wide Band
VAA Virtual Antenna Array
VP Vertical Polarization
WLAN Wireless Local Area Network
WPAN Wireless Personal Area Network
SAR Specific Absorption Rate
xvi
Chapter 1: INTRODUCTION
1 Chapter One:
INTRODUCTION
1.1 Introduction
New technology and wireless communication systems have developed over the last two decades.
This development requires a wider bandwidth, higher data rate, and more compact devices [1]. In order
to achieve the desired requirements, future wireless communication systems are likely to work in the
millimeter and sub-terahertz (THz) range [2, 3]. The Millimeter-wave (mmW) and Sub-Terahertz (S-
THz) frequency ranges offer a number of unique advantages over other spectra for a large number of
emerging applications. These applications include high-resolution images, ultra-high-speed short-
distance communication systems, bio-medical, pharmaceutical, security, sensing, radar detectors and
spectroscopy. This indicates that wireless devices are required to support different technologies and
operate in different frequency bands. Remarkable progress has been made toward the development of
high-performance technology platforms for the practical realizations of these applications over recent
years. A low-cost and low-loss integrated circuit and system technology platform is essential for
general purpose applications. The millimeter band ranges from 30 GHz to 0.3 THz and the terahertz
waves offer bands in the range from 0.3 THz to 10 THz. The design of antennas in these ranges is
considered a challenging task since mastering the fabrication process and the measurement setup are
still under investigation worldwide [4-8].
One of the most promising applications in these ranges is 5G mobile communications. The 5G
denotes the next major phase of mobile telecommunication standards beyond the 4G standards. The
5G technology will change the way the higher bandwidth users access their phones. The second
application in these ranges is the on-chip systems for indoor applications because it provides high
speed for a short distance which can be used for video streaming, broadcasting and networking. In
contrast, the on-chip antennas suffer from low gain, low efficiency and complicated technology [2, 6,
9-12].
Another application is an automotive radar detector for short, medium and long-range radar.
Currently, there are several manufacturers worldwide of the automotive radar system at 24 GHz and
77/94 GHz that offers a maximum range of 250 m for detection depending on the type of radar. This
range of the radar is not sufficient for the train to detect any moving or static objects at the railways’
cross-sections ahead of time to stop the train safely to avoid collisions at the intersections. So, another
critical factor to mention in the radar system is the resolution required to detect the object in a very
large range with a very small angle of detection [5, 13-17].
1
1.2 Objectives
The main objective of our thesis is to design and implement antennas capable of achieving the
required specifications of popular applications in the millimeter and THz bands. This thesis aims to
introduce antennas for 5G, automotive radars, short-range communication and sub-THz applications;
all the designed antennas are verified by different methodology and analyses.
1.3 Features of mmW and S-THz

The millimeter ranges occupy the frequency from 30 GHz to 300GHz. The design of antennas
in these ranges is a challenging task since mastering the fabrication process in addition to the
measurement setup are still under investigation worldwide. One of the main restrictions on millimeter-
wave communication is atmospheric absorption. The signal attenuation is very high due to the rain,
fog, and any moisture in the air that reduce the transmission distances. The international
telecommunication union (ITU) introduced a model for the signal attenuation due to the atmospheric
gas, and this model is applicable for frequencies from 1 GHz to 1000 GHz and is used for polarized
and non-polarized fields (ITU recommendation number (ITU-R P 676-10)) [18]. The atmospheric
attenuation can be calculated from the following formula:
𝛾𝛾 = 0.182𝑓𝑓𝑁𝑁 ′′ (𝑓𝑓). (1.1)
where 𝛾𝛾 is the atmospheric attenuation, and quantity N"() is the imaginary part of the complex
atmospheric refractivity. From Eq. (1.1) and details of quantity (N) in [18], the atmospheric attenuation
versus frequency can be introduced, as shown in Figure 1.1. We note that the signal attenuated
significantly at frequency [13, 18-20]. First of all, it is clear that in the microwave range (up to 30
GHz), the atmospheric attenuation is rationally low at a few tenths of dB/km. In contrast, the
atmospheric attenuation has a large peak around 60 GHz, limiting this band's communication before
falling to about 0.3 dB/km around 80 GHz. Following that, the raising in attenuation becomes the
widest behaviour. From the graph, we can deduce that the 28/38 GHz is suitable for 5G applications
and the 30GHz bandwidth from 70GHz to 100GHz has low attenuation for automotive radar.
2
15
4
10 10
Atmospheric Gas Attenuation (dB/km)
2 0
10
20 40 60 80 100
0
10
-2
10
1 2 3
10 10 10
Frequency (GHz)
Figure 1.1 Atmospheric attenuation versus frequency.

The rain, fog, and atmospheric gas can be considered as one of the main limiting factors for
radar systems when operating above 5 GHz according to the relations of path losses introduced in [18].
Figure 1.2 shows the amount of path losses at different ranges against the frequency range. We notice
that path loss is directly proportional to the distance and the frequency.
200
180
160
Path Loss (dB)
140
120
Range: 100 m
Range: 1 km
100
Range: 10 km
1 2 3
10 10 10
Frequency (GHz)
Figure 1.2 Path loss for different ranges (R=100 m, R=1 km, R=10 km)
1.4 Applications of mmW and S-THz

Many studies are introduced in the literature to solve the problems of technology used in mmW
range and sub-terahertz range because of the mmW and S-THz technology is still under investigation
[7, 20-23]. Whereas there are some recommendations for each application through these bands. This
thesis focuses on different applications in these bands, as shown in Figure 1.3.
3
Figure 1.3 Proposed applications in the thesis (spectrum of Millimeter and sub-THz applications)
1.4.1 Automotive Radar Applications

Nowadays, there has been growing interest in automotive radar systems to measure the variable
of the surrounding environment, such as distance to target and velocity of a target to avoid the collision.
Therefore, radar sensors have drawn more attention because of their ability to make driving more
comfortable, stable and safer. The automotive radars are classified according to the operating range
into long-range radar (LLR) (10- 250 or 300 m), medium-range radar (MRR) (1-100 m), and short-
range radar (SRR) (0.15-30 m), where both LLR and MRR are required to detect the forward obstacles
[5, 13, 19, 24]. High gain and more efficient antennas are practical components in the automotive
radar system.
1.4.2 5G Applications
The International Telecommunication Union (ITU) has created several groups to achieve all
5G standards before 2020. The ITU releases the applicable frequencies for the new mobile generation
(5G) between 24 GHz to 86 GHz. Even though the range of 5G is still under review, there are several
candidate bands [25]. The range from 28 GHz to 38 GHz is highly recommended. In order to design
an effective antenna for 5G mobile phone, several fundamental challenges need to be considered. One
of these challenges is the free space loss (FSL) and atmospheric absorption (AA) that have high values
due to the higher frequency of millimeter ranges[1]. Also, FSL and AA allow for the reuse of the
spectrum due to the limit of interference amount between adjacent cells. Although relatively lower
losses and ease of technology can be achieved at lower microwave frequencies, these frequencies
suffer from a lower data rate, high latency, and vice versa for millimeter.
4
Nevertheless, most of the antennas in this side are limited to the linearly polarized antennas,
while in the real case, the mobile terminal will encounter different sorts of movements in Euler areas
in addition to the antenna operating in the MM-wave bands. Therefore, the miss-polarization among
the transmitter and the receiver antenna is one of the main significant loss factors in this
communication system. The circular polarization antenna loses half of the power in the transmitter or
receives linear polarization. Therefore, for full utilization of power in 5G systems, the antenna of dual-
polarization candidates to solve the problems of power losses and increase the bit error rate of the
communication systems. So, the antenna with different polarization (polarization diversity) plays an
essential key to solving the mentioned problems and improving channel capacity. On the other hand,
some advanced antenna techniques were reported to solve the problem of high FSL in MM-wave [9,
26, 27]. The researchers still introduce different studies to obtain the optimum antenna specifications
that can be used in this range.
1.4.3 Short Range Communications

Nowadays, low-frequency bands are very crowded, and with the rapid growth of communication
technologies, high-speed short-range wireless communications require wideband, higher data rates,
and compact size. In order to achieve the aforementioned requirements, the Millimeter-wave band at
60 GHz has towed more and more attention because it offers unlicensed bandwidth (from 57 GHz to
64 GHz) for several applications such as video streaming, wireless gaming, short-distance
communication and wireless personal area network (WPAN) [28, 29]. So, the complementary metal-
oxide-semiconductor (CMOS) technology is considered a good solution for cost and circuit integration
issues at this frequency. However, the CMOS substrate is inherited losses due to its high permittivity
(εr=11.9) and low resistivity (σ=10S/m). Additionally, CMOS antennas at 60 GHz require more
enhancements of antenna efficiency and antenna gain [28, 30-32].
1.4.4 S-THz Applications

The terahertz waves band ranges from 0.3 THz to 10 THz; the terahertz frequency range offers
new specifications over another spectrum for many applications, such as high-resolution imagers,
ultra-high-speed, short-distance communication systems, biomedical, pharmaceutical, security,
sensing, and spectroscopy.
This indicates that wireless devices are required to support different technologies and operate in
different frequency bands [22]. Several studies have been performed to produce an antenna structure
5
able to satisfy the demands for THz applications. A key factor for THz applications is a technology
platform for better performance of this band.
1.5 Original Contribution

Four major applications in this thesis are illustrated in the previous section. The works were
undertaken in this thesis aim to introduce antenna designs compatible with these four independent
applications one by one to solve the problems of each application. This thesis possesses four original
contributions, which are listed as follows:
1. Virtual antenna array (VAA) to enhance the angular resolution of the radar with a minimum
number of antenna elements compared with the conventional planar antenna array (PAA)
proposed for the automotive radar sensors. This VAA can be used to increase the radar range
and decrease the number of antennas in the antenna array. The proposed VAA is firstly
simulated and evaluated on a simple structure. Furthermore, the concept of a hybrid antenna
configuration is introduced as another solution to increase the range of the automotive radar.
The detailed illustrations of the VAA designs, fabrication, and measurements are exhibited in
Chapter 3 (Antenna Design for Automotive Radar), [33, 34].
2. Novel dual linear polarized metasurface antenna based on the characteristic mode analysis
(CMA) is proposed for the 5G applications. The dual-polarization is used to solve the problems
of isolation and channel capacity for MIMO smartphone designs. Furthermore, the theory of
characteristic mode (TCM) or characteristic mode analysis (CMA) is introduced as an accurate
analysis for a few unit cells of the metasurface. The detailed illustrations of the dual-polarized
antenna for handheld 5G systems with the comprehensive study of the interaction of an antenna
with the human body and vice versa are exhibited in Chapter 4 (5G mobile applications), [35].
3. Novel two end-fire antennas based on the hybrid technique to increase the radiation
characteristics are proposed of the on-chip systems for short communications at 60 GHz. The
proposed two antennas solve the problems of low efficiency and low gain in traditional on chip
antennas. Accurate results are introduced by comparison with different simulation tools. The
first one, a 60 GHz Yagi-Uda antenna and the second one is a 60 GHz Vivaldi on-chip antenna.
The two antennas are presented on standard 0.18 µm CMOS technology. The detailed
6
illustrations of the two antenna designs are presented in Chapter 5 (Short Communications),
[36-38].
4. A novel disk resonator antenna (DRA) fed by coplanar waveguide (CPW) technique with
compact size and high gain using silicon on glass (SOG) technology platform is proposed. The
CPW feed is patterned on the backside of the Si wafer before the bonding process from the
Pyrex side. In addition, the dielectric waveguide (DWG) is matched with the disc dielectric
antenna using CPW feed. The DRA covers the band from 325 GHz to 600 GHz and can work
in broadside and end-fire radiation. This antenna has high efficiency and low cost. The detailed
illustrations of the DRA and the antenna array designs are introduced in Chapter 6 (Sub-THz
applications), [39, 40].
1.6 Software Packages Used

In this thesis, we used different software packages based on different techniques. The first package
is CST Microwave Studio (CST-MS) that introduce a wide variety of solvers for different problems.
We used four solvers in this thesis namely: time domain solver, Eigen-mode solver, and multilayer
solver (MLS) or characteristic mode analysis (CMA). The CST-MS that is based on time-domain
solver is used to simulate the VAA and the hybrid antenna array in chapter three. In addition to
simulating two dual-polarized 5G antenna in chapter four, two end-fire on-chip antenna in chapter 5
and S-THz antennas in chapter six. On-other hand, the CST-MS that is based on eigenmode solver is
used to analyze the operating modes for the DRA in chapter six. Also, the CST-MS based on CMA is
used in chapter five to analyze the characteristic mode of metasurface that is used for 5G antenna. A
second package is COMSOL software based on multi-physics analyze is used in chapter six to analyze
the modes of DRA. The third software package is HFSS (High Frequency Structure Simulation) based
on the finite element method is used to ensure the results of on-chip antennas.
1.7 Thesis Organization

The work undertaken in this thesis is organized as follows:
1. Chapter 1 presents a brief overview of the proposed research background, in addition to present
the motivations, objectives and the original contributions for this thesis.
2. Chapter 2 focuses on a survey about the different applications in the millimeter and sub-THz
frequency range. The complete literature survey about the automotive radar sensors at 24 GHz
and 77 GHz is introduced. A survey about the 5G antennas and the new guidelines for the new
7
generation of mobile communications are presented. Furthermore, a literature review based on

the technology that can be used for short communications at 60 GHz and the on-chip antenna
is presented. Finally, the different types of antennas and technology that can be used in sub-
THz are given.
3. Chapter 3 presents the concept of VAA which apply on the antenna array for the automotive
radar. The usage of VAA concept aims to reduce the number of antenna elements used in the
array and decrease the number of channels required in the transceiver system of automotive
radar. The VAA and unequal power divider are introduced to achieve flat-shoulder shape (FSS)
radiation pattern for covering the long range radar and the short range radar. The number of
the VAA elements are defined according to the integration between the method of moment
(MOM) and the genetic algorithm (GA) to achieve the optimum number of elements.
Furthermore, the hybrid antenna array is introduced with 16 elements to achieve the
requirement of the Long range radar (LRR) at 77 GHz. The element of this antenna array
consists of a hybrid radiator and dielectric resonator. The hybrid radiator is a circular patch that
is fed by aperture method and the dielectric resonator is a ring that is fed by the circular patch
to operate at 77 GHz with high gain. The electromagnetic band gap (EBG) structure is
implemented on the top layer to widen the proposed band, gives high gain, reduces surface
waves and gives good isolation between antenna array elements. Details of these designs,
fabrications, and measurements are introduced in this chapter.
4. Chapter 4 introduces a novel dual-polarized MIMO antenna for 5G mobile handset. Also, it
presents a complete study for metasurface (MTS) that has a low profile, lightweight, easy
integration and low loss in contrast to metamaterial. So, the CMA is used to analyze the
proposed antenna with MTS. The isolation coefficients, envelope correlation coefficient
(ECC), channel capacity loss (CCL) of the MIMO are also calculated. Furthermore, this chapter
presents a comprehensive study on the performance of compact antenna design. Return loss,
radiation patterns, specific absorption rate (SAR), and efficiency of this antenna are computed
in free space, in the presence of handset as well as in the presence of head and hand. The peak
SAR in the head is compared with SAR limits in the safety standards and so the maximum
radiated power of each antenna is determined.
5. Chapter 5 introduces two different end-fire antenna configurations to enhance the radiation
characteristics of on-chip antennas that is used in short communications at 60 GHz. A hybrid
technique that depends on reducing the backward radiation and reduce the surface waves is
8
used to solve the problems of low radiation for on-chip antennas. Also, comprehensive study
of the on-chip antenna is introduced.
6. Chapter 6 presents one element, two elements and four elements disc resonator antenna (DRA)
with compact size and low profile based on the silicon on glass technology platform. The
proposed antenna consists of a silicon straight section waveguide segment connected in series
with disc resonator which acts as radiating element. The CPW power divider with compact size
is used to the disc resonator. The proposed antenna in this chapter is introduce to cover S- THz
band from 325 GHz to 600 GHz. Furthermore, the end-fire and broadside antennas are
introduced. The antenna has more compact size when compared to other published antennas.
7. Chapter 7 gives the final conclusion of the presented works as well as suggestions for promising
future works.
9
Chapter 2: Millimeter and S-THz Applications
2 Chapter Two:
ANTENNAS FOR MILLIMETER
AND SUB-THZ APPLICATIONS
2.1 Introduction
This chapter introduces a literature review on the antennas that are used for the different
applications through the mmW and S-THz ranges. We focus here on the antennas for automotive
radars, 5G applications, short-range communications and S-THz applications.
2.2 Automotive Radar Sensors (ARS)

2.2.1 ARS Bands
Nowadays, there has been growing interest in the automotive radar system to measure the
variables of the surrounding environment, such as the distance to target and velocity of a target to
avoid the collision. So, radar sensors have drawn more attention because of their ability to make driving
more comfortable, stable and safer.
Figure 2.1 Automotive radar frequency bands.
A 24 GHz and 77 GHz are two frequencies that are predominantly used for automotive sensors, as
shown in Figure 2.1. In 1999, the automotive radar systems appeared in the international market that
was designed to operate as short-range radar (SRR) at 24 GHz and 76 GHz for long-range radar (LRR),
and from 77 GHz to 81 GHz for medium and short-range sensors. So, the radar can be classified
according to its range to LRR, MRR and SRR. Since then, many different radar systems have been
developed; the first radar that operates in 77 GHz was introduced by Daimler S class; after this year,
other companies such as Jaguar, Nissan and BMW followed [13].
The 77 GHz band has various benefits that forced the designers to shift the radar applications
toward this band. One of these reasons is revolved around 24 GHz radar disadvantages as:
10
• It includes an ISM band (Industrial, Scientific and Medical) with 200 MHz starting from 24.05
GHz and ending to 24.25 GHz, called the narrowband (NB). Furthermore, it includes 5 GHz to
be an ultra-wideband (UWB). For the SRR, the NB and UWB have been used in legacy
automotive sensors for the 24-GHz band.
• It doesn’t support the long-range radar.
• According to the European-Telecommunications-Standards-Institute (ETSI) and the Federal-
Communications-Commission (FCC), the 24-GHz UWB band will not be used from January
1, 2022, and this is called “sunset date” [41, 42].
The other reasons include the benefits of 77 GHz radar compared with the 24 GHz radar, such as
shown in Figure 2.2. The 77 GHz radar has advantages such as wideband and small size. On the other
hand, it has considerable effort in its design and implementation. Furthermore, the radar cross-section
of Pedestrians are small and usually tend to change their directions. So, the radar needs to have a high
resolution to track the direction changes and avoid collisions with the pedestrians. The 77 GHz radars
include two sub-bands:76 − 77𝐺𝐺𝐺𝐺𝐺𝐺 and 77 − 81𝐺𝐺𝐺𝐺𝐺𝐺 (also called 79 GHz band). The automotive
radar sensors at 77-GHz are classified according to the radar distance to three types, as shown in Table
2. 1 and Table 2. 2 [5].
Table 2. 1 Comparison between two bands of SRR

Type SRR (24 GHz) SRR (77 GHz)
Operating Band 24 GHz – 24.25 GHz Temporary Band 77 GHz – 81 GHz Permanent Band
Range Resolution 75 cm 4 cm
Impedance BW 17 % 5%
Beam Width Wide More focused
Sensor Size 3X X
Angular Resolution 3X X
Table 2. 2 Automotive radar classification [5]
Type LRR MRR SRR
ERIP 55 dBm -9 dBm/MHz -9 dBm/MHz
Frequency Band 76-77 GHz 77 -81 GHz 77-81 GHz
Bandwidth 600 MHz 600 MHz 4 GHz
11
Main Aspect Detection Range Detection Range Range accuracy
Range 10 -250 m 1-100 m 0.15-30 m
Range Resolution 0.5 m 0.5 m 0.1 m
Range Accuracy 0.1 m 0.1 m 0.02 m
Velocity Resolution 0.6 m/s 0.6 m/s 0.6 m/s
Angular Accuracy 0.10 0.50 0.50
HPBW in Azimuth ±150 ±400 ±800
HPBW in Elevation ±50 ±50 ±100
Dimensions 74 × 77 × 58 mm 50 × 50 × 50 mm 50 × 50 × 20 mm
Figure 2.2 (a) The beam coverage of the radar modes and (b) the distance covered in meter [5].
The angular resolution of the radar means the distinction between the two targets. It depends
on the two main parameters: wavelength and aperture size of the antenna. The angular accuracy of the
radar means the accuracy of angle measurement based on the wavelength, aperture size, and signal to
noise ratio. The angular resolution and angular accuracy can be calculated from equation (2.1) and
equation (2.2) using the Rayleigh criterion [5].
∆𝜑𝜑 = 1.22 𝑑𝑑
𝜆𝜆 (2.1)
𝛿𝛿𝛿𝛿 =
∆𝜑𝜑 (2.2)
√2 𝑆𝑆𝑆𝑆𝑆𝑆
Where d antenna aperture size, 𝜆𝜆 wavelength, and SNR signal to noise ratio. Figure 2.3 shows the
angular resolution and angular accuracy with the frequency variation, when d=30 mm, and SNR=10
12
dB. We noted that ∆𝜑𝜑 =(29.130, 9.070) and 𝛿𝛿𝛿𝛿 =(6.510, 2.030) at 24 GHz and 77 GHz, respectively.
In conclusion, the radar at 24 GHz needs an antenna with three times larger than that used at 77 GHz
to achieve the same angular resolution.
35
Angular Resolution
X: 24
Y: 29.13 Angular Accuracy
30
25
20
Angle(degree)
15
X: 77
Y: 9.078
10
X: 24
Y: 6.513
5 X: 77
Y: 2.03
0
20 30 40 50 60 70 80 90 100
F(GHz)
Figure 2.3 Angular resolution and angular accuracy versus frequency,

d=30 cm and SNR=10 dB.
The automotive radar sensors (ARS) are used to eliminate the possibility of collisions occurring or
risky situations. So, it is used to alert the driver, control the vehicle to prevent an accident, rearview
traffic crossing alert, or blind spot detection. More than one radar sensor is used to detect the obstacles
and the relative speed of the target. To avoid the collision and reduce the risk, the appropriate action
should be taken by the processing unit; this action depends on the reflected signal from the target.
The functions of any automotive radar system should include the following objectives:
• Detect the obstacles surrounding the vehicle

• The relative position of the target to the vehicle
• The relative speed of the target to the vehicle
Then by decision-maker unit can take one or more actions from the following:
• Alert the driver about the dangerous situation.

• Prevent collision by the control of the vehicle in risk situations
• Adaptive Cruise Control (ACC)
• Assist the driver for car parking
13
So, the automotive radar can be described as another driver with you, as shown in Figure 2.4.
Figure 2.4 Description of automotive radar [43].
There are six main parameters that affect the performance of the automotive radar as follows:
• Detection-Range
• Speed-Detection-Range
• Range-Precision
• Velocity-Precision
• Angular-Resolution
• Angular-Width-of-View
2.2.2 Commercial Sensors

In 2008, the Denso Company introduced the third generation of its long-range radar sensor,
including digital beamforming for the first time in its radar. The digital beamforming is achieved by
one transmit antenna and five switched received antennas. The antenna itself is an array formed of
slotted waveguides [44].
In 2009, researchers from BOSCH Company introduced a new long-range transceiver chip as
shown in Figure 2.5, to operate in the band 76-77 GHz with a radar range of up to 250m, range accuracy
of 0.1 m and its relative speed of -75 to 60 m/s with speed accuracy 0.12 m/s [5, 45]. It has four
patches antennas with a dielectric lens to give high gain for long-range radar and to create four slightly
offset beams. All four antennas are receiving antennas, with the middle two antennas also
14
simultaneously transmitting, as shown in Figure 2.5. In 2015, BOSCH introduced the fourth generation
of the automotive sensor to enhance the radar performance with lens antenna [24].
(a)LRR3[5], (b)LRR4[24]
Figure 2.5 BOSCH LRR3 [5] and LRR4 structure [24].
On the other hand, Conti Company introduced ARS 300 as another radar in 2009 that operates
for the long and medium-range by using different patterning of the spindle [24]. The ARS system
consists of the reflect-array antenna that provides auto alignment by grooved rotating drum, as shown
in Figure 2.6. Table 2. 3shows the characteristics and performance parameters for different commercial
automotive radar sensors taken from datasheets.
Figure 2.6 ARS 300 sensor with opened cover [24]
15
Table 2. 3 Commercial radar sensors

Sensor F(GHz) Dimensions Range BW Azimuth Multi- Accuracy
(mm) (m) Angle range
(degree)
Bosch LRR3 77 74 x 70 x 78 250 1 GHz 30 Single 0.1m, 0.12 m/s
Delphi ESR 77 173 x 90 x 49 174 - 20 Multi 1.8m, 0.12m/s
Cont. ARS30x 77 120 x 90 x 49 250 1 GHz 17 Multi 1.5% , 0.14 m/s
Denso 77 78 x 77 x 38 150 1 GHz 20 Single 1.8, 0.12m/s

DNMWR004
SMS UMRR 24 212 x 154 x 40 250 250 MHz 36 Single 2.5% , 0.28
type 40
TRW AC100 24 460 x 460 x 50 150 100 MHz 16 Single ---
2.2.3 Types of Radars

2.2.3.1 Types of Radars According to Configuration
The radar system can be classified according to the transmitter and receiver position relative to the
target into two types: Monostatic and Bistatic radar. In the monostatic configuration, a single antenna
is used for transmitting the signal and receiving it, whereas the transmitter and receiver are collected
in the same device as a transceiver. In this thesis, the monostatic radar configuration will be used (see
Figure 2.7(a)). In other words, when the antennas of the transmitter and the receiver are closed to each
other and at the same location, the radar is called monostatic. In the configuration of bistatic, two
antennas are used, one for the transmitter and the other for the receiver with the displaced distance
between them. The antennas are physically separated, as shown in Figure 2.7(b). If Bistatic radar has
more than one receiver antenna, then it is known as multi-static radar [19, 24, 46].
(a)Monostatic
(b)Bistatic
Figure 2.7 Radar Configuration
16
For one way radar, the received power at the target can be calculated from equation (2.3):
𝑃𝑃𝑡𝑡 𝐺𝐺𝑡𝑡 𝐺𝐺𝑟𝑟 𝜆𝜆2 (2.3)
𝑃𝑃𝑟𝑟 =
(4𝜋𝜋𝜋𝜋)2
The antenna gain can be expressed as a function of an effective area Ae as presented in equation (2.4)
4𝜋𝜋𝐴𝐴𝑒𝑒 (2.4)
𝐺𝐺 =
𝜆𝜆2
The target reflects a portion of power in a reverse way, in the direction of the radar. This portion of
power depends on the Radar Cross Section (RCS) of the target. The RCS describes the target
characteristics such as its size and dimension as seen by the radar. For the radar target, the amount of
reflected power by the target is equal to the re-radiated power of the antenna with an effective area
equal to the RCS of the target. Therefore, the receiving antenna's effective area (Ae) is replaced by the
RCS (σ). So, the reflected power from the target can be expressed as:
𝑃𝑃𝑡𝑡 𝐺𝐺𝑡𝑡 𝜆𝜆2 (4𝜋𝜋𝜋𝜋) 𝑃𝑃𝑡𝑡 𝐺𝐺𝑡𝑡 (4𝜋𝜋𝜋𝜋) (2.5)
𝑃𝑃𝑟𝑟 = =
(4𝜋𝜋𝜋𝜋)2 𝜆𝜆2 (4𝜋𝜋𝜋𝜋)2
The reflected power back to the radar receiver is:
𝑃𝑃𝑡𝑡 𝐺𝐺𝑡𝑡 (4𝜋𝜋𝜋𝜋) 𝐺𝐺𝑟𝑟 𝜆𝜆2 𝑃𝑃𝑡𝑡 𝐺𝐺𝑡𝑡 𝐺𝐺𝑟𝑟 𝜎𝜎𝜆𝜆2 (2.6)
𝑃𝑃𝑟𝑟 = =
(4𝜋𝜋𝜋𝜋)2 (4𝜋𝜋𝜋𝜋)2 (4𝜋𝜋)3 𝑅𝑅 4
𝜆𝜆 2 4𝜋𝜋𝜋𝜋 𝜆𝜆 2 (2.7)
𝑃𝑃𝑟𝑟 = (𝑃𝑃𝑡𝑡 𝐺𝐺𝑡𝑡 𝐺𝐺𝑟𝑟 ) � � � 2 ��
4𝜋𝜋𝜋𝜋 𝜆𝜆 4𝜋𝜋𝜋𝜋
𝜆𝜆 4
So, the free space loss (FSL) of the monostatic radar 𝐹𝐹𝐹𝐹𝐹𝐹 = �4𝜋𝜋𝜋𝜋� .
Where
• Pr : Received power in watts.
• Pt : Peak transmitted power in watts.
• Gt : Transmitter Gain.
• Gr : Receiver Gain .
• λ: Wavelength (m).
• σ: RCS of the target (m2 ) .
• R: Range between radar and target (m).
2.2.3.2 Types of Radar According to Operation

1. Pulsed Radar
The pulsed radar depends on measuring of the time delay between transmission and reception pulse
where the radar transmits a number of pulses then calculate the delay time and the change in pulse
17
width. Due to the delaying of the reflected pulse and the change of pulse width, we can calculate the
distance between the sensor and the target in addition to the speed of the target relative to the speed of
the vehicle.
Typically, the pulsed radars have a blind speed and ambiguous range issues. In addition,
transmitting a narrow pulse in the time domain means that a large amount of power must be transmitted
in a short period of time. In order to avoid this issue, spread spectrum techniques may be used.
In pulsed radar, the generated radio signal at a constant frequency 𝑓𝑓0 passes through the pulse
shaping device that converts it to a train of pulses. Suppose the propagation speed c of the
electromagnetic wave in the medium is known and the round trip delay is t. In that case, we can
calculate the distance between the target and the radar from the simple following equation:
𝑐𝑐𝑐𝑐 (2.8)
𝑅𝑅 =
2
In case of the motion of target, the relative velocity can be determined from the Doppler shift
of received signal frequency𝑓𝑓𝑟𝑟 as shown in equation (2.9). The Doppler shift is the difference between
the transmitted and received frequency𝑓𝑓𝑑𝑑 = 𝑓𝑓𝑟𝑟 − 𝑓𝑓0 .
𝑐𝑐𝑓𝑓𝑑𝑑 (2.9)
𝑣𝑣𝑟𝑟 =
2𝑓𝑓0
The maximum range for pulsed radar depends on the pulse repetition rate (PRR) of the
transmitted pulses 𝑇𝑇𝑝𝑝 as shown in Figure 2.8(a) and can be determined from equation (2.10). In other
words, it is defined as the maximum range of pulse that can send from the transmitter before the next
pulse is emitted. Figure 2.8(b) illustrates that the system can receive echo pulses after sending the other
pulse. In this case, the range to the target is calculated by false information to be ∆𝑡𝑡3 instead of the real
range that equal 𝑇𝑇𝑝𝑝 + ∆𝑡𝑡3 . The range accuracy of the radar depends on the operating bandwidth as
demonstrated in equation (2.11) [43].
(a)
18
(b)
Figure 2.8 Pulsed radar signal (a) Transmitted signals, (b) Received signals
𝑐𝑐𝑇𝑇𝑝𝑝 (2.10)
𝑅𝑅𝑚𝑚𝑚𝑚𝑚𝑚 =
2
𝑐𝑐𝜏𝜏𝑝𝑝 𝑐𝑐 (2.11)
∆𝑅𝑅 = ≈
2 2𝐵𝐵
Where B is the transmitted bandwidth.
2. Continuous-Wave Radar (CW)
In the CW radars, the speed of the target can be estimated by calculating the Doppler frequency
which is the difference between the frequency of the transmitted signal and the frequency of the
received signal. These systems are incapable of detecting the target range, and they cannot distinguish
between objects moving toward or away from the transmitter [43]. Table 2. 4 shows the comparison
between the advantages of CW and pulsed radar.
Table 2. 4 Comparison between CW and pulsed radar

Parameter CW Radar Advantages Pulsed Radar Advantages
Hardware complexity Low High
Average Power High Low
Short range target detection Good Poor
Moving target discrimination Good Poor
Target range determination Poor Good
High transmitter receiver isolation Poor Good
3. Frequency-Modulated Continuous-Wave (FMCW) Radar
FMCW radar is the same as CW radar; in contrast, the FMCW radar can change its operating
frequency during the measurements that mean the transmission signal is modulated in frequency
19
(frequency modulation). The amplitude of the transmitted signal is proportional to the

instantaneous frequency. The generated signal is then transmitted and by measuring the round-trip
delay and the frequency difference, the detector can estimate the velocity and the distance of the
moving object. A sawtooth function is considered the simplest signal and most often used. The
transmitted and received signals of FMCW are shown in Figure 2.9(a); this figure shows the
variation of signal frequency versus time. The reflected signal is received with a delayed time ∆𝑡𝑡.
In case of the target has the relative speed to the transmitter, the received signal shift by𝑓𝑓𝑑𝑑 . The
frequency difference between the transmitted signal and the reflected signal is defined as the
intermediate frequency (IF) from the mixer output. Figure 2.9(b) illustrates the absolute difference
between the transmitted and received signals. The radar range, the relative velocity, and range
resolution can be calculated based on f1 and f2 [47].
(a) Transmitted and received FMCW signal
(b)IF signal at mixer output
Figure 2.9 Ranging with FMCW radar [47].
𝑐𝑐𝑇𝑇𝑝𝑝 𝑓𝑓1 + 𝑓𝑓2 (2.12)

𝑅𝑅 =
2𝐵𝐵 2
𝑐𝑐 𝑓𝑓1 −𝑓𝑓2
𝑣𝑣𝑟𝑟 = 2𝑓𝑓 (2.13)
𝑐𝑐 2
𝑐𝑐
∆𝑅𝑅 = 2𝐵𝐵 (2.14)
Where fc is the center frequency between f1 and f2.
20
2.2.4 Antenna Design for ARS

High gain and more efficient antennas in the automotive radar system are effective
components. Consequently, the extensive literature review reports several kinds of antennas for
automotive radar sensors. Low profile, low cost, compact size, and ease of integration are the main
keys to solve the problems of automotive radar antennas design. This section focuses on the state of
the art of radar antennas, especially 77GHz automotive radar antennas. In the automotive sensors, the
direction of arrival (DOA) should be improved [48, 49].
2.2.4.1 Horn Antenna

In the past years, the horn and parabolic antennas were often used as a first selection for the
automotive radar system, as shown in Figure 2.10. Figure 2.10 shows the early radars for the
automotive application introduced in the 1970s [47, 50]. The used antennas were huge and bulky to fit
the proposed automotive radar. Nevertheless, these radars were remarkable steps to introduce the
current professional radars. Due to the high gain of horn antennas become one of the most well-known
antennas used in the radar systems to provide narrow HPBW and high power handling, but the gain of
horn antennas is directly related to their size. So, the horn antenna with high gain is generally massive.
(a) (b)
Figure 2.10 Early automotive radar system (a) parabolic antenna[50], (b) horn antenna[47].
2.2.4.2 Lens Antenna

The lens antenna is one of the famous antennas that has been the first choice for commercial,
automotive radar applications to provide high directive beam. The lens antennas with beam switching
capability have higher opportunities to the traditional phased antenna array due to excessive metallic
and phase shifter losses, specifically at a higher frequency band (77 GHz)[51]. Recently, most
commercial sensors used either a rotationally symmetric lens fed by a small array of patches or a large
2-D patch antenna array to overcome the problems of the feeding network of the antenna array. In
2005, Colome et al. [52] introduced cylinder lens fed by microstrip patches with dual feed to increase
21
the isolation between transmitter and receiver for Bi-Static radar at 24 GHz. This antenna achieves a
gain of 15.3dB and HPBW 21.30, 37.80 for E-plane and H-plane, respectively. Based on the work
in[52], Weing et. al [53] introduced the same idea by placing the uniform series antenna array along
the focus line of the lens. Figure 2.11 depicts the configuration of the lens antenna. In the elevation
(YZ) plane, the lens is fed by a column of a series microstrip patch antenna. But in the azimuth (XZ)
plane, the receiving antenna consists of a lens added to the uniform antenna array that is placed on the
focal line of this lens. By 2017, Saleem et al. [4, 51]introduced an integrated lens antenna that consists
of 6 layers of cylindrical Luneburg, where the Luneburg lens is the lens that has varying permittivity
in the radial direction. Seventeen sources feed the lens; these sources are planar log-periodic dipoles.
In the last few days, the fourth generation of commercial sensors introduced by BOSCH uses a lens
antenna with a short focal length[54].
Figure 2.11 Design of cylinder lens antenna in the elevation plane (left) and azimuth plane (right)[53].
2.2.4.3 Planar Antennas

Several types of radar sensors use planar antennas due to their simple structure, low profile,
ease of integration and low manufacturing cost. The most common types of planar antennas that are
used in the automotive radar are series-fed microstrip patch arrays, grid antennas, slotted substrate
integrated waveguide, comb line, etc. The antenna array consists of patches that are combined in series
and/or parallel arrangements to provide high directivity. Figure 2.12 depicts two different antenna
22
arrays operating at 77 GHz[55, 56]. The first type is a four-column; each column consists of 10
rectangular series patches with an overall size of 20.43× 7.83 mm2. This antenna was printed on
Rogers substrate with a thickness of 0.127 mm and a dielectric constant of 3. The second type is printed
on the same substrate of the first type and consists of four columns, and each column consists of 12
leaf patches. One additional patch at the end of the feed line is printed for matching. Each six-leaf
patch is arranged on one side of the microstrip feed line, and they are inclined by 450 from the
microstrip feed line. This antenna has an overall size of 16.72 × 10.75 mm2. The two antenna types
have similar gain 19.8 dBi and angular width 11.60, 190 in azimuth-plane, elevation plane, respectively.
In contrast, the first type has a smaller size than the second type. Lizuka et al. [57] developed the series
antenna array, as shown in Figure 2.13. Vasanelli et al. in [58] introduce a low radar cross-section
antenna array by using an artificial magnetic conductor (AMC) set around the series antenna to reduce
the reflection in the direction of the car fascia, as depicted in Figure 2.14. The AMC has the capability
to eliminate the reflected wave from the antenna.
(a)First type (b) Second type
Figure 2.12 Series antenna array[55, 56].
23
Figure 2.13 Photograph of developed series antenna[57].
Figure 2.14 Series microstrip antenna array integrated with AMC[58].
2.3 Antennas for 5G Mobile Handset

The spectrum of the mobile communications systems below 3 GHz bands has been occupied
through the last decade. This spectrum range suffers from high shortages and cannot maintain up with
the fast growth in the communications systems rate for the near future. A necessary solution for 5G
wireless communication is the use of mmW bands to enhance communication quality [1, 9, 59-66].
There are two candidate spectrums for 5G applications, lower spectrum and higher spectrum bands.
The lower candidate bands are 3.3-4.2 GHz, 4.4-5 GHz and 6-7 GHz. Otherwise, the higher candidate
bands are (24.25-27.5 GHz, 26.5-27.5 GHz, 26.5-29.5 GHz, 27.5-28.28 GHz, and 37-39 GHz) and
some frequencies above 60.0 GHz are 53.3–66.5 GHz, 55.4–66.6 GHz, 56.6–64.8 GHz, 57.0–64.0
GHz and 57.0–65.0 GHz [67].
24
So, an antenna with wideband, stable radiation features and high gain is desired to overcome
high propagation loss within mmW bands. In recent years, some papers have been published to
introduce antennas for 5G terminal applications such as phased array [68], switchable antennas[59],
dual circular-polarized antennas and dual linear-polarized antennas [69].
2.3.1 Beam Steerable Antenna

The phased array antennas have significant challenges to implement inside the mobile handset
because of size limitations [70]; the phased array is integrated with a phase shifter and digital
beamforming to provide the same functions as a steerable antenna. Furthermore, many studies are
introduced to solve this issue which include using patch [64], slot [71] and dipole antenna arrays [72].
These methods mainly use one-dimensional linear arrays, with a fan-shaped beam pattern, on separate
substrates, positioned in the cellular handset to achieve a broad beam coverage along with high gain
within the restricted mobile size [73-80].
Figure 2.15 Beam steering idea for a talk mode and browsing mode [9]
(a) (b)
Figure 2.16 Steerable antenna array with full board (b) feeding details [9]
Lt=67.1, wt=17.28, ht=7.1, hs=1, α=600, Ls=5.43, d=5.45, ws=0.26, Lf=0.5, Lg=1(units (mm))
25
Recently, Bang et al. [9] introduced a dual-mode scenario of the proposed antenna arrays for
the talking and data modes as shown in Figure 2.15 and Figure 2.16. These two modes are introduced
for beam-steering to provide high gain and wide coverage. The suggested antenna by the author
consists of two subarrays, each array with eight rotated slot antenna elements. The antenna is printed
on the top of the upper frame and portion of the handset's back cover. According to the operating mode,
the subarray is selected. The first subarray is positioned on the handset's back cover to reduce the
effect of the antenna on the user's head and is operated when the handset is in talking mode. In contrast,
the second antenna is placed on the front frame of the handset to operate in the browsing mode or data
mode because the browsing mode needs a radiation pattern like the hemispherical. Also, Zangh et al.
[11] provided an antenna array consisting of two passive parasitic elements and one active element.
Two switches are utilized in this design to control the steering beam, as shown in Figure 2.17. Two
short circuit microstrip transmission lines with different lengths are connected with the switches. Two
printed antenna array is printed on the sidewall of the mobile chassis to provide an 1800 coverage
angle. However, this antenna provides a good coverage angle with each state of switches but suffers
from high complexity, 3D structure, and high loss in switches.
(a) (b)
(c) (d)
26
(e)
Figure 2.17 Geometry of antenna array (a) 3-D view, (b) detailed view, (c) exploded view, (d) back view with
surface copper (e) back view without surface copper (units (mm)) [11]
2.3.2 Switchable Phased Array

The phased array covers only 3600, so the switchable phased array is introduced to cover all
angles using more than one antenna. Figure 2.18 shows the different radiation pattern shapes that can
be used for 4G and 5G mobile phones. According to the shape of the radiation pattern and to cover all
angles three antennas will be needed at least for a 5G portable handset. Ojaroudiparchin et al. [64]
introduced three printed antenna arrays on the mobile substrate's three different sides, as shown in
Figure 2.19. Each sub-array consists of eight rectangular patch antennas with a half-wavelength
distance between the patches elements and fed by coaxial probes. This antenna operates from 21 GHz
to 22 GHz. To obtain the desired direction for the beam, the feed switches between the three sub-
arrays. To select between the three sub-arrays, a microwave switch kit connects the feeding source to
a power divider that connects to the phase shifter kit before each array.
27
(a) (b)
Figure 2.18 Proposed radiation pattern of mobile phone (a)4G and (b)5G [64]
Figure 2.19 Switchable phased arrays (a) side view with full PCB, (b) 3 antenna array, (c) top layer view of one
array, (d) bottom layer view of one array [64]. Wsub=55, Lsub=110, hsub=0.787, Wp=4.32, Lp=2, d=6.5, d1=4.5, W=4.574,
L=3.787, W1=1.72, L1=3, W2=0.5 (units (mm)).
2.3.3 Dual Polarized Antenna

Due to mastering of the dual-polarized antennas to introduce a solution in enhancing the
isolation and channel capacity, this makes these antennas a good candidate for MIMO smartphone
28
designs [60, 65, 69, 81-86]. In [87] Yang Li et al., introduced a hybrid eight-ports orthogonal dual-
polarized antenna for 5G smartphones; this antenna consists of 4 L-shaped monopole slot elements
and 4 C-shaped coupled fed elements. The 4 L-shaped elements are printed at the corners and the 4 C-
shaped elements are printed at the middle on a thick 1mm FR-4 substrate. This design achieves 12.5
dB, and 15 dB for the isolation and the cross-polarization, respectively. Over the past months, Zaho
et. al [88] presented a 5G/WLAN dual-polarized antenna based on the integration between inverted
cone monopole antenna and cross bow-tie antenna for VP and HP, respectively. A 90◦ phase difference
feeding network feeds the cross bow-tie antenna, so, the separated power divider and phase shifter are
introduced to be used as a feeding network. In [89], Huang et al. introduced a dual-polarized antenna
that consists of a main radiator, an annulus, and a reflector. The main radiator consists of two pairs of
differentially-driven feedlines to transmit the energy to the coplanar patch. This structure achieves 26
dB and 35dB for the isolation and the cross-polarization, respectively. Eight-ports dual-polarized
antenna array is reported in[90], the proposed antenna array is composed of four square loops, and
each loop is excited by two orthogonal fed coupled feeding strips. Recently, Parchin et al. [65]
introduced eight-port MIMO antennas using four square ring slot antennas, as shown in Figure 2.20.
Each square ring slot is fed by two microstrip lines to achieve dual-polarization. The antennas are
positioned at the four corners of the PCB to provide full coverage with dual-polarization. Two rings
are printed with each antenna and operate as parasitic elements to provide isolation between the two
ports of the antenna.
(a)3-D view (b)Front view (c)Back view

Figure 2.20 Dual-polarized square ring slot antenna [65]
2.4 Short Range Communications (SRC)

Currently, the applied wireless systems such as Bluetooth and WLANs that operate at lower
frequency bands (2.45, 5.2, 5.8 GHz) below 6 GHz suffer from low data rates. Therefore, these
technologies became unable to provide similar information rates like the wired standard rates such as
gigabit Ethernet and high-definition-multimedia-interface (HDMI). Therefore, with the advent of the
29
unlicensed 60 GHz band, researchers make the best use of the 7 GHz band from 57 GHz to 64 GHz
[91-99].
The SRC becomes one of the dominant communication facilities during the last two decay
because of its features such as Multi-Gigabit-per-second (MGbps) rate, high video, high streaming,
and networking. The availability of broadband at 60 GHz in addition to the high attenuation at this
band open the attention doors to the researchers to use this frequency for SRC such as point to point
communications and point to multi-points communications. Antenna-on-chip (AoC) integration with
other circuits will ensure low-cost SoC because of the removal of costs associated with external
antennas [100-102].
2.4.1 CMOS Technology

The AoC that is printed on CMOS technology at 60 GHz which use silicon substrate, has low
radiation efficiency and low gain. Because of the high permittivity of the silicon substrate, most of the
power is confined within the silicon substrate. So, the low resistive silicon substrates are preferred to
improve the radiation features of the AoC. In this thesis, we use 180 nm (0.18 𝜇𝜇m) CMOS technology
which consists of a base substrate from silicon with thickness of 200𝜇𝜇𝜇𝜇 and thin silicon oxide layer
(Sio2) with thickness of 10.34𝜇𝜇𝜇𝜇 as shown in Figure 2.21. The Sio2 layer consists of six metal layers,
the thickness of layers M1 to M5 is 0.53 𝜇𝜇𝜇𝜇 and the thickness of M6 is 2.34𝜇𝜇𝜇𝜇.
Figure 2.21 Structure of 0.18 µm CMOS
2.4.2 AoC Designs

Several studies have been introduced in the literature about the CMOS technology to meet the
requirements of the system on-chip such as dipole [103-105], monopole[106, 107], triangular [31, 108-
30
110], Yagi-Uda [29, 96, 98], Vivaldi [111, 112], bow-tie [32], and printed inverted F antenna (PIFA)
[113]. Figure 2.22 illustrates different configurations of on-chip antennas that are introduced in the
literature.
Dipole [103] Triangular [110] Vivaldi [111]
Yagi-Uda [96] Bowe-Tie [32] PIFA [113]

Figure 2.22 Different configurations of on-chip antenna designs.
2.5 Sub-THz Applications

Compared to traditional microwave engineering, the sub terahertz field is comparatively new. It is
now described as the sector that includes techniques, manufacturing methods and devices that operate
in the 100 GHz to 1 THz frequency band. This band includes wavelengths of submillimeter varying
from 3 mm to 300 μm [3]. Owing to countless molecular spectral lines in this band carrying vital
information for multi-gigabyte communications, astronomy as well as atmospheric research, the sub-
THz frequencies take more and more attention. Unquestionably, astronomy applications played a
significant role as a driver for the growth of THz technology and components. But soon, atmospheric
window research in these bands as well as fast advances in the processing capacities of semiconductors,
opened up a range of new THz applications for communication, defect detection, safety and biomedical
imaging. Although there are several works in the biomedical at the microwave range, they suffer from
limited resolution [114-117]. Thus the fast development seen over the last 20 years in THz frequencies
made THz a field in communications systems [22, 118-124]. Therefore, a short review of distinct THz
31
antennas is provided in this section. In this section, we focus on the antenna designs for sub-THz
applications introduced in the literature. The number of antenna designs in the range from 0.1 THz to
1 THz is limited because the fabrication process in this range is still in progress. A key factor for THz
applications is a technology platform for better performance of this band. Different technology
platforms can be used for better performance of this band, such as CMOS, flip-chip, and hybrid
techniques [3, 22, 121, 125-128].
2.5.1 THz Applications

2.5.1.1 Astronomy and Atmospheric
Certainly, astronomy and atmospheric sciences have been the first to explore the potential of
THz technology for universe exploration as well as the earth features. The powerful water vapour
absorption lines at 183 GHz and 557 GHz (as shown in Figure 1.1) have been used in universe water
exploration and potentially life. When measuring the spectral lines of these molecules gave
information about the surrounding temperature, pressure, gas velocities as well as magnetic fields
within the observational region, the absorption of THz radiation by water, oxygen and other gasses led
to many space missions.
2.5.1.2 Communications
The frequency band of 100 GHz to 1000 GHz, which has not yet been assigned for particular
uses, is of particular concern for future wireless devices with information rates of more than 100 Gb/s.
Despite the presence of different types of terahertz antenna, Koch [129] suggested waveguide horn
and planar antenna to be used for next communication systems. Where, the horn antenna provides
high efficiency and low loss. The horn-based imaging and communication systems were discussed in
detail in [120, 123] at terahertz frequency. However, there is a higher potential for the planar antenna
structure that has integration compatibility with planar systems. In [130] a 4 x 4 antenna array is printed
on polypropylene substrate with 𝜀𝜀𝑟𝑟 = 2.35 𝑎𝑎𝑎𝑎𝑎𝑎 𝑡𝑡𝑡𝑡𝑡𝑡𝑡𝑡 = 0.0005 at 300 GHz is introduced to
achieve peak gain of 18.1 dBi. In [131] three series patches are introduced to operate at 0.1 THz and
achieved 12 dBi of gain for short communication systems. This antenna is printed on thin Rogers
substrate with height 0.127 mm and dielectric constant 2.2. The same research group improved this
antenna by using polymer substrate with thickness 0.025 mm instead of Rogers substrate [132]. This
antenna achieved 16 dBi of gain but with five series patches instead of three patches in the previous
antenna. All antennas are shown in Figure 2.23.
32
(a)Planar antenna array [130] (b)Three series patches antenna array [131]
(c)Five series patches [132]

Figure 2.23 Different configurations of antennas for THz short communications.
In order to increase the antenna gain and directivity, different designs are introduced in [22, 133,
134]. In [134] an array of glass lens antennas arranged on a silicon (Si) substrate is introduced based
on planar metallic rectangular waveguide structure. In [133], the authors presented a two tapered
dielectric antenna that is designed and implemented in the suspended SOG waveguide platform. [135].
33
Figure 2.24 Real time focal plane antenna array camera [134].
Figure 2.25 Imaging system based on horn antenna at 200 GHz [123].
34
2.5.1.3 THz Imaging

The THz imaging is estimated as an attractive technique for reducing, or possibly eliminating,
the effect of atmospheric circumstances with low visibility. In THz imaging, there are several
contributing factors to attract the researchers to it such as:
• Low penetration depths of imaging in optical and IR bands are not attractive for non-destructive
applications. Although, the THz image allows elevated capacity for penetration compared to
the optical and IR.
• The THz band enhance far-field spatial resolution compared to the millimeter waves and
reduced Rayleigh scattering compared to the infrared (IR). The THz imaging is considered as
a one of non-ionizing radiation system that has benefits over ionized radiation systems such as
positron emission tomography (PET), magnetic resonance imaging, planar X-rays, and X-ray
CT scan.
• It has a low scattering loss compared to the infrared.
For optimum detection, the imaging systems should be wide-band and highly efficient. Therefore,
Trichopoulos et. al [136] introduced a real-time THz imaging camera for broadband focal plane array,
as Figure 2.24. The THz camera includes a hemispherical lens and objective lens. The hemispherical
lens consists of a planar slot antenna array that redirects the receiving THz signals to high permittivity
lens. On the other hand, Kim et.al [123] introduced a THz imaging system as shown in Figure 2.25. In
this system the horn antenna is used to improve the imaging system's resolution. The waveguide
antenna is used in this system instead of aperture antenna to reduce the power loss with small size
(1.3x0.648 mm2) to have a cutoff frequency of 115 GHz.
2.6 Conclusion
This chapter focused on the main applications in mm-wave and S-THz ranges. The literature
review for automotive radar, 5G mobile, short-range communications and S-THz antennas are
investigated. The bands, commercial sensors and the different antenna configurations that used for
automotive radar are introduced to present a guide lines in the next chapter. Furthermore, the different
techniques, antenna configurations and candidate bands of 5G applications are introduced as a one of
the master applications in mm-wave ranges. The third part in this chapter presents the on-chip
technology and its applications for short-range communications. Finally, the antenna designs in S-THz
ranges and their applications such as astronomy, communications, and imaging are introduced.
35
Chapter 3: Antennas Design for Automotive Radars
3 Chapter Three:
ANTENNA DESIGN FOR
AUTOMOTIVE RADARS
3.1 Introduction
This chapter introduces a novel antenna array for an automotive radar system based on the concept
of a virtual antenna array (VAA). The proposed VAA is introduced to serve medium-range radar
(MRR) and long-range radar (LRR) by the same antenna at the same time that has a flat shoulder shape
(FSS) radiation pattern. Furthermore, an unequal power divider is introduced to feed the VAA and its
excitation coefficient based on the method of moment and the genetic algorithm. The proposed VAA
consists of two linear antenna arrays with a total number of elements equal to 16 patches and an overall
size of 𝟑𝟑𝟑𝟑 × 𝟒𝟒𝟒𝟒 mm2 to cover the band from 23.55 GHz to 24.7 GHz. The VAA is fabricated and
measured; it presents a good agreement between all the simulated, synthesized and measured results
is found. The second part of this chapter introduces an antenna array for LRR that operates at 77 GHz.
This antenna depends on the hybrid radiator to provide wide bandwidth and high gain. Furthermore,
the AMC technique reduces the surface wave and enhances the isolation between elements. The
antenna is simulated by CST version 2018 and HFSS version 16 to verify its results.
3.2 MIMO/Phased Array

The phased array is considered one of the main important antennas widely used for radar in
civilian and military applications. The phased array consists of multiple TX antennas and multiple Rx
antennas that are often co-located. Another type of multi-antenna is the Single-Input-Multiple-Output
(SIMO) radar. The radar has one TX antenna and multiple Rx antennas. The number of Rx antennas
directly affects the angular resolution, where the angular resolution enhances with increasing the
number of Rx antennas. Whilst, this type of radar limits the increasing number of Rx antennas because
each antenna requires an individual receiver [137, 138].
Also, in Multiple-Input-Multiple-Output (MIMO) radar, the radar consists of multiple TX antennas
and multiple RX antennas. But compared to the phased-array radars, the MIMO radars have more
degrees of freedom that enhance the angular resolution, the characteristic identifiable, and give more
flexibility for transmitting beam-pattern design. Furthermore, the MIMO monostatic radars can
synthesize a more extensive virtual array to provide more enhancement of angular resolution and the
number of targets that can be detected. In summarizing, the collected MIMO can divide into two
categories: the bi-static MIMO radar, the radar that does not share any antenna between transmitter
36
and receiver, and the monostatic radar, it’s the radar that shares the same antenna array for transmitter
and receiver. As an example, the SIMO radars that have one TX antenna and 𝑀𝑀𝑇𝑇𝑇𝑇 × 𝑁𝑁𝑅𝑅𝑅𝑅 RX antennas
are equivalent to the MIMO radars that have 𝑀𝑀𝑇𝑇𝑇𝑇 TX antennas and 𝑁𝑁𝑅𝑅𝑅𝑅 RX antennas. So, the MIMO
radar achieves a low cost and high angular resolution compared to the other system [139].
(a)SPCM (b)MPCM
Figure 3.1 Modes of MIMO radar antenna.
Figure 3.2 MIMO radar model [139].
On the other hand, the MIMO radar can be divided into two kinds: the first one is the radar
with single-phase-centre-multibeam (SPCM) and the other one is the multiple-phase-centre-multibeam
(MPCM) as shown in Figure 3.1. In the SPCM, the data are split/divided according to the angular
position in the azimuth direction. This technique gives freedom to the sampling rate of each channel.
In contrast, in the MPCM, the radar transmits broad multi-beam. This technique applies in the case of
the requirement of broad beams.
Figure 3.2 shows the general case of the MIMO. In this case, we consider a MIMO radar that
has M transmit antenna to transmit M orthogonal waveforms. The echoes signals will be received by
37
N receive antenna. The antenna that used on the receive may or may not be the same antenna that used
on transmit [138, 139].
Figure 3.3 Example of MIMO system [138]
From Figure 3.3, the MIMO received signal at each receiving antenna is the weighted
summation of all the transmitted waveforms.
𝑀𝑀
𝑆𝑆𝑟𝑟𝑟𝑟 (𝑡𝑡) = � ℎ𝑛𝑛,𝑚𝑚 𝑆𝑆𝑡𝑡𝑡𝑡 (𝑡𝑡) 𝑓𝑓𝑓𝑓𝑓𝑓 𝑚𝑚 = (1,2, … … , 𝑀𝑀) 𝑎𝑎𝑎𝑎𝑎𝑎 𝑛𝑛 = (1, 2, … … . 𝑁𝑁) (3.1)
𝑚𝑚=1
Where𝑆𝑆𝑟𝑟𝑟𝑟 (𝑡𝑡) is the received signal at antenna number (n), ℎ𝑛𝑛,𝑚𝑚 is the channel response/coefficient for
channel number (n,m), and 𝑆𝑆𝑡𝑡𝑡𝑡 (𝑡𝑡) is the transmitted signal of antenna number (m). When the
transmitted signals are orthogonal, the following relation should be achieved
𝛿𝛿(𝑡𝑡), 𝑚𝑚 = 𝑚𝑚′
� 𝑆𝑆𝑡𝑡𝑡𝑡 (𝑡𝑡)𝑆𝑆𝑡𝑡𝑚𝑚′ (𝑡𝑡)∗ 𝑑𝑑𝑑𝑑 = � (3.2)
0 𝑚𝑚 ≠ 𝑚𝑚′
The transmitted signals are designed to be orthogonal signals and these signals are extracted
by M matched filter as shown in Figure 3.4 at each Rx antenna. So, the total number of extracted
signals equal MN. The channel response is assumed to be unity (ideal channel) in all analysis in this
chapter.
3.3 Linear Virtual Antenna Array

Consider the M -TX antennas and N- RX antennas that are parallel and collocated for each
other, where the distance between the antennas is 𝑑𝑑𝑡𝑡𝑡𝑡 , 𝑑𝑑𝑟𝑟𝑟𝑟 for the transmitter and the receiver,
respectively. Where the mth TX antenna is located at 𝑋𝑋𝑋𝑋, 𝑚𝑚 = 𝑑𝑑𝑡𝑡𝑡𝑡 . Also, the nth RX antenna is
located at 𝑋𝑋𝑋𝑋, 𝑛𝑛 = 𝑑𝑑𝑟𝑟𝑟𝑟 . Suggested that the proposed target in the far-field point, so we can represent
the steering vectors of the transmitted signal 𝑎𝑎(𝜃𝜃𝑠𝑠 ) and the received signal 𝑏𝑏(𝜃𝜃𝑠𝑠 ) by:
𝑎𝑎(𝜃𝜃𝑠𝑠 ) = �𝑒𝑒 𝑗𝑗𝑘𝑘𝑘𝑘𝑡𝑡1 sin 𝜃𝜃𝑠𝑠 , 𝑒𝑒 𝑗𝑗𝑘𝑘𝑘𝑘𝑡𝑡2 sin 𝜃𝜃𝑠𝑠 , … … … . , 𝑒𝑒 𝑗𝑗𝑘𝑘𝑘𝑘𝑡𝑡𝑡𝑡 sin 𝜃𝜃𝑠𝑠 �
𝑇𝑇 (3.3)
38
𝑏𝑏(𝜃𝜃𝑠𝑠 ) = �𝑒𝑒 𝑗𝑗𝑑𝑑𝑟𝑟1 sin 𝜃𝜃𝑠𝑠 , 𝑒𝑒 𝑗𝑗𝑑𝑑𝑟𝑟2 sin 𝜃𝜃𝑠𝑠 , … … … . , 𝑒𝑒 𝑗𝑗𝑘𝑘𝑘𝑘𝑟𝑟𝑟𝑟 sin 𝜃𝜃𝑠𝑠 �
𝑇𝑇 (3.4)
The MIMO received signal can be expressed as

𝑆𝑆𝑟𝑟 (𝑡𝑡) = ℎ 𝑆𝑆𝑡𝑡 (𝑡𝑡)𝑎𝑎(𝜃𝜃𝑠𝑠 )𝑏𝑏 𝑇𝑇 (𝜃𝜃𝑠𝑠 ) (3.5)
𝑣𝑣(𝜃𝜃𝑠𝑠 ) = 𝑎𝑎(𝜃𝜃𝑠𝑠 )𝑏𝑏 𝑇𝑇 (𝜃𝜃𝑠𝑠 ) (3.6)
𝑒𝑒 𝑗𝑗𝑘𝑘𝑘𝑘𝑡𝑡1 sin 𝜃𝜃𝑠𝑠
⎡ 𝑗𝑗𝑘𝑘𝑘𝑘𝑡𝑡2 sin 𝜃𝜃𝑠𝑠 ⎤
⎢ 𝑒𝑒 ⎥
)
𝑣𝑣(𝜃𝜃𝑠𝑠 = ⎢ ⋮ ⎥ �𝑒𝑒 𝑗𝑗𝑘𝑘𝑘𝑘𝑟𝑟1 sin 𝜃𝜃𝑠𝑠 , 𝑒𝑒 𝑗𝑗𝑘𝑘𝑘𝑘𝑟𝑟2 sin 𝜃𝜃𝑠𝑠 , … … … . , 𝑒𝑒 𝑗𝑗𝑘𝑘𝑘𝑘𝑟𝑟𝑟𝑟 sin 𝜃𝜃𝑠𝑠 �1×𝑁𝑁 (3.7)
⎢ ⋮ ⎥
⎣𝑒𝑒 𝑗𝑗𝑘𝑘𝑘𝑘𝑡𝑡𝑡𝑡 sin 𝜃𝜃𝑠𝑠 ⎦𝑀𝑀×1
𝑒𝑒 𝑗𝑗𝑗𝑗(𝑑𝑑𝑡𝑡1 +𝑑𝑑𝑟𝑟1 ) sin 𝜃𝜃𝑠𝑠 𝑒𝑒 𝑗𝑗𝑗𝑗(𝑑𝑑𝑡𝑡1 +𝑑𝑑𝑟𝑟2 ) sin 𝜃𝜃𝑠𝑠 ⋯ 𝑒𝑒 𝑗𝑗𝑗𝑗(𝑑𝑑𝑡𝑡1 +𝑑𝑑𝑟𝑟𝑟𝑟 ) sin 𝜃𝜃𝑠𝑠
𝑗𝑗𝑗𝑗(𝑑𝑑𝑡𝑡2 +𝑑𝑑𝑟𝑟1 ) sin 𝜃𝜃𝑠𝑠
𝑣𝑣(𝜃𝜃𝑠𝑠 ) = � 𝑒𝑒 𝑒𝑒 𝑗𝑗𝑗𝑗(𝑑𝑑𝑟𝑟2 +𝑑𝑑𝑡𝑡2 ) sin 𝜃𝜃𝑠𝑠 … 𝑒𝑒 𝑗𝑗𝑗𝑗(𝑑𝑑𝑡𝑡2 +𝑑𝑑𝑟𝑟𝑟𝑟 ) sin 𝜃𝜃𝑠𝑠 �
(3.8)
⋮ ⋮ … ⋮
𝑗𝑗𝑗𝑗(𝑑𝑑𝑡𝑡𝑡𝑡 +𝑑𝑑𝑟𝑟1 ) sin 𝜃𝜃𝑠𝑠 𝑗𝑗𝑗𝑗(𝑑𝑑𝑡𝑡𝑡𝑡 +𝑑𝑑𝑟𝑟2 ) sin 𝜃𝜃𝑠𝑠 𝑗𝑗𝑗𝑗(𝑑𝑑𝑡𝑡𝑡𝑡 +𝑑𝑑𝑁𝑁 ) sin 𝜃𝜃𝑠𝑠
𝑒𝑒 𝑒𝑒 … 𝑒𝑒 𝑀𝑀×𝑁𝑁
In other words, the steering vector can be expressed as the Kronecker product between the two
steering vectors 𝑣𝑣(𝜃𝜃𝑠𝑠 ) = 𝑎𝑎(𝜃𝜃𝑠𝑠 )⨂𝑏𝑏(𝜃𝜃𝑠𝑠 ), where ⨂ denotes the Kronecker Product [139].
In this example, we introduce an example of Kronecker Product:
Assume two matrices A, B with any dimensions
𝑎𝑎11 ⋯ 𝑎𝑎1𝑛𝑛 𝑏𝑏11 ⋯ 𝑏𝑏1𝑛𝑛
A=� ⋮ ⋱ ⋮ � , 𝐵𝐵 = � ⋮ ⋱ ⋮ �
𝑎𝑎𝑛𝑛1 ⋯ 𝑎𝑎𝑛𝑛𝑛𝑛 𝑏𝑏𝑛𝑛1 ⋯ 𝑏𝑏𝑛𝑛𝑛𝑛
(3.9)
𝑎𝑎11 𝐵𝐵 ⋯ 𝑎𝑎1𝑛𝑛 𝐵𝐵
𝐴𝐴⨂𝐵𝐵 = � ⋮ ⋱ ⋮ �
𝑎𝑎𝑛𝑛1 𝐵𝐵 ⋯ 𝑎𝑎𝑛𝑛𝑛𝑛 𝐵𝐵
To study different cases of virtual MIMO, we assume that 𝑑𝑑𝑡𝑡 = 𝛽𝛽𝛽𝛽𝑟𝑟
3.3.1 Equal Distance

• Case 1: Uniform Array
We assume that the antenna is linear uniform with equal distances and an equal number of
elements between TX antennas and Rx antennas. So, in this case, 𝛽𝛽 = 1, 𝑀𝑀 = 𝑁𝑁 = 𝐿𝐿, 𝑑𝑑𝑡𝑡 = 𝑑𝑑𝑟𝑟 = 𝑑𝑑.
We assume that the first element is the reference element at origin for TX and RX to solve this case.
The steering vector of the TX is the same steering vector of the RX and can be expressed as:
𝑇𝑇
𝑎𝑎(𝜃𝜃𝑠𝑠 ) = 𝑏𝑏(𝜃𝜃𝑠𝑠 ) = �1, 𝑒𝑒 𝑗𝑗𝑗𝑗𝑗𝑗 sin 𝜃𝜃𝑠𝑠 , 𝑒𝑒 𝑗𝑗2𝑘𝑘𝑘𝑘 sin 𝜃𝜃𝑠𝑠 , ⋯ , 𝑒𝑒 𝑗𝑗(𝐿𝐿−1)𝑘𝑘𝑘𝑘 sin 𝜃𝜃𝑠𝑠 � (3.10)
39
1 𝑒𝑒 𝑗𝑗𝑗𝑗𝑗𝑗 sin 𝜃𝜃𝑠𝑠 ⋯ 𝑒𝑒 𝑗𝑗(𝐿𝐿−1)𝑘𝑘𝑘𝑘 sin 𝜃𝜃𝑠𝑠

𝑗𝑗𝑗𝑗𝑗𝑗 sin 𝜃𝜃𝑠𝑠
𝑒𝑒 𝑗𝑗2𝑘𝑘𝑘𝑘 sin 𝜃𝜃𝑠𝑠 𝑒𝑒 𝑗𝑗𝑗𝑗𝑗𝑗𝑗𝑗 sin 𝜃𝜃𝑠𝑠 � (3.11)
𝑣𝑣(𝜃𝜃𝑠𝑠 ) = � 𝑒𝑒 …
⋮ ⋮ … ⋮
𝑒𝑒 𝑗𝑗(𝐿𝐿−1)𝑘𝑘𝑘𝑘 sin 𝜃𝜃𝑠𝑠 𝑒𝑒 𝑗𝑗(𝐿𝐿)𝑘𝑘𝑘𝑘 sin 𝜃𝜃𝑠𝑠 … 𝑒𝑒 𝑗𝑗(2𝐿𝐿−2)𝑘𝑘𝑘𝑘 sin 𝜃𝜃𝑠𝑠 𝐿𝐿×𝐿𝐿
From equation (3.11), we noted that the number of effective virtual phase centers (EVFC) =2L-
1. The following expression in (3.12) introduces the length/number of EVPC.
𝟏𝟏 𝟐𝟐 … 𝐋𝐋 − 𝟏𝟏 𝐋𝐋 𝐋𝐋 − 𝟏𝟏 … 𝟐𝟐 𝟏𝟏 𝑳𝑳 = 𝑴𝑴
𝟏𝟏 … ⋮ … 𝟏𝟏 𝑳𝑳 = 𝑴𝑴 − 𝟏𝟏
.
⋱ … ⋮ … .. ⋮
.
⋱ ⋮ .. ⋮ (3.12)
𝟏𝟏 𝟐𝟐 𝟑𝟑 𝟐𝟐 𝟏𝟏 𝑳𝑳 = 𝟑𝟑
𝟏𝟏 𝟐𝟐 𝟏𝟏 𝑳𝑳 = 𝟐𝟐
𝟏𝟏 𝑳𝑳 = 𝟏𝟏
For more investigation, we introduce this example with the assumption that 𝑀𝑀 = 𝑁𝑁 = 4. We
assume that the positions of the transmitter, receiver, virtual antenna array are 𝑋𝑋𝑡𝑡 , 𝑋𝑋𝑟𝑟 , 𝑋𝑋𝑣𝑣 , respectively.
𝑋𝑋𝑡𝑡 = [1111], 𝑋𝑋𝑟𝑟 = [1111] are the position of Tx and Rx elements, can be considered as a function of
distance as 𝑋𝑋𝑡𝑡 = 𝑋𝑋𝑟𝑟 = [0, 𝑑𝑑, 2𝑑𝑑, 3𝑑𝑑]. So,
1 𝑒𝑒 𝑗𝑗𝑗𝑗𝑗𝑗 sin 𝜃𝜃𝑠𝑠 𝑒𝑒 𝑗𝑗2𝑘𝑘𝑘𝑘 sin 𝜃𝜃𝑠𝑠 𝑒𝑒 𝑗𝑗3𝑘𝑘𝑘𝑘 sin 𝜃𝜃𝑠𝑠
𝑗𝑗𝑗𝑗𝑗𝑗 sin 𝜃𝜃𝑠𝑠
𝑣𝑣(𝜃𝜃𝑠𝑠 ) = � 𝑒𝑒𝑗𝑗2𝑘𝑘𝑘𝑘 sin 𝜃𝜃 𝑒𝑒 𝑗𝑗2𝑘𝑘𝑘𝑘 sin 𝜃𝜃𝑠𝑠 𝑒𝑒 𝑗𝑗3𝑘𝑘𝑘𝑘 sin 𝜃𝜃𝑠𝑠 𝑒𝑒 𝑗𝑗4𝑘𝑘𝑘𝑘 sin 𝜃𝜃𝑠𝑠 �
𝑒𝑒 𝑠𝑠 𝑒𝑒 𝑗𝑗3𝑘𝑘𝑘𝑘 sin 𝜃𝜃𝑠𝑠 𝑒𝑒 𝑗𝑗4𝑘𝑘𝑘𝑘 sin 𝜃𝜃𝑠𝑠 𝑒𝑒 𝑗𝑗5𝑘𝑘𝑘𝑘 sin 𝜃𝜃𝑠𝑠
𝑒𝑒 𝑗𝑗3𝑘𝑘𝑘𝑘 sin 𝜃𝜃𝑠𝑠
𝑒𝑒 𝑗𝑗4𝑘𝑘𝑘𝑘 sin 𝜃𝜃𝑠𝑠 𝑒𝑒 𝑗𝑗5𝑘𝑘𝑘𝑘 sin 𝜃𝜃𝑠𝑠 𝑒𝑒 𝑗𝑗6𝑘𝑘𝑘𝑘 sin 𝜃𝜃𝑠𝑠 4×4 (3.13)
= [1, 2𝑒𝑒 𝑗𝑗𝑗𝑗𝑗𝑗 sin 𝜃𝜃𝑠𝑠 , 3𝑒𝑒 𝑗𝑗2𝑘𝑘𝑘𝑘 sin 𝜃𝜃𝑠𝑠 , 4𝑒𝑒 𝑗𝑗𝑗𝑗𝑗𝑗 sin 𝜃𝜃𝑠𝑠 , 3𝑒𝑒 𝑗𝑗𝑗𝑗𝑗𝑗 sin 𝜃𝜃𝑠𝑠 , 2𝑒𝑒 𝑗𝑗𝑗𝑗𝑗𝑗 sin 𝜃𝜃𝑠𝑠 , 𝑒𝑒 𝑗𝑗𝑗𝑗𝑗𝑗 sin 𝜃𝜃𝑠𝑠 ]
So, the virtual vector can be expressed as 𝑋𝑋𝑣𝑣 = [1, 2, 3, 4, 3, 2, 1]. Figure 3.5 depicts the corresponding
EVPC of this example. We note that the number of virtual elements=7.
3
T
x
R
x
2.5
2
Amplitude
1.5
0.5
0
1 1.5 2 2.5 3 3.5 4 4.5 5 5.5 6
Position
(a)Positions of TX and RX elements
40
4
Amplitude
0
1 2 3 4 5 6 7 8 9
Position
(b)EVPC
Figure 3.4 Virtual phase center of uniform array (M=4, N=4)
• Case 2: Non-uniform array
We assume that the antenna is linear uniform with an equal number of elements between Tx
antennas and Rx antennas. So, in this case, 𝛽𝛽 = 1, 𝑀𝑀 = 𝑁𝑁 = 𝐿𝐿, 𝑑𝑑𝑡𝑡𝑡𝑡 = 𝑑𝑑𝑟𝑟𝑟𝑟 = 𝑑𝑑𝑚𝑚 . The steering vector
of the TX is the same steering vector of the RX and can be expressed as:
𝑇𝑇
𝑎𝑎(𝜃𝜃𝑠𝑠 ) = 𝑏𝑏(𝜃𝜃𝑠𝑠 ) = �𝑒𝑒 𝑗𝑗𝑘𝑘𝑘𝑘1 sin 𝜃𝜃𝑠𝑠 , 𝑒𝑒 𝑗𝑗𝑗𝑗𝑑𝑑2 sin 𝜃𝜃𝑠𝑠 , ⋯ , 𝑒𝑒 𝑗𝑗𝑘𝑘𝑘𝑘𝑀𝑀 sin 𝜃𝜃𝑠𝑠 � (3.14)
𝑒𝑒 𝑗𝑗𝑗𝑗(2𝑑𝑑1 ) sin 𝜃𝜃𝑠𝑠 𝑒𝑒 𝑗𝑗𝑗𝑗(𝑑𝑑1 +𝑑𝑑2 ) sin 𝜃𝜃𝑠𝑠 ⋯ 𝑒𝑒 𝑗𝑗𝑗𝑗(𝑑𝑑1 +𝑑𝑑𝑀𝑀 ) sin 𝜃𝜃𝑠𝑠
𝑗𝑗𝑘𝑘(𝑑𝑑2 +𝑑𝑑1 ) sin 𝜃𝜃𝑠𝑠
𝑣𝑣(𝜃𝜃𝑠𝑠 ) = � 𝑒𝑒 𝑒𝑒 𝑗𝑗𝑗𝑗(𝑑𝑑2 +𝑑𝑑2 ) sin 𝜃𝜃𝑠𝑠 … 𝑒𝑒 𝑗𝑗𝑗𝑗(𝑑𝑑2 +𝑑𝑑𝑀𝑀 ) sin 𝜃𝜃𝑠𝑠 � (3.15)
⋮ ⋮ … ⋮
𝑗𝑗𝑗𝑗(𝑑𝑑𝑀𝑀 +𝑑𝑑1 ) sin 𝜃𝜃𝑠𝑠 𝑗𝑗𝑗𝑗(𝑑𝑑𝑀𝑀 +𝑑𝑑2 ) sin 𝜃𝜃𝑠𝑠 𝑗𝑗𝑗𝑗(2𝑑𝑑𝑀𝑀 ) sin 𝜃𝜃𝑠𝑠
𝑒𝑒 𝑒𝑒 … 𝑒𝑒
It can be detected that the maximum number of virtual phase that can be achieved in the case of non-
uniform array 𝐿𝐿𝑣𝑣 = 𝐿𝐿(𝐿𝐿 + 1)/2. As example suppose that M=4, N=4. In this case we assume that
the position of transmitter and receiver are𝑋𝑋𝑡𝑡 , 𝑋𝑋𝑟𝑟 , respectively. 𝑋𝑋𝑡𝑡 =[0 1 0 1 0 0 1 0 0 0 1], 𝑋𝑋𝑟𝑟 =[0 1 0 1
0 0 1 0 0 0 1], the position of Tx and Rx elements can be considered as 𝑋𝑋𝑡𝑡 = 𝑋𝑋𝑟𝑟 = [𝑑𝑑, 3𝑑𝑑, 6𝑑𝑑, 10𝑑𝑑]
𝑒𝑒 𝑗𝑗𝑗𝑗(2𝑑𝑑) sin 𝜃𝜃𝑠𝑠 𝑒𝑒 𝑗𝑗𝑗𝑗(4𝑑𝑑) sin 𝜃𝜃𝑠𝑠 𝑒𝑒 𝑗𝑗𝑗𝑗(7𝑑𝑑) sin 𝜃𝜃𝑠𝑠 𝑒𝑒 𝑗𝑗𝑗𝑗(11𝑑𝑑) sin 𝜃𝜃𝑠𝑠
𝑗𝑗𝑗𝑗(4𝑑𝑑) sin 𝜃𝜃𝑠𝑠
𝑣𝑣(𝜃𝜃𝑠𝑠 ) = � 𝑒𝑒 𝑗𝑗𝑗𝑗(7𝑑𝑑) sin 𝜃𝜃 𝑒𝑒 𝑗𝑗𝑗𝑗(6𝑑𝑑) sin 𝜃𝜃𝑠𝑠 𝑒𝑒 𝑗𝑗𝑗𝑗(9𝑑𝑑) sin 𝜃𝜃𝑠𝑠 𝑒𝑒 𝑗𝑗𝑗𝑗(13𝑑𝑑) sin 𝜃𝜃𝑠𝑠 �
𝑒𝑒 𝑠𝑠 𝑒𝑒 𝑗𝑗𝑗𝑗(9𝑑𝑑) sin 𝜃𝜃𝑠𝑠 𝑒𝑒 𝑗𝑗𝑗𝑗(12𝑑𝑑) sin 𝜃𝜃𝑠𝑠 𝑒𝑒 𝑗𝑗𝑗𝑗(16𝑑𝑑) sin 𝜃𝜃𝑠𝑠
𝑒𝑒 𝑗𝑗𝑗𝑗(11𝑑𝑑) sin 𝜃𝜃𝑠𝑠
𝑒𝑒 𝑗𝑗𝑗𝑗(13𝑑𝑑) sin 𝜃𝜃𝑠𝑠 𝑒𝑒 𝑗𝑗𝑗𝑗(16𝑑𝑑) sin 𝜃𝜃𝑠𝑠 𝑒𝑒 𝑗𝑗𝑗𝑗(20𝑑𝑑) sin 𝜃𝜃𝑠𝑠 (3.16)
𝑒𝑒 𝑗𝑗𝑗𝑗(2𝑑𝑑) sin 𝜃𝜃𝑠𝑠 , 2𝑒𝑒 𝑗𝑗𝑗𝑗(4𝑑𝑑) sin 𝜃𝜃𝑠𝑠 , 1𝑒𝑒 𝑗𝑗𝑗𝑗(6𝑑𝑑) sin 𝜃𝜃𝑠𝑠 , 2𝑒𝑒 𝑗𝑗𝑗𝑗(7𝑑𝑑) sin 𝜃𝜃𝑠𝑠 , 2𝑒𝑒 𝑗𝑗𝑗𝑗(9𝑑𝑑) sin 𝜃𝜃𝑠𝑠 , 2𝑒𝑒 𝑗𝑗𝑗𝑗(11𝑑𝑑) sin 𝜃𝜃𝑠𝑠 ,
=� �
𝑒𝑒 𝑗𝑗𝑗𝑗(12𝑑𝑑) sin 𝜃𝜃𝑠𝑠 , 𝑒𝑒 𝑗𝑗𝑗𝑗(13𝑑𝑑) sin 𝜃𝜃𝑠𝑠 , 𝑒𝑒 𝑗𝑗𝑗𝑗(14𝑑𝑑) sin 𝜃𝜃𝑠𝑠 , 2𝑒𝑒 𝑗𝑗𝑗𝑗(16𝑑𝑑) sin 𝜃𝜃𝑠𝑠 , 𝑒𝑒 𝑗𝑗𝑗𝑗(20𝑑𝑑) sin 𝜃𝜃𝑠𝑠
The position of virtual elements can be expressed as 𝑋𝑋𝑣𝑣 = [0 0 1 0 2 0 1 2 0 2 0 2 1 1 0 0 2 0 0 0 1],
So, 𝐿𝐿𝑣𝑣 = 10. Figure 3.5 (a) shows the positions of TX and RX antennas
41
3
T
x
R
x
2.5
2
Amplitude
1.5
0.5
0
0 2 4 6 8 10 12
Position
(a)positions of TX and RX antennas

4
3.5
2.5
Amplitude
1.5
0.5
0
2 4 6 8 10 12 14 16 18 20 22
Position
(b) EVPC
Figure 3.5 virtual phase center of non-uniform elements (M=N)
3.3.2 Unequal Distance

In this section we focus on case of 𝛽𝛽 = 𝑁𝑁;
We assume that, the transmitter and receiver are overlapped and start at the same point. 𝛽𝛽 = 𝑁𝑁,
number of the transmitter antennas equal 𝑀𝑀, and the number of receiver antennas equal N. The
distance between receiver elements is denoted by d and distance between transmitter elements
denotes by Nd.
𝑇𝑇
𝑎𝑎(𝜃𝜃𝑠𝑠 ) = �1, 𝑒𝑒 𝑗𝑗𝑗𝑗𝑗𝑗𝑗𝑗 sin 𝜃𝜃𝑠𝑠 , … … … . , 𝑒𝑒 𝑗𝑗(𝑀𝑀−1)𝑁𝑁𝑁𝑁𝑁𝑁 sin 𝜃𝜃𝑠𝑠 � (3.17)
𝑇𝑇
𝑏𝑏(𝜃𝜃𝑠𝑠 ) = �1, 𝑒𝑒 𝑗𝑗2𝑘𝑘𝑘𝑘 sin 𝜃𝜃𝑠𝑠 , … … … . , 𝑒𝑒 𝑗𝑗(𝑁𝑁−1)𝑘𝑘𝑘𝑘 sin 𝜃𝜃𝑠𝑠 � (3.18)
1 𝑒𝑒 𝑗𝑗𝑗𝑗𝑗𝑗 sin 𝜃𝜃𝑠𝑠 ⋯ 𝑒𝑒 𝑗𝑗(𝑁𝑁−1)𝑘𝑘𝑘𝑘 sin 𝜃𝜃𝑠𝑠

𝑗𝑗𝑗𝑗𝑗𝑗𝑗𝑗 sin 𝜃𝜃𝑠𝑠 𝑗𝑗(𝑁𝑁+1)𝑘𝑘𝑘𝑘 sin 𝜃𝜃𝑠𝑠
𝑣𝑣(𝜃𝜃𝑠𝑠 ) = � 𝑒𝑒 𝑒𝑒 ⋯ 𝑒𝑒 𝑗𝑗(2𝑁𝑁−1)𝑘𝑘𝑘𝑘 sin 𝜃𝜃𝑠𝑠 �
(3.19)
⋮ ⋮ ⋮ ⋮
𝑒𝑒 𝑗𝑗((𝑀𝑀−1)𝑁𝑁)𝑘𝑘𝑘𝑘 sin 𝜃𝜃𝑠𝑠 𝑒𝑒 𝑗𝑗((𝑀𝑀−1)𝑁𝑁+1)𝑘𝑘𝑘𝑘 sin 𝜃𝜃𝑠𝑠 ⋯ 𝑒𝑒 𝑗𝑗(𝑁𝑁𝑁𝑁−1)𝑘𝑘𝑘𝑘 sin 𝜃𝜃𝑠𝑠
𝑀𝑀×𝑁𝑁
42
As an example, we assume that M=3, N=4, the positions of transmitter and receiver are𝑋𝑋𝑡𝑡 , 𝑋𝑋𝑟𝑟
respectively. 𝑋𝑋𝑡𝑡 =[1 0 0 0 1 0 0 0 1 0 0 0 1], 𝑋𝑋𝑟𝑟 =[1 1 1 1], the positions of Tx and Rx elements can be
consider as 𝑋𝑋𝑡𝑡 = [0, 4𝑑𝑑, 8𝑑𝑑], 𝑋𝑋𝑟𝑟 = [0, 𝑑𝑑, 𝑑𝑑, 3𝑑𝑑] with the first element as reference in Tx and Rx.
1 𝑒𝑒 𝑗𝑗𝑗𝑗𝑗𝑗 sin 𝜃𝜃𝑠𝑠 𝑒𝑒 𝑗𝑗2𝑘𝑘𝑘𝑘 sin 𝜃𝜃𝑠𝑠 𝑒𝑒 𝑗𝑗3𝑘𝑘𝑘𝑘 sin 𝜃𝜃𝑠𝑠
𝑣𝑣(𝜃𝜃𝑠𝑠 ) = �𝑒𝑒 𝑗𝑗4𝑘𝑘𝑘𝑘 sin 𝜃𝜃𝑠𝑠 𝑒𝑒 𝑗𝑗5𝑘𝑘𝑘𝑘 sin 𝜃𝜃𝑠𝑠 𝑒𝑒 𝑗𝑗6𝑘𝑘𝑘𝑘 sin 𝜃𝜃𝑠𝑠 𝑒𝑒 𝑗𝑗7𝑘𝑘𝑘𝑘 sin 𝜃𝜃𝑠𝑠 � (3.20)
𝑒𝑒 𝑗𝑗8𝑘𝑘𝑘𝑘 sin 𝜃𝜃𝑠𝑠 𝑒𝑒 𝑗𝑗9𝑘𝑘𝑘𝑘 sin 𝜃𝜃𝑠𝑠 𝑒𝑒 𝑗𝑗10𝑘𝑘𝑘𝑘 sin 𝜃𝜃𝑠𝑠 𝑒𝑒 𝑗𝑗11𝑘𝑘𝑘𝑘 sin 𝜃𝜃𝑠𝑠
So, the positon’s vector of the virtual elements can be expressed as 𝑋𝑋𝑣𝑣 = [1 1 1 1 1 1 1 1 1 1 1 1 ]
3
T
x
R
x
2.5
2
Amplitude
1.5
0.5
0
0 1 2 3 4 5 6 7 8 9 10 11
Position
(a)
3
2.5
2
Amplitude
1.5
0.5
0
0 2 4 6 8 10 12 14
Position
(b)
Figure 3.6 EVPC of M=3, N=4, 𝜷𝜷 = 𝟒𝟒 overlapped
The same number of EVPC can be achieved with no-overlapped as shown in Figure 3.7.
43
3
T
x
R
x
2.5
2
Amplitude
1.5
0.5
0
0 2 4 6 8 10 12 14 16
Position
(a)
3
2.5
2
Amplitude
1.5
0.5
0
0 2 4 6 8 10 12 14 16 18
Position
(b)
Figure 3.7 EVPC of M=3, N=4, 𝜷𝜷 = 𝟒𝟒 No-overlapped
3.3.3 ULA Vs. VAA

In this case study, the comparison between conventional antenna array and Virtual array is introduced.
We assume that we have a uniform linear array (ULA). The antenna array consists of 8 isotropic
elements on Z-axis. The array factor can be considered as:
𝐴𝐴𝐴𝐴𝐶𝐶 = 1 + 𝑒𝑒 −𝑗𝑗𝑗𝑗𝑗𝑗𝑗𝑗𝑗𝑗𝑗𝑗(𝜃𝜃) + 𝑒𝑒 −𝑗𝑗2𝑘𝑘𝑘𝑘𝑘𝑘𝑘𝑘𝑘𝑘(𝜃𝜃) + 𝑒𝑒 −𝑗𝑗3𝑘𝑘𝑘𝑘𝑘𝑘𝑘𝑘𝑘𝑘(𝜃𝜃) + 𝑒𝑒 −𝑗𝑗4𝑘𝑘𝑘𝑘𝑘𝑘𝑘𝑘𝑘𝑘(𝜃𝜃) + 𝑒𝑒 −𝑗𝑗5𝑘𝑘𝑘𝑘𝑘𝑘𝑘𝑘𝑘𝑘(𝜃𝜃)
(3.21)
+ 𝑒𝑒 −𝑗𝑗6𝑘𝑘𝑘𝑘𝑘𝑘𝑘𝑘𝑘𝑘(𝜃𝜃) + 𝑒𝑒 −𝑗𝑗7𝑘𝑘𝑘𝑘𝑘𝑘𝑘𝑘𝑘𝑘(𝜃𝜃)
The Virtual MIMO (VMIMO) provides the same effect of 8 elements with only 6 elements
with the following specification 𝐿𝐿𝑣𝑣 = 𝑀𝑀𝑀𝑀, 𝑑𝑑𝑡𝑡 = 𝑁𝑁𝑑𝑑𝑟𝑟 , where 𝐿𝐿𝑣𝑣 is the total number of elements in case
of VMIMO. The virtual array factor can be expressed as
44
𝐴𝐴𝐴𝐴𝑉𝑉 = �1 + 𝑒𝑒 −𝑗𝑗4𝑘𝑘𝑘𝑘𝑘𝑘𝑘𝑘𝑘𝑘(𝜃𝜃) � (1 + 𝑒𝑒 −𝑗𝑗𝑗𝑗𝑗𝑗𝑗𝑗𝑗𝑗𝑗𝑗(𝜃𝜃) + 𝑒𝑒 −𝑗𝑗2𝑘𝑘𝑘𝑘𝑘𝑘𝑘𝑘𝑘𝑘(𝜃𝜃) + 𝑒𝑒 −𝑗𝑗3𝑘𝑘𝑘𝑘𝑘𝑘𝑘𝑘𝑘𝑘(𝜃𝜃) ) (3.22)

𝐴𝐴𝐴𝐴𝐶𝐶 = 𝐴𝐴𝐴𝐴𝑉𝑉 If and only if 𝑑𝑑𝑡𝑡 = 𝑁𝑁𝑑𝑑𝑟𝑟
Figure 3.8 shows the comparison between the array factor of uniform linear array (ULA) and VMIMO.
We noted that the AF of the ULA and VMIMO are the same under the previous conditions.
Figure 3.8 Array factor of linear uniform array and VMIMO

The relation between the number of antennas of LAA and VMIMO can be described as following:
𝑁𝑁 ∗ 𝑀𝑀|𝑈𝑈𝑈𝑈𝑈𝑈 = 𝑁𝑁 + 𝑀𝑀|𝑉𝑉𝑉𝑉𝑉𝑉𝑉𝑉𝑉𝑉 If and only If 𝑑𝑑𝑡𝑡 = 𝑁𝑁𝑑𝑑𝑟𝑟 in VMIMO case

(3.23)
3.4 Virtual Antenna Array Design

The basic concept of VAA is introduced in [137-141]. Suppose that an M element transmitting
linear array distributed along the x-direction and an N element receiving linear antenna array
distributed along the y-direction. This distribution constitutes the basic building elements of the virtual
antenna array. The concept of a virtual antenna array is illustrated in Figure 3.9. It appears clearly in
radar applications where the received signal depends mainly on the multiplication of the radiation
patterns for the transmitting and receiving antennas. As will be illustrated, the radiation pattern of the
planar antenna shown in the right-hand side is equivalent to the multiplication of the radiation patterns
of the two linear antenna arrays in the left-hand side of Figure 3.9.
45
Figure 3.9 Virtual Antenna array concept

The array factor of the transmitter linear antenna array is calculated according to equation (3.24):
𝐴𝐴𝐴𝐴𝑇𝑇 (𝜃𝜃, ∅) = ∑𝑀𝑀
𝑚𝑚=1 𝑎𝑎𝑚𝑚 𝑒𝑒
𝑗𝑗𝑗𝑗𝑑𝑑𝑚𝑚 sin 𝜃𝜃 cos ∅ (3.24)
Where 𝑎𝑎𝑚𝑚 is the mth complex excitation coefficient. The gain of the transmitting antenna is
proportional to the square of the array factor such that
� 𝑇𝑇 (𝜃𝜃, ∅)�
𝐺𝐺𝑇𝑇 (𝜃𝜃, ∅) = 𝛼𝛼𝑇𝑇 �𝐴𝐴𝐴𝐴
2 (3.25)
� 𝑇𝑇 (𝜃𝜃, ∅) is the normalized transmitter array factor and similarly, the receiver array
Where 𝐴𝐴𝐴𝐴
factor is written as:
𝐴𝐴𝐴𝐴𝑅𝑅 (𝜃𝜃, ∅) = ∑𝑁𝑁
𝑛𝑛=1 𝑎𝑎𝑛𝑛 𝑒𝑒
𝑗𝑗𝑗𝑗𝑑𝑑𝑛𝑛 sin 𝜃𝜃 sin ∅ (3.26)
And the receiver gain is also written as

� 𝑅𝑅 (𝜃𝜃, ∅)�
𝐺𝐺𝑅𝑅 (𝜃𝜃, ∅) = 𝛼𝛼𝑅𝑅 �𝐴𝐴𝐴𝐴
2 (3.27)
� 𝑅𝑅 (𝜃𝜃, ∅) is the normalized receiving array factor.

𝐴𝐴𝐴𝐴
Suppose that an object of radar cross section 𝜎𝜎 is positioned in front of the transceiver system then the
receiving power is calculated according to [49] so that:
𝑃𝑃𝑡𝑡 𝐺𝐺𝑅𝑅 (𝜃𝜃, ∅)𝐺𝐺𝑇𝑇 (𝜃𝜃, ∅)𝜎𝜎𝜆𝜆2 (3.28)
𝑃𝑃𝑟𝑟𝑟𝑟 = ℎ(𝜃𝜃, ∅)
(4𝜋𝜋)3 𝑅𝑅 4
Where 𝑃𝑃𝑟𝑟 : Received Power in watts, 𝑃𝑃𝑡𝑡 :Peak transmitted power in watts, 𝐺𝐺𝑇𝑇 : Transmitter Gain,
𝐺𝐺𝑅𝑅 : Receiver Gain, 𝜆𝜆: Wavelength (m), 𝜎𝜎: RCS of the target (m2), R: Range between radar and target
(m) and ℎ(𝜃𝜃, ∅) is the channel response/coefficient for the wave impinging from the transmitter and
reflected from the scattered under investigation and then back to the receiving point.
In case of normal radar operation where one antenna is utilized for both transmission and reception,
the used antenna may be a planar array of MxN elements of array factor equals:
𝐴𝐴𝐴𝐴𝑃𝑃 (𝜃𝜃, ∅) = ∑𝑀𝑀
𝑚𝑚=1 𝑎𝑎𝑚𝑚 𝑒𝑒
𝑗𝑗𝑗𝑗𝑑𝑑𝑚𝑚 sin 𝜃𝜃 cos ∅
⊗ ∑𝑁𝑁
𝑛𝑛=1 𝑎𝑎𝑛𝑛 𝑒𝑒
𝑗𝑗𝑗𝑗𝑑𝑑𝑛𝑛 sin 𝜃𝜃 sin ∅ (3.29)
46
Where ⨂ denotes to the Kronecker Product [142]. From equations (3.24), (3.26) and (3.29), it is noted
that the array factor of the planar array in either receiving or transmitting modes equals the
multiplication of the array factor of the transmitter and the array factor of the receiver of the virtual
array. So that, the planar array factor is written as,
𝐴𝐴𝐴𝐴𝑃𝑃 (𝜃𝜃, ∅) = 𝐴𝐴𝐴𝐴𝑇𝑇 (𝜃𝜃, ∅) ⊗ 𝐴𝐴𝐴𝐴𝑅𝑅 (𝜃𝜃, ∅) (3.30)
The gain of the array factor equals
� 𝑃𝑃 (𝜃𝜃, ∅)�2
𝐺𝐺𝑃𝑃 (𝜃𝜃, ∅) = 𝛼𝛼𝑃𝑃 �𝐴𝐴𝐴𝐴 (3.31)
� 𝑃𝑃 is the normalized array factor of the planar array.

Where 𝐴𝐴𝐴𝐴
By noticing equation (3.25), (3.27), (3.29) and (3.31), the gain of the planar antenna array could be
written in terms of the gain of the virtual array transceiver as follows
𝐺𝐺𝑃𝑃 (𝜃𝜃, ∅) = 𝐺𝐺𝑇𝑇 (𝜃𝜃, ∅) ⊗ 𝐺𝐺𝑅𝑅 (𝜃𝜃, ∅) (3.32)
In case of the planar array, the received power is calculated as follows,
𝑃𝑃𝑟𝑟𝑟𝑟 =
𝑃𝑃𝑡𝑡 𝐺𝐺𝑃𝑃2 (𝜃𝜃,∅)𝜎𝜎𝜆𝜆2
ℎ(𝜃𝜃, ∅) (3.33)
(4𝜋𝜋)3 𝑅𝑅 4
Comparing equation (3.28) with equation (3.33) taking into consideration equation (3.32) the
following relation is held;
𝑃𝑃𝑟𝑟𝑟𝑟 = 𝐺𝐺𝑃𝑃 (𝜃𝜃, ∅) ⊗ 𝑃𝑃𝑟𝑟𝑟𝑟 (3.34)
This means that the received power using the planar antenna array is greater than that of the virtual
array by a factor of 𝐺𝐺𝑃𝑃 (𝜃𝜃, ∅). Then what is the benefit of the virtual array? The reply to this question
is introduced in the following section.
The second step, is to apply the concept of VAA to a two-dimensional antenna array that is created
by placing two orthogonal linear antenna arrays (LAA) to each other. Each LAA consists of 10
rectangular patch microstrip antennas as shown in Figure 3.9. The number of elements in two LAA
are the same to achieve the same angular resolution in x and y planes. The distance between the
elements in X, and Y directions is half air wavelength from operating frequency to avoid the grating
lobe (d=6 mm). At the same time we consider the PAA that consists of 100 elements. The patch is
designed to resonate at 24 GHz on Rogers RO4003 substrate with dielectric constant 3.38 and
thickness 0.2 mm. The patch width W=4.3 mm and length L=3.3 mm. The patch is designed to cover
the band from 23.55 GHz to 24.7 GHz for short range automotive radar as shown in Figure 3.10.
47
Figure 3.10 Reflection coefficient of the rectangular patch antenna, 𝜀𝜀𝑟𝑟 = 3.38, ℎ = 0.2 𝑚𝑚𝑚𝑚.
We introduce three different cases to compare between the PAA and VAA as following:
• Case I:
We consider the PAA that consists of 100 elements in TX mode as compared with the VAA that
consist of M=10, N=10 (only 20 elements) orthogonal as shown in Figure 3.9. We notice that the
radiation patterns are close together for the main lobe and the sidelobe and the difference in the back
lobe is due to the mutual coupling effect in case of PAA that isn’t present in the VAA, as shown in
Figure 3.11. The comparisons are introduced at different angles (𝜑𝜑 = 00 , 300 , 450 , 600 , 𝑎𝑎𝑎𝑎𝑎𝑎 900 ). So,
we can summarized that the VAA with only 20 elements introduce the same radiation pattern of PAA
with 100 elements and provide enhancement in the back lobe.
Phi Case I
=0
0
-20
Normalized Gain (dB)
-40
00 -60
-80
PAA (10x10)
VAA M=10, N=10
-100
0 20 40 60 80 100 120 140 160 180
Theta (degree)
48
=30
0
-20
-40

-60
300
-80
-100 PAA (10x10)

VAA M=10, N=10
-120
0 20 40 60 80 100 120 140 160 180
Theta (degree)
=45
0
-50
450 -100
-150 PAA N=10x10

VAA Nx=10, Ny=10
0 20 40 60 80 100 120 140 160 180

Theta (degree)
=60
0
-20
-40
-60
600
-80
-100 PAA (10x10)

VAA M=10, N=10
-120
0 20 40 60 80 100 120 140 160 180
Theta (degree)
=90
0
-20
-40
900 -60
-80
PAA (10x10)
VAA M=10, N=10
-100
0 20 40 60 80 100 120 140 160 180
Theta (degree)
Figure 3.11 Comparison between VAA and PAA, case I.
49
• Case II
In the RX side, the equation of the reflected power from the target can be expressed as:
𝑃𝑃𝑡𝑡 𝐺𝐺𝑡𝑡 (4𝜋𝜋𝜋𝜋)
𝑃𝑃𝑟𝑟𝑟𝑟𝑟𝑟= (3.35)
(4𝜋𝜋𝜋𝜋)2
The reflected power back to the radar receiver is expressed as:
𝑃𝑃𝑡𝑡 𝐺𝐺𝑡𝑡 (4𝜋𝜋𝜋𝜋) 𝐺𝐺𝑟𝑟 𝜆𝜆2 𝑃𝑃𝑡𝑡 𝐺𝐺𝑡𝑡 𝐺𝐺𝑟𝑟 𝜎𝜎𝜆𝜆2
𝑃𝑃𝑟𝑟 = (4𝜋𝜋𝜋𝜋)2 (4𝜋𝜋𝜋𝜋)2
= (4𝜋𝜋)3 𝑅𝑅 4
(3.36)
Where 𝑃𝑃𝑟𝑟 : Received Power in watts, 𝑃𝑃𝑡𝑡 :Peak transmitted power in watts, 𝐺𝐺𝑡𝑡 : Transmitter Gain, 𝐺𝐺𝑟𝑟 :
Receiver Gain, 𝜆𝜆: Wavelength (m), 𝜎𝜎: RCS of the target (m2), R: Range between radar and target (m).
In equation (3.36), we notice that the received power consists of gain of TX and RX antennas(𝐺𝐺𝑡𝑡𝑡𝑡 =
𝐺𝐺𝑡𝑡 𝐺𝐺𝑟𝑟 ). Therefore in case II we compare between 𝐺𝐺𝑡𝑡𝑡𝑡 of the PAA and the 𝐺𝐺𝑟𝑟 of the VAA as shown
inFigure 3.12. One can notice that the half power beam width (HPBW) of VAA is wider than that of
PAA. Also, the side lobes and back lobes of VAA have larger values than that of PAA. So, we present
case III that introduces the optimum number of M and N that gives the same HPBW of PAA
Phi Case II
=0
0
-50
00
-100
PAA (10x10)
VAA M=10, N=10
-150
0 20 40 60 80 100 120 140 160 180
Theta (degree)
=30
0
-50
-100
300 -150
-200
PAA (10x10)
VAA M=10, N=10
-250
0 20 40 60 80 100 120 140 160 180
Theta (degree)
50
=45
0
-100

450
-200
PAA (10x10)
VAA M=10, N=10
-300
0 20 40 60 80 100 120 140 160 180
Theta (degree)
=60
0
-50
-100
600
-150
PAA N=10x10
VAA Nx=10, Ny=10
-200
0 20 40 60 80 100 120 140 160 180
Theta (degree)
=90
0
-50
900 -100
PAA (10x10)
VAA M=10, N=10
-150
0 20 40 60 80 100 120 140 160 180
Theta (degree)
Figure 3.12 Comparison between VAA and PAA, case II.
• Case III
In this case, we need to introduce the VAA that is equivalent to the PAA in the receiving mode.
So, we optimize the number of elements in VAA by using genetic algorithm (GA) (CST
optimization tools). The comparison between VAA and PAA in the receiving mode with the same
HPBW is depicted in Figure 3.13 . However, the VAA achieves the same HPBW of PAA but its
side lobe and back lobe still have high values than that of PAA. Therefore, we need to use non-
uniform excitation for VAA to give the same levels of side lobes and back lobes of PAA.
51
Phi Case III
=0
0
-50

00 -100
PAA Tx, Rx (10x10)

VAA M=14, N=14
-150 3dB
0 20 40 60 80 100 120 140 160 180

Theta (degree)
=30
0
-50
-100
300 -150
PAA Tx, Rx N=10x10

-200
VAA M=14, N=14
3dB
-250
0 20 40 60 80 100 120 140 160 180
Theta (degree)
=45
0
-50
450
-100
PAA Tx, Rx (10x10)
VAA M=14, N=14
3dB
-150
0 20 40 60 80 100 120 140 160 180
Theta (degree)
=60
0
-50
600
-100
PAA Tx, Rx N=10x10
VAA Nx=14, Ny=14
3dB
-150
0 20 40 60 80 100 120 140 160 180
Theta (degree)
52
=90
0
-50

900
-100
PAA Tx, Rx (10x10)
VAA M=14, N=14
3dB Line
-150
0 20 40 60 80 100 120 140 160 180
Theta (degree)
Figure 3.13 Comparison between VAA and PAA, case III.

In this work, the MoM/GA, a semi-analytical technique, is utilized to synthesize the two columns
of the virtual array to mimic the planar array. Consider the configuration of isotropic sources antenna
array along the Z-axis with equal distance between them [143]. The MoM/GA summarizes the
synthesis problem to the following equation:
[𝑍𝑍]𝑀𝑀×𝑀𝑀 [𝐼𝐼]𝑀𝑀×1 = [𝑉𝑉]𝑀𝑀×1 (3.37)
Where the elements of the matrix [𝑍𝑍]𝑀𝑀×𝑀𝑀 are given by,

𝜋𝜋
𝑧𝑧𝑚𝑚𝑚𝑚 = ∫0 𝑒𝑒 𝑗𝑗(𝑑𝑑𝑛𝑛 −𝑑𝑑𝑚𝑚)𝑘𝑘𝑘𝑘𝑘𝑘𝑘𝑘(𝛩𝛩) 𝑑𝑑𝑑𝑑 (3.38)
and the elements of the vector [𝑉𝑉]𝑀𝑀×1 are given by,

𝜋𝜋
𝑉𝑉𝑚𝑚 = ∫0 𝐴𝐴𝐴𝐴𝑑𝑑 (𝛩𝛩) 𝑒𝑒 −𝑗𝑗𝑗𝑗𝑑𝑑𝑚𝑚𝑐𝑐𝑐𝑐𝑐𝑐(𝛩𝛩) 𝑑𝑑𝑑𝑑 (3.39)
The excitation coefficients 𝑎𝑎𝑛𝑛 are determined by solving the linear system of equation (3.37).
Where 𝑎𝑎𝑛𝑛 are the elements of the matrix [𝐼𝐼]𝑀𝑀×1 , where [𝐼𝐼]𝑀𝑀×1 = [𝑎𝑎1 , 𝑎𝑎2 , 𝑎𝑎3 , … … , 𝑎𝑎𝑀𝑀 ]𝑇𝑇
In order to get the same received power for both the planar and the virtual array, the number of elements
for the transmitter and the receiver array should be adjusted so that
𝐺𝐺𝑃𝑃 (𝜃𝜃, ∅) = �𝐺𝐺𝑇𝑇𝑇𝑇𝑇𝑇𝑇𝑇 (𝜃𝜃, ∅) ⊗ 𝐺𝐺𝑅𝑅𝑅𝑅𝑅𝑅𝑅𝑅 (𝜃𝜃, ∅) (3.40)
From equation (3.40) and applying the MOM/GA to the VAA with only 18 elements we can have the
same equivalent radiation pattern of the PAA with 100 elements as shown in Figure 3.14. The
excitation coefficients of the first 9 elements of proposed VAA in this case according to the MOM/GA
are [0.15, 0.257, 0.363, 0.462, 0.571, 0.667, 0.781, 0.868, and 1.0].
53
0
PAA (10x10)
VAA (18 elements)
-20
Array Factor (dB)
-40
-60
-80
0 20 40 60 80 100 120 140 160 180
Theta (Degree)
Figure 3.14 Comparision between radiation pattern of PAA and VAA
3.5 Design Antenna Array for LRR and MRR

The automotive radars are classified according to the operating range into; long- range radar (LLR)
(10- 250 or 300 m), medium-range radar (MRR) (1-100 m), and short range radar (SRR) (0.15-30 m).
Where both LRR and MRR are required to detect the forward obstacles. So, most of the radars use two
transmitters with two unique radiation patterns to give the LRR mode and MRR mode. The
performance of this system isn’t efficient for the automotive radar due to the switching performance
(time response and losses) between the two modes. Recently, Xu et.al [16, 144], developed the idea of
the shaped beam antenna that was used in other applications such as satellites and radars to introduce
a single antenna that meets the requirements of LRR and MRR in automotive radar applications. The
authors introduce two papers in this direction using substrate integrated waveguide (SIW) power
divider to feed the planar patches array and to feed the slot SIW array.
3.5.1 Proposed Radiation Pattern of LRR and MRR

In this study, we use the same transceiver for the LRR and MRR modes with the same antenna
but we need to determine the antenna gain difference between two modes (LRR and MRR). We assume
that the minimum received power at the radar is constant in the two modes 𝑃𝑃𝑟𝑟𝑟𝑟 , because the sensitivity
of the receiver is constant in the two modes. 𝑅𝑅𝐿𝐿 , and 𝑅𝑅𝑚𝑚 are considered the radar ranges for LRR and
MRR, respectively. Also, we assume that 𝑅𝑅𝐿𝐿 = 300𝑚𝑚 and 𝑅𝑅𝑚𝑚 = 100𝑚𝑚, so, 𝑅𝑅𝐿𝐿 = 𝐾𝐾𝑅𝑅𝑚𝑚 . Moreover,
𝐺𝐺𝑇𝑇𝐿𝐿 , 𝐺𝐺𝑇𝑇𝑀𝑀 are the transmitting antenna gain for LRR and MRR, respectively. The received power is
expressed as
54
𝑃𝑃𝑡𝑡 𝐺𝐺𝑡𝑡 𝐺𝐺𝑟𝑟 𝜎𝜎𝜆𝜆2 𝑃𝑃𝑡𝑡 𝐺𝐺𝑡𝑡𝑀𝑀 𝐺𝐺𝑟𝑟 𝜎𝜎𝜆𝜆2 𝑃𝑃 𝐺𝐺 𝐿𝐿 𝐺𝐺 𝜎𝜎𝜆𝜆2

𝑃𝑃𝑟𝑟𝑟𝑟 = (4𝜋𝜋)3 𝑅𝑅 4
= (4𝜋𝜋)3 𝑅𝑅𝑚𝑚 4
𝑡𝑡 𝑡𝑡 𝑟𝑟
= (4𝜋𝜋)3 (𝐾𝐾𝑅𝑅 4
. (3.41)
𝑚𝑚 )
We can determine the gain difference 𝐺𝐺𝑑𝑑 between the two operating mode from equation (3.41) as
[𝐺𝐺𝑑𝑑 ] = [𝐺𝐺𝑡𝑡𝐿𝐿 ] − [𝐺𝐺𝑡𝑡𝑚𝑚 ] = 40 log10 𝐾𝐾 (3.42)
In case 𝐾𝐾 ≥ 3, 𝐺𝐺𝑑𝑑 ≥ 19 𝑑𝑑𝑑𝑑, 𝐺𝐺𝑑𝑑 is define as the difference gain between the two scenario
modes as show in Figure 3.15 that depicts the expected radiation pattern of proposed system. This
shape is called flat-shoulder shape and shows the suggested ideal radiation pattern of the proposed
antenna to support the two expected scenarios mode, where 𝜃𝜃𝐿𝐿 = 150 at -3dB as half power beam
width in case of LRR and 𝜃𝜃𝑚𝑚 = ±400 at 𝐺𝐺𝑑𝑑 level as beam width in case of MRR.
Figure 3.15 The suggested radiation pattern of the antenna to support MRR and LRR (LMRR)
3.5.2 Excitation Coefficients

To achieve the flat shoulder shape (FSS) pattern, we propose radiation pattern by combination
of two Dolph-Chebyshev antenna arrays [145]. The radiation pattern (FSS) in 3. 15 is applied to the
MoM/GA algorithm as a desired radiation pattern. The synthesis process using equations (3.37 to 3.39)
results in the excitation coefficients that guarantee the occurrence of the FSS pattern when exciting an
8-element antenna array whose elements are half-wavelength spaced from each other.
As the FSS pattern is considered a broadside radiation pattern, the phase shift between elements
should vanish. After applying the MoM/GA algorithm, it is noticed that the excitation coefficients
pattern is symmetric around the center of the array. The left-handed and right-handed four elements
are excited by the proposed excitation coefficient as shown in Table 3. 1 Then according to these
coefficients, we need to design unequal 8 × 1 power divider.
55
3.5.3 Power Divider Design

We designed unequal Willikson 8x1 power divider to feed the proposed antenna as shown in Figure
3.16 according to the excitation coefficients from MOM/GA. The Willikson power divider is designed
on Rogers RO4003 substrate with dielectric constant 3.38 and thickness 0.2 mm. The dimensions of
the proposed power divider is based on the equations presented in [146]. We use the quarter wave
length transformer and stepped impedance matching to achieve the required excitation coefficient at
frequency 24 GHz. Here, we use 24 GHz as an example of automotive radar frequency instead of 77
GHz because of the limitation of our measurements and fabrications facilities in our laboratory. The
proposed power divider consists of 3 stages to introduce 8 excitation coefficients. PD1 is an equal
power divider, PD2, PD3 and PD4 are unequal power dividers. All the simulated S-parameters
magnitude and phase of the proposed power divider are depicted in Figure 3.17. We notice that all
ports achieve the required magnitude and phase according to the output from the MOM/GA technique.
Furthermore, the comparison between the required values of magnitude and phase from MOM/GA
and CST are shown in Table 3. 1 with equal phase between the power divider ports.
Table 3. 1 Amplitude of power divider outputs
Synthesis (Ideal) Simulated (CST)
Amplitude Phase Amplitude Phase

Power
(dB) (Degree) (dB) (Degree)
P1 -3 (0,0) -3.21 (-96.15, -96.15)

P2 -10.838 (0,0) -10.62 (-58.75, -58.75)
P3 -3.788 (0,0) -4.765 (-57.939, -57.939)
A1 -17.122 (0,0) -17.72 (101.544,101.544)
A2 -11.04 (0,0) -11.6 (103.324,103.324)
A3 -7.642 (0,0) -8.19 (102.73,102.73)
A4 -6.38 (0,0) -6.88 (105.253,105.253)
56
(a)Reference power divider
(b)The Proposed power divider
Figure 3.16 The configuration of Wilkinson power divider with impedance distributions
Table 3. 2 Impedance of power dividers (optimization values), (all values in Ω)
PD Zf Z1 Z2 Z3 Z4 R
1 50 70.7 70.7 50 50 100
2 26.2 76.1 33.46 68.6 28.5 135
3 26.79 76.217 33.311 66.494 25.77 68
4 33.31 65.1 39.94 55.94 27.5 135
57
-5
S
-10 11
S
21
-15
S
31
-20 S
41
S-Parameters (dB)
S
-25 51
S
61
-30
S
71
-35 S
81
S
-40 91
-45
23 23.2 23.4 23.6 23.8 24 24.2 24.4 24.6 24.8 25
F(GHz)
(a)Magnitude (dB)
200
150
S
21
100
S
S-parameters (degree)
31
S
50 41
S
51
S
0 61
S
71
-50 S
81
S
91
-100
23 23.2 23.4 23.6 23.8 24 24.2 24.4 24.6 24.8 25
F(GHz)
(b)Phase (degree)
Figure 3.17 S-parameters of power divider
3.5.4 Antenna Array Design

Figure 3.18 (a) shows geometry of the rectangular patch linear antenna array with half wave
length as a distance between the elements. The FSS radiation pattern is achieved as shown in Figure
3.18 (b) and there are good agreements between the simulated FSS pattern from CST and required FSS
pattern from MOM/GA. The concept of VAA is applied to the proposed antenna as shown in Figure
3.19 (a). The VAA is fabricated on Rogers RO4003C substrate with dielectric constant 3.38 and
thickness 0.2 mm as shown in Figure 3.19 (b). We notice that the VAA operates at 24.25 GHz with
400 MHz bandwidth and it has a very good isolation coefficient between the transmitter and receiver
antennas (more than 43 dB) as shown in Figure 3.20. The achieved isolation is one of the main factors
58
that support our proposed VAA. Therefore, we don’t need to integrate circulator to the system because
we use two antennas with very good isolation. The second advantage in our design is that the VAA
needs simple feeding network comparable with the same structure in case of the PAA. Also, Figure
3.20 shows the comparison between the simulated and measured reflection coefficient of VAA. The
VAA antenna with its feeding network that is based on a cascaded network of Wilkinson power
dividers printed on the same substrate of overall dimensions of 30x48x0.2 mm3.
(a)Geometry of LAA
0
CST
MOM/GA
-10
-20
-30
-90 -60 -30 0 30 60 90
Theta (Degree)
(b) Normalized XZ plane gain pattern
Figure 3.18 Linear antenna array (Lp=3.2mm, Wp=3.45 mm, S1=24 mm, S2= 12mm, and S3=6 mm)
59
(a)VAA configuration
(b) Photo of VAA
Figure 3.19 Geometry and photo of fabricated VAA
60
-10
S (Meas.)
11
-20 S (Meas.)
21
S (CST)
S- Parameters (dB)
11
-30
S (CST)
21
-40
-50
-60
-70
22 22.5 23 23.5 24 24.5 25 25.5 26
F(GHz)
Figure 3.20 Simulated and measured S-parameters of VAA
The radiation pattern measurements have two stages; the first stage, we measure the TX antenna's
radiation pattern as common radiation pattern measurements in the anechoic chamber. The second
stage is to measure the receiving mode radiation pattern of VAA. Figure 3.21 (a) shows the setup
structure of our system to measure the radiation pattern of VAA; the measurement system is
homemade. The VAA antenna is positioned on the plate. Two DC motors rotate this plate
in two planes (azimuth and elevation plane). The DC motors are controlled by a microcontroller kit
(Arduino kit), and the VAA is connected to the VNA through two channels to send and receive the
power. We use the 2-D reflector as a target. Figure 3.21(c) shows a good agreement between the
simulated and measured radiation pattern for E-Plane and H-Plane.
(a)Proposed measurements system
61
(b)Photo inside anechoic chamber

0
0
=90 (CST)
0
=0 (CST)
0
=90 (Meas.)
-20
0
=0 (Meas.)
Normalized Radiation Pattern (dB)
-40
-60
-80
-150 -100 -50 0 50 100 150
Theta(Degree)
(c) Radiation patterns
Figure 3.21 Setup system for measurements and radiation patterns.

Table 3. 3 shows the comparison between our proposed VAA for automotive radars and the
antennas that are presented in the literature. As far as we know, only two papers in the literature
combine between the LRR mode and MRR mode [16, 144], but these papers have a complicated
feeding SIW network with antenna size 13.2𝜆𝜆0 ×5.1𝜆𝜆0 and 3.9 𝜆𝜆0 ×7.9𝜆𝜆0 excluding the feeding
network size for antenna in [16] and antenna in [144], respectively. Our design has small size compared
to the aforementioned antennas and achieves the same radiation properties. Furthermore, the proposed
design used VAA concept that reduces the number of antenna elements to eight patches for Tx and
eight patches for Rx compare with the 60 patches presented in [16] and 96 slot antennas presented in
62
[144]. Furthermore, the proposed design achieved the requirements of LRR and MRR beam angles in
E-plane and H-plane. Otherwise, the automotive radar antennas that are introduced in literatures have
a large size, low gain and multiple number of elements.
Table 3. 3 Comparison between proposed antenna and antennas in literature

Paper/Year F Rada Antenna HPBW Gain BW Size Substrate
(GHz) r description (dBi) (GHz) 𝜺𝜺𝒓𝒓
E Plane H
plane
[147]/2010 77 LRR RP (8× 8) ±90 - 20 1 5.9𝜆𝜆0 ×6.4𝜆𝜆0 3.38
[148]/2012 77 LRR RP(5× 14) 100 - 20.5 1.5 8.1𝜆𝜆0 ×2.95𝜆𝜆0 Silicon
11.8
[149]/2013 77 LRR VP (8× 18) 4.80 18.30 20.8 1 17.96𝜆𝜆0 ×7.7𝜆𝜆0 2.2
[150]/2014 80 LRR VP ±70 - 25.6 1.5 (WFN) 3.38

(16× 16) RX, 5.9𝜆𝜆0 ×6.4𝜆𝜆0
(2× 16) TX
[151]/2015 77 LRR Varying patch ±150 - 18.5 2.4 10.3𝜆𝜆0 ×10.3𝜆𝜆0 3.38
(2× 1)
[152]/2016 77 LRR MC (32× 32) 100 -- 24.7 0.7 10.8𝜆𝜆0 ×19.2𝜆𝜆0 3.38
[58]/2018 77 LRR 10-element 200 -- 4 - (WOFN) 3

series, AMC 9.7𝜆𝜆0 ×3.6𝜆𝜆0
[144]/2018 77 LRR/ SIW (6× 16) ±7.50, -- 21.7 2.9 (WOFN) 2.2
MRR ±400 3.9 𝜆𝜆0 ×7.9𝜆𝜆0
[16]/2017 77 LRR/ Microstrip ±7.50, -- 20 9.5 (WOFN) 2.2
MRR +SIW (6*10) ±400 13.2𝜆𝜆0 ×5.1𝜆𝜆0
[153]/2015 24 SRR Grid 33 7 900 13.8 6 1.44𝜆𝜆0 ×11.7𝜆𝜆0 3
elements
[15]/2019 23.7 SRR Patches with 10 1500 11 1 2.9𝜆𝜆0 ×5.3𝜆𝜆0 3.38
mushroom
This Work 24.1 LRR/ 16 RP VAA ±70 ±70 17 1.15 (WFN) 3.38
MRR ±380 ±380 3.6𝜆𝜆0 ×2𝜆𝜆0
*RP: rectangular patch, VP: varying patch, MC: microstrip combline, WOFN: without feeding network, WF: with feeding
network, VAA: virtual antenna array.
3.6 Hybrid Antenna for 77 GHz Automotive Radar

This section introduces a linear antenna array for LRR automotive radar to operate at 77 GHz. The
proposed antenna consists of a hybrid radiator and dielectric resonator. The hybrid radiator is a circular
patch that feed by aperture method and the dielectric resonator is ring that is fed by the circular patch
to operate at 77 GHz. The EBG structure is implemented on the top layer to widen the proposed band,
gives high gain and to reduce surface waves. The compact size of one element 3mm×3mm×1.19mm
to operate from 75.3 GHz to 80 GHz with gain of 12.3 dBi.
63
3.6.1 Antenna Design

1. Configuration
Figure 3.22 shows the geometry of the proposed antenna. All the labeled dimensions are
tabulated in Table 3. 4 Antenna parameters (all dimensions in mm). The antenna consists of three
layers. The 50-ohm microstrip feed line is designed on a first layer, this layer is made from silicon
material with εr = 11.7 and H1=0.278 mm. The second layer is Rogger RT5880 material with εr =
2.2, and H2=0.278 mm then the third layer is a ring dielectric resonator from silicon material with εr =
11.7 and H3=0.635mm.
(a) Top view (b)AMC unit cell
(c)side view (d)structure details

Figure 3.22 Antenna geometry
64
Table 3. 4 Antenna parameters (all dimensions in mm)

L1 W1 R1 R2 R3 A1 A2 A3 A4
3 3 0.4 0.7 0.9 0.35 0.3 0.22 0.12
A5 Lf Ls Ws Wf H1 H2 H3
0.1 1.8 0.85 0.08 0.12 0.278 0.278 0.635
2. Analysis of Dielectric Ring

To design dielectric ring resonator, the hybrid mode (hybrid electromagnetic (HEMmnδ)) should be
calculated as a first step in our design. The hybrid mode is calculated in first time by Cohn [154] in
1968 and after this developed in [155-157]. The ring resonator parameter can be calculated from the
following equations:
2
𝑐𝑐 𝑋𝑋𝑚𝑚𝑚𝑚 2 𝛿𝛿𝛿𝛿
𝑓𝑓𝑚𝑚𝑚𝑚𝛿𝛿 = �� +� � (3.43)
2𝜋𝜋 �𝜀𝜀𝑒𝑒𝑒𝑒𝑒𝑒 𝑟𝑟 2ℎ𝑒𝑒𝑒𝑒𝑒𝑒
ℎ𝑒𝑒𝑒𝑒𝑒𝑒 = ℎ2 + ℎ3 (3.44)
2
𝑟𝑟𝑜𝑜𝑜𝑜𝑜𝑜 ℎ𝑒𝑒𝑒𝑒𝑒𝑒
𝜀𝜀𝑒𝑒𝑒𝑒𝑒𝑒 =
𝑟𝑟𝑖𝑖𝑖𝑖 2 ℎ3 (𝑟𝑟𝑜𝑜𝑜𝑜𝑜𝑜 2 − 𝑟𝑟𝑖𝑖𝑖𝑖 2 ) ℎ3 𝑟𝑟𝑜𝑜𝑜𝑜𝑜𝑜 2 ℎ2 (3.45)
+ +
𝜀𝜀𝑟𝑟𝑟𝑟𝑟𝑟𝑟𝑟 𝜀𝜀𝑟𝑟_𝑟𝑟𝑟𝑟𝑟𝑟𝑟𝑟 𝜀𝜀𝑟𝑟_𝑠𝑠𝑠𝑠𝑠𝑠
2ℎ𝑒𝑒𝑒𝑒𝑒𝑒
𝛿𝛿 = 𝛽𝛽 (3.46)
𝜋𝜋
𝑋𝑋𝑚𝑚𝑚𝑚 2
𝛽𝛽 = �𝐾𝐾02 𝜀𝜀𝑒𝑒𝑒𝑒𝑒𝑒 − � � (3.47)
𝑟𝑟
𝐽𝐽′ (𝑥𝑥) = 0 n odd
𝑋𝑋𝑚𝑚𝑚𝑚 = � 𝑚𝑚 (3.48)
𝐽𝐽𝑚𝑚 (𝑥𝑥) = 0 𝑛𝑛 𝑒𝑒𝑒𝑒𝑒𝑒𝑒𝑒
Where 𝑓𝑓𝑚𝑚𝑚𝑚𝛿𝛿 resonant frequency, m,n are order of resonant mode, 𝛿𝛿 value between 0 and 1 to identify
the number of half wavelength changing in z direction, c speed in free space, 𝑋𝑋𝑚𝑚𝑚𝑚 zero of derivative
Bessel function and Bessel function for n odd and n even, respectively.
65
300
HEM
11
250 HEM
12
HEM
13
200
F(GHz)
150
100
50
0
1 1.5 2 2.5 3 3.5 4
R (mm)
3
Figure 3.23 Outer radius of dielectric resenator versus frequency for the first three modes
(𝛿𝛿 = 0.6, 𝑅𝑅2 = 𝑅𝑅3 − 0.6, 𝐻𝐻3 = 0.635 𝑚𝑚𝑚𝑚, 𝐻𝐻2 = 0.278 mm).
The antenna is designed to operate with the first hybrid mode HEM11δ . Figure 3.23 shows the
relation between the different values of DR outer radius versus the resonant frequency. We notice that
at R3=0.9 mm the resonant frequency of first mode is around 77 GHz. The antenna consists of two
radiators; the first radiator is circular patch to operate at 77 GHz, and the second radiator is the ring
dielectric antenna that is optimized to operate at 77 GHz to give wide bandwidth, and high gain. The
patch is designed on top of second layer and it is fed by the aperture slot on the ground plane between
first and second layer. The design process; in the first, the circular patch is coupled to the feeder
through aperture without ring resonator as conventional techniques and adjust its resonant frequency
at 77 GHz. The resonant frequency can be controlled by the slot length which is equal to the half-
guided wavelength. A hybrid technique is used and adjusts the dimensions of ring resonator to operate
at 77 GHz.
(a)Simulation setup of AMC (b) phase of AMC

Figure 3.24 Artificial Magnetic conductor result
66
3.6.2 Results and Discussion for Single Element
A structure of unit cell of EBG with periodic boundary is simulated to model an infinite
periodic surface. The wave port is positioned a half wavelength above periodic surface of the structure,
and normal plane waves are launched to illuminate it. The periodic surface is chosen as the phase
reference plane. Figure 3.24(a) shows the simulation setup for the planar artificial magnetic conductor
(AMC). With this setup, the observation plane and periodic surface are in different locations, so the
reflection phase has to be translated to the periodic surface.
The bandwidth of AMC performance is generally defined in the range from 900 to -900 in this range
the BW = 10 GHz with center frequency at 77 GHz as shown in Figure 3.24(b). The EBG structures
are added to act as an artificial magnetic conductor, AMC, within its stop bands. The proposed EBG
configuration reveals stop bands at 77 GHz for automotive radar applications. This means that it has
high surface impedance within this band, where the tangential magnetic field is small, even with a
large electric field along the surface. The EBG structure is positioned perpendicular to the antenna.
0
CST
HFSS
-10
Reflection Coefficient(dB)
-20
-30
-40
-50
70 71 72 73 74 75 76 77 78 79 80
F(GHz)
Figure 3.25 Simulated reflection coefficient of the proposed antenna
After applying EBG, the size of antenna is reduced with the final dimensions shown in Table
3. 4. Figure 3.25 shows the return loss of the proposed antenna. Taking the -10 dB return loss as a
reference, the antenna operates from 75.3 GHz to 80 GHz. Figure 3.26 shows the radiation pattern of
the antenna with high gain 12.3 dBi at 77 GHz. The radiation pattern is directive in the perpendicular
plane of the antenna with HPBW =350 in XZ plane and YZ plane. The gain of the proposed antenna
and the radiation efficiency are shown in Figure 3.27.
67
(a)YZ (b) XZ
Figure 3.26 Radiation pattern at 77 GHz
Figure 3.27 Gain and radiation efficiency of the proposed antenna
3.6.3 Antenna Array

The recent automotive radar needs to detect the objective at longer distance than 250m. So, the
system needs an antenna with high gain, narrow beamwidth and sometimes beam scanning. Therefore,
in this section, the antenna array with 8 and 16 elements in series and in parallel are introduced to
achieve the requirements of LRR as shown in Figure 3.28 and Figure 3.29, respectively. The antenna
arrays have a total size 20.5mm×3mm and 40.5× 3mm for 8 elements and 16 elements, respectively.
The thickness of two arrays are the same thickness as one element 1.19 mm with 0.4𝜆𝜆0 between
elements at 77 GHz. The EBG is used to reduce the mutual coupling which has the advantages of
reducing the distance between elements to be less than the half wavelength. In series antenna array,
the optimized values of slot length and width are (Ls=0.9 mm and Ws= 0.1 mm) but in the parallel
antenna arrays are the same of the aforementioned single element. The EBG structure is used to give
high gain, wide bandwidth and isolation between elements to reduce mutual coupling.
68
(a)Front of series antenna array (b)Back of serise antenna array
(c)Front of parallel antenna array
(d)Back of parallel antenna array

Figure 3.28 Geometry of the proposed antenna arrays with 8 elements
69
(a)Front of series antenna array (b)Back of serise antenna array
(c)Front of parallel antenna array
(d)Back of parallel antenna array

Figure 3.29 Geometry of the proposed antenna arrays with 16 elements
The S-parameters of the proposed antenna arrays are shown in Figure 3.30, we notice that the
antenna arrays still operate in the proposed band for LRR and the isolation coefficients in case of
parallel antenna arrays are below 30 dB. The advantage of series antenna arrays is that they don’t need
feeding network, with the same number of elements and the same size.
70
Table 3. 5 presents the comparison between the performances of the proposed antenna arrays. We
notice that the two antenna arrays have similar bandwidth but the series antenna arrays have small
gain, and small HPBW than that of parallel antenna arrays because the power coupling for the last two
or three elements in series antenna array is less than its value for the start elements. The HPBW in
parallel less than in series, therefore, longer range is achieved in parallel than in series. The antenna
array gain is larger in parallel than in series. 16 elements has narrower BW in parallel than in series.
(a)parallel N=8
(b)parallel N=16
71
(c)series
Figure 3.30 S-Parameters of proposed antenna array configurations.
The 3-D radiation patterns of the proposed different configurations of antenna array are shown in
Figure 3.31. Noted that the parallel configuration has high gain than that of series configuration by
about 2.4dBi. Figure 3.32 depicts the comparison between 2-D radiation patterns of antenna arrays in
XZ plane and YZ plane. The two configurations have little difference in gain and HPBW. The two
antenna arrays have average gain and average efficiency (17.5 dBi, 84.5%) and (19.5 dBi, 85%) for 8
elements series and parallel configurations, respectively. On the other hand, the gain and efficiency of
antenna arrays of 16 elements are (20.2 dBi, 82%) and (21.5 dBi, 84.5%) for series and parallel
configurations, respectively. The comparison between the parameters of the two configurations are
summarized in Table 3. 5. We notice that the HPBW of parallel configurations are better than that of
series configuration by about 0.70. Therefore, the parallel configuration can give long range in the
radar applications than the series configuration but it still needs a feeding network that will reduce its
efficiency.
72
Figure 3.31 3-D radiation pattern of proposed antenna array configurations.
20
N=8
)
i
0
Radiation Pattern (dB
-20
0
XZ Plane( =0 ), Parallel
0
-40 YZ Plane( =90 ), Serise
0
YZ Plane( =90 ), Parallel
0
XZ Plane( =0 ), Serise
-60
-150 -100 -50 0 50 100 150
Frequency (GHz)
(a)N=8
73
30
20 N=16
10
)
i
0
-10
Radiation Patteren (dB

-20
-30
-40 0
XZ Palne( =0 ), Parallel
-50 0
YZ Palne( =90 ), Serise
-60 0
YZ Palne( =90 ), Parallel
-70 0
XZ Palne( =0 ), Serise
-80
-150 -100 -50 0 50 100 150
Frequency (GHz)
(b)N=16
Figure 3.32 2-D radiation pattern of proposed antenna array configurations.
22
21
20
)
i
19
Gain (dB
N=8 Parallel
18 N=8 Serise
N=16 Parallel
17 N=16 Serise
16
70 72 74 76 78 80 82 84 85
Frequency (GHz)
(a)Gain
90
85
80
Efficiency (%)
75
N=8 Parallel
70
N=8 Serise
N=16 Parallel
65 N=16 Serise
60
70 72 74 76 78 80 82 84 85
Frequency (GHz)
(b)Efficiency
Figure 3.33 Gain and efficiency of proposed different antenna array configurations.
74
Table 3. 5 Comparison between antenna arrays performance
Number of Elements N=8 N=16
Geometry Series Parallel Series Parallel
Size (mm2) 3×20.5 20.5×3 3×40.5 40.5×3
BW (GHz) 73-82 74-80 73-82 72-80
Gain (dBi) 17.5 22.2 19.7 24.7
Efficiency (%) 86 87 83 84
HPBW(XZ) 32.50 4.50 30.50 2.80
HPBW(YZ) 5.30 320 3.40 28.70
Feeding network No need Need No need Need
3.7 Conclusion
In this chapter, a novel concept in the design of an automotive radar antenna system is
introduced. The VAA concept is utilized to have a simple, highly isolated and a high efficient antenna
array that competes for the PAA in most of its characteristics. The proposed concept is analyzed and
verified using an analytical solution and used the MoM/GA algorithm to get the optimum
characteristics of the VAA compared to the PAA antenna. The work ended by a design of the VAA
antenna with its feeding network that is based on a cascaded network of Wilkinson power dividers on
the same substrate of overall dimensions of 30x48x0.2 mm3. The antenna has an FSS radiation pattern
that supports both the MRR and the LRR application simultaneously. The experimental results agree
well with simulated results. Furthermore, two configurations of antenna arrays (series and parallel
configurations) for LRR at 77 GHz are introduced. These antenna arrays are used to achieve high
resolution by introducing small HPBW. Each antenna element consists of two hybrid resonators and
AMC surrounding it. The two resonators are used to achieve wide bandwidth in addition to high gain
and AMC pattern to achieve high isolation between elements in addition to high gain and wide
bandwidth.
75
Chapter 4: Antenna Design for 5G
4 Chapter Four:
Antenna Design for 5G
4.1 Introduction
The researchers still introduce different studies to obtain the optimum specifications of the
antenna that can be used for 5G applications, especially for the range from 28 GHz to 38 GHz.
Therefore, the antenna design for 5G applications introduced in this chapter has a dual-polarized
property to avoid the high attenuation in the millimeter range and give high data rate communications.
Furthermore, the MIMO antenna configuration is presented with a comprehensive study for all
parameters. The characteristic mode analysis is used here to analyse the antenna performance in
addition to using metasurface to enhance the proposed antenna. Section 4.2 introduces the
characteristic mode theory, while section 4.3 presents a dual-polarized antenna for a 5G application.
Section 4.4 discusses the metasurface, while the antenna using these metasurfaces is given in section
4.5. The conclusion is given in section 4.7.
4.2 Characteristics Mode Theory

Over the previous few years, significant progress has been introduced in the field of antennas
engineering. To overcome various of difficulties faced in antennas designs, many different methods
and design ideas have been introduced. One of the exciting breakthroughs in antenna engineering is
the characteristic mode theory (CMT) or theory of characteristic mode (TCM) or characteristic mode
analysis (CMA), which focuses on its comprehensive implementation in many critical antenna models.
Its promising potential has always attracted antenna engineers ' attention [158-167].
The characteristic modes (CMs) are set of current modes for arbitrarily shaped conducting
bodies that are presented numerically. They are independent of any excitation and depend only on the
conductive object's size and shape. For PEC, the CM theory is widely defined in [168, 169]. After this,
it has been spread for composite metal-dielectric objects, where the results of the characteristic modes
are achieved using a surface integral equation-based MoM.
A set of currents (i.e. Current distributions) is detected when TCM is applied to an object; each
current is unique in its distribution. The other significant benefits of TCM in comparison to other
computational electromagnetic methods are better described. While there are benefits for all
computational electromagnetic solvers, TCM is unique as unlike any other. This is partly due to the
76
fact that TCM does not require excitation sources to be placed when determining the electromagnetic
characteristics of an analyzed object. These solvers usually involve a physical excitation or feeding
component to excite an antenna/object. Once solutions are provided from these kinds of computational
solvers, only single current distribution will be achieved, which generates a specific pattern of
radiation.
The CMT has a history that is brief but complex. In 1965 [168], Garbacz introduced the idea
that the electromagnetic characteristics of an object can be defined by a linear combination of modal
field patterns, which are determined only by the shape and material properties of the object. This initial
definition used a scattering matrix that describes an object's interaction with an exciting wave as a
mathematical representation. This definition offered theoretical proof of the possibility of
decomposing any excitable current on an object into an infinite set of radiating currents, with the most
significant having the lowest magnitude of its eigenvalue.
Each eigenvalue is described in terms of the radiation resistance of the respective current.
However, this theoretical introduction to CMs needed a previous understanding of the distinctive
patterns of far-field radiation and their related values. Garbacz [170, 171] has provided two possible
solutions to find the unknown modes; both methods are difficult to implement and cannot be used on
all geometries of objects. There has been no published study on this subject for another six years with
this important drawback.
The researches and studies of theoretical or experimental applications of CMT have been developed
and grown significantly over the last 10 years. Most of these applications are summarized as a synthesis
of the antennas, small/compact antennas, MIMO systems, pattern synthesis, and scattering problems
[158-167, 172-176].
4.2.1 Analysis of CMs

As described in [169], the characteristic modes or characteristics currents are obtained by solving a
particular eigenvalue equation that is derived from the Method of Moments (MoM) impedance matrix,
𝑖𝑖
𝑍𝑍𝑍𝑍 = 𝐸𝐸𝑡𝑡𝑡𝑡𝑡𝑡 where 𝑍𝑍 = 𝑅𝑅 + 𝑗𝑗𝑗𝑗 (4.1)
From (4.1), the characteristic modes (CMs) are introduced based on the coefficient matrix’s
generalized values by Harrington [177]:
𝑍𝑍�𝐽𝐽��⃗ ��⃗
𝑛𝑛 � = 𝜈𝜈𝑛𝑛 𝑅𝑅�𝐽𝐽𝑛𝑛 �
(4.2)
Where 𝜈𝜈𝑛𝑛 :eigenvalues, R is a real part of the impedance matrix/operator for the MoM, ��⃗
𝐽𝐽𝑛𝑛 is an
eigencurrent (eigenfunction) and from equation (4.1) and (4.2)
77
𝑅𝑅�𝐽𝐽��⃗ ��⃗ ��⃗

𝑛𝑛 � + 𝑗𝑗𝑗𝑗�𝐽𝐽𝑛𝑛 � = 𝜈𝜈𝑛𝑛 𝑅𝑅�𝐽𝐽𝑛𝑛 �
(4.3)
𝑋𝑋�𝐽𝐽��⃗ ��⃗
𝑛𝑛 � = 𝜆𝜆𝑛𝑛 𝑅𝑅�𝐽𝐽𝑛𝑛 �
(4.4)
𝜆𝜆𝑛𝑛 =
𝜈𝜈𝑛𝑛 −1
, 𝜈𝜈𝑛𝑛 = 1 + 𝑗𝑗𝜆𝜆𝑛𝑛 (4.5)
𝑗𝑗
Where 𝜆𝜆𝑛𝑛 corresponding characteristic values to eigenvalues, sometimes called eigenvalues

because they have a direct relation with 𝜈𝜈𝑛𝑛 . R gives an orthogonal radiation pattern of CMs. To solve
the MoM equation, the CMs are used as a basis function to expand the unknown total current, J, on the
surface of the metal as
𝑉𝑉 𝑖𝑖 𝐽𝐽
𝑛𝑛 𝑛𝑛
𝐽𝐽 = ∑𝑛𝑛 1+𝑗𝑗𝜆𝜆 , The excitation coefficient can be expressed as 𝑉𝑉𝑛𝑛𝑖𝑖 = ∑𝑛𝑛 𝐸𝐸 𝑖𝑖 𝐽𝐽𝑛𝑛 (4.6)
𝑛𝑛
Where 𝑉𝑉𝑛𝑛𝑖𝑖 the excitation coefficient on the conductor is surface and 𝐸𝐸 𝑖𝑖 is the impressed E-field.
. The eigenvalues have a range from −∞ < 𝜆𝜆𝑛𝑛 < +∞ that can be divided into three regions:
• −∞ < 𝜆𝜆𝑛𝑛 < 0 , CM has capacitive and store electric energy.
• 𝜆𝜆𝑛𝑛 = 0, CM has a resonate frequency and radiates efficiently.
• 0 < 𝜆𝜆𝑛𝑛 < +∞, CM has inductive and store magnetic energy.
The first step in the analysis of the CMs, is to get the eigenvalues because they introduce the data
on how the related modes (𝐽𝐽𝑛𝑛 ) radiate and how they relate to the resonance. In CMA, many modes can
be determined dependent or equal to the number of unknown numbers in the equation of MoM at every
frequency. The eigencurrents, unaffected by the type of source or excitation method, depend only on
the shape and dimensions of the structure and its operating frequency. Also, the total current on an
antenna's surface can be calculated as a summation of eigencurrents of the antenna. Therefore, the
eigenvalues are used as an indicator to know the resonant frequency for each characteristic mode.
The multilayer solver and integral equation solver from CST microwave studio is chosen to
implement the CMT with very high quality [158, 159, 166]. These solvers are used to calculate the
characteristic modes and those related parameters.
The second parameter for CMA is a characteristic angle (𝛼𝛼𝑛𝑛 ) which is used to describe the antenna
operation and performance. The Characteristic angle calculates the difference in phase between the
characteristic currents (𝐽𝐽𝑛𝑛 ) and its related characteristic field. The characteristics angle can be
calculated from the following relation
𝛼𝛼𝑛𝑛 = 1800 − 𝑡𝑡𝑡𝑡𝑡𝑡−1 (𝐽𝐽𝑛𝑛 ) (4.7)
The characteristic angles values are within this range 00 ≤ 𝛼𝛼𝑛𝑛 ≤ 3600 that can be divided into three
regions:
• 𝜆𝜆𝑛𝑛 > 0, 𝛼𝛼𝑛𝑛 < 1800 (CM has inductive and store magnetic energy)
78
• 𝜆𝜆𝑛𝑛 < 0, 𝛼𝛼𝑛𝑛 > 1800 (CM has capacitive and store electric energy)
• 𝜆𝜆𝑛𝑛 = 0 (CM is in resonance)
The third parameter for CMT is a model significance (MS). Equation (4.5) demonstrates the main
parameters that affect the significance of each CM to a radiated field and from equation (4.6) the term
1
�1+𝑗𝑗𝜆𝜆 � seems more compatible to express the variation of the 𝐽𝐽𝑛𝑛 rather than the variation of 𝜆𝜆𝑛𝑛 . This
𝑛𝑛
term represents the inherent normalized amplitude for each current mode 𝐽𝐽𝑛𝑛 and it is named the modal
significance. If its value close to 1, the mode meets the resonance condition.
1
𝑀𝑀𝑀𝑀𝑛𝑛 = �1+𝑗𝑗𝜆𝜆 � (4.8)
𝑛𝑛
From MS equation (4.8), we are able to calculate the resonance of the CM in addition to the operating
bandwidth of CM. The CM that has a resonance must be at 𝜆𝜆𝑛𝑛 = 0 and 𝑀𝑀𝑀𝑀𝑛𝑛 = 1 and the CM that
doesn’t contribute to the radiated field is at 𝑀𝑀𝑀𝑀𝑛𝑛 = 0. Furthermore, the half-power radiating bandwidth
can be calculated from the following approximation formula [178]:
𝐹𝐹ℎ (𝑀𝑀𝑀𝑀𝑛𝑛 = 0.707) − 𝐹𝐹𝑙𝑙 (𝑀𝑀𝑀𝑀𝑛𝑛 = 0.707) (4.9)
𝐵𝐵𝐵𝐵𝑛𝑛 ≈
𝐹𝐹𝑟𝑟𝑟𝑟𝑟𝑟 (𝑀𝑀𝑀𝑀𝑛𝑛 = 1)
𝐹𝐹ℎ and 𝐹𝐹𝑙𝑙 are the edges of the high and low frequency bands of any maximum where the MS is equal
to or higher than 0.707. 𝐹𝐹𝑟𝑟𝑟𝑟𝑟𝑟 is the resonant frequency where 𝑀𝑀𝑀𝑀𝑛𝑛 = 1 or the place of the highest MS.
Modal bandwidth is often a significant parameter in many CMT antennas, as it helps to determine the
radiation characteristics of the CM. It is important to understand, however, that the modal bandwidth
corresponds to the half-power bandwidth of the radiated pattern (for single-mode excitation) and not
an exciting structure's impedance bandwidth.
4.3 Dual Polarized Antenna for 5G

Due to the mastering of the dual-polarized antennas to introduce a solution in enhancing the
isolation and channel capacity, makes these antennas a good candidate for MIMO smartphone designs
[69, 87-90, 179, 180]. In [87] Yang Li et al., introduce a hybrid eight ports orthogonal dual-polarized
antenna for 5G smartphones; this antenna consists of 4 L-shaped monopole slot elements and C-shaped
coupled fed elements. The 4 L-shaped are printed at the corners and the 4 C-shaped are printed in the
middle on a thick 1mm FR-4 substrate. This design achieves 12.5 dB, 15 dB for the isolation and the
cross-polarization, respectively. Over the past months, Zaho et al. [88] presented a 5G/WLAN dual-
polarized antenna based on the integration between inverted cone monopole antenna and cross bow-
79
tie antenna for VP and HP, respectively, where a 90◦ phase difference feeding network feeds the cross
bow-tie antenna, so, the separated power divider and phase shifter are introduced to be used as a
feeding network. In [89] Huang et al. introduce a dual-polarized antenna that consists of a main
radiator, an annulus, and a reflector. The main radiator consists of two pairs of differentially-driven
feedlines to transmit the energy to the coplanar patch. This structure achieves 26 dB and 35dB for the
isolation and the cross-polarization, respectively. Eight-ports dual-polarized antenna array is reported
in [90], the proposed antenna array is composed of four square loops and each loop is excited by two
orthogonal fed coupled feeding strip.
4.3.1 Antenna Design

To achieve a dual-polarized antenna, two slot antennas are introduced and etched on the same
substrate as shown in Figure 4.1. The proposed antenna consists of two orthogonal slots to achieve
pure dual-polarization and gives two pattern diversity.
The proposed antenna is implemented on a Rogers 4003C substrate with a dielectric constant 3.38,
tangential loss of 0.0027 and thickness h=0.2 mm. The thin substrate is selected to reduce the losses at
the millimeter band and to be compatible with the end launch connector (1.85 mm) with a very thin
pin. All the simulation, optimization and CMA were conducted with the computer simulation
technology (CST): time-domain solver and multilayer solver.
The slot antenna is one of the popular antennas used for smartphones in the last few years [65,
181-188] due to its simple structure and its multiple operating modes. Figure 4.1(a and b) shows the
proposed slot antenna (Ant. I), where a slot is etched on the top ground plane and is fed by the 50-ohm
microstrip line. The feed line is printed on the opposite side of the ground plane to feed the slot antenna.
The second antenna (Ant. II) is a slot antenna fed by the CPW line, as depicted in Figure 4.1 (c and d).
As a result of the thin layer substrate used, the width of the CPW-feeding line is wider than the inner
pin of the end-launch connector (The width of the CPW feed line without bottom ground is Wf2=2.5
mm, while the diameter of the end-lunch pin is 0.18 mm). Therefore, we used a CPW with a ground
plane to decrease the width of the CPW-feeding line and achieve 50-ohm input impedance and be
compatible with the end-launch connector. After this, the CPW-feeding line is tapered to match the
aperture/slot impedance.
80
(a) Front view of Ant. I (b) Back view of Ant. I
(c) Front view of Ant. II (d) Back view of Ant. II
(e) Front view of Ant. III (f) Back view of Ant. III
Figure 4.1 Geometry of proposed antenna
The third antenna (Ant. III: proposed antenna) consists of the integration between Ant. I and
Ant. II configurations to achieve dual-polarization from the two antennas. The feed lines of antenna I
and antenna II are printed on a different face of the substrate as shown in Figure 4.1(e and f),
furthermore, the feed lines are orthogonal together. Moreover, the two slots are used to prevent the
coupling between the two feed lines. Table 4. 1 gives the geometric dimensions of the proposed
antenna (all dimensions in mm).
81
Table 4. 1 Geometric parameters of the proposed antennas (mm)

Parameters An. I Ant. II Ant. III
L 20 20 10
W 20 20 20
Ls1 5 --- 5
Ws1 0.6 --- 0.6
Y1 1.6 --- 1.6
Wf1 0.428 --- 0.428
Lf1 11.6 --- 6.6
Ls2 --- 3.8 3.7
Ws2 --- 1.2 1.2
Wt --- 0.225 0.225
Lt --- 0.4 0.4
Lc1 --- 3.5 3.5
Lc2 --- 4 4
G --- 0.2 0.2
Wc --- 0.45 0.45
Wg --- 3.5 3.5
D --- --- 0.9
4.3.2 CM Analysis of Antennas

The CMA is introduced as a first step to provide a method to discover the physical essence of
the proposed antennas. In addition to enhancing the bandwidth of the proposed antennas by creating
multiple resonant modes using CMA. Figure 4.2 shows the characteristic mode analysis of Ant. I, we
notice that the slot antenna achieves multiple operating modes around 28 GHz. Any mode can be
considered as a resonant mode when achieve MS=1 (MS>0.707) in addition to achieve 𝜆𝜆𝑛𝑛 = 0
and 𝛼𝛼𝑛𝑛 = 1800 . However, modes number 4-8 don’t have eigenvalue equal to zero but they are very
closed to the zero and their modal significant values more than 0.98, therefore they don’t consider as
a pure resonant mode. Therefore, these modes consist of a small capacitive loaded and a small amount
from electric energy. Figure 4.3 depicts the characteristic mode analysis of Ant. II, one can notice that
the CPW slot antenna gives a wide bandwidth for resonant modes than that of Ant. I. Figure 4.4 shows
the characteristic mode analysis for Ant. III, we can notice that their modes are a combination between
modes of Ant. I and Ant. II. Moreover, this antenna achieves wide bandwidth according to their MS
values. Table 4. 2 shows a comparison between the resonant frequencies for each characteristic mode
of the aforementioned antennas. We notice that Ant. III has a large number of resonant modes within
the simulated range.
82
(a) Eigen values (b) Characteristic angles
(c) Modal Significance

Figure 4.2 Characteristic mode analysis of Ant. I
2
270
n )(Degree)
0 1 2 3 4 5
)
240
n
6 7 8 9 10
-2
210
Eigen Values (
1 2 3 4 5
Characteristic Angle (
-4
6 7 8 9 10
180
-6 150
-8 120
22 24 26 28 30 32 34 36 22 24 26 28 30 32 34 36
Frequency (GHz) Frequency (GHz)
83
) n
0.8
Modal Significance (MS

0.6
MS MS MS MS MS
1 3 5 7 9
0.4 MS MS MS MS MS
2 4 6 8 10
0.2
22 24 26 28 30 32 34 36
Frequency (GHz)

Figure 4.3 Characteristic mode analysis of Ant. II
0
)(Degree)
1 2 3 4 5
250
) n
-5 6 7 8 9 10
n
Eigen Values (
-10 1 2 3 4 5 200
Characteristic Angle (
6 7 8 9 10
-15
150
-20
22 24 26 28 30 32 34 36 22 24 26 28 30 32 34 36

1
n)
0.8
0.6
0.4 MS MS MS MS MS
1 3 5 7 9
MS MS MS MS MS
2 4 6 8 10
0.2
0
22 24 26 28 30 32 34 36
Frequency (GHz)

Figure 4.4 Characteristic mode analysis of Ant. III
84
Table 4. 2 Comparison between resonant frequencies for CM of three antennas

Modes Ant. I Ant. II Ant. III
1 28.6 27.03 28
2 25.04 29.1 27.6
3 29 31.07 29.28
4 32.96 29.45 30.6
5 X 36 36
6 X 25.25 29.69
7 X X 31.4
8 X X X
9 X X X
10 32.3 X 23
*X: mean that this mode doesn’t have resonant frequency through the simulated bandwidth.
4.3.3 Slot Antenna Designs

Figure 4.5 shows the current and the electric field distributions of the slot antennas I and II. In
the case of Ant. I, the magnitude of surface current on the ground plane are the strongest above the
microstrip line thus the slot executes the highest disruptive impact on this current. The current is
completely impeded near the center of the slot and induces a charge build-up on the long sides of the
slot and this like the capacitor. Moreover, the current near to the parts of the slot bends around its ends
to continue to pass and this like the inductor, Therefore, the slot antenna is equivalents to two shunt
transmission lines shunted by a parallel integration of an inductor and a capacitor. Also, a radiation
resistor can be added to the equivalent circuit of slot antenna for long slots.
(a)Current distribution and electric field for Ant. I
85
(b) Current distribution and electric field for Ant. II
(c)Simulated electric field Ant. I (d) Simulated electric field Ant. II

Figure 4.5 Expected and simulated electric field and current distributions for Ant. I and Ant. II.
𝜆𝜆
The proposed slot antenna operates with a length equivalent to � 𝑔𝑔�2� from resonant frequency
at the working:
𝑐𝑐 (4.10)
𝐿𝐿𝑠𝑠 =
2𝑓𝑓𝑟𝑟 �𝜖𝜖𝑒𝑒𝑒𝑒𝑒𝑒
where 𝐿𝐿𝑠𝑠 is slot length, c is the velocity of light free space, 𝑓𝑓𝑟𝑟 is the resonant frequency, and 𝜖𝜖𝑒𝑒𝑒𝑒𝑒𝑒 is
the relative effective permittivity of proposed antenna. Figure 4.6 shows the reflection coefficient of
Ant. I at different values of length and width of the slot. For Ant. II, the wide bandwidth is achieved
due to the multiple resonance mode are excited by the combination of the CPW and aperture antenna.
The resonant frequency and BW are tuned by the length and width of aperture antenna (Ls2, Ws2).
Figure 4.7 shows the effect of length and width of the aperture antenna on the operating bandwidth.
86
-20
| (dB)
-40
11
Ls =4 mm Ls =5 mm
1 1
|S
Ls =4.5 mm Ls =5.5 mm
-60 1 1
24 26 28 30 32 34
Frequency (GHz)
(a)Reflection coefficient at different values of slot length

0
-20
| (dB)
Ws =0.2 mm Ws =0.8 mm
1 1
11
-40 Ws =0.4 mm Ws =1 mm
|S
1 1
Ws =0.6 mm
1
-60
24 26 28 30 32 34
Frequency (GHz)
(b) Reflection coefficient at different values of slot width
Figure 4.6 Reflection coefficient of Ant. I at different values of slot width and slot length.
87
-10
| (dB)
-20 Ls =2.6 mm Ls =3.8 mm

2 2
11
Ls =3 mm Ls = 4mm
2 2
|S
-30 Ls =3.4 mm Ls =4.2 mm

2 2
Ls =3.6 mm
2
-40
24 26 28 30 32 34 36
Frequency (GHz)
(a)Reflection coefficient at different values of slot length

0
-10
| (dB)
-20
11
Ws =0.8 mm Ws =1.6 mm
|S
2 2
-30 Ws =1.2 mm Ws = 2 mm
2 2
24 26 28 30 32 34 36
Frequency (GHz)
(a)Reflection coefficient at different values of slot width

Figure 4.7 Reflection coefficient of Ant. II at different values of slot width and slot length.
Ant. I has vertical polarization and Antenna II has horizontal polarization. Ws1, Ws2 are the
dimensions of slot widths to control the matching of vertical and horizontal modes, respectively.
Furthermore, y1 is a tuning parameter for matching port 1, and Wt, Lt are parameters to optimize the
matching at port 2. To consider the practical case, we consider the end launch connector in our designs
as shown in Figure 4.8. Therefore, the feed lines are increased by 5 mm in x and y direction to avoid
the interconnection between the two connectors.
88
(a)Front view (b)3-D view

Figure 4.8 Proposed antenna design with connectors.
A high separation between the two ports can be achieved due to the orthogonality
characteristics and symmetric/antisymmetric characteristics of the three modes of CPW (with the
ground).
Figure 4.9 Simulated S-parameters of two ports antenna.
The reflection coefficients and isolation coefficients of the proposed two ports antenna are
shown in Figure 4.9. One can notice that the isolation coefficients between the two ports have a high
value through the operating bandwidth (more than 45 dB) and the antenna has good matching for (S11
and S22). The antenna achieves 2.2 GHz as a wide bandwidth from 27 GHz to 29.2 GHz when port 1
89
is excited and 5 GHz from 25.6 GHz to 31.6 GHz for port 2. The proposed antenna achieved common
bandwidth (2.2 GHz) to cover the 5G applications at 28 GHz.
The Ant. III is redesigned without ground in the bottom (the width of CPW line without ground
is recalculated) to make the antenna with only one common ground in the top. The S-parameters of
this design is shown in Figure 4.10. The results still ensure that the antenna has good matching and
high isolation between its ports.
0
S
11
S
22
-10
S
21
S
12
-20
S-Parameters (dB)
-30
-40
-50
-60
-70
22 24 26 28 30 32 34
Frequency (GHz)
Figure 4.10 Simulated S-parameters of the two ports antenna (CPW without ground in the bottom).
Figure 4.11 and Figure 4.12 show the surface current and electric field distributions for two
ports. The surface currents and electric field of the proposed antenna at 28 GHz for two ports are
presented to ensure that the proposed antenna achieves dual-polarization between their ports. It is clear
to note that the surface current and electric field flow along the y-axis when port 1 is fed. While they
flow along the x-axis when port 2 is excited. Therefore, dual-polarization is achieved.
90
Figure 4.11 Current distribution of proposed antenna at 28 GHz from two ports
Figure 4.12 Electric field distributions from two ports.

The radiation patterns of the proposed antenna from port 1 and port 2 in both XZ and YZ planes at 28
GHz are shown in Figure 4.13. We can observe that the cross-polarization levels in both planes are
less than 40 dB as compared with the co-polarizations. The antenna achieved gain is 6.23 dBi and 6.85
dBi as shown in Figure 4.14.
91
Port XZ Plane (Phi=0) YZ Plane (phi=90)

Port 1
Cross-Polarized Co-Polarized Co-Polarized Cross-Polarized

(a) (b)
Port 2
0
30 330
60 300
0 -20
-40 -
60 -8
0 -1 00
90 270
120 240
150 210
180
Co-Polarized Cross-Polarized Cross-Polarized Co-Polarized

(c) (d)
Figure 4.13 Co-Polarized and Cross Polarized for port1 and port 2.
92
8.5
Port 1
8 Port 2
7.5
7
Gain (dBi)
6.5
5.5
5
22 24 26 28 30 32 34 36
Frequency (GHz)
Figure 4.14 Gain of proposed antenna.
4.4 Metasurface Design

Recently, the metasurface (MTS), which is a 2-D structure of metamaterial, has been used to
improve the bandwidth and give a low profile of microstrip antennas by placing the metasurface unit
cells in manipulating electromagnetic waves direction. The MTS is considered one of radiating
surfaces that can radiate electromagnetic waves by exciting it through a nearby external antenna.
Extensive works have been introduced in order to analyze the characteristics of MTS such as the
equivalent circuit, dispersion diagram, periodic structures, surface impedance tensor, transmission and
scattering coefficients, theory of effective medium, and reflection phase diagram [189-191]. The CMA
has been used in the past few years to predict the characteristics and behaviour of MTS in the accurate
form [192-195].
4.4.1 Analyze Metasurface using CMA

This section introduces the analysis of MTS and the comparison between the antenna and MTS unit
cells. The metasurface of 7x7 unit cells (square patches) is proposed at 28 GHz, as shown in Figure
4.15. The square patches are printed on Rogers RO 4003C substrate with dielectric constant 3.38 and
a thickness of 0.2 mm. the CMA solver from CST microwave studio is used with free-space boundary
conditions to analyze this structure. To examine the metasurface modal behaviors, the first ten modes
are calculated (Eigenvalues, characteristic angles, and modal significances) in the range from 18 GHz
93
to 38 GHz. Over the band, we notice that only the modes from 1 to 9 have resonant frequencies. In
this design, we have two groups from degenerate modes (J1/J2 and J7/J8) resonant around 28 GHz and
31 GHz, respectively. In this design, vertical and horizontal polarizations are desired. Therefore, we
need two similar modes (one is VH and the other is HP) over the proposed band. In this design, we
have two degenerate modes (J1/J2 and J7/J8) that can be used, but the other modes are not considered
(6). On the other hand, J7/J8 are at the end of the operating band (31 GHz). Figure 4.16 (a)-(c) shows
that the modes J1/J2 are the only two modes with pure resonant at 28 GHz. Also, J1\J2 have a
characteristic angle equal to 1800 at 28GHz in addition to zero Eigenvalue at 28GHz.
Figure 4.15 Metasurface structure (W1=1.7 mm, g=0.2 mm, Wm=13.3 mm).
94
0
2
-10
) n
-20 -2
Eigen Values (
-4
-30
25 30 35
-40 1 2 3 4 5
6 7 8 9 10
-50
18 20 22 24 26 28 30 32 34 36 38
Frequency (GHz)
(a)Eigen values
(b)Characteristic angles
95
1
J
5
) J J ,J
n 3 1 2
0.8
J
4
0.6
J
J 10
0.4 9
J ,J
J 7 8
6
0.2 MS MS MS MS MS
1 3 5 7 9
MS MS MS MS MS
2 4 6 8 10
0
20 25 30 35
Frequency (GHz)
(c)Modal Significance
Figure 4.16 CMA parameters of MTS.
The modal electrical surface currents are shown in Figure 4.17 and the current directions of
each mode are indicated by black arrows. All the field and current in this section are calculated at 28
GHz. As can be noticed, first modal current (J1) is in phase through the MTS and its polarization in
the y-direction. Also, the second modal current (J2) is typical as (J1) but with 900 out of phase. In other
words, J2 directs in the x-direction through the MTS. Therefore, J1/J2 are a pair of orthogonal modes.
As a result of all currents of first and second modes are in phase, they have pure broadside radiation
as shown in Figure 4.18. J3 and J5 have symmetrical distribution about the y-axis, and x-axis,
respectively, with null current at the center of MTS and null along the z-axis in the radiation pattern
as shown in Figure 4.18. The current of the third mode (J3) flows as a closed loop with null at the
center and thus it is like the behavior of inductive, which can be verified from its characteristic angle
about 28 GHz which is below 1800. The currents of mode J4 and mode J6 are self-symmetrical about y
and x, respectively. J7 and J8 are 900 out of phase and symmetrical around 450 from the x-axis and y-
axis, respectively. The last two modes have quasi-quadrature symmetric about the x-axis and y-axis at
the same time. Clearly, the only modes J1 and J2, have good main lobes, whereas the other modes have
96
split main lobes. Therefore, these are unacceptable modes according to (6) and the theory of mode
expansion.
97
Figure 4.17 Modal surface current of MTS.
98
Figure 4.18 Modal radiation patterns of MTS.
99
4.5 Proposed Antenna with MTS

In this section, the MTS is used to provide high gain, wide bandwidth, reduce the antenna size
and reduce the back radiation from slot antenna. The integration between the proposed antenna and
MTS is as following:
• The dual polarized antenna is based on two slots are introduced to operate at 28 GHz.
• Then we optimize the dimensions of metasurface unit cell to have the pair of orthogonal
modes (J1-J2) with broadside radiation. Moreover, the others modes of MTS are out of the
focused band and they have spilt the main lobe.
• This section introduces the integration between the dual slots antenna and the MTS. The
slots that are adjusted to operate at 28 GHz are used to excite the modes J1 and J2 of the
MTS that have pure resonant at 28 GHz.
The MTS is fed by the two slot antennas; therefore, the two small slots are etched from the MTS and
aligned to the slots of the antenna to increase the coupling between the antenna and MTS. Figure 4.19
delicates the configuration of proposed antenna with MTS layer. The optimized dimension of the
antenna after integrated with MTS are shown in Table 4. 3. The overall dimensions of antenna is
extended by 5 mm in x and y to be compatible with the end launch connector (1.85 mm). The proposed
antenna printed on Rogers 4003C with dielectric constant 3.38 and thickness 0.2 mm. The prototype
of the proposed antenna is shown in Figure 4.19 (c).
(a)3-D structure (b)Top view
100
(c)MTS layer (d)Slots layer (e)Bottom layer

Figure 4.19 Configuration of proposed antenna with MTS.
Table 4. 3 Dimensions of proposed MTS antenna (mm)

L W Ls1 Ws1 Y1 Wf1 Lf1 Ls2 Ws2 Wt Lt Lc1
20 20 4.8 0.6 1.6 0.428 6 3.2 1.1 0.21 0.4 3
Lc2 G Wc Wg d Wm W1 gm Wm1 Wm2 Lm1 Lm2
3.5 0.2 0.45 3.5 0.9 13.3 1.7 0.2 0.4 1 4.8
(a)Front view (b)Back view

Figure 4.20 Prototype of the proposed antenna with MTS.
Figure 4.21 shows the measured and simulated reflection coefficients and isolation coefficients
of the ports for the proposed antenna. The results confirmed that the proposed antenna achieves wide
bandwidth from two ports (26.5-29.5 GHz for port 1 and 25.5 – 30 GHz for port 2) that satisfy the
requirements of 5G in term of bandwidth. The measured operating frequency of the proposed antenna
is at 28 GHz, which is in good agreement with the simulated result. The proposed antenna achieves
good isolation between its ports (more than 40 dB).
101
-10
-20
S (S) S (M)
11 11
S-Parameters (dB)
-30 S (S) S (M)

22 22
S (S) S (S)
21 12
-40 S (M) S (M)
21 12
-50
-60
-70
20 22 24 26 28 30 32 34 36
Frequency (GHz)
Figure 4.21 Simulated and measured S-parameters of single antenna (Ports 1 and 2).
The normalized radiation patterns of MTS antenna are shown in Figure 4.22 for port 1 and port 2 in
the x-z plane and y-z plane at 28GHz. The co and cross components are introduced with more than 40
dB as a difference between them. Furthermore, the MTS achieved low back radiation at the x-z and y-
z planes. The radiation efficiency of the two ports is illustrated in Figure 4.23. The efficiency of the
proposed MTS antenna is around 95% within the whole band. The gain of the proposed antenna is
shown in Figure 4.24; one can notice that the MTS is used to increase the gain of the proposed antenna
by about 4 dBi.
102
Port XZ Plane (Phi=0) YZ Plane (phi=90)

Port 1 0
30 330
60 300
0
-20
-40
-60
-80
90 -100 270
120 240
150 210
180
Cross-Polarized Co-Polarized Co-Polarized Cross-Polarized

(a) (b)
Port 2 0
0
30 330
30 330
60 300
60 300
0
0 -20
-20 -40
-40 -60
-60 -80
-80 90 270
90 270
120 240 120 240
150 210
150 210
180
180
Co-Polarized Cross-Polarized Cross-Polarized Co-Polarized

(c) (d)
Figure 4.22 Co-Polarized and Cross-Polarized for port1 and port 2 of MTS antenna at 28GHz.
103
100
Port 1
95 Port 2
90
85
Efficiency (%)
80
75
70
65
60
55
22 24 26 28 30 32 34 36
Frequency (GHz)
Figure 4.23 Efficiency of proposed antenna.

11
10.5
10
Gain (dBi)
Port 1
9.5 Port 2
8.5
22 24 26 28 30 32 34 36
Frequency (GHz)
Figure 4.24 Gain of proposed MTS antenna.
4.6 MIMO Antenna Design

4.6.1 Fabrication and Measurements
The multiple-input-multiple-output (MIMO) system is preferred for 5G smartphone
applications to meet the high demand to maximize throughput and quality of service. In other words,
the MIMO antenna technology is one of the most significant components of future wireless
104
communication schemes as it improves throughput without raising input power and bandwidth.
However, the incorporation of the MIMO antenna scheme into the same board for handheld devices
that have a small size is challenging owing to the high mutual coupling between the adjacent antenna
components, particularly when they are spaced less than a half-wavelength apart. In our proposed
MIMO antenna, the antenna elements are positioned at the corner of the handset board with a total
dimension 100 × 60 𝑚𝑚𝑚𝑚2 as shown in Figure 4.25 for design configurations and prototypes of MIMO
antenna with MTS.
(a)Front geometry (b)Back geometry
(c)Front photo (d)Back photo

Figure 4.25 MIMO configuration and prototype of proposed antenna with MTS.
105
-5
-10
Reflection Coefficients (dB)
-15 S S
11 55
S S
-20 22 66
S S
33 77
-25 S S
44 88
-30
-35
-40
-45
22 24 26 28 30 32 34
Frequency (GHz)
(a)Simulated reflection coefficients (b)Simulated mutual coupling
0 -30
-10 -40
Mutual Coupling (GHz)
S S
11 55
-20 -50
S S
22 66 S S
21 61
S S
33 77 S S
-30 -60 31 71
S S S S
44 88 41 81
S
51
-40
-70
22 24 26 28 30 32 34 36
22 24 26 28 30 32 34 36
(c)Measured reflection coefficients (d)Measured mutual coupling

Figure 4.26 S-parameters of proposed antenna with MTS.
Some of the antennas in smartphones require a common ground plane between its MIMO
elements. Therefore, the MIMO antenna with printed common ground plane on the top layer is
presented in Figure 4.27. The common ground plane does not have any significant changes on the
reflection coefficients of the MIMO elements, in contrast, it reduces the isolation between ports by
small significant amount as shown in Figure 4.28. One can notice that the worst isolation coefficient
is higher than 37 dB between port 1 and port 2. Furthermore, all ports have good matching and achieve
the required BW for 5G applications.
106
(a) (b) (c)

Figure 4.27 The configuration of MIMO antenna with common ground (a)Front view, (b)Back view and (c)3-D
view.
0 -30
S S
11 55
S S -40
22 66
-10
S S
33 77
Mutual Coupling (dB)
-50
S S
44 88
-20 S S
21 31
-60
S S
41 51
-30 S S
-70 61 71
S
81
-40 -80
22 24 26 28 30 32 34 36 22 24 26 28 30 32 34 36
(a) Reflection coefficients (b) Isolation coefficients

Figure 4.28 S-parameters of proposed MIMO antenna with common ground plane.
4.6.2 Envelope Correlation Coefficient

The envelope correlation coefficient (ECC) is one of the main parameters to evaluate the MIMO
performance. Where the ECC is used to calculate the similarity between the antenna performances
and evaluate the diversity between the elements of MIMO. The acceptable value of ECC should be
less than 0.5 [196-199]. Whereas the lower values of ECC mean that the two antennas are good
isolated. The ECC can be calculated as [200]:
∗ ∗
|𝑆𝑆𝑛𝑛𝑛𝑛 𝑆𝑆𝑛𝑛𝑛𝑛 + 𝑆𝑆𝑚𝑚𝑚𝑚 𝑆𝑆𝑚𝑚𝑚𝑚 |2
𝜌𝜌𝑛𝑛𝑛𝑛 = (4.11)
�1 − (|𝑆𝑆𝑛𝑛𝑛𝑛 |2 + |𝑆𝑆𝑚𝑚𝑚𝑚 |2 )� �1 − (|𝑆𝑆𝑚𝑚𝑚𝑚 |2 + |𝑆𝑆𝑛𝑛𝑛𝑛 |2 )�
Where 𝜌𝜌: ECC, S:S-parameter, S*: complex conjugate of S-parameters, m, and n are number of
antenna m,n =1,2,….,8.
Figure 4.29 shows the ECC between MIMO elements from simulated and measured data. It is
obvious from the figure that the ECC is less than 0.001 within the operating band.
107
-3
10
2.5
12 13 14 15 16
2
17 18 23 27 28
1.5
ECC
0.5
0
24 26 28 30 32 34
Frequency (GHz)
(a)Simulated ECC
-3
10
3
12 13 14 15 16
2.5
17 18 23 27 28
2
1.5
ECC
0.5
0
24 26 28 30 32 34
Frequency (GHz)
(b)Measured ECC
Figure 4.29 ECC parameter of proposed antenna with MTS.
The ECC can be calculated based on the radiation pattern as:
4𝜋𝜋 2
��⃗
�∬0 �𝐹𝐹 ��⃗
m (𝜃𝜃, ∅) × 𝐹𝐹n (𝜃𝜃, ∅)�𝑑𝑑Ω�
𝜌𝜌𝑚𝑚𝑚𝑚 = 2 2
(4.12)
4𝜋𝜋 4𝜋𝜋
��⃗
∬0 �𝐹𝐹 ��⃗
m (𝜃𝜃, ∅)� 𝑑𝑑Ω ∬0 �𝐹𝐹n (𝜃𝜃, ∅)� 𝑑𝑑Ω
Where 𝜌𝜌: ECC, F(𝜃𝜃, ∅): radiation patterns of antenna #m or #n, m and n are number of the antenna
m,n =1,2,….,8.
108
Figure 4.30 shows the different ECC values between the MIMO elements (two elements each time)
based on the 3-D radiation pattern of each element. The values of ECC is very small due to the different
polarization between the neighbour antennas. It is observed that the values of ECC is less than 0.02
and this means that the MIMO antenna has a good diversity performance.
0.02
0.018 12 13 14 15 16
0.016 17 18 23 27 28
0.014
0.012
0.01
ECC
0.008
0.006
0.004
0.002
0
26 27 28 29 30 31 32
Frequency (GHz)
(a)Simulated ECC
0.018
12 13 14 15 16
0.016
17 18 23 27 28
0.014
0.012
0.01
ECC
0.008
0.006
0.004
0.002
26 27 28 29 30 31 32
Frequency(GHz)
(b)Measured ECC
Figure 4.30 ECC parameter of the proposed antenna with MTS.
4.6.3 Smartphone Modeling

(A) Impacts of Housing and Phone Components
To simulate the real environment of the smartphones, the antenna is integrated with the housing
and the components of mobile as shown in Figure 4.31. The screen, speaker, camera, battery, and other
components are considered with the proposed MIMO antenna and all components are covered by
plastic material. The materials of each part are tabled in Table 4. 4. The module of liquid crystal display
(LCD) consists of two parts; the LCD panel and the LCD shield that have the same size of PCB. The
109
battery cell is placed inside the battery shield as shown in Figure 4.31. There are top and bottom fillers
to fix the board. Four plastic holders are used to fix the LCD panel and the LCD shield on the filler.
The dimensions of all components are compatible with commercial smartphones. The MIMO antenna
is tested inside the phone taking the housing and the components into considerations. The S-parameters
of the proposed antenna are shown in Figure 4.32. One can notice that the reflection coefficients from
all elements are affected due to the existence of the housing. The reflection coefficient of the ports are
slightly shifted but are still have good matching and achieve the requirements for millimeter 5G. On
the other hand, there is a high isolation between ports. The 3-D radiation patterns of MIMO elements
are presented in Figure 4.33.We can notice that the radiation patterns are in different directions due to
the diversity between the elements.
Table 4. 4 Materials of smartphone

Part Material Properties
Housing Plastic 𝜀𝜀𝑟𝑟 =2.2, 𝑡𝑡𝑡𝑡𝑡𝑡𝑡𝑡=0.005
Camera Glass(Pyrex) 𝜀𝜀𝑟𝑟 =4.82, 𝑡𝑡𝑡𝑡𝑡𝑡𝑡𝑡=0.0054
LCD shield Metal(Copper) 𝜎𝜎=5.8e7 (S/m)
LCD panel Glass(Pyrex) 𝜀𝜀𝑟𝑟 =4.82, 𝑡𝑡𝑡𝑡𝑡𝑡𝑡𝑡=0.0054
Holder Plastic 𝜀𝜀𝑟𝑟 =2.2, 𝑡𝑡𝑡𝑡𝑡𝑡𝑡𝑡=0.005
Connectors Plastic 𝜀𝜀𝑟𝑟 =2.2, 𝑡𝑡𝑡𝑡𝑡𝑡𝑡𝑡=0.005
Battery cell Metal(Copper) 𝜎𝜎=5.8e7 (S/m)
Battery shell Plastic 𝜀𝜀𝑟𝑟 =1.5
Filler Plastic_HDPE 𝜀𝜀𝑟𝑟 =2.3
Speaker Plastic 𝜀𝜀𝑟𝑟 =2.2, 𝑡𝑡𝑡𝑡𝑡𝑡𝑡𝑡=0.005
Antenna PCB Rogers 4003C 𝜀𝜀𝑟𝑟 =3.38, 𝑡𝑡𝑡𝑡𝑡𝑡𝑡𝑡=0.0027
*HDPE: High Density Polyethylene
(a)Top view without cover and LCD (b)Bottom view without cover
110
(c)Top view (d)Bottom view

Figure 4.31 Mobile modelling with the components.
-20
S-Parameters (dB)
-40
S S S S S
11 44 77 31 61
-60
S S S S S
22 55 88 41 71
S S S S S
33 66 21 51 81
-80
24 26 28 30 32 34
Frequency (GHz)
Figure 4.32 S-Parameters of the proposed MIMO antenna inside housing.
111
Figure 4.33 3-D radiation pattern of all ports at 28 GHz (with housing).
The Research on health risk from the electromagnetic waves produced from wireless terminals
is introduced in the literature. The Specific absorption rate (SAR) is a figure of merit for evaluating
the power absorbed by the human tissues. For the frequencies used by current mobile communications
networks of second, third and fourth generation (2 G, 3 G and 4 G), basic constraints on RF-EMF
exposure are defined in terms of the Specific Absorption Rate (SAR) to avoid, broad safety margins,
adverse health effects associated with excessive localized tissue heating and heat stress of the whole
body [201-211]. The SAR quantifies the absorbed energy per unit of tissue volume. The SAR values
should follow one of two standards: the American standard (1.6 w/kg) for each 10 g and the European
standard (2 w/kg) for each 1g [212-215].
112
Figure 4.34 SAR distribution from MIMO elements.
Table 4. 5 SAR values (W/kg)

Port 1 2 3 4 5 6 7 8
SAR (1g) 0.69 0.71 0.64 0.73 0.75 0.78 0.67 0.67
SAR (10g) 0.48 0.49 0.501 0.52 0.54 0.496 0.57 0.45
For the millimeter wave range, there is two approaches to calculate the electromagnetic exposure to
the human:
1. SAR: Some papers in the literature evaluate electromagnetic exposure by the same previous
definition of SAR [9].
2. Power density: the term to calculate the electromagnetic exposure into the human body
changed from SAR to the term of power density (Pd) because the absorption becomes more
superficial due to the fact that penetration is very low at higher frequencies [216-219].
Figure 4.34 shows SAR distribution from 8 elements for 10g standard. The SAR values for the two
standards are summarized in Table 4. 5. The antenna is proximity close to the human head model with
0.5 mm distance and inclined as take mode by (600). The reference power of the proposed antenna
elements at 28 GHz is set to 24 dBm for each element.
113
Table 4. 6 Power density limits from different standards
ICNIRP [213] FCC[212] IEEE[214, 215]

F(GHz) 10-300 6-100 3-30 30-100
10, (20 cm2) 10, (100 𝜆𝜆2) 10, (100 cm2)

Pd (W/m2), A 10, (1 cm2)
200, (1 cm2) 18.56f0.699 200, (1 cm2)
Table 4. 7 Power density values at 28 GHz from different ports according to different standards.
Port 1 Port 2 Port 3 Port 4 Port 5 Port 6 Port 7 Port 8
ICNIRP 1.32 1.32 1.35 1.32 1.28 1.25 1.35 1.35
FCC 1.97 1.96 1.99 1.99 1.98 1.98 1.97 1.97
IEEE 1.71 1.69 1.78 1.77 1.68 1.69 1.71 1.71
In the second approach, IEEE, FCC and International Commission on Non- Ionizing Radiation Protection
(ICNIRP) introduced frequency limits at which the definition of SAR calculation shifts to power density calculation as
shown in Table 4. 6. The conversion frequency at which this shift in exposure metric is 3 GHz, 6 GHz and 10 GHz, for
IEEE, FCC, and ICNIRP, respectively. In other words, at mm-Wave frequencies, PD is currently preferred due
to the difficulty of determining a reasonable volume for SAR assessment when penetration depths are
very low [212-215]. The power density exposure into the human model is calculated as shown in Table
4. 7 for all ports and compare its values from different standards. We noted that all the power densities
satisfy the safety guidelines. Table 4. 8 lists two comparison sections; the first section makes a
comparison between the proposed antenna and the referenced dual-polarized antennas, and the second
section makes a comparison between the proposed antenna and the referenced MIMO antennas of
smartphones. High isolation, low profile, low complexity, compact size, high efficiency, high gain,
high cross-polarization are achieved in the proposed antenna.
114
Table 4. 8 Comparison between referenced antennas and the proposed antenna

G
ai
X-
Isolation n Freq. Compli
Ref Size (𝝀𝝀𝟎𝟎 𝟑𝟑 ) (dB) (d
pol
(GHz)
Eff. (%)
cated
Remarks
(dB)
Bi
)
• Six layers.
8.16-
[220] 1.37×1.37×0.222 39 13 42 NA High • Based on integrated
11.15
cavity.
30.1- • Multilayer organic
[221] 3.2×3.2×0.1 20 3.8 25 High
30.9 buildup substrates.
• H-shaped slot antenna.
1.86-
[86] 0.93×0.93×0.004 26 4.5 28 NA High • 900 phase shift feeding
2.97
network.
• 2-substrate
• Dual-pol.
2.4- Mediu
[222] 1.75×1.75×0.02 35 8.6 20 NA • Circular dipole
4.12 m
• Microstrip line Balun
feed
• Fed by parallel strip
1.88-
[223] 1.73×1.03×0.144 30 9.4 20 NA High line balun.
2.9
• Bulk structure
Mediu
[224] 1.3×1.3×0.05 27 9.6 29 4.8-5 •
m
• SIW horn antenna
7.4 27.5- • Works as a waveguide
[225] 2.7×2.54×0.10 18 10 High
8 29.5 antenna.
• 3 layers
• Low profile
Proposed
0.83×0.83×0.03 40 11 40 25.5-30 92 Low • Two orthogonal slots.
antenna
• Dual feed.
MIMO Antenna for 5G Smartphones
Phone Isola Dual-
Ref. H0 MIMO Gain Eff.
Board ECC tion Pol. Remarks
(mm) Order (dBi) (%)
(mm2) (dB) (X.Pol.)
• C-shaped coupled-fed
[87] 136×68 5 8 0.15 12.5 yes (15) 55 and L-shaped
monopole slot.
• 3-D folded monopole.
[226] 150×75 6.2 8 NA 0.08 11 No 42 • Dual band @3.5 GHz,
and 5 GHz.
[183] 150×80 0.8 8 NA 0.05 17.5 No 62 • Open slot antenna
• 3 layers.
Yes
[227] NA 1.93 8 7 NA 20 90 • Yagi-uda
(18.3)
• Endfire radiation
• Low profile
Proposed Yes
100×60 0.4 8 11 0.001 40 90 • Two orthogonal slots.
antenna (40)
Dual feed.
4.7 Conclusion
This chapter introduces a dual-polarized MIMO antenna with eight elements for a 5G smartphone.
The MIMO configuration is based on the diversity between elements. The dual-polarization antenna
115
is introduced to overcome the high attenuation in 5G communication system and give high data rates.
Furthermore, the orthogonal polarization between the antenna ports is used to achieve high isolation
between antenna ports. The antenna achieved a good matching bandwidth of more than 2GHz at center
frequency of 28GHz. The antenna is combined with MTS to increase its gain and bandwidth. CMT
analyzes the MTS and all the parameters are investigated. The antenna is fabricated and measured.
The electromagnetic exposures into the human model from the proposed antenna at 28 GHz are
investigated and analyzed in terms of SAR and power density.
116
Chapter 5: Antenna Design for Short Range Communications
5 Chapter Five:
Antenna Design for Short
Range Communications
5.1 Introduction
Nowadays, low-frequency bands are very crowded and with the rapid growth of communication
technologies, high-speed short-range wireless communications require a wide band, higher data rate,
and compact size. In order to achieve the desired requirements, the millimeter-wave (mmW) band at
60 GHz has more and more attention because it offers unlicensed bandwidth (from 57 GHz to 64 GHz)
for several applications such as video streaming, wireless gaming, short-distance communication
WPAN [228, 229]. The complementary metal-oxide-semiconductor (CMOS) technology is considered
a good solution to cost and circuit integration issues at this frequency. However, the CMOS substrate
is inherited losses due to its high permittivity (εr=11.9) and low resistivity (σ=10 s/m). Additionally,
CMOS antennas at 60 GHz require more enhancements of antenna efficiency and antenna gain [228,
230, 231].
The inherent losses in CMOS substrate is a key factor in RF CMOS designs. So, several studies have
been performed to solve the problem of inherent losses in CMOS substrate due to its high permittivity
and low resistivity causes performance degradation. Different methodologies are presented to improve
the antenna on-chip performance, such as micromachining [232] and proton implantation [233].
Nevertheless, these techniques suffer from reeducation of system level integration and increase the
overall cost. On the other hand, Barakat et al. [31], introduce an Artificial Magnetic Conductor (AMC)
and High Impedance Surface (HIS) to improve radiation characteristics in the broadside antennas, also
introduce a shield plane inserted between the AOC and the lossy CMOS substrate to minimize the loss
[234].
This chapter focuses on end-fire antennas. Thus, the Yagi-Uda antenna and Slot Tapered antennas
are common end-fire antennas reported in previous works [29, 112, 235]. The previous 60-GHz Yagi
antenna designs using CMOS technology suffer low radiation efficiency and low gain communications
to replace the metal interconnects between chips [29, 235]. Bao et-al. [235], presented 60-GHz
differential Yagi antenna using 0.18-μm CMOS technology combine with AMC to improve the
radiation performance with overall size 2.45mm ×1.8mm and feed by differential feeding G-S-G-S-
D. However, the achievable gain is - 2.64 dB and the F/B ratio is 16.6 dB. Recently, El-saidy et-al
[29], improved gain of Yagi-Uda antenna by changing antenna location from -1.7 dBi at center of
117
CMOS to -0.7 dBi at 1700 µm far from the center. Moreover, in [112], the Vivaldi antenna is
introduced in order to give a gain of -0.4 dBi and radiation efficiency of 32% with an overall size of
785µm×930µm.
This chapter presents a solution for the low gain of the on-chip antenna systems and the poor
efficiency of this system with outdoor systems. The high gain antennas on the two side walls of the
on-chip system (OCS) are introduced to communicate between the on-chip antenna (OCA) and the
outdoor systems (ODS). Where the high transmission between the OCS and the ODSs is achieved by
using high gain antennas to transmit between them. So, a Quasi Yagi Antenna (QYA) and a Tapered
Slot Vivaldi Antenna (TSVA) are introduced to enhance the radiation properties of the end-fire radiator
in millimetre wave range for OCS. The proposed two antennas are introduced to use for point to point
communications. The antennas are designed using standard 0.18µm six metal-layer CMOS
technology. The first antenna is a Quasi Yagi-Uda consisting of a T-shaped meandered line that
operates as a driven dipole element connected to a Coplanar Wave Guide (CPW) through a Coplanar
Slot (CPS) line transition. A meandered parasitic strip on front of the driver operates as a director and
a planar arc is used as a reflector to reduce the back radiation. The overall size of the Antenna on Chip
(AOC) is 0.72×0.85mm2. The proposed antenna gives an end-fire radiation pattern with 0.3 dBi
simulated average gain and 45% radiation efficiency. The second introduced, antenna is a Vivaldi
antenna with three techniques to enhance the radiation properties. The first technique is the insertion
of an elliptical patch parasitic radiating element in the Vivaldi aperture to enhance the coupling
between arms and produce strong radiation in the end-fire direction. The second technique is the
addition of corrugation at the antenna edge to improve the antenna characteristics. The third technique
is the insertion of a planar reflector at the backend of the antenna, which greatly improves the front-
to-back (F/B) ratio. The overall size of the antenna on-chip is 0.5×0.87mm2. The proposed antenna
reveals an end-fire radiation pattern with 0.8 dBi simulated average gain and 37% radiation efficiency.
We need to introduce a 3-D mm-wave system in our proposed system, as shown in Figure 5.1. The
high gain antennas are required to be on the sides to communicate with another system at a long
distance. It is designed to operate at the same proposed band (57-64 GHz). In this chapter, we focus
on the design of the on-chip antennas to serve the connection between layers.
118
Figure 5.1 Build 3-D system Packaging
5.2 Yagi-Uda Antenna

The proposed antenna is designed using 0.18µm standard six metal layers CMOS (5×5 mm2 standard
dimensions). The Quasi-Yagi antenna is designed in metal six with overall size of 720µm×850µm as
shown in Figure 5.2. The antenna dimensions are tabulated in Table 5.1. The proposed Yagi antenna is
composed of a driven element, a parasitic director and a reflector. The driven element is a T-shaped
dipole with meander line shape in order to increase the path over which the surface current flows. It is
fed by a CPW transmission line so the need for a transition from CPW to CPS appears since the antenna
is fed by a CPS line. In the millimetre wave circuit, the CPW is used to be suitable with the Ground-
Signal-Ground G-S-G feeding standard. The analysis of this transition is introduced in [236].
Generally, in Yagi antenna design, a metallic strip is always used as a director to improve directivity
by directing the wave into the end-fire direction and to enhance the impedance matching in the high-
frequency band. The ground plane acts as a reflector in conjunction with a planar arc that acts as a
second reflector to prevent any radiation in the back side and direct all the radiated power to the front
direction.
119
(a)Top view (CST) (b)Top view (HFSS)
(c)Transition from CPW to CPS
(d) Side view

Figure 5.2 Yagi antenna geometry
120
Table 5.1 Antenna dimensions (µm)
Lg Lp Lc Lf L1 L2 L3 L4
150 40 100 30 400 160 100 300
Wg Wc W1 W2 W3 W4 W5 D
180 60 30 85 30 30 10 80
R1 R2 S
300 350 85
The proposed antenna is simulated using commercial CST Microwave Studio 2017 and HFSS
version 16. All the results are verified by the aforementioned softwares. The antenna performance is
studied in two cases, with using planar arc as a main reflector and using ground as a main reflector
without arc. Figure 5.3 shows the return loss of the antenna to cover band from 50 GHz to 80 GHz
with good matching in two different cases; case 1, the ground is used as reflector with Wg=350 µm
and in case 2, the planar arc used as reflector with Wg=180µm. The return losses from CST and HFSS
are close together.
-5
-10
-15
Return Loss (dB)
-20 Ground As Reflector (CST)

Planar Arc Reflector (CST)
Ground As Reflector (HFSS)
-25 Planar Arc Reflector (HFSS)
-30
-35
50 55 60 65 70 75 80
F(GHz)
Figure 5.3 Return loss of the Yagi antenna.
Figure 5.4 and Figure 5.5show the gain and radiation efficiency of the proposed antenna with and
without arc, respectively. We notice that the arc enhance the value of gain by 0.8 dBi because it
reflects the back radiation from the Yagi antenna. Furthermore, the gain of the antenna is verified
by using HFSS in the two cases and there are good agreement between the results. Furthermore, the
antenna efficiency is enhanced by using the arc to be about 45%.
121
0.5
0
Gain(dBi)
-0.5
Gain without Arc (CST)

-1 Gain without Arc (HFSS)
Gain with ARC (CST)
Gain wiht ARC (HFSS)
-1.5
50 55 60 65 70 75 80
F(GHz)
Figure 5.4 Gain of the Yagi-Uda antenna.

50
45
40
35
Radiation Efficiency (%)
30
25
20
Rad. Eff. with Ground As A Reflector (CST)
15 Rad. Eff. with Arc As A Reflector (CST)

Rad. Eff. with Ground As A Reflector (HFSS)
10 Rad. Eff. with Arc As A Reflector (HFSS)
5
50 55 60 65 70 75 80
F(GHz)
Figure 5.5 Radiation efficiency of the Yagi-Uda antenna.
Due to the inherent losses of the substrate, the antenna position on the substrate effects on the
antenna performance. So, the performance of the Yagi antenna is presented at different three positions,
as shown in Figure 5.6. The gain and the radiation efficiency are increased for P2 and P3, as shown in
Table 5.2 due to decreasing the effect of substrate. Figure 5.7 shows the radiation pattern of the antenna
in the XY plane and ZY plane at 60 GHz and 65 GHz. There is a good agreement between the simulated
radiation pattern from CST and HFSS, as depicted in Figure 5.7. Also, we notice that the radiation
pattern of the antenna is in the direction of the Y-axis to ensure that the antenna has end-fire radiation.
122
Figure 5.6 Yagi antenna positions.
Table 5.2 Comparison between different positions of the antenna

Positions Return Loss (dB) Gain(dBi) Rad. Eff.
P1 -24 -0.18 30
P2 -18 0.2 36
P3 -17 0.31 45
(a) (b)
Figure 5.7 The radiation pattern of the antenna in (a)XY plane and (b)YZ plane.
5.3 Tapered Slot Antenna

The Vivaldi antenna is designed using 0.18µm CMOS technology with the standard dimensions.
The CMOS consists of the silicon substrate (5×5 mm2 with height 200 µm, high permittivity (εr=11.9)
and low resistivity (σ=10S/m)) and a thin Sio2 layer (5×5 mm2 with thickness 10.4 µm and dielectric
constant εr=4) as shown in Figure 5.2 (d). The Vivaldi antenna is designed in a metallic six-layer with
an overall size of 500µm×870µm, as shown in Figure 5.8(a). The antenna dimensions are tabulated in
Table 5.3. The proposed Vivaldi antenna consists of two arms flared in opposite directions and
123
symmetrically rotated around the antenna aperture axis. The antenna is fed by the transition from
CPW to CPS according to the theory and the analysis introduced in [236] and the transition is
optimized as shown in Figure 5.8(c). This transition is introduced to make the feeding method suitable
with the millimeter circuit. Three different techniques are (as shown in Figure 5.9) introduced to
enhance the radiation characteristics of the antenna. The first technique is the parasitic elliptical patch
in the aperture area between the Vivaldi arms to increase the coupling between the two arms and to
produce strong radiation in the end fire. The second technique is the Sin corrugation of the two arm
outer edges. The corrugated edges are defined by 𝐴𝐴𝑐𝑐 𝑠𝑠𝑠𝑠𝑠𝑠(𝑚𝑚𝑚𝑚), m is fraction of variable y-axis and 𝐴𝐴𝑐𝑐
amplitude. The corrugation enlarges the effective aperture size to improve the gain. The final technique
is the addition of a planar reflector to prevent back radiation and to improve the front to back ratio.
(a)Top view (CST) (b)Top view (HFSS)
124
(c) Transition from CPW to CPS

Figure 5.8 Vivaldi antenna geometry
Table 5.3 Antenna dimensions (µm)
L W Lg L1 L2 W1 W2 A B
870 500 150 120 50 50 30 100 500
Ac Wc Ws Wt Wf S Lt R1 R1
50 25 55 22 12 20 110 80 40
125
Figure 5.9 Design steps of Vivaldi antenna
The transition from CPW to CPS is designed as the first step to show the amount of
transmission losses. The analysis of the transition from CPW to CPS is introduced in [236]. Figure 5.10
shows the S-parameters of transition from CPW to CPS on CMOS technology to ensure that the
transmission coefficient is acceptable. The simulated curves of return loss are shown in Figure 5.11 to
ensure that the antennas operate from 50 GHz to 70 GHz with good matching. The gain of four
antennas are introduced in Figure 5.12 to see the effect of ellipse shape is about 2dBi. Moreover, the
sin corrugation and the ground reflector increase the gain by 0.4 dBi and 0.9 dBi, respectively. The
radiation efficiency for the four types of antenna are presented in Figure 5.13. Figure 5.14 shows the
radiation pattern of the antenna in the XY plane and ZY plane at 60 GHz to ensure that it is end-fire.
-5
S (CST)
11
-10 S (CST)
21
S (HFSS)
-15 21
S-Parameters (dB)
S (HFSS)
11
-20
-25
-30
-35
-40
40 50 60 70 80 90 100
F(GHz)
Figure 5.10 S-Parameters of transition from CPW to CPS
126
-5
Antenna 1 (CST)
Antenna 2 (CST)
-10 Antenna 3 (CST)
Antenna 4 (CST)
Reflection Coefficient (dB)
-15
-20
-25
-30
50 52 54 56 58 60 62 64 66 68 70
F(GHz)
(a)S11 using CST

-5
Antenna 1 (HFSS)
-10 Antenna 2 (HFSS)
Antenna 3 (HFSS)
Reflection Coefficient (dB)
Antenna 4 (HFSS)
-15
-20
-25
-30
50 52 54 56 58 60 62 64 66 68 70
F(GHz)
(b)S11 using HFSS

Figure 5.11 Reflection coefficient of the proposed Vivaldi antenna for four cases using CST and HFSS
-1
Antenna 1 (CST)
Gian (dBi)
-2 Antenna 2 (CST)
Antenna 3 (CST)
-3 Antenna 4 (CST)
Antenna 1 (HFSS)
Antenna 2 (HFSS)
-4
Antenna 3 (HFSS)
Antenna 4 (HFSS)
-5
50 52 54 56 58 60 62 64 66 68 70
F(GHz)
Figure 5.12 Gain of Vivaldi antennas
127
40
35
Radiation Efficiency (%)
30
Antenna1 (CST)
Antenna 2 (CST)
25 Antenna 3 (CST)
Antenna 4 (CST)
Antenna 1 (HFSS)
20 Antenna 2 (HFSS)
Antenna 3 (HFSS)
Antenna 4 (HFSS)
15
50 55 60 65 70 75 80 85 90
F(GHz)
Figure 5.13 Radiation efficiency of Vivaldi antennas
(a) (b)
Figure 5.14 The radiation pattern of the Vivaldi antenna in (a)XY plane and (b)YZ plane
The characteristics of proposed antennas are compared with other published papers, as shown
in Table 5.4. We notice that the proposed two antennas introduce high gain and efficiency compared
with the antenna in literature due to using different techniques for each antenna to enhance its
performance. Furthermore, the gain of the Vivaldi antenna is best than that of the Yagi-Uda antenna;
in contrast, the efficiency of Yagi-Uda is better than that of Vivaldi because the metallic loss in the
Yagi-Uda is small compared with the Vivaldi antenna.
128
Table 5.4 Comparison with previous CMOS antennas

F/B ratio
Reference CMOS (µm) F(GHz) Size (mm2) Gain (dBi) Eff. (%)
(dB)
[234] 0.18 60 2.45×1.8 -2.64 16.8 16.66
[235, 237] 0.18 60 1×0.87 -0.746 21.4 11.01
[29] 0.18 60 0.785×0.94 -0.4 32 NA
[96] 0.18 60 1.1× 0.95 -8 10 9
[238] 0.13 60 2.2× 1.3 -0.2 19.6 NA
[239] CMOS 60 0.5×0.15 -20 NA NA
[240] 0.18 9.45 2×2.1 -29 21.1 NA
[91] CMOS 60 1.86×1.86 -1 NA NA
[241] 28 33 0.66×0.85 -1.8 30 NA
[242] 65 24 2.5× 2.5 -1 41 10
[243] Bi-CMOS 165 1.26×0.92 0.3 32 NA
Proposed Antenna
0.18 60 0.631× 0.46 0.35 45 16.3
(QYA)
Proposed Antenna
0.18 60 0.5× 0.87 0.8 37 18
(TSVA)
5.4 Design of MIMO on-chip Antenna

In recent years, Multiple-Input-Multiple-Output (MIMO) have been developed to be essential
in most of wireless communications systems to make best use of the multipath scattering phenomena
either to increase the system capacity, to enhance the overall channel gain or to mitigate the fading
effect according to the type of channel. Furthermore, the MIMO technology is used to improve
reliability, excessive increase of data rates and enhancement of capacity without exhaustion of
transmitted power and bandwidth [244]. The antenna designers confront two main problems in any
MIMO design; the first one is the implementation of multiple antennas in the close size of the portable
devices whereas the second one is the isolation between elements to reduce mutual coupling in the
proposed band. Therefore, the performance of the antenna is deteriorated with increasing the mutual
coupling between the elements.
5.4.1 Design of Two Elements

This part introduces three different configurations of the MIMO On-chip (MOC) for two elements.
Configuration I (Conf. I) introduces two elements side by side configuration with gap=120𝜇𝜇𝜇𝜇, Conf.
II introduces two elements side by back configuration and Conf. III introduces two elements back by
129
back configuration. The three configurations of the MIMO antenna structure have two elements that
provide better isolation between elements without using any additional technique or decoupling circuit.
The optimized geometry of the proposed three different configurations based on achieving maximum
isolation is shown in Figure 5.15. Here, the ground plane and the arc play a significant role in the
isolation performance of the proposed antenna. Furthermore, the diversity of the antenna positions
helps to reduce coupling and achieve better isolation among them. The CST microwave studio and
HFSS are used together to verify the simulated results for the S-parameters as illustrated in Figure
5.16. In this figure, only S11, and S21 are simulated because of the symmetrical arrangement of antenna
elements in the structure. A good agreement between the simulated results of CST and HFSS is
obtained. Table 5.5 shows a comparison between the three aforementioned configurations. One can
notice that all three configurations offer good matching and high isolation because of using arc as
reflector between the elements. The isolation between elements is more than 40 dB that provides an
enhancement in the radiation properties of antennas. We notice that Conf. II has low isolation because
the distance between its ports (1 and 2) is small compared to the other configurations but it still has
isolation of 40 dB. On the other hand, Conf. II, and Conf. III achieve diversity in the radiation pattern
and this diversity is the main factor in the MIMO designs.
2 2
1
1 2 1
(a)Conf. I (b)Conf. II (c)Conf. III
Figure 5.15 Different configurations of two elements MIMO Yagi-Uda antenna

0
-10
-20 S (CST)
11
S (CST)
S-Parameters (dB)
21
-30 S (HFSS)
11
S (HFSS)
21
-40
-50
-60
40 45 50 55 60 65 70 75 80
F(GHz)
130
(a)Conf. I
0
-10
-20 S (CST)
11
S (CST)
S-Parameters (dB)
31
-30 S (HFSS)
11
S (HFSS)
31
-40
-50
-60
40 45 50 55 60 65 70 75 80
F(GHz)
(b)Conf. II
0
-10
-20 S (CST)
11
S (CST)
S-Parameters (dB)
21
-30 S (HFSS)
11
S (HFSS)
21
-40
-50
-60
40 45 50 55 60 65 70 75 80
F(GHz)
(c)Conf. III
Figure 5.16 S-parameters of different configuration of two elements MIMO Yagi-uda antenna.
Table 5.5 Comparison between three configurations of MIMO antennas
Parameters Conf. I Conf. II Conf. III
|S11 | (dB) (at 60 GHz) 32 27 27
|S21 | (dB) (at 60 GHz) 43 40 43
Diversity No Yes Yes
BW (GHz) 51-67 51.5-67 51.5-67
131
5.4.2 Four Elements MIMO

This section introduces four elements MIMO antenna as shown in Figure 5.17. The provided
MIMO consists of two back by back and two side by back antennas. This configuration is introduced
to provide high isolation between all elements. Moreover, the antennas in the proposed configuration
are orthogonal together, giving diversity in the radiation patterns. These properties of the proposed
MIMO indicate that antenna can be used for spatial multiplexing or pattern diversity. Figure 5.18
illustrates the S-parameters of the proposed MIMO antenna to ensure that the antenna covers the band
from 51 GHz to 67 GHz with good matching and high isolation. All the results are verified by CST in
addition to HFSS. From the 3-D radiation pattern of four elements that is introduced in Figure 5.19, we
notice that the radiation pattern of 4 elements is in different directions because of the diversity between
elements.
1
4 2
3
Figure 5.17 Configuration of four elements MIMO antenna.
132
-10 S (CST)
11
S (CST)
21
-20 S (CST)
31
S (CST)
S-Parameters (dB)
41
-30 S (HFSS)
11
S (HFSS)
21
-40 S (HFSS)
31
S (HFSS)
41
-50
-60
40 45 50 55 60 65 70 75 80
F(GHz)
Figure 5.18 S-parameters of proposed four elements MIMO antenna
(a)Port 1 (b)Port 2
(c)Port 3 (d)Port4
Figure 5.19 Radiation pattern of proposed MIMO antenna at different ports
5.4.3 MIMO Parameters

The MIMO have four different parameters that should be tested to ensure that the MIMO gives
a good performance:
• Envelope Correlation Coefficient (ECC)
The ECC is one of the main parameters used to characterize the performance of MIMO antenna.
Where it measures the similarity between the antennas performance especially its radiation patterns.
The ECC can be calculated from the following formula [200]:
133
∗ ∗
|𝑆𝑆𝑛𝑛𝑛𝑛 𝑆𝑆𝑛𝑛𝑛𝑛 + 𝑆𝑆𝑚𝑚𝑚𝑚 𝑆𝑆𝑚𝑚𝑚𝑚 |2
𝜌𝜌𝑛𝑛𝑛𝑛 = (5.1)
�1 − (|𝑆𝑆𝑛𝑛𝑛𝑛 |2 + |𝑆𝑆𝑚𝑚𝑚𝑚 |2 )� �1 − (|𝑆𝑆𝑚𝑚𝑚𝑚 |2 + |𝑆𝑆𝑛𝑛𝑛𝑛 |2 )�
Where 𝜌𝜌: ECC, S:S-parameter, S*: complex conjugate of S-parameters, m, and n are number of
antenna m,n =1,2,3,4.
The value of ECC should be less than 0.5 over the operating band according to the published
standards [196-199]. Whereas the lower values of ECC mean that the two antennas are good isolated.
Figure 5.20 shows the ECC between MIMO elements. It is obvious from the figure that the ECC is
less than 0.0003 within the operating band.
-4
10
3.5
12
(CST)
3
(CST)
13
2.5 (CST)
14
(HFSS)
12
2
(HFSS)
12
ECC
1.5 (HFSS)
12
0.5
0
40 45 50 55 60 65 70 75 80
F(GHz)
Figure 5.20 ECC of four elements proposed MIMO antenna
• Diversity Gain (DG)

The second parameter is a diversity gain, where the diversity gain (DG) can be expressed as
𝐷𝐷𝐷𝐷 = 10√1 − 𝐸𝐸𝐸𝐸𝐸𝐸 2 (5.2)
As shown in Figure 5.21, the DG has 10 dB as a high value during the operating band due to the fact
that ECC is extremely low (𝐸𝐸𝐸𝐸𝐸𝐸 ≅ 0).
• Total Active Reflection Coefficient

The total active reflection coefficient (TARC) is the third parameter that indicates the coupling
between ports. Its minimum value is 0 which means that all incident power is radiated whereas the
maximum value is 1 which means that all the incident power is reflected. The TARC affects greatly
the operating bandwidth of the MIMO antenna system [200]. The TARC is calculated according to the
following equations [200]:
134
(5.3)
�∑N
i=1|br |
2
Γa t =
�∑N
i=1|a i |
2
[b] = [S][a] (5.4)
Where ai, br are the incident signals and reflected signals, respectively. [S], [a] and [b] represent
scattering matrix, excitation vector, and scattered vector of the antenna, respectively.
Figure 5.22 shows the TARC curves upon exciting port one at 1𝑒𝑒 𝑗𝑗0 when others ports have the same
amplitude but with different excitation phases. Some of possible cases are introduced as shown in the
Figure. The values of the TARC is referred to the effective BW of MIMO system. We can observe that
the operating BW of the proposed antenna is not affected by different excitation phase of the other
ports.
• Channel Capacity Loss

The final parameter is the channel capacity loss (CCL). The standard of CLL is CCL<0.4 b/s/Hz
[197]. The capacity of MIMO system grow up with the increase of antenna numbers
𝐶𝐶𝐶𝐶𝐶𝐶 = −𝑙𝑙𝑙𝑙𝑙𝑙2 det(𝜓𝜓𝑅𝑅 ) (5.5)
𝜌𝜌11 𝜌𝜌12 𝜌𝜌13 𝜌𝜌14
𝜓𝜓𝑅𝑅 = �𝜌𝜌 𝜌𝜌 𝜌𝜌 𝜌𝜌 �, 𝜌𝜌𝑚𝑚𝑚𝑚 = 1 − |∑4𝑛𝑛=1 𝑆𝑆𝑚𝑚𝑚𝑚
∗
𝑆𝑆𝑛𝑛𝑛𝑛 |, 𝜌𝜌𝑚𝑚𝑚𝑚 = −�∑4𝑛𝑛=1 𝑆𝑆𝑚𝑚𝑚𝑚
∗
𝑆𝑆𝑛𝑛𝑛𝑛 �, for m,p=1,2.3 or 4
31 32 33 34
Figure 5.23 shows the simulation and the measurement of the CCL for MIMO antenna. One can notice
that the CCL values within the proposed band is less than 0.3 b/s/Hz.
10
9.9999999
9.9999998
DG (CST)
12
DG (dB)
9.9999997 DG 13
(CST)
DG 14
(CST)
9.9999996 DG (HFSS)
12
DG 13
(HFSS)
9.9999995 DG (HFSS)
14
9.9999994
40 45 50 55 60 65 70 75 80
F(GHz)
Figure 5.21 DG of four elements proposed MIMO antenna
135
-10
-15
0 0 0 0
0 , 30 , 60 , 90
0 0 0 0
0 ,0 , 30 , 50
0 0 0 0
-20 0 ,0 , 90 , 90
0 0 0 0
0 , 180 ,0 , 180
TARC(dB)
0 0 0 0
0 , 90 , 180 , 270
0 0 0 0
-25 0 , 30 , 270 , 300
-30
-35
40 45 50 55 60 65 70 75 80
F(GHz)
Figure 5.22 TARC of proposed four elements MIMO antenna

1.6
CCL (CST)
1.4 CCL (HFSS)
1.2
1
CCL (b/s/Hz)
0.8
0.6
0.4
0.2
0
40 45 50 55 60 65 70 75 80
F(GHz)
Figure 5.23 CCL of four elements proposed MIMO antenna
5.5 Conclusion
A Yagi-Uda and Vivaldi antennas are presented to support impedance bandwidth from 51 GHz
to more than 67 GHz. The proposed antennas are designed using 0.18 µm CMOS technology with
substrate size 5×5 mm2 and overall antennas size 0.72×0.85 mm2 and 0.5×0.87 mm2 for Yagi-Uda and
Vivaldi, respectively. These antennas are introduced to support point to point communications.
Different techniques are used to enhance the radiation pattern properties of the Yagi Uda and the
Vivaldi antennas. The antennas have a maximum gain of 0.4 dBi, 0.7 dBi and maximum radiation
efficiency of 38%, 37% for the QYA and TSVA, respectively. The simulated antennas radiation pattern
shows that the antennas have end-fire radiation characteristics and F/B ratio of 16 dB for Yagi-uda and
18 dB for Vivaldi. Furthermore, three configurations of two elements MIMO antenna and four
elements MIMO antenna are introduced in this chapter to provide a high-performance antenna that can
136
be used for short-range communications or indoor networks. The ECC, CCL, DG and TARC are
introduced to investigate that the proposed antenna has good performance over the proposed band.
137
Chapter 6: THz Antennas
6 Chapter Six:
Antenna Designs for
THz Applications
6.1 Introduction
This chapter introduces a disc resonator antenna array with compact size and wide bandwidth for
THz applications. The disc antenna is design based on modified silicon on glass (SOG) technology
platform from high resistivity Si. A Dielectric Waveguide (DWG) is matched with the disc dielectric
antenna using CPW feed. The CPW feed is designed on the Pyrex side of the Si wafer bonded to the
Pyrex. The proposed antenna is designed to operate from 325 GHz to 500 GHz with good return loss.
The end-fire and broadside antennas are introduced with a high gain of about 17 dBi. The antenna has
high efficiency and low cost. Also, the antenna array is introduced with compact size 1
mm× 𝟎𝟎. 𝟕𝟕𝟕𝟕 𝐦𝐦𝐦𝐦 with H=0.11 mm for endfire and H=0.31 mm for the broadside.
6.2 THz
The terahertz waves offer bands from 0.3 THz to 10 THz. The terahertz frequency range offers
new specifications over other spectrum for a large number of applications, such as high-resolution
imagers ultra-high-speed short-distance communication systems, bio-medical, pharmaceutical,
security, sensing, and spectroscopy. This indicates that wireless devices are required to support
different technologies and operate in different frequency bands [3, 125].
In order to increase the antenna gain and directivity, several methods are introduced in [134, 245,
246]; In [134], an array of glass lens antennas arranged on a silicon (Si) substrate is introduced based
on planar metallic rectangular waveguide structure. In [245], the authors present a two tapered
dielectric antenna designed and implemented in the suspended SOG waveguide platform.
6.3 The transition from MWG to DRW

According to the dimensions of the metallic waveguide (MWG) in the x and y-axis, it is excited by
TE10 mode or TE01 mode. In this case, we assume that a>b and the proposed MWG is excited by TE10
mode with the main component of electric field Ey, as shown in Figure 6.1. A part of the dielectric rod
waveguide (DRW) with length Li is inserted into the waveguide without any modification for
impedance matching between the dielectric rod and the metallic waveguide, as shown in Figure 6.1
(b). Part of excited power is transformed into the surface wave power at the end of feed, which is
138
y
TE11 mode and the analysis of this coupling method and converting the mode from MWG to DRW are
introduced in [247-252].
(a)3-D
(b)2-D without tapered section (c)2-D with tapered section

Figure 6.1 Geometry of the DRW fed by MWG
In the MWG, two interfaces I1 and I2, partly reflect the excited TE10 mode, as shown in Figure
6.1 (b and c). The first interface, I1, is the interface inside the MWG between the air-filled and dielectric
rod filled, and the second interface, I2, is the interface at the edge of MWG between the MWG and the
dielectric rod. The minimum reflection coefficient from the conversion between the MWG and DRG
can be calculated as follows:
139
1 Γ −Γ2
Γ𝑚𝑚𝑚𝑚𝑚𝑚 = 1−Γ (6.1)
1 Γ2
Where Γ1 and Γ2 are reflection coefficients at interface I1 and interface I2, respectively.
Z −Z
Γ1 = Z𝑎𝑎+Z𝑑𝑑 (6.2)
𝑎𝑎 𝑑𝑑
y
To calculate Γ2 , we need to calculate wave impedance of TE11
𝑘𝑘𝑘𝑘
Z 𝑇𝑇𝑇𝑇 = 𝛽𝛽
(6.3)
Where Z 𝑇𝑇𝑇𝑇 mode impedance, 𝛽𝛽 propagation constant, 𝜂𝜂 intrinsic impedance, and k wave number.
Z −Z
Γ2 = Z𝑑𝑑 +Z𝑇𝑇𝑇𝑇 (6.4)
𝑑𝑑 𝑇𝑇𝑇𝑇
To minimize the total reflection coefficient at interfaces, the taper section is used to increase the
excitation of DRW mode as shown in Figure 6.1 (c).
6.4 Design of DRW

In this part, the transition between metallic waveguide (MWG) and the dielectric rod waveguide
(DRW) is studied; for this transition, there are two main points that should be considered:
• The TE (transverse electric) field distribution of the high dielectric rod has contrast in the core
relative to the outside of the rod in contrast to the MWG that have equal electric field
distribution.
• The continuity of field from the MWG to the DRW should be considered to be a smoothly
transition between them.
Firstly, the DWG is designed to show the amount of losses. The taper section of the DRW can be
matched between the DWG and the metal rectangular waveguide used for excitation without any
additional structures. The fundamental mode TE10 of the metal waveguide (WR2.2) with standard
dimensions excites the 𝑇𝑇𝑇𝑇11 mode in the high-permittivity DWG, providing good matching and
transmission. The proposed DRW is designed using high-resistivity Si with a relative permittivity of
𝜀𝜀𝑟𝑟 = 11.7 and a conductivity of 𝜎𝜎 = 0.01 S/m. Figure 6.2 shows the design of DRW and excited with
WR-2.2 for band from 325 GHz to 500 GHz and WR-1.5 for the band from 500 to 700 GHz. Figure
6.3 shows the transmission coefficient of the DRW with small losses in the proposed band from 325
GHz to 700 GHz. CST and HFSS verify all the results; results are closed together. The losses at this
frequency band are very small compared with the microstrip line. Figure 6.4 shows the 2-D simulation
using COMSOL and 3-D simulation using CST software in order to show the single-mode operation
140
of DRW; it is easy to note that the field confines in the DRW and there is no leakage outside the DRW
cross-section. Therefore, this dielectric rod is used as a transmission line in our proposed design. The
width (a) and height (b) of DRW is selected to ensure that there is one dominant mode at the proposed
frequency 𝑎𝑎�𝑏𝑏 ≈ 0.5.
(a) DRW with metallic waveguide
(b)DRW
Figure 6.2 Geometry of the proposed DRW
0
-0.2
Transmission Coefficient (dB)
-0.4
WR-2 (CST)
-0.6
WR-1.5 (CST)
WR-2 (HFSS)
WR-1.5 (HFSS)
-0.8
-1
350 400 450 500 550 600 650 700
F(GHz)
Figure 6.3 Transmission coefficient of the DRW
141
(a) 2-D using COMSOL (b) 3-D using CST

Figure 6.4 First mode in DRW
6.5 The Transition from CPW to DRW

In this section, we study the transition from CPW to DRW. The geometry of the CPW transition is
shown in Figure 6.5. The CPW transmission line is the best candidate line to realize the transition from
CMOS and the components to the dielectric waveguide structures. The dielectric waveguide is tapered
at one end to transition from DRW to WR-2.2 rectangular metallic waveguide. At the other end of the
DRW, the DRW to CPW transition is patterned over the DRW. The transition from DRW to CPW is
designed to couple the fundamental mode of the dielectric waveguide to the CPW line. The two ground
planes are extended gradually to the sides of the DRW. The two extended ground planes, forces the
electric field to align horizontally along the CPW line. All the dimensions of this transition are shown
in Figure 6.5. Figure 6.6 shows the electric field distribution of the CPW transition, most of the electric
field is coupled to the CPW line where a small portion of the energy is still traveling through the
dielectric waveguide. The simulated S-parameters is shown in Figure 6.7 from the s-parameters one
can note that the insertion loss of the transition is between 0.2-1.3 dB. The return loss is better than 18
dB over the frequency range. The HFSS and CST simulated results are introduced to verify the structure.
142
Figure 6.5 Transition from CPW to DRW
(a)Electric field contrast
(b)Electric field lines
Figure 6.6 Electric field distribution in transition from CPW to DRW
143
0
S (CST)
21
S (CST)
11
-10
S (HFSS)
21
S (HFSS)
21
-20
S-Parameters (dB)
-30
-40
-50
340 360 380 400 420 440 460 480 500
F(GHz)
Figure 6.7 S-parameters of transition from CPW to DRW
6.6 Design of One Element

The proposed antenna consists of a silicon straight section waveguide segment connected in
series with a disc resonator acting as a radiating element. The CPW is used to couple the power from
the DWG to the disc resonator. The CPW is tapered at the input of the antenna for providing a smooth
transition to the disc antenna, as shown in Figure 6.8.
(a)3-D view (b)Back-side view

Figure 6.8 Antenna Geometry
The proposed disc dimensions are calculated according to the equations that introduced in chapter 3
(from eq. (3.43) to eq. (3.48)). Furthermore, in 1976 Yih [253] introduces an approximate relation to
calculate the endfire tapered dielectric rode antenna.
144
𝜆𝜆0 (6.5)
𝑅𝑅 =
𝜋𝜋
� (𝜖𝜖𝑟𝑟 − 1)
2
(a)First mode of Disc resenator (b)Electric field in the CPW feed

Figure 6.9 Electric field distribution.
Figure 6.9 (a) shows the 2-D simulation using COMSOL software in order to show the single
mode operation of DRA and Figure 6.9 (b) shows the 3-D simulation using CST microwave studio to
show the electric field distribution on CPW and DRA. The antenna operates from 400 GHz to 500 GHz
with height H=0.11mm, and with H=0.31 mm the antenna operates from 325 GHz to 500 GHz with the
same height of DWG (0.11 mm). The antenna is simulated by CST and HFSS simulators and the return
loss of antenna with two different height of disc is shown in Figure 6.10. The most common type of
dielectric antenna is the tapered antenna, which is inherently long and radiates in the end-fire direction
[121, 128, 133, 134, 245, 246, 254, 255]. The radiation pattern of the proposed antenna with H=0.11
mm is end-fire radiation, while with H=0.31 mm is broadside with high gain as shown in Figure 6.11.
The radiation patterns at 490 GHz is selected as an example because this frequency has the best
matching for the two antennas. At the higher frequency the dielectric resonators have hybrid modes
HEmn or EHmn which are described by a combination of two linear modes TE mode and TM mode.
The HE or EH is described by the dominant linear mode TE or TM, respectively. The variation in the
radiation pattern between two heights is due to the variation of propagation hybrid mode in each case
as shown in Figure 6.12. Figure 6.13 shows the gain of the two antennas over the operating band, we
notice that the antennas have average gain more than 11 dBi over the operating bands.
145
-5
-10
-15
-20
| (dB)
-25
11
|S
-30
H=0.11 mm (CST)
-35
H=0.31 mm (CST)
H=0.11 mm (HFSS)
-40 H=0.31 mm (HFSS)
-45
300 350 400 450 500 550 600
F(GHz)
Figure 6.10 Return losses of the one element antenna

`
(a) H = 0.11 mm (b) H=0.31 mm
Figure 6.11 3-D radiation pattern of the antenna at 490 GHz.
(a) EH12 at H=0.11 mm (b)EH25 at H=0.31 mm
Figure 6.12 Hybrid mode distribution for two antennas at 490 GHz
146
Figure 6.13 Gain of the proposed antenna
6.7 Design of Two Elements

The antenna array with two elements is introduced with CPW power divider with compact size as
shown in Figure 6.14. The parameters of this power divider is optimized using built particle swarm
optimization (PSO) tools in CST microwave studio. The proposed antenna consists of a silicon straight
section waveguide segment connected in series with a disc resonator, which acts as a radiating element.
The antenna array operates from 400 GHz to 500 GHz with height H=0.11mm, and with H=0.31 mm
the antenna operates from 325 GHz to 500 GHz with the same height of DWG (0.11 mm) as shown in
Figure 6.15. On the other hand, the CPW with curvature at the end of CPW operates as the Vivaldi
antenna at a higher frequency, so at H=0.11 mm, the radiation pattern is end-fire. Still, with H=0.31
mm, the broadside radiation pattern is achieved, as shown in Figure 6.16. The gain of the antenna array
is increased, as shown in Figure 6.17.
(a)3-D structure (b)Back view

Figure 6.14 Geometry of the antenna array
147
Figure 6.15 Return loss of the antenna
(a)H=0.11 mm (b)H=0.31 mm
Figure 6.16 3-D radiation pattern of antenna array at 490 GHz
(a)Gain
148
(b)Rradiation Efficiency
Figure 6.17 Radiation properties of antenna array.
6.8 Conclusion
A new dielectric antennas for one element disc and two elements are presented to support wide band
for THz wireless communication applications. The proposed antenna consists of DRW section and disc
resonator antenna feeding with CPW. The antenna array introduces more directivity and narrow beam
width. The antenna array has gain of 16 dBi with efficiency 80%. The endfire and broadside radiation
patterns are achieved with different disc height.
149
Chapter7: Conclusion
7Chapter Seven:
Conclusion and Future
work
7.1 Thesis Conclusions
The study in this thesis is provided to design, develop and implement different antennas for the
millimeter and Sub-THz applications. All research work carried out aims at fulfilling the objectives of
this thesis as outlined in section 1.2 and overcoming the limitations of existing designs for the
millimeter and sub-THz applications as well as the challenges faced by each application. This chapter
summarizes the contributions introduced in this thesis and indicates some future directions for further
studies. The work in this thesis is divided to cover the most of millimeter and S-THz applications and
to provide most of the technologies that can be used in these frequency ranges.
Antennas are one of the most important main parts of communications, radars, and imaging systems
because the system's achievement depends on their performance. Therefore, the research work carried
out in this thesis aims to introduce antennas for automotive radars, 5G mobile handsets, short
communications, and multi gigs communications.
After the comprehensive studies that are introduced in chapter 2 for the main applications in the
millimeter and sub-THz ranges, the thesis provides a complete analysis, verification and design for the
concept of VAA. The VAA is introduced to solve the problems of large size, low isolations, low
efficiency, low resolution and small range for the antenna arrays of automotive radar. The performance
of the VAA is compared with the performance of the PAA. The VAA is designed and implemented
with an overall size 30 × 48 × 0.2 mm3. The antenna achieved a high gain of 17 dBi, FSS radiation
pattern shape to support LRR and MRR with beam width ±70 and ±370 , respectively. The
experimental results agree well with the simulated results. Furthermore, the second section of chapter
3 introduced a hybrid antenna at 77 GHz, consisting of a disc patch fed by an aperture coupled and
ring dielectric resonator to give wide bandwidth from 75 GHz to 80 GHz. Furthermore, the AMC is
used to provide a high gain of the antenna and isolate between the elements in case of the antenna
array. Two configurations of antenna arrays (series and parallel configurations) for LRR at 77 GHz
are introduced. These antenna arrays are used to achieve high resolution by introducing small HPBW.
The second contribution in this thesis is introducing the antenna for the future mobile communications
(5G). We introduce a concept of characteristic mode analysis to give complete analysis for the
proposed antenna. The proposed 5G antenna combined between two slot antennas, one slot antenna is
150
fed by micrstrip line from the opposite direction and the other slot antenna is fed in the same layer by
CPW. The proposed antenna for 5G achieves dual polarization to overcome the problems of high
attenuation and propagation losses at 28 GHz. Furthermore, the proposed antenna is combined with
metasurface to enhance the bandwidth and the gain. Finally, a MIMO antenna with four elements is
introduced. The interaction between the antenna and the human body for this antenna is taken in our
consideration. The antenna is fabricated and measured. Good agreement is found between simulated
and measured results.
The on-chip technology especially CMOS is used in this thesis as one of the popular technology
in this band of frequency (especially for unlicensed band 57-64 GHz). Therefore, two different antenna
configurations are introduced using CMOS technology to solve the problems of low efficiency and
low gain associated to this technology. The first antenna is a Quazi-Yagi-uda antenna that composed
of a driven element, a parasitic director and a reflector. The second antenna is a tapper slot Vivaldi
antenna with different technique to enhance its radiation characteristics. The two antennas are designed
using 0.18 µm CMOS technology with substrate size 5×5 mm2 and the antennas size are 0.72×0.85
mm2 and 0.5×0.87 mm2 for Yagi-uda and Vivaldi, respectively. The two antennas achieved a gain of
0.4 dBi, 0.7 dBi and maximum radiation efficiency of 38%, 37% for the QYA and TSVA, respectively.
The simulated antennas radiation pattern shows that the antennas have end-fire radiation characteristics
and F/B ratio of 16 dB for Yagi-uda and 18 dB for Vivaldi. Finally, to solve the inherent propagation
losses and increase the data rate for this application, a four elements MIMO antenna is introduced that
can be used for short-range communications or indoor networks. The ECC, CCL, DG and TARC are
introduced to prove that the proposed antenna has good performance over the proposed band.
The antenna for the S-THz applications is also presented with the technology of dielectric
waveguide that can be suitable for this range of frequencies. The antenna array with two elements is
introduced. The antenna consists of a dielectric waveguide (DWG) and disc resonator. The CPW
power divider is used to convert the feeding method from a metallic waveguide to traditional feed
method that is compatible with PCB designs. The antenna array achieved a gain of 16 dBi with high
efficiency of 80% in addition to the fact that the antenna achieved end-fire and broadside radiation
characteristics depending on the height of the disc resonator.
7.2 Suggestions for Future Works

Although the research studies carried out in this dissertation have effectively solved a number
of significant problems in introducing effective antennas for the aforementioned applications, some
proposed works may be recommended to further enhance for these applications:
151
1. Apply the antennas with the radar system and introducing an antenna for LRR, MRR and
SRR is a challenge and can be considered a future work of this part. Furthermore, the VAA
can be applied for different applications such as radars, satellites, and communication
systems.
2. The work in the 5G applications is still under investigation to select the best mechanisms
for it. Therefore, an intensive effort still needs to exert in this direction. The antenna with
Omni-direction can be introduced. The work for the 5G base stations is needed for more
studies.
3. The antenna arrays using CMOS with metamaterial can be used to enhance the indoor
applications at 60 GHz.
4. The dielectric wave guide technology can be used to introduce compact antennas and
sensors that can be used for imaging and security applications by introducing imaging
algorithms to analyze the reflected signals from them.
152
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AIN SHAMS UNIVERSITY

Antenna Designs for Recent Millimeter and THz

Prof. Dr. Esmat Abdel-Fattah Abdallah

Antenna Designs for Recent Millimeter and THz Applications

Doctor of Philosophy in Electrical Engineering

This dissertation is submitted as a partial fulfillment of the degree of Doctor of Philosophy in

Student Name: Kamel Salah Kamel Sultan

• Assoc. Prof. Mohamed Basha

1. K. S. Sultan, H. H. Abdullah, E. A. Abdallah, and H. El-Hennawy “MOM/GA-based virtual

1. K. S. Sultan, H. H. Abdullah, E. A. Abdallah, M.A. Basha and H. El-Hennawy “A 60-GHz

RESEARCHER DATA ........................................................... ERROR! BOOKMARK NOT DEFINED.

ABSTRACT ................................................................................................................................................ III

ACKNOWLEDGEMNT .......................................................... ERROR! BOOKMARK NOT DEFINED.

TABLE OF CONTENTS ..................................................................................................................... VII

LIST OF TABLES ................................................................................................................................ XIV

LIST OF ABBREVIATIONS .............................................................................................................XV

1 CHAPTER ONE: INTRODUCTION .............................................................................................. 1

1.1 Introduction ........................................................................................................................................................ 1

1.2 Objectives ........................................................................................................................................................... 2

1.3 Features of mmW and S-THz ............................................................................................................................... 2

1.4 Applications of mmW and S-THz.......................................................................................................................... 3

1.5 Original Contribution ........................................................................................................................................... 6

1.6 Software Packages Used ...................................................................................................................................... 7

1.7 Thesis Organization ............................................................................................................................................. 7

2 CHAPTER TWO: ANTENNAS FOR MILLIMETER AND SUB-THZ

2.1 Introduction ............................................................................................................................................ 10

2.2 Automotive Radar Sensors (ARS) .................................................................................................... 10

2.3 Antennas for 5G Mobile Handset...................................................................................................... 24

2.4 Short Range Communications (SRC) .............................................................................................. 29

2.5 Sub-THz Applications........................................................................................................................... 31

2.6 Conclusion ............................................................................................................................................... 35

3 CHAPTER THREE: ANTENNA DESIGN FOR AUTOMOTIVE RADARS................ 36

3.1 Introduction ............................................................................................................................................ 36

3.2 MIMO/Phased Array ............................................................................................................................. 36

3.3 Linear Virtual Antenna Array ........................................................................................................... 38

3.4 Virtual Antenna Array Design .......................................................................................................... 45

3.5 Design Antenna Array for LRR and MRR ..................................................................................... 54

3.6 Hybrid Antenna for 77 GHz Automotive Radar........................................................................... 63

3.7 Conclusion ............................................................................................................................................... 75

4 CHAPTER FOUR: ANTENNA DESIGN FOR 5G .................................................................. 76

4.1 Introduction ............................................................................................................................................ 76

4.2 Characteristics Mode Theory ............................................................................................................ 76

4.3 Dual Polarized Antenna for 5G ......................................................................................................... 79

4.4 Metasurface Design .............................................................................................................................. 93

4.5 Proposed Antenna with MTS ........................................................................................................... 100

4.6 MIMO Antenna Design ....................................................................................................................... 104

4.7 Conclusion ............................................................................................................................................. 115

5 CHAPTER FIVE: ANTENNA DESIGN FOR SHORT RANGE

5.1 Introduction .......................................................................................................................................... 117

5.2 Yagi-Uda Antenna................................................................................................................................ 119

5.3 Tapered Slot Antenna ........................................................................................................................ 123

5.4 Design of MIMO on-chip Antenna .................................................................................................. 129

5.5 Conclusion ............................................................................................................................................. 136

6 CHAPTER SIX: ANTENNA DESIGNS FOR THZ APPLICATIONS ....................... 138

6.1 Introduction .......................................................................................................................................... 138

6.2 THz ........................................................................................................................................................... 138

6.3 Transition from MWG to DRW ........................................................................................................ 138

6.4 Design of DRW ...................................................................................................................................... 140

6.5 Transition from CPW to DRW ......................................................................................................... 142