600 W Half Bridge LLC Eval Board
600 W Half Bridge LLC Eval Board
600 W Half Bridge LLC Eval Board
Application Note
Intended audience
This document is intended for design engineers who wish to evaluate high performance alternative
topologies for medium to high power SMPS converters, and develop an understanding of the design process
and how to apply the somewhat complex LLC design methods to their own system applications.
600 W half bridge LLC eval board with 600 V CoolMOS™ C7 and
digital control by XMC™
Table of contents
1 Introduction ................................................................................................................................... 4
2 HB LLC converter principles of operation ...................................................................................... 7
2.1 Tank configuration and operational modes ...................................................................................... 7
2.2 Analysis of the basic tank characteristics using the FHA method ................................................... 11
2.3 Tank Q values and m inductance ratio: system implications .......................................................... 12
3 A LLC design methodology for specifc application requirements starting from component
technologies ................................................................................................................................ 13
3.1 Design flow ........................................................................................................................................ 13
3.1.1 Input design data ........................................................................................................................ 14
3.1.2 Select operating frequency range for design targets ................................................................ 15
3.1.3 Select LLC primary switch based on system requirements and technology trade-offs ........... 15
3.1.4 1st goal: design the isolation transformer for Vin and Vo for efficiency targets at target fo
operating point ........................................................................................................................... 19
3.1.5 2nd goal: design the transformer Lm for the selected switch technology................................... 25
3.1.6 3rd goal: evaluate the chosen Lm against the SIMPLIS peak gain curve nomagraph and system
gain requirements - select the working Ln value [(Lr+Lm)/Lr] needed for system gain without
capacitive mode. ......................................................................................................................... 27
3.1.7 4th goal: resonant tank design & verification: calculate Lr based on Ln ratio and Lm; calculate
total Cr value, and verify boost up gain target and Fmin set values, Cr AC RMS stress ............... 30
3.2 Synchronous rectification stage design considerations.................................................................. 34
4 Board description ........................................................................................................................ 36
4.1 General overview............................................................................................................................... 36
4.2 Infineon BOM ..................................................................................................................................... 38
4.2.1 Primary HV MOSFETs CoolMOSTM IPP60R180C7......................................................................... 38
4.2.2 XMC4200 microcontroller ........................................................................................................... 38
4.2.3 Half bridge gate drive 2EDL05N06PF.......................................................................................... 39
4.2.4 Advanced dual channel gate drive 2EDN7524F ......................................................................... 40
4.2.5 Bias QR flyback controller ICE2QR2280Z ................................................................................... 40
4.2.6 SR MOSFETs OptiMOSTM BSC010N04LS ...................................................................................... 41
4.3 Board schematics .............................................................................................................................. 42
4.3.1 LLC switching power stage and output synchronous rectification ........................................... 42
4.3.2 Primary side controller board schematics ................................................................................. 44
4.3.3 Biasboard schematic .................................................................................................................. 45
4.3.4 PCB configuration ....................................................................................................................... 46
5 Operation of 600 V CoolMOSTM C7 technology in the 600 W LLC evaluation board with digital
control by XMC™ ........................................................................................................................... 47
5.1 Introduction....................................................................................................................................... 47
5.2 The voltage controlled oscillator ...................................................................................................... 47
5.3 Full ZVS area ...................................................................................................................................... 49
5.4 Burstmode operation ........................................................................................................................ 49
5.5 Adaptive dead time ........................................................................................................................... 52
5.6 Synchronous rectification operation................................................................................................ 54
5.7 Critical LLC operations - hard commutation and capacitive load mode ........................................ 56
5.8 Efficiency plot .................................................................................................................................... 57
600 W half bridge LLC eval board with 600 V CoolMOS™ C7 and
digital control by XMC™
8 References .......................................................................................................... 62
9 List of abbreviations..................................................................................................................... 63
600 W half bridge LLC eval board with 600 V CoolMOS™ C7 and
digital control by XMC™
1 Introduction
The reduction of size in power converters by increasing switching frequency and reducing magnetics
component size is a goal that has been persued for decades. The development of resonant converters with
Zero Voltage Switching (ZVS) has been a cornerstone of this effort. It has at times been rightly said that
resonant converters are a way to make good power from mediocre semiconductors. With the advent of
CoolMOS™ high performance silicon switches based on the superjunction concept, the improvements in
Figure of Merit (FoM) lessened the need for resonant topologies for many years. Now, the industry
requirements for high efficiency in converter performance, which drove a trend towards resonant switching
square wave converters such as the phase shift full bridge converter, are creating the need for a closer look
at the somewhat more difficult to design multiresonant LLC converter. Fundametal to practical density
improvement is efficiency optimization, with the attendant reduction of thermal dissipation.
Classically, fully resonant converters have had a nominal disadvantage in conduction losses compared with
soft switching square wave converters like the Phase Shift Full Bridge(FSFB), due to the difference in peak
versus RMS current for sinusoidal current waveforms versus trapezoidal. However, with the advent of the
multi-resonant converter, and its boost mode of operation, it is possible with modern MOSFETs and their
excellent FoM to achieve highly optimized results with the LLC converter. This is largely due to the fact that
the square wave converter is optimized at maximum duty cycle, which is only achieved at low line condition.
Hence, to provide operational capability with typical PFC front ends, and some converter hold up time
capability, they will typically need to be optimized for DC input as low as 325 V or 300 V, wherein they will
normally operate at 380 V with a less favorable crest factor and higher net RMS current.
In contrast, an LLC converter can be optimized for the nominal DC input voltage, and use the boost mode
below the main resonance to achieve low line regulation, with proper design. As these operational
conditions are transistory, usually only for tens of milliseconds, the efficiency and thermal impact of higher
RMS losses are minimal. Combine this with a favorable silicon BOM situation compared with a Phase Shift
Full Bridge (PSFB) for the mid power range, and the proper design approach, and a high performance
converter is readily in reach.
The main benefits of the LLC are due to its full resonant behavior allowing soft voltage and current
transitions, which intrinsically help to minimize losses in both the power devices and magnetic components.
Figure 1 is a summary of the main differences between the two most popular soft switching topologies in
server SMPS arena, the HB LLC (red bars) and the ZVS PSFB (blue bars). The FoMs are assigned based on
common practical rules well known to SMPS designers.
The selection of the most suitable topologiy is always a trade–off between the performance target and
personal preference/experience: according to Figure 1, the overall average FOM is higher for the HB LLC than
the ZVS PSFB.
600 W half bridge LLC eval board with 600 V CoolMOS™ C7 and
digital control by XMC™
Figure 1 Comparison of several Figure of Merit(FoM) metrics based on cost and performance between
the ZVS PSFB converter (blue) and the LLC HB (red).
This application note will describe the design of an LLC isolated DC-DC stage designed to be part of an 80+
Titanium converter, with an efficiency of 97.5% at 50% load or higher. When configured with a high
performance PFC stage operating at 230 VAC with efficiency of 98.5% or more (no more than 9 W loss), this
combination will meet the 80+ Titanium requirements at half load.
600 W half bridge LLC eval board with 600 V CoolMOS™ C7 and
digital control by XMC™
Figure 2 LLC converter efficiency and target limits for complete Titanium Std. converter (incl. PFC)
In the next section the principles of operation for the LLC converter will be examined, and the main tank
design concepts reviewed. This will be followed by a brief overview of the design methodologies using both
First Harmonic Approximation (FHA), and supplemented by an alternative design path using simulation
based nomagraphs and component analysis. A variety of issues will be explored, including minimization of
losses through optimum transformer design and operating frequency selection. Then, the 600 W evaluation
board circuitry and component BOM will be described in detail.
This document will describe an analog controlled 600 W half bridge (HB) LLC converter fully designed using
Infineon products.
The evaluation board can be ordered on line (ISAR) using the following ordering code:
EVAL_600W_12V_LLC_C7_d
600 W half bridge LLC eval board with 600 V CoolMOS™ C7 and
digital control by XMC™
The LLC is a resonant converter - that means it operates with frequency modulation, instead of Pulse Width
Modulation (PWM) - the traditional approach to power conversion. The LLC is a multi-resonant converter in
that there are two resonant modes that impact the overall voltage gain. One is the series resonant
combination formed by Cr and Lr; the other is a resonant mode at a lower frequency in which the inductive
component is the combination of Lr and Lm, the magnetizing inductance of the isolation transformer. The
overall transfer function gain “G” is defined by:
VO 1 N
G= (
= ´ K Q,Ln,Fx ´ S
Vin 2 NP
) (1)
Where the gain factor is modified by ½ for a half bridge configuration, and 1 for a Full Bridge, and K(Q,Ln,Fx)
is a function defining the tank gain as a function of the Q of the tank and the reflected output load, Ln is the
ratio of Lr to Lm, and Fx is the normalized frequency, being 1 at the series tank resonance.
When operating at the primary resonance (fo) of the series resonant tank components Lr and Cr, the highest
efficiency can be achieved because load current (I_Lr in Figure 4) can be switched under ZCS conditions,
optimizing for lowest switching losses at turn-off for the LLC primary side switches, S1 and S2. Furthermore,
the magnetizing current of the transformer primary, Lm, can be sized so that it provides resonant ZVS
600 W half bridge LLC eval board with 600 V CoolMOS™ C7 and
digital control by XMC™
transitions for the LLC switchs S1 and S2, largely eliminating turn-on losses except for Epassive losses in the
MOSFET epitaxial and edge structure, which may be thought of as an equivalent series resistance connected
with Coss.
Figure 4 Fully resonant operating mode, at the resonant point for Cr and Lr, with near ZCS turn-off of the
primary side MOSFETs.
Over resonant mode results in buck operation, or reduction of the output voltage, to a degree dependent on
the resonant circuit components, the Ln ratio, and the degree of output loading. Turn-off switching is no
longer ZCS, and losses increase in this mode depending upon the switching point on the primary resonant
current.
Under resonant mode results in boost operation until the resonant frequency is reached, based on the tank
components Cr, Lr + Lm, and Reff, the effective loading reflected to the primary side. Boost gain comes in this
mode, but the primary to secondary current transfer is discontinuous (Fig. 6, 7). Additionally, operating at
the lower frequency increases the I-Lm current value, and as this current is not transferred to the output, it
only contributes to increased conduction losses. Lower I-Lm results in a lower increase in conduction loss,
but also lower boost up gain.
600 W half bridge LLC eval board with 600 V CoolMOS™ C7 and
digital control by XMC™
Figure 5 Over resonant operation, above Cr-Lr resonance, for both half cycles, showing tank current
waveforms and non-ZCS turnoff of the primary side MOSFETs
600 W half bridge LLC eval board with 600 V CoolMOS™ C7 and
digital control by XMC™
Figure 6 Under resonant DCM operation, (between resonant point of Cr and Lr, versus resonant point of
Cr and Lr+Lm) half cycle 1
Figure 7 Under resonant DCM mode operation, (between resonant point of Cr and Lr, versus resonant
point of Cr and Lr+Lm) 2nd half cycle
600 W half bridge LLC eval board with 600 V CoolMOS™ C7 and
digital control by XMC™
2.2 Analysis of the basic tank characteristics using the FHA method
The starting point in a resonant converter design is the definition of an energy transfer function, which can
be seen as a voltage gain function. In other words, a mathematical relationship between the input and
output voltage of the converter. Trying to get this function in an “exact” way involves several nonlinear
circuit behaviors governed by complex equations requiring difficult mathematical techniques for closed
form solutions [2, 3]. However, under the assumption that the LLC operates in the vicinity of the series
resonant frequency some important simplifications can be introduced.
In fact, under this assumption, the current circulating in the resonant tank can be considered purely
sinusoidal, ignoring all of the higher order harmonics: this is the so called First Harmonic Approximation
method (FHA), which is the most common approach to the design of an LLC converter. This approach is
quite valid for high Q factors with substantial loading, near the primary resonance, but falls off in accuracy
at lower Q factors and lighter loading, and away from the primary resonance.
Using the FHA method the voltage gain is calculated with reference to the following equivalent resonant
circuit, shown in Figure 8, with an assumed drive based on sine wave excitation; i.e. the first harmonic. This
is a transformation of the circuit of Figure 8, in which the output transformer and rectifier + filter are
replaced with an equivalent load Rac effective, which is the output loading of the converter transformed back
thorugh the converter transformer.
The mathematical expression of the gain K is given in terms of a normalized resonant frequency Fx:
(
K Q,Ln,Fx = ) (
Fx 2 Ln -1 )
(2)
( Ln ´ Fx -1) + Fx ´ ( Fx -1) ´ ( Ln -1) ´ Q
2 2 2
2 2 2 2
where:
Lr
Lr Lm fr
1
Fx
fs
Rac
8 Np ²
Ro ; Cr (3)
Ln ; ; ; Q ;
Lr Lr Cr fr ² Ns ² Rac
Using this method, families of curves can be calculated by modeling the variation in the Q on the primary
side derived from the reflected AC load, or Rac, derived from the output load Ro.
600 W half bridge LLC eval board with 600 V CoolMOS™ C7 and
digital control by XMC™
FHA can be very useful for visualizing trends and understanding the basic operating concepts in a format
that is adaptable to calculation using math programs or spreadsheets. Due to the approximations used, FHA
has some accuracy issues, which are greatest in the Q factor range where power supplies are typically
designed, from 0.5 and below. [1, 2, 3]. Exact form calculations are quite difficult, and so there is a trend
towards using simulation with a tool such as SIMPLIS. The POP (Periodic Operating Point) analysis and
variable stepping make detailed simulation investigation of LLC peak gain curves in an exact sense
reasonably feasible. A detailed explanation on the usage of FHA is presented in [4], and is not duplicated
here. An alternative design process based on SIMPLIS generated peak gain curves in nomagraph [5] will be
described.
600 W half bridge LLC eval board with 600 V CoolMOS™ C7 and
digital control by XMC™
Design Transformer Lm
2nd Goal: Design transformer Lm for Minimize Conduc' on Loss
switch technology: ZVS switching at fo, fmax
Dead ' me target period based on Qoss/Co(tr)
600 W half bridge LLC eval board with 600 V CoolMOS™ C7 and
digital control by XMC™
From the table above, the first important design parameters can be derived:
600 W half bridge LLC eval board with 600 V CoolMOS™ C7 and
digital control by XMC™
While this seems a superficially easy parameter to specify, in practice it has considerable bearing on other
design targets and component characteristics. Published LLC designs have fo operating ranges all the way
from 40 kHz to 1 MHz and more - a separate paper could be written about the challenges and advantages to
different approaches, especially considering whether the target is purely maximizing efficiency, or whether
power density (which still must rely on efficiency even with forced air cooling) is the main target.
Power density target – how much useful improvement is possible through shrinking the size of the
magnetic components? At what point is conduction efficiency compromised?
Cooling technology – heat must be removed, and high density high frequency designs can make this
more difficult
Transformer design, which tends to move in stepped parameter groups, due to granularity of options
such as core sizes, and practical turns winding steps due to steep turns ratio for low voltage outputs
Semiconductor technology - there is a range of performance capability within the families of super-
junction MOSFETs, and new technologies such as SiC and GaN switches will open additional
possibilities in the near future. Here, the output Qoss is a limiting factor with regards to the energy
required for ZVS transitions, followed by switching turn-off losses and turn-on Epassive losses. In
particular, fo operation may not be a problem, but assuring safe and adequately efficient operation
at the resulting fmax for protection, no load, and soft start requires some evaluation.
EMC compatibility – traditional SMPS design usually strives to keep the fundamental frequency below
the 150 kHz lower measurement for conducted EMI, but the low harmonic signature of a well
designed LLC converter gives some latitude for selecting a higher operating frequency
With superjunction MOSFET types suited to LLC applications, a reasonable initial range to consider for the
target power range of 600 W is 100 – 160 kHz. A lower switching frequency range might permit incremental
improvement of the efficiency, but probably only with larger core designs than would be cost effective.
3.1.3 Select LLC primary switch based on system requirements and technology
trade-offs
First, the choice of LLC switch is based on the electrical characteristics that influence switching behavior
under normal conditions operating at fo. Key device parameters include Qoss, which describes the output
charge needed to transition the drain to source voltage passively (when the MOSEFET is not turned on) and
describes the behavior during ZVS switching. Qoss is not usually given in high voltage MOSFET data sheets,
but the parameter for time related output capacitance Co(tr) is given, and is derived from Qoss. The lower the
value of the effective Co(tr), the less current is required for a given drain to source transition time, and the this
allows a higher value of magnetizing inductance for the transformer, which in turn lowers parasitic losses on
the primary side. Also important are Qgd, which describes the charge required for gate to drain switching,
and Rg, which describes the limiting internal gate resistance. Combined, these two parameters give an
indication of turn-off capability and losses, and hence the maximum operating frequency.
600 W half bridge LLC eval board with 600 V CoolMOS™ C7 and
digital control by XMC™
The LLC converter has two operating modes that impose particular stresses on the primary side power
MOSFETs. One is always present, operating at higher switching frequencies up to fmax, as a result of
regulation requirements, initial soft start, and over current protection. The main requirement under these
conditions is assuring that sufficient I-Lm current is available to complete ZVS turn-on transitions in the
allocated dead time td, and that the turn-off behavior is sufficiently fast so that losses are not excessive
when turning off under non-ZCS conditions, and that turn-off time is accomplished quickly within the total
period allocated for dead time.
The other challenging operating condition is capacitive region operation in the boost mode below fo, which
in a properly designed LLC converter should be avoided at all times, yet in a few conditions may be
unavoidable, albeit briefly. In this operating mode, the MOSFET body diode will be conducting current and
then hard commutated when the other MOSFET on the primary side of the LLC converter turns on. Hard
commutation leads to high di/dt and high dV/dt through the primary side loop, which as well as stressing
the body diode and parasitic bipolar transistor of the MOSFET, may lead to hard avalanche operation
simultaneously. Device characteristics affecting this mode include the reverse recovery charge Qrr (the
lower, the better), and the maximum allowable diode commutation speed, which is a measure of the
MOSFET robustness under this operating condition.
Three current Infineon MOSFET technologies may be considered most suitable for the LLC application, but
even here there is a spread of characteristics that should be considered, taking into account the parameters
that may be most important for a particular design implementation.
CFD2 650 V is a MOSFET technology using a platinum doping lifetime killing process derived from CoolMOS™
C6, but with a number of enhancements, including a gate threshold range of 3.5 to 4.5 V optimized for bridge
topology applications, and a much lower internal gate resistance. The lifetime killing process reduces
reverse recovery charge by about 10:1, that dramatically improves Trr and lowers the peak Irrm in hard
commutated applications. OTOH, the Qgd charge is fairly high, which contributes to stable control in hard
commutation in conventional bridge converters, but which drawback for the high operating frequency of an
LLC. Effective output capacitance is moderately high, but with no abrupt corner region, so ZVS dV/dt, while
slower, is well controlled. This may be the component to choose if the target design is known to have
parameters leading to capacitive mode operation, as it will be the most robust choice under those
conditions.
P6 600 V is also derived from C6, but with a focus on higher switching frequency SMPS applications. The
super-junction structure and cell structure were optimized more for high frequency switching applications,
though keeping the same overall pitch geometry, and much of the diode robustness of the C6 series. The
600 W half bridge LLC eval board with 600 V CoolMOS™ C7 and
digital control by XMC™
gate threshold voltage range of 3.5 to 4.5 V is also better suited to bridge applications than standard
MOSFETs (to avoid CdV/dt turn-on when in the off state). It uses substantially lowered Qgd, which is only
about 35% of the CFD2 technology, which speeds drain to source switching time. The diode characteristics
are conventional as regards Qrr, but robust as regards the high safe diode commutation speed that is
permissible. Due to the internal Rg, switching speed is not quite as fast as C7, but the robustness of diode
technology and low Qgd places it well in the midrange for performance.
C7 600 V is derived from the best in class 650 V C7, but with emphasis on further improvements in FoM as
regards gate charge and output capacitance relative to RDS[on], and further lowering both hard switching and
soft switching turn-on losses. Body diode robustness has been substantially improved, raising the maximum
diode commutation speed from 55 A/s to 350 A/s, allowing some capacitive mode capbility. Co(tr) is even
slightly higher than for CFD2, but this is a tricky issue, due to the difference in the shape of the capacitance
versus drain to source voltage. Figure 12 compares the capacitance characteristics of all three technologies
for 190/180 mΩ class parts. Note that the X-axis and Y-axis are not the same for each graph. C7 600 V has the
lowest capacitance above 50 V, but the highest overall under 50 V.
This has a decided impact on the ZVS turn-on behavior and dV/dt, as can be seen in this SIMetrix simulation
comparing the 65R190CFD, the 60R190P6, and the 60R180C7 (Figure 13). The greater turn-off delay of the
65R190CFD and higher zero voltage Coss delays the onset of the ZVS transition, and higher Coss reduces the
dV/dt of the mid region transistion. Note that both simulations and measurements must be evaluated
carefully, because normal production tolerances can result in 20% or more difference for some device
capacitances from lot to lot.
600 W half bridge LLC eval board with 600 V CoolMOS™ C7 and
digital control by XMC™
Figure 13 Comparison of dead time ZVS transition simulation at peak I-Lm of 1.4 A
Also, as expected due to the low Coss above 50 V, the dV/dt in the mid voltage region between 50 and 350 V
with C7 is much higher than the other two technologies. Apart from capacitively coupled common mode
EMC, this is also a potential concern depending on the driver technology used. A sufficiently high CMTR
capability should exist in the driver for peak I-Lm at fo and below, when I-Lm will be highest, along with the
dV/dt. If needed, this may be counteracted using low value capacitors in the range of 47 to 180 pF
connected from drain to source with each C7 MOSFET.
This document will describe the performance of a 600 W HB LLC evaluation board using CoolMOSTM C7 600 V
technology.
600 W half bridge LLC eval board with 600 V CoolMOS™ C7 and
digital control by XMC™
3.1.4 1st goal: design the isolation transformer for Vin and Vo for efficiency targets
at target fo operating point
The target efficiency of this design is fixed by the 80+ Titanium standard; that means fixing certain minimum
requirements for the HV DC-DC stage at 10%, 20%, 50%, 100% load conditions.
The most critical condition for the main transformer is the full load, mainly due to thermal reasons. The
selection of the core size and material is performed according to this condition along with the power density
(thus switching frequency) target and the available airflow.
Keeping due margin room in the design, the minimum efficiency requirement at full load is fixed for the HB
LLC converter to 97%, which means the goal is to keep the total dissipated power in that condition below 18
W.
In order to guarantee a balanced spread of power and heating, a good rule of thumb in the design of the LLC
converter is to keep the total power dissipated in the main transformer below 1/6 of the total dissipated
power, which means the maximum dissipated power shall be 3 W. This is our first important design input.
The max operating temperature is 55°C, as is common in typical server applications. Due to transformer
safety insulation approvals, the max operating temperatiure of the transformer must be lower than 110°C,
so:
(8)
From (7) and (8) the required max thermal resistance of the core shape can easily be derived:
(9)
So, the selected core shape must have thermal resistence lower than 18.3°C/W.
This requrement can be fulfilled with different choices: the preferred method will allow maximizing the ratio
between available winding area and effective volume, of course compatibilty with eq. (18).
Also considering the power density target (in the range of 20 W/in³), the most suitable selection is
PQ 35/35, shown in Figure 4, as the PQ40 offers little benefit with the increase in size.
The related coil former shows a minimum winding area of 1.58 cm² and a thermal resistence of 16.5°C/W, so
lower than (18) and thus able to dissipate up to 3.33 W by keeping the ΔTMAX <55°C.
Once verified that the thermal equations are fulfilled, we can proceed with the design of the primary and
secondary windings and the core material selection, with some important goals:
Fitting the geometry/overall dimensions of the core
Fulfilling the condition (7)
Try to split the losses as equally as possible between core and windings: ideally “fifty-fifty” should be
achieved at full load, but any percentage close to it would be acceptable.
600 W half bridge LLC eval board with 600 V CoolMOS™ C7 and
digital control by XMC™
The selected core material is the ferrite TDK PC95, showing a very interesting plot of core losses (PCV) vs.
flux density vs. frequency (see Figure 14 below):
3C92 is an improved type when compared with 3C90, offering steadily improving core loss at high
temperature up to 90-95°C. For efficiency over a wide temperature range, 3C95 and 3C97 offer the flattest
temperature vs core loss curves, with PC95/3C95 being the best at temperatures below 85°C, typically found
in server and telecom applications.
600 W half bridge LLC eval board with 600 V CoolMOS™ C7 and
digital control by XMC™
possible value. Other factors such as the influence of fringing flux must be considered also, but are
secondary or tertiary effects.
Two design examples will be shown, along with the influence design choices will have for the chosen
operating frequency.
First, we will consider an initially conservative approach with regards to material selection, core loss, and
operating frequency. By this, we will first target an operating frequency in the range of 100 kHz or slightly
more, and the use of well established and relatively low cost core material, such as 3C90 or Magnetics Inc.
type R, for the selected PQ35 core type. This is based on the early prototype developed for this project.
Core physical parameters Ferroxcube PQ3535 (vendors vary slightly in specified parameters):
For this version, the chosen transformer turns ratio n=15, the tank resonant frequency fr = 115 kHz, and the
primary magnetizing inductance target was 180 uH. The voltage in regulation on the primary side is
established from the output voltage, rectifier drop, and turns ratio n:
( ) ( )
VP = n´ VO +V f = 15´ 12+0.2 = 183V
(11)
The actual minimum operating frequency is determined by relationship to fr:
N P _min =
(
n´ VO +V f ) = 23.265
2´ fmin´ Ae ´ DB (13)
Given the desired turns ratio of 15, and the necessity for whole turns on the secondary, this results in a
primary winding of 30T and a secondary of 2T.
Using this configuration the working ∆B can be established, and the approximate core loss estimated for the
fr working point using the Steinmetz coefficients for this core material at 100 – 200 kHz:
1 æ 0.5 ö
DB = ´ ç VO ´ n´ = 0.138T
N p ´ Ae è fr ÷ø
(14)
600 W half bridge LLC eval board with 600 V CoolMOS™ C7 and
digital control by XMC™
c d
æ fr ö æ DB ´10 ö 10-3
Pcore = a ´ ç 3 ÷ ´ ç ´Ve ´ -6 = 0.513W
è 10 ø è 2 ÷ø 10 (15)
The next step is to do a first cut estimation for winding losses, based on splitting the available winding
window (An) between one primary winding and two secondary windings, using a realistic k fill factor for
copper winding allocation, and estimating the conductor resistance based on conductor length and area
derived from the available winding area and MLT (mean length turn) for this core/bobbin type.
Winding area:
k
An_ p = An ´ = 7.6´10-6
2 (18)
k
An_ s = An ´ = 3.8´10-6
2´ N sec
(19)
Estimated DC winding resistance (not considering yet the number of conductor strands or form factor
needed to target the equivalent AC winding resistance):
r ´ lwire.pri
Rdc _ pri = = 0.153W
Awire.pri
(22)
r ´ lwire.sec
Rdc _ sec = = 1.361´10-3 W
Awire.pri
From the basic analysis of the converter, and knowing the sum of the primary magnetizing current and the
primary side load current, the winding conduction losses can be calculated, and the total transformer loss
estimated:
600 W half bridge LLC eval board with 600 V CoolMOS™ C7 and
digital control by XMC™
From this we can see that there is a real problem with the winding loss and the total power dissipation in the
transformer considering the Rth of this core. With sufficient forced-air cooling the design can work, but it will
fall short of the efficiency target.
There is not a smooth granularity of design options – given the turns ratio n and the need for complete
windings on the secondary, the only option (besides a larger core and larger window area to increase the
copper cross section area) is to reduce the number of turns (from 2 to 1) on the secondary, and adjust the
operating frequency to a range which this core geometry can support with reasonable losses. It is likely that
this will require a better core material. We will now review that for the final design, retaining the PQ35 core
form factor.
The proposed alternative design raises the switching frequency to 155 kHz. It also adjusts the turns ratio n =
16, so that the converter operating point is better optimized at the nominal DC input of 380 V. This results in
primary turns of 16, which increases the ∆B core losses at the minimum operating frequency.
( ) (
VP = n´ VO +V f = 15´ 12+0.2 = 195V ) (27)
N P _min =
(
n´ VO +V f ) = 15.4
2´ fmin´ Ae ´ DB (29)
At fr, the calculated core loss for the 3C90 material is almost 2.5 W with this low turns primary. For this
reason, a higher performance material such as PC95 or 3C95 is needed. Then the calculated core loss at fr is
reduced to about 1.3 W. This is still substantially higher than the original design, so lets look at the
estimated winding loss next.
The available winding area is the same, but the number of turns is about half in each case, which both cuts
the winding length in half and allows roughly doubling the working conductor cross section. As a result, the
winding resistance drops substantially:
600 W half bridge LLC eval board with 600 V CoolMOS™ C7 and
digital control by XMC™
r ´ lwire.pri
Rdc _ pri = = 0.044W
Awire.pri
(30)
r ´ lwire.sec
Rdc _ sec = = 0.34 ´10-3 W
Awire.pri
(31)
Though these loss estimates do not include possible issues with AC winding resistance and fringing flux,
known design techniques can keep these effects down to reasonable levels. From this, it appears that it is
possible to meet the target loss goals for the transformer design.
So the primary is realized in a “sandwich” technique using 16 turns of 4 layers of Litz wire with 45 strands of
0.1mm diameter. This minimizes the AC losses due to skin and proximity effects. The secondary uses a
copper band of 20x0.5 mm.
The final structure of the main transformer is shown in Figure 16 below. This has been developed in
cooperation with the partner company ICE Transformers s.r.l., Loreto Aprutino (PE) – Italy.
With the calculated turns ratio, there are only two likely possibilities for the turns structure. For any given
core, if two turns are used on the secondary instead of one, this will roughly quadruple the DC losses on the
secondary. A factor of two results because of double the MLT (mean length of turns) and another factor of
two results because the wire cross-section must be halved in order to fit in the available window area.
A number of popular core types are capable of supporting the possible frequency range and volt seconds
required, such as the PQ3230, ETD39, ETD33, PQ35 and PQ40. The winding window ends up being the
deciding factor for achieving low I2R losses. In all cases there is an optimum gap range whether using a 32:2
winding or 16:1 winding. The choice is based on the minimum between core losses dominating for small air
gap dimensions (lower frequency) and higher proximity losses for large gap lengths.
With the 16:1 winding structure, the PQ35 and PQ40 show the lowest losses by 20-25% overall, due to the
window area and lowest core loss, with an optimum operating frequency range between 150 and 250 kHz.
With a 32:2 winding structure, the PQ40 core will return the best performance by about 10%, at an optimum
frequency in the range of 50-75 kHz, but will be approximately 10% higher in losses than the PQ35 or PQ40 in
the 150-200 kHz range with a optimized 16:1 winding structure.
With this choice, at full load condition the total copper losses will be (primary + secondary, DC+AC
components) 1.1 W, the core losses are 1.8 W, so overall:
Ptrafo = Pcopper + Pcore = 2.9W < Ptrafo_Max (35)
600 W half bridge LLC eval board with 600 V CoolMOS™ C7 and
digital control by XMC™
Figure 16 Winding structure of the PQ 35/35 LLC transformer (ICE Transformers s.r.l.)
An important transformer parameter in an LLC design is the primary or magnetizing inductance Lm. This
value is obtained with a distributed air-gap on the side legs of the PQ core: this construction is preferred
since it minimizes the effect of the (so called) “fringing flux” whch generates additional losses in the
windings close to the inner limb.
3.1.5 2nd goal: design the transformer Lm for the selected switch technology
Given a target for minimum switching frequency for the PQ35 design in the range of 150-160 kHz, the desired
magnetizing inductance Lm must be determined next. This also has a significant interaction with the
transformer design relating to the operating gap and core losses, at the target operating frequency fo. Once
again, all three possible Infineon MOSFET technologies will be examined, in the interest of producing a
result with broad applicability.
The nominal switching period from 160 kHz: fo = 160 kHz; TS = 1/fo = 6.37 s
The target fmax will be defined as 250 kHz: fmax = 250 kHz; TS2 = 1/fmax = 4s
For the fo switching period Ts, it is suggested to use a dead time interval in the range of 1/18 to 1/20 of the
overall period; longer deadtime intervals will start to compromise the fo efficiency by raising the RMS loss
for a given transferred power. Using these criteria, it is suggested to set the dead time td to 350 ns.
600 W half bridge LLC eval board with 600 V CoolMOS™ C7 and
digital control by XMC™
Given that MOSFET capacitance and magnetic components have some variability in production, a guard
band should be employed to assure ZVS switching for components in a production environment.
Experience or rule-of-thumb suggests a total guard band of 30% be applied. Calculating for all three
Infineon MOSFET technologies,
TS2 ´t d
LmfmaxCFD = = 200.32mH (38)
16´ CotrCFD ´1.3
TS2 ´t d
Lmfmax P6 = = 255mH (39)
16´ CotrP6 ´1.3
TS2 ´t d
Lmfmax C7 = = 192mH (40)
16´CotrC 7 ´1.3
Based on these calculations, a suggested value for Lm lies in the range of 190 to 200 mH.
The next step for Lm determination is to check back against the starting transformer design and determine if
the target Lm matches with a core gapping and operating frequency choice that will meet the design
efficiency goals based on calculated losses. Comparing how this specific solution looks in comparison to
nearby design points is a useful way to evaluate for insight.
Table 3 Estimated Lm & loss for 16:1 transformer design “spread” using PQ35 core
Gap Lm fo Calc. PRI/SEC RMS loss * Calc core loss
target
Design 1 ~0.2 mm ~250 H 125 kHz 0.5 W/0.3 W 3.0 W
Design 2 ~0.3 mm ~200 H 160 kHz 0.5W/0.3 W 2.2 W
Design 3 ~0.4 mm ~130 H 230 kHz 0.5W/0.3 W 1.8 W
*Assumes primary winding with 90 strands 0.1 mm wire; secondary of 20 mm x 0.4 mm copper tape
Not discussed are skin and proximity effect and fringing losses, which are generally only feasible to estimate
with FEM tools. With the proposed winding structure, secondary effects should have a low impact the
winding losses; the key factor is choice of gap, primary inductance, and the resulting core losses. With more
advanced semiconductors with lower switching loss and lower Qoss, such as GaN, a case could be made for
raising the operating frequency for this core and construction to 250 kHz nominal fo. Certainly an Fmax of 250
kHz should be no problem.
600 W half bridge LLC eval board with 600 V CoolMOS™ C7 and
digital control by XMC™
3.1.6 3rd goal: evaluate the chosen Lm against the SIMPLIS peak gain curve
nomagraph and system gain requirements - select the working Ln value
[(Lr+Lm)/Lr] needed for system gain without capacitive mode.
The earlier calculation when system parameters were entered, established the system regulation gain
needed to cover the range from low line at 350 V to maximum input at 410 V. Now the input Lr value will be
selected to meet the meet both the system regulation requirements and the system gain, while avoiding
operating near the capacitive mode region.
Typically the Ln value of the Lr to Lm ratio would be evaluated by estimation using FHA, as described earlier;
but the region of preferred operating Q for power supplies is unfortunately the region in which FHA is least
accurate, including in the critical boost up mode, where it usually underestimates the gain. Exact calculation
is quite difficult to do, so instead an interactive nomagraph of sorts can be prepared with pre-calculated
peak gain curves plotted as a function of Ln and full load Qmax. SIMPLIS is used to simulate the LLC converter
transfer function open loop, using the Periodic Operating Point mode and variable stepping to relatively
quickly simulate and measure the range of conditions (Figure 17). The wide range of measurement
functions in SIMPLIS facilitates quick acquisition of a variety of design verification data for the LLC
converter, including power dissipated, AC coupled RMS voltage, gain and phase, etc.
R4
100m Circuit Notes:
IDQ1 1) Designed for 380V to 12V conversion, output power set = 150W
{CrHB}
Power(V4) Power(Q1) 2) All circuit values parameterized in F11 Window, press F11 to edit
CR1
Mean=-717.64884W Mean=9.9598019W 3) Transformer model is ideal, with turns ratio "N", defined in F11 window.
IRF840
V4 Q1 4) Output Load set by parameter "Ro", which is in turn a function of Lr, Cr, Q, and N
{VIN} R1
5) Direct setting of Ro is possible in F11 Window, Currently set to 12.5A
5
6) Output voltage is controlled control via “Fsw” parameter in F11 window, Nominal Fo = 155kHz
C1
VGSQ1 100p Mean=5.8136708W
VSW Mean=594.74568W
N:1:1 Power(D2)
{Lr} Power(R3)
Ir Ip Is1
LR Im {N} TX1 BYW81P-100 R3
Mean=10.000821W IDQ2 D2 {Ro}
Vs Mean=14.608266W
P1 S1
Power(Q2) LP {Co}
{Lp} Co
IRF840
R2 Q2
{CrHB} Cr-RMS
Power(Co)
5 CR2 S2
Mean=43.902899W
VGSQ2 Power(CR) Power(D1)
Mean=23.86802mW
Is2
D1
U1 Maximum=143.12266A @6.8591209uSecs BYW81P-100
Deadtime={TDEAD} Mean=26.7119A
HS_SOURCE =OUT/IN
V2 Mean=53.841312? Mean=208.69565m?
HS_GATE AC 1
RAC Ro
POP_TRIGGER V6 V5 V7
{Fsw/100k} {RAC} {Ro}
LLC_Modulator_Open_Loop
X2
Frequency=91kHertz Frequency
600 W half bridge LLC eval board with 600 V CoolMOS™ C7 and
digital control by XMC™
0.9
0.8
( )
G1x3 Ln 0.7
G1x6( Ln)
0.6
G2x0( Ln)
0.5
G2x5( Ln)
0.3
0.2
0.1
0
1 3 5 7 9 11 13 15 17 19
Ln
Figure 18 Peak gain curves Q vs Ln from SIMPLIS, showing constant gain gain curves as a function of Ln
and Q at 1.3x, 1.6x, 2.0x, 2.5x, and 3.0x
The peak gain nomagraph (Figure 18) illustrates the requirements for Ln ratio and Q to achieve peak system
gains. Note how for lower gain curves high Q is possible for the tank loading conditions (better efficiency),
and as expected, achieving high tank gains with high Ln ratios requires very low tank Q, and high circulating
current and the attendent losses.
To get some truly useful information from this nomagraph, it is necessary to plot the Lm curve for the
application with the Q calculated as a function of Lm and the Ln ratio. In this case, the necessary application
data for the Lm curve calculation is now available:
QTARGET = 0.25 ® 0.3
Lm = 195´10-6
fo = 155´103
RL = 0.24
n = 16
Where Rl is the effective output load resistance for 12 V at 50 A; and QTARGET is the preferred initial target range
for full load tank Q, based on efficiency goals and a typical desired Ln range between 9 and 14.
600 W half bridge LLC eval board with 600 V CoolMOS™ C7 and
digital control by XMC™
Figure 19 Peak gain curves Q vs Ln from SIMPLIS, with calculated curve vs Ln for Lm being evaluated, and
desired FL Qmax range (2.5 to 3.0)
Lower Ln gives a smaller frequency range for span of regulation, but accompanied with higher tank
current and conduction losses
A higher Ln gives a smaller series inductor, but this in turn requires a larger Cr for the same fo; this can
be a problem for capacitor technology and RMS voltage withstanding at high frequencies for Ln > 14.
Also, a larger Cr value contributes to the likelihood of capacitive mode operation at start up and
during burst mode and increases the duration in capacitive mode.
The optimal trade-off of efficiency and control span is typically with Ln in the range of 9-14
While operating with Qmax just reaching the required peak gain at fmin gives the best efficiency in
theory, in practice this makes it quite difficult to consistently avoid capacitive mode operation under
dynamic regulation events. Component tolerances also stack up, and mandate having design margin
to achieve the required gain and avoid capacitive mode at the same time. This leads to another rule-
of-thumb, that is is usually effective to buffer system gain at Qmax/FL by at least 20%.
In summary, target higher peak gain, as a rule-of-thumb use 35-45% higher fmin set point for minimum
frequency operation compared with fmin at nominal FL Qmax, and completely avoid capacitive mode
operation.
In this nomagraph calculation, we can see that Lm = 195 H is on the peak gain for 1.6x, which gives a
reasonable buffer margin for the system gain requirement of approx. 1.1. If overload margin was not
a concern, Ln could be reduced to a lower value, with a higher value of Lr and smaller Cr, but this could
lead to margin issues for operation up to the overload OCP protection point. Additionally, a larger Lr
has a significant cost and density impact.
600 W half bridge LLC eval board with 600 V CoolMOS™ C7 and
digital control by XMC™
3.1.7 4th goal: resonant tank design & verification: calculate Lr based on Ln ratio
and Lm; calculate total Cr value, and verify boost up gain target and Fmin set
values, Cr AC RMS stress
Using Ln = 13, we can use FHA to plot and visualize the actual fmin at several load points (Q plotted at 25%
load steps) and determine a usable fmin set point for the minimum frequency operation for the LLC controller
(Figure 20). Q curves are plotted beyond full load, to 115% (suggested OCP) and 150%.
Figure 20 System gain visualisation using Kmax and Kmin gain values, and determining functional fmin set
point versus calculated fmin at each load condition
The FHA plot makes the fmin set point strategy much clearer; in this case, a 90 kHz limit will be used for the
lowest operating frequency programmed for the controller. Ideally, we would like to see the gain curves all
intersect the Kmax boundary before hitting the fminSetPoint. In actuality, using SIMPLIS to spot check gains,
they do; the FHA gain calculation underestimates the actual value.
Lm 195mH
Using Ln = 13, then Lr = = = 15uH (18)
Ln 13
In the case of the C7 based LLC converter, the chosen value is 15.5 H, but as part of this is realized by the
leakage inductance of the power transformer, the working design value for the resonant inductor is 14 H.
600 W half bridge LLC eval board with 600 V CoolMOS™ C7 and
digital control by XMC™
drwabacks, such as difficult controllability of the Lr value in mass production, and negative impacts on the
power transfer.
In this case, an external Lr is used. As the evaluation board was developed for test / benchmarking and high
power density is not in the main focus, having the resonant inductance available externally allows
experimentation with the resonant tank.
The external resonant choke of 14 H is realized using a RM-12 core and a winding construction as
illustrated in Figure 21 below and implemented by the partner company ICE Transformer s.r.l., Loreto
Aprutino (PE) - Italy.
Now that we have the working value for Lr, calculation of the nominal value for Cr is straight forward.
1
Cr = = 66.7nF (19)
4 ´ p ´ Lr ´ fO2
2
For the LLC converter, the value of 66 nF will be used, but for two reasons it will be split in to a pair of 33 nF
capacitors, using the half bridge configuration shown in Figure 22. This configuration, rather than the single
capacitor (the usual way the LLC converter is drawn), has some practical advantages as regards the dynamic
properties at start up and in burst mode. The connection point for the transformer will initially start out in a
more balanced ‘between the rails’ position, which minimizes the chance and duration of operation in
capacitive mode at start up and when exiting the pause in burst mode.
600 W half bridge LLC eval board with 600 V CoolMOS™ C7 and
digital control by XMC™
IPP60R180C7
IPP60R180C7
Figure 22 HB Cr capacitor configuration to reduce capactive mode operation and distribute AC voltage
stress between two capacitors
Note the configuration for the LLC half bridge should be based on a low inductance working loop for the
primary side power transistors and the connection for the local bypass capacitor C7. C7 also assures a low
impedance connection at high frequencies for both Cr capacitors C9 and C10 with reference to the primary
side ground and the positive bulk voltage.
Another reason for using the split capacitor arrangement is the limited frequency and RMS voltage handling
capability for film capacitors. The larger the value of the capacitor for a given technology, the lower the
frequency cut off point. Higher voltage capacitors extend the AC frequency capability, but with much larger
physical packaging.
Figure 23 B32652 film capacitor recommended AC operating voltage and frequency limits
600 W half bridge LLC eval board with 600 V CoolMOS™ C7 and
digital control by XMC™
Y2 Y1
15 600
Peak to Peak
Voltage
Across Cr
10 Tank Capacitor
500
5
400
300
A
-5
200
-10
100
-15
0
-20
Time/uSecs 2uSecs/div
Figure 24 Peak boost up gain at 90 kHz and 171 VRMS AC voltage stress on Cr
Worst case RMS voltage on Cr occurs under conditions that are normally only transitory; this includes
overload and boost up operation for low Vin during hold up. SIMPLIS can be easily used to investigate these
conditions In Figure 24, we can see that at the OCP threshold range of load at minimum operating frequency
with maximum boost up, the simulation predicts a worst case AC coupled RMS voltage of ~170 V RMS across
Cr. Under normal full load conditions at 155 kHz, the “measured” AC coupled RMS voltage is ~48 V RMS,
which is closely in line with the capabilities of the B32652 630 VDC/250 VAC film capacitor (Figure 22). A slightly
more robust solution for higher frequencies would be the 1 kVDC/250 VAC version of this series.
600 W half bridge LLC eval board with 600 V CoolMOS™ C7 and
digital control by XMC™
600 W half bridge LLC eval board with 600 V CoolMOS™ C7 and
digital control by XMC™
[1] @ 10% Pmax [2] Conduction [3] Gate driving losses [4] Total SR losses (Pcond +
losses Pcond Pgate Pgate)
[5] 1 x BSC010N04LS [6] 31 mW [7] 367 mW [8] 398 mW
per SR branch
[9] 2 x BSC010N04LS [10] 15 mW [11] 734 mW [12] 749 mW
per SR branch
[13] 3 x BSC010N04LS [14] 10 mW [15] 1102 mW [16] 1112 mW
per SR branch
[17] @ 50% Pmax [18] Conduction [19] Gate driving losses [20] Pcond + Pgate
losses Pcond Pgate
[21] 1 x BSC010N04LS [22] 771 mW [23] 367 mW [24] 1138 mW
per SR branch
[25] 2 x BSC010N04LS [26] 386 mW [27] 734 mW [28] 1120 mW
per SR branch
[29] 3 x BSC010N04LS [30] 257 mW [31] 1102 mW [32] 1359 mW
per SR branch
[33] @ Pmax [34] Conduction [35] Gate driving losses [36] Pcond + Pgate
losses Pcond Pgate
[37] 1 x BSC010N04LS [38] 3084 mW [39] 367 mW [40] 3451 mW
per SR branch
[41] 2 x BSC010N04LS [42] 1542 mW [43] 734 mW [44] 2276 mW
per SR branch
[45] 3 x BSC010N04LS [46] 1028 mW [47] 1102 mW [48] 2130 mW
per SR branch
On one hand, at 10% Pmax and still partly at 50% Pmax, using 3 x BSC010N04LS per SR branch can penalize the
efficiency due to excessive gate driving losses. On the other hand, at Pmax, the design suffers from very high
SR conduction losses if using a single BSC010N04LS per SR branch, with as well the risk of making the
MOSFET overheat.
Consequently, if considering the bias losses, using 2 x BSC010N04LS per SR branch would provide the best
balancing of the efficiency at low, mid and full load.
600 W half bridge LLC eval board with 600 V CoolMOS™ C7 and
digital control by XMC™
4 Board description
4.1 General overview
The overall converter electrical configuration is shown in Figure 24. The main control for the LLC converter is
located on the primary side, using either the analog ICE2HS01G or digital XMC4200 control board. The LLC
controller board generates both the primary side LLC control signals for the IPP60R180C7 primary side
switches, and the secondary side synchronous rectifier control signals for the OptiMOS™ BSC010N04LS
synchronous rectifiers, using a high-speed coupler with safety isolation to transmit the SR control. A PID
controller for voltage regulation and a fault controller monitoring current through a low ohmic shunt are
also located on the secondary side, and communicate to the primary side LLC controller board through
conventional opto-couplers. Control circuitry on both sides of the isolation barrier is powered by a bias
module using a CoolSET™ ICE2QR2280Z, which integrates a quasi-resonant flyback controller with a flyback
power transistor and depletion mode startup cell.
LR
IPP60R180C7
IPP60R190P6
BSC010N04LS
CR
ACPL-K73L
Fault
PWM Control Sync Rec
Controller
Primary Side : Secondary Side
PID
LLC Control Controller
ICE2HS01
Or
+12V Secondary Side
XMC4200 Bias Module ICE2QR2280Z
+12V Driver and Control
600 W half bridge LLC eval board with 600 V CoolMOS™ C7 and
digital control by XMC™
Figure 3 is the top view, bottom view and the assembly of 600 W HB LLC evaluation board. Key components
are: (1) heatsink assembly of primary side switches IPP60R190P6 (2) Resonant capacitor
(3) LLC analog controller ICE2HS01G (4) Resonant inductor (5) Main DC-DC transformer (6) PCB assembly of
the auxiliary circuit with bias QR Flyback controller ICE2QR2280Z (7) Heatsink assembly for cooling the
synchronous rectifier (8) Output capacitor (9) Output inductor (10) Half-Bridge MOSFET gate driver
2EDL05N06PFG, (11) Synchronous Rectifier OptiMOS™ BSC010N04LS and (12) Dual Channel Gate Drive
2EDN7524F used for Synchronous Rectifier MOSFETs.
12
10
11
1 2 3 4 5 6 7 8 9
600 W half bridge LLC eval board with 600 V CoolMOS™ C7 and
digital control by XMC™
Suitable for hard and soft switching (PFC and high performance LLC)
Increased MOSFET dv/dt ruggedness to 120 V/ns
Increased efficiency due to best in class FoM RDS(on)*Eoss and RDS(on) *Qg
Best in class RDS(on)/package
Qualified for industrial grade applications according to JEDEC (J-STD20 and JESD22)
600 W half bridge LLC eval board with 600 V CoolMOS™ C7 and
digital control by XMC™
600 W half bridge LLC eval board with 600 V CoolMOS™ C7 and
digital control by XMC™
Off line gate clamping function for safety when not powered
Asymmetric undervoltage lockout thresholds for high side and low side
Insensitivity of the bridge output to negative transient voltages up to -50 V, given by SOI-technology
Ultra fast low capacitance bootstrap diode
Main features
Industry-standard pinout
Two independent low-side gate drivers
5 A peak sink/source output driver at VDD = 12 V
True low-impedance rail-to-rail output (0.7 Ω and 0.5 Ω)
Enhanced operating robustness due to high reverse current capability
-10 VDC negative input capability against GND-Bouncing
Very low propagation delay (19 nS)
Typ. 1 ns channel to channel delay matching
Wide input and output voltage range up to 20 V
Active low output driver even on low power or disabled driver
High flexibility through different logic input configurations
PG-DSO-8, PG-VDSON-8 and TSSOP-8 package
Extended operation from -40°C to 150°C (junction temperature)
Particularly well suited for driving standard MOSFETs, superjunction MOSFETs, IGBTs or GaN power
transistors
600 W half bridge LLC eval board with 600 V CoolMOS™ C7 and
digital control by XMC™
ultra-low power consumption during standby mode operation and low output voltage ripple. The numerous
protection functions give full protection of the power supply system in potential failure situations.
The key features of the ICE2QR2280Z for use as an auxiliary converter of this LLC evaluation board are:
High voltage (650 V/800 V) avalanche rugged CoolMOS™ with startup cell
Quasi-resonant operation
Load dependent digital frequency reduction
Active burst mode for light load operation
Built-in high voltage startup cell
Built-in digital soft-start
Cycle-by-cycle peak current limitation with built-in leading edge blanking time
Foldback point correction with digital sensing and control circuits
VCC undervoltage and overvoltage protection with autorestart mode
Over load /open loop protection with autorestart mode
Built-in over temperature protection with autorestart mode
Adjustable output overvoltage protection with latch mode
Short-winding protection with latch mode
Maximum on time limitation
Maximum switching period limitation
600 W half bridge LLC eval board with 600 V CoolMOS™ C7 and
digital control by XMC™
IPP60R180C7
IPP60R180C7
600 W half bridge LLC eval board with 600 V CoolMOS™ C7 and
digital control by XMC™
600 W half bridge LLC eval board with 600 V CoolMOS™ C7 and
digital control by XMC™
600 W half bridge LLC eval board with 600 V CoolMOS™ C7 and
digital control by XMC™
600 W half bridge LLC eval board with 600 V CoolMOS™ C7 and
digital control by XMC™
600 W half bridge LLC eval board with 600 V CoolMOS™ C7 and
digital control by XMC™
5.1 Introduction
The operation of CoolMOSTM C7 in the 600 W LLC evaluation board controlled by the Infineon ICE2HS01G
analog controller is already described in [8]; moreover, [7] and [8] give an overview about the design and
performance of the converter featured with the above-mentioned analog controller.
This document is focused on the operation of the same 600 W LLC converter, but featuring a digital control
by the Infineon XMC4200 microcontroller located on the converter’s primary side on a dedicated daughter
board pin-to-pin compatible with the analog one.
Because it uses a VCO input pin for regulation (as does the ICE2HS01G), full digital control loop can not be
implemented within the XMC4200; the system design used relies upon a loop control built around the
industry standard TL431.
The organization and functionality of the XMC4200 for LLC is easily understood for those familiar with the
ICE2HS01G. Furthermore, the header block settings for key variables facilitate adjusting the nominal
minimum bridge frequency, the maximum bridge frequency, and the dead time (in multiples of 12.5ns),
replicating the resistor programming functionality of the ICE2HS01G.
In the following paragraphs, some fundamental operating modes of the LLC converter are analysed and the
benefits of the digital control are highlighted with regard to some specific conditions.
600 W half bridge LLC eval board with 600 V CoolMOS™ C7 and
digital control by XMC™
600 W half bridge LLC eval board with 600 V CoolMOS™ C7 and
digital control by XMC™
Ch1: VDS_LS
Ch2: VGS_LS
Ch4: IDS_LS
600 W half bridge LLC eval board with 600 V CoolMOS™ C7 and
digital control by XMC™
Ch1: VDS_LS
Ch3: VOUT
Ch4: ISD_LS
Ch2: VGS_LS
Figure 37 Burst mode opeartion at no-load and very light load condition with ICE2HSO1G
Figures 38 and 39 show the implementation of the burst mode in the version with digital control by the
XMC4200.
From the comparison between Figures 37 and 38 one can appreciate – due to the same time division (1S/div)
- that the burst frequency is significantly higher in the digitally controlled version. This allows the Vout
regulation and ripple requirements to be fulfilled in the digital version even in burst mode operation, as
shown by the purple waveform in both Figure 36 and 37 (VOUT). This appears to be flat without the variation
of 2.62 V that can be seen in Figure 35.
In the digital control, the burst mode is entered or exited according to the monitored value of the feedback
voltage of the error amplifier. The goal is to achieve a limitation of power transfer in order to keep Vout in
regulation and remove possible regulation problems due to the natural flattening of the gain curves at high
switching frequencies.
According to this strategy, the burst mode is entered when VFREQ < 0.38 V (fsw = 245 kHz), which and exited
when VFREQ> 0.45 V (fsw = 230 kHz).
The SR function is disabled during burst mode.
600 W half bridge LLC eval board with 600 V CoolMOS™ C7 and
digital control by XMC™
Figure 38 Burst mode operation at no-load and very light load condition with XMC4200
Figure 39 Burst mode operation at no-load and very light load condition with XMC4200
600 W half bridge LLC eval board with 600 V CoolMOS™ C7 and
digital control by XMC™
a. b.
c. d.
Four typical / possible settings are illustrated: the condition (a) represents the correct setting, where the
MOSFET is switched on exactly after its output capacitance has been completely discharged. In (b) the ZVS is
not achieved for lack of either resonant energy or not enough dead time. In (c) the ZVS is achieved, but the
dead time is too long. (d) is the worst case scenario, where the dead time is so long that the device is
switched on after the current has changed polarity, when the output capacitance has started to be charged
again, resulting in a non-ZVS turn-on.
In our LLC design, the magnetizing current slightly changes as a function of the load. One can expect a
certain increase of magnetizing current when the load increases and/or the input voltage decreases. If a
fixed dead time is used, it must be set according to the lowest value of the magnetizing current, which
happens at very light load, with the goal to achieve a ZVS similar to that shown in Figure 40(a). Since the
magnetizing current is expected to increase with the load, thus reducing the time needed to discharge the
MOSFET output capacitance, setting a constant dead time might lead to the situation shown in Figure 40(c).
The best way to prevent it is to set an adaptive dead time as a function of the load and input voltage.
Figure 41 shows the adaptation implemented in the digital control of our 600 W LLC.
Figure 42 illustrates the practical implementation, showing a difference of 30 nS in the dead time set
respectively at Iout=35 A and Iout=50 A (being Vin=Vin_nom=380 VDC): this will allow an optimal setting, through
minimizing the MOSFET body diode conduction.
600 W half bridge LLC eval board with 600 V CoolMOS™ C7 and
digital control by XMC™
Figure 42 Adaptive dead time measured at Iout=35 A and Iout=50 A respectively (VIN=380 VDC)
600 W half bridge LLC eval board with 600 V CoolMOS™ C7 and
digital control by XMC™
600 W half bridge LLC eval board with 600 V CoolMOS™ C7 and
digital control by XMC™
600 W half bridge LLC eval board with 600 V CoolMOS™ C7 and
digital control by XMC™
5.7 Critical LLC operations - hard commutation and capacitive load mode
In an LLC converter, hard commutation of the body diode may occur during the start-up, burst mode,
overload and short circuit conditions. See also [7], [10] and [11]. Hard commutation happens in an LLC
during the commutation period of the body diode. During this time, the resonant current is flowing through
the body diode of the MOSFET creating a ZVS condition until this MOSFET turns on. When the current is not
able to change direction prior to the turn-on of the other MOSFET, more charge will be stored in the P-N
junction of that MOSFET. When the other MOSFET turns on, a large shoot-through current will flow due to
the reverse-recovery current of the body diode. This results into a high reverese recovery peak current IRRM
and high reverse recovery dV/dT that could sometimes result in a MOSFET breakdown.
Another critical LLC operation is the capacitive load mode, which the converter enters when resonant
current “leads” the voltage in the HB midpoint. In that condition, each of the two HB MOSFETs is turned on
while the current is still forward circulating in the body diode of the opposite MOSFET. This turns into a
stress on both MOSFETs similar to what seen during hard commutation. This condition can be minimized in
the design by the proper selection of resonant components and properly setting the minimum and
maximum operating frequencies.
The digital control offers the possibility to prevent or, at least, minimize the occurrence of these two critical
operations through dedicated algorithms in combination with some additional sensing information from
the HW.
A typical condition that could possibly trigger hard commutation is the start-up: an effective way to prevent
this is to guarantee that the first “complete” switching sequence starts only when the resonant capacitance
is charged at Vin/2, preventing any initial transformer flux imbalance. This can be achieved by not using 50%
duty cycle at start-up, but actually conditioning the MOSFET turn-on time to the zero crossing of the
resonant current (see also [10]).
In the 600 W LLC evaluation board HW, originally designed for primary side analog control by the
ICE2HS01G2, the primary current zero crossing detection information is not available. Therefore, the hard
commutation prevention algorithm at start-up (including the burst mode) consists of stepping the pulse
width on HS and LS devices gradually to reach 50%: 0%, 16% , 33%, 50%.
The Figure 45 below shows the 600 W start-up condition with the analog control (a) and digital control by
XMC (b). A reduction in the drain current peak from 13 A to 3 A is achieved by moving from analog to digital
control, with a consequent reduction of the stress on the MOSFET.
600 W half bridge LLC eval board with 600 V CoolMOS™ C7 and
digital control by XMC™
In the 600 W LLC evaluation board the capacitive load mode is prevented by design, as explained in detail in
chapter 3 of this document. However an algorithm to prevent it is also implemented in the source code.
For the prevention of capacitive load mode, resonant current zero crossing detection would be needed in
order to measure the phase difference with the voltage in the HB midpoint.
A valuable alternative solution used in our design consists of the measurement of the instantaneous value of
the resonant current, which is available in the circuit implemented in Figure 46.
The conversion of the ADC is synchronized with the rising edge of the high side MOSFET PWM signal: a
voltage higher than 0.2 V measured on ADC input during that rising edge is a clear indicator of capacitive
load mode. If this happens, the SW will immediately shut down the converter and resume operation with a
soft start procedure.
Our 600 W LLC is able to fulfill the 80+ Titanium Standard requirements for the HV DC-DC stage.
600 W half bridge LLC eval board with 600 V CoolMOS™ C7 and
digital control by XMC™
5.9 Summary
In the version of the 600 W LLC design with digital control, all of the features of the analog design using the
ICE2HS01G have been implemented.
The final efficiency result is very similar for both versions.
However, additional features have been introduced in order to make the design more flexible and reliable:
these are typical expectations for a digitally controlled SMPS application and they have been completely
fulfilled in our design.
In order to make our evaluation board user friendly, a Graphical User Interface (GUI) would be a natural
evolution of our digital controll. This is planned as next step, by using serial communication between the
MCU and the board via an RS232/TTL or USB/TTL interface.
The GUI will allow the user/designer to quickly set some design parameters and to check in real time some
typical indicators (e.g. input/output current/voltage) and the status of the converter, including possible fault
conditions.
600 W half bridge LLC eval board with 600 V CoolMOS™ C7 and
digital control by XMC™
6 Test/power-up procedure
Test Test procedure Condition
1. Auxiliary circuit turn- Apply 30 VDC on the input. Vin: ~30 VDC
on
Orange LED will light up
2. LLC converter turn-on Apply 350 VDC. Converter will give Vin: 350 VDC
Vout =12 VDC.
Vout:12 V
3. Operational switching Using voltage probe, monitor Vin:380 Vdc
frequency switching frequency at following
Vout:12 V
test conditions:
@5 A Output load 10% load - ~ Iout: 5 A
155 kHz*
@25 A Output load 50% load - ~ Iout: 25 A
142 kHz*
@50 A Output load 100% load - ~ Iout: 50 A
132 kHz*
(*measure freq. at “Pri_LS_VGS“-
connector)
[* +-10 kHz]
4. Fan enable Switch the load from 50 A to 5 A. Vin =380 Vdc
Increase the output load current
Iout= 5 A
from 11-14 A, fan should turn on.
ÞFan is off
Vin =380 Vdc
Iout= 11-14 A
ÞFan is on
5. Switch off input start- Switch off the input Vin= 0 Vdc
up at no load
Iout= 0 A
Switch at 380 Vdc on no load Vin =380 Vdc
output. Operation should be in
Iout = 0 A
burst mode.
Vout = 11,5 – 12,5
6. Switch off input; Start- Switch off the input Vin= 0 Vdc
up at full load
Iout= 0 A
Apply 380 Vdc with full load @50 A Vin =380 Vdc
output. Vout is in between 11.8 –
Vout: 11,8 – 12,3 Vdc
12.2 Vdc*
(*measure on the board- Iout = 50 A
600 W half bridge LLC eval board with 600 V CoolMOS™ C7 and
digital control by XMC™
connector)
7. Running no load -> Switch off load from 380 Vdc 50 A Vin =380 Vdc
output short circuit to 380 Vdc 0 A.
Short circuit the load using the (after short circuit) Vout = 0
short circuit function of the e- Vdc
load. Converter should latch.
Iout = 0 A
8. Switch off input & Switch off the input. Vin= 0 Vdc
remove short circuit
9. Running full load -> Remove short circuit function on Iout= 0 A
over current protection the load.
Apply 380 Vdc 50 A with full load Vin =380 Vdc
output. Increase the current on
Iout = 50 A
the output 1 A each step until the
converter goes into protection
starting from 50 A. OCP occurs OCP = between 55 A – 62 A
between 55 A and 62 A.
10. Running full load -> Apply 380 Vdc 50 A with full load Iout= 0 A
output short circuit output. Short circuit the load
Vin =380 Vdc
using the short circuit functions
of the load. Converter should Iout = 50 A
latch.
(after short circuit) Vout=0
Vdc
11. Switch off input; Switch off the input. Vin= 0 Vdc
start-up -> output short
Iout= 0 A
circuit
Apply 380 Vdc with output load Vin =380 Vdc
short circuit. Converter should be
Iout = short circuit
in hiccup/latch mode.
Vout = 0 V short circuit
(hiccup/latch)
12. Switch off input & Switch off the Input. Vin= 0 Vdc
remove short circuit
Remove short circuit function on Iout= 0 A
the load.
13. Dynamic loading Apply 380Vdc. Set the electronic Vin =380 Vdc
load to dynamic loading mode
with the following settings:
CCDH1: Iout 5 A Iout = 5 A…50 A
CCDH2: Iout 50 A Vout = 11,5 – 12,5 Vdc
Dwell time: 10 mS
Load slew rate: 1 A/µS
600 W half bridge LLC eval board with 600 V CoolMOS™ C7 and
digital control by XMC™
In the following links, you can find more detailed information about the devices used from Infineon and the
magnetic components.
Microcontroller XMC4200
http://www.infineon.com/dgdl/Infineon-XMC4100_XMC4200-DS-v01_02-
en.pdf?fileId=db3a30433afc7e3e013b3cf9b2816573
600 W half bridge LLC eval board with 600 V CoolMOS™ C7 and
digital control by XMC™
8 References
[1} “PFC Demoboard – System Solution: High Power Density 800W 130kHz Platiinum Server Design”,
Infineon Technologies May 2015.
[2] J. F. Lazar, R. Martinelli, “Steady-State Analysis of the LLC Series Resonant Converter”, IEEE APEC 2001,
Volume 2, pp 728-735
[3] R. Nielsen, “LLC and LCC resonance converters: Properties, Analysis, Control”, www.runonielsen.dk
[4] C. Oeder, A. Bucher, J. Stahl, T. Duerbaum, “A comparison of Different Design Methods for the
Multiresonant LLC Converter with Capacitive Output Filter”, (COMPEL), 2010 IEEE 12th Workshop on Control
and Modeling for Power Electronics.
[5] S. Abdel-Rahman, “Resonant LLC Converter – Design and Modeling”, Infineon Technologies AN2012-09,
September 2012.
[6] Y. Liu, “High Efficiency Optimization of LLC Resonant Converter for Wide Load Range,” MSEE Thesis
Virginia Polytechnic, December 2007.
[7] F. Di Domenico, A. Steiner, J. Catly, “Design of a 600W HB LLC Converter using 600V CoolMOS™ P6”,
Infineon Technologe AN- August 2015
[8] A. Steiner, F. Di Domenico, J. Catly, F. Stückler, “600W half bridge LLC Evaluation Board with 600V
CoolMOS™ C7”, Infineon Technology AN – June 2015
[9] T. Fujihira: “Theory of Semiconductor Superjunction Devices”, Jpn. J. Appl. Phys., Vol.36, pp. 6254-6262,
1997
[10] Lawrence Lin, Gary Chang: “Consideration of Primary side MOSFET Selection for LLC topology”,
Infineon Technologies AN, 2014
[11] F. Stückler, S. Abdel-Rahman, K. Siu: “ 600V CoolMOS™ C7 Design Guide”, Infineon Technologies AN
May 2015
[12] Anders Lind: “LLC Converter Design Note”, Infineon Technologies AN 2013-03
[13] Liu Jianwei, Li Dong: “Design Guide for LLC Converter with ICE2HS01G”, Infineon Technologies AN V1.0,
July 2011
[14] Dr. Arshad Mansoor, Brian Fortenbery, Peter May-Ostendop and others: “Generalized Test Protocol for
Calculating the Energy Efficiency of Internal Ac-Dc and Dc-Dc Power Supplies”, Revision 6.7, March 2014
600 W half bridge LLC eval board with 600 V CoolMOS™ C7 and
digital control by XMC™
9 List of abbreviations
BOM……………………………………………………………………………………………………..Bill Of Materials
BM……………………………………………………………………………………………………………Burst Mode
CGD ...................................................................................................internal gate drain capacitance CGD=Crss
Ciss.......................................................................................................................input capacitance Ciss=CGS+CGD
Co(er) .................................................................................................effective output capacitance, energy related
Co(tr) ................................................................................................effective output capacitance, time related
Cr .......................................................................................................................................resonant capacitance
di/dt .........................................................................................steepness of current slope at turn off / turn on
DUT ...........................................................................................................................................device under test
dv/dt .........................................................................................steepness of voltage slope at turn off / turn on
Eoff................................................................................................................................energy losses at switch off
Eon..........................................................................................................................energy losses loss at switch on
Eoss........................................................................stored energy in output capacitance (Coss) at typ. VDS=400V
FHA..........................................................................................................First Harmonic Approximation Method
FOM.................................................................................................................................................Figures of Merit
fr ............................................................................................................................................resonant frequency
GUI………………………………………………………………………………………………Graphic User Interface
ID........................................................................................................................................................ drain current
IRMS..................................................................................................................effective root mean square current
Imag .........................................................................................................................................magnatizing current
Im,pk.................................................................................................................................peak magnetizing current
K............................................................................................................................................................. gain factor
Lr ........................................................................................................................................... resonant inductance
Lm..................................................................................................................................... magnetizing inductance
m .......................................................................................................................................... inductance factor
Np ............................................................................................................................................. primary winding
NS.........................................................................................................................................secondary winding
n .................................................................................................................................... transformer turn ratio
MOSFET...........................................................................metal oxide semiconductor field effect transistor
Pcond_SR .........................................................................................synchronous rectification conduction losses
PFC..............................................................................................................................power factor correction
PNP.............................................................................................................. bipolar transistor type (pnp vs. npn)
QOSS .........................................................................................................................Charge stored in the COSS
Q ...................................................................................................................................................quality factor
Rac.....................................................................................................................................total equivalent resistor
RDS(on)........................................................................................................... ……drain-source on-state resistance
Rg,tot .......................................................................................................................................total gate resistor
Ro ....................................................................................................................................................output resistor
Rth ...............................................................................................................................................thermal resistance
SMPS………………………………………………………………………………………Switch Mode Power Supply
tdead .....................................................................................................................................................dead time
600 W half bridge LLC eval board with 600 V CoolMOS™ C7 and
digital control by XMC™
Revision history
Major changes since the last revision
Page or Reference Description of change
-- First Release
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