WO2013049224A2

WO2013049224A2 - Power transformer active flux balance circuit with driver dead time control

Info

Publication number: WO2013049224A2
Application number: PCT/US2012/057359
Authority: WO
Inventors: Milan Dragojevic
Original assignee: Murata Power Solutions
Priority date: 2011-09-26
Filing date: 2012-09-26
Publication date: 2013-04-04
Also published as: WO2013049224A3

Abstract

A converter including a DC voltage source; a transformer; a bridge circuit including first and fourth transistors in a first leg and second and third transistors in a second leg and controlled such that, when current flows in the first leg, current does not flow in the second leg and such that, when current flows in the second leg, current does not flow in the first leg; and a flux balance circuit arranged to measure the currents in the first and second legs and to control a dead time of the first, second, third, and fourth transistors based on the measurement of the currents in the first and second legs so that the currents flowing in the first and second legs are equal or substantially equal.

Description

POWER TRANSFORMER ACTIVE FLUX BALANCE CIRCUIT WITH DRIVER DEAD

TIME CONTROL

BACKGROUND OF THE INVENTION

1. Field of the Invention

[0001] The present invention relates to power conversion. More specifically, the present invention relates to isolated DC/DC converters.

2. Description of the Related Art

[0002] Conventional full-bridge and push-pull converters suffer from a power

transformer's tendency toward flux unbalance and core saturation. Conventional full-bridge and push-pull topologies require peak-current mode control to achieve transformer flux balance and to avoid core saturation. However, true peak-current mode control is challenging to implement with digital control, so the voltage mode control is used as a control method with conventional digital controllers. Conventional voltage mode control requires DC blocking capacitors to avoid transformer saturation. The transformer saturation can still be a problem during transients. The DC blocking capacitors conduct full power and can be unreliable.

[0003] The most common conventional control method is peak-current mode control, applicable for both push-pull and full-bridge topologies.

[0004] Conventional control methods add a DC blocking capacitor in series with the transformer primary winding, which suffers from various disadvantages. DC blocking capacitors can only be used in bridge-type topologies and cannot be used in push-pull topologies. The DC blocking capacitor conducts full power and can be unreliable. The DC blocking capacitor can also create unequal stress on the secondary rectifiers.

[0005] The typical conventional full-bridge converter with voltage mode control is shown in Fig. 1. The converter includes four primary switches Ql, Q2, Q3, Q4, two half-bridge drivers DRV1, DRV2, a transformer Tl, two output rectifier diodes Dl, D2, an output inductor LI, an output capacitor CI, and a DC blocking capacitor CIO. Each leg of the full-bridge circuit is driven by a half-bridge driver DRV1, DRV2. The full-bridge circuit is controlled by two out-of-phase PWM signals, PWM_A, PWM_B. The PWM controller is located on the primary side.

Transformer flux balance is achieved by the capacitor CIO being connected in series with the primary winding. This passive flux balance technique is applicable with converters that use voltage mode control.

[0006] An active flux balance technique is used in the full-bridge converter with peak- current mode control shown in Fig. 2. The peak current during the conduction of primary switches Ql, Q2 is controlled and is identical to the peak current during the conduction of primary switches Q3, Q4. Half-bridge driver DRVl controls primary switches Ql, Q4, and half- bridge driver DRV2 controls primary switches Q2, Q3. Each of the half-bridge drivers DRVl, DRV2, includes the following input/output pins:

1) IN_H pin receives the PWM signal PWM_A or PWM_B that will be used to drive the high-side switch (i.e., primary switch Ql or Q3);

2) IN L pin receives the PWM signal PWM_B or PWM_A that will be used to drive the low-side switch (i.e., primary switch Q4 or Q2);

3) DRV_H pin outputs a drive signal that drives the high-side switch (i.e., primary

switch Ql or Q3);

4) DRV_L pin outputs a drive signal that drives the low-side switch (i.e., primary switch Q4 or Q2);

5) DRV_H_RTN pin provides the return path for the high-side switch (i.e., primary

switch Ql or Q3);

6) DRV_L_RTN pin provides the return path for the low-side switch (i.e., primary switch Q4 or Q2);

7) DT_H pin provides dead time control for the high-side switch (i.e., primary switch Ql or Q3); and

8) DT_L pin provides dead time control for the low-side switch (i.e., primary switch Q4 or Q2).

[0007] Resistors R3, R4 are connected to the DT_H pin to control the dead time of the high-side switch (i.e., primary switch Ql, Q3), and resistors R2, R5 are connected to the DT_L pin to control the dead time of the low-side switch (i.e., primary switch Q4, Q2).

[0008] The measured primary current has two components: the reflected load current and the magnetizing current. Because the peak current and the reflected load current are identical, the magnetizing current is identical for each cycle. The flux in the transformer is directly proportional to the magnetizing current. Therefore, the flux in the transformer is symmetrical. The current measurement and the pulse termination are instantaneous. The current is continuously monitored and as soon as it reaches a predetermined programmed level, the gate signal to terminate the active switches is generated. This is an active flux balance technique because the duration of the switch conduction time is controlled to achieve identical primary peak current. It is a forward type because the predetermined peak current is programmed into the controller, and as soon as the current reaches the predetermined programmed peak value, the signal to terminate the active switches is generated. The controller can also provide short- circuit protection for the primary switches Ql, Q2, Q3, Q4. The peak-current mode control is not applicable for use with digital control because of the need for continuous current measurement. In digital (discrete time) control, the measurements are not continuous as in analog control. The output voltage, currents, temperature, etc. are sampled at discrete times. It is possible to make a controller that uses continuous current measurement. In that case, the controller is not a pure digital controller, but is a mixed-mode controller.

[0009] The conventional volt-second balance loop technique is used, for example, in Analog Device's digital controller ADP1043 shown in Fig. 3. The average current for each leg of the full-bridge circuit of this controller is measured. The volt-second balance loop technique reduces (or increases) the conduction of the pulse width modulation (PWM) signal to make the average current for each leg of the full-bridge circuit identical. This technique requires that the current measurement signal cross the safety isolation barrier because the controller for this technique is located on the secondary side, which results in an arrangement that is bulky and expensive because of crossing the safety isolation barrier.

[0010] A conventional feedback-type active flux balance technique is shown in Fig. 3. This technique is used in Analog Device's digital controller ADP1043A shown in Fig. 3. The ADP1043A has a dedicated circuit to maintain volt-second balance in the main transformer when a converter is operating in a full-bridge topology. This means that a DC blocking capacitor is not necessary. [0011] The dedicated circuit monitors the DC current flowing in both halves of the full bridge and stores this information. The dedicated circuit also compensates and/or adjusts the PWM drive signals to ensure that equal current flows in both halves of the full bridge. The current measurement is through the CS1 pin. Several switching cycles are required before the circuit operates effectively because of feedback control and sampling and because of the need to average the primary current.

SUMMARY OF THE INVENTION

[0012] To overcome the problems described above, preferred embodiments of the present invention provide a flux balance circuit that eliminates the need for DC blocking capacitors and that is located on the primary side, and that does not need any control or measurement signal to cross the safety isolation barrier.

[0013] According to a preferred embodiment of the present invention, a converter includes a DC voltage source; a transformer; a bridge circuit including first and fourth transistors in a first leg and second and third transistors in a second leg and controlled such that, when current flows in the first leg, current does not flow in the second leg and such that, when current flows in the second leg, current does not flow in the first leg; and a flux balance circuit arranged to measure the currents in the first and second legs and to control a dead time of the first, second, third, and fourth transistors based on the measurement of the currents in the first and second legs so that the currents flowing in the first and second legs are equal or substantially equal.

[0014] The dead times of the first, second, third, and fourth transistors are preferably controlled based on a comparison of the currents in the first and second legs. The converter further preferably includes error amplifiers that perform the comparison of the currents in the first and second legs. The converter further preferably includes a PWM controller arranged to provide a pulse width modulation signal to the first, second, third, and fourth transistors. The PWM controller is preferably located on a secondary side of the transformer.

[0015] The flux balance circuit preferably includes a first current-sensing resistor connected to the first leg and arranged to measure the current flowing in the first leg and a second current-sensing resistor connected to the second leg and arranged to measure the current flowing in the second leg. The flux balance circuit preferably includes a current-sensing resistor connected to the first and second legs and arranged to measure the currents flowing in the first and second legs. The converter further preferably includes an unfolding circuit connected to the first and second legs and controlled such that the current-sensing resistor measures at least one of the current flowing in the first leg and the current flowing in the second leg.

[0016] The flux balance circuit preferably includes a current-sensing transformer. The converter further preferably includes an unfolding circuit connected to the first and second legs and controlled such that the current-sensing transformer measures at least one of the current flowing in the first leg and the current flowing in the second leg. The current-sensing transformer is preferably connected in series with the DC voltage source. The current-sensing transformer is preferably connected in series with a primary winding of the transformer. The converter further preferably includes a current-sensing resistor connected to the current- sensing transformer. The converter further preferably includes first and second current-sensing resistors connected to the current-sensing transformer.

[0017] Preferably, no DC blocking capacitors are included. The transformer preferably defines an isolation barrier, and preferably, no control or measurement signal crosses the isolation barrier.

[0018] The converter further preferably includes a rectification circuit connected to the transformer. The rectification circuit preferably includes diodes. The rectification circuit preferably includes MOSFETs.

[0019] The converter further preferably includes a first half-bridge driver connected to the first and fourth transistors and a second half-bridge driver connected to the second and third transistors.

[0020] The above and other features, elements, characteristics and advantages of the present invention will become more apparent from the following detailed description of preferred embodiments of the present invention with reference to the attached drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

[0021] Fig. 1 shows a conventional full-bridge converter with voltage mode control. [0022] Fig. 2 shows a conventional full-bridge converter with peak-current mode control.

[0023] Fig. 3 shows a conventional digital controller that uses a feedback-type active flux balance technique.

[0024] Fig. 4 shows a converter with a control circuit using two current-sensing resistors according to a first preferred embodiment of the present invention.

[0025] Fig. 5 shows a converter with a control circuit using a single current-sensing resistor according to a second preferred embodiment of the present invention.

[0026] Fig. 6 shows a converter with a control circuit using a current-sensing transformer according to a third preferred embodiment of the present invention.

[0027] Fig. 7 shows a converter with a control circuit using a current-sensing transformer according to a fourth preferred embodiment of the present invention.

[0028] Fig. 8 shows a converter with a control circuit using a current-sensing transformer according to a fifth preferred embodiment of the present invention.

DETAILED DESCRIPTION OF PREFERRED EMBODIMENTS

[0029] The converter of the first preferred embodiment of the present invention is shown in Fig. 4. The converter inverts a DC voltage supplied by DC source V6 to an AC voltage using transformer Tl and primary switches Ql, Q2, Q3, Q4 and then converts the AC voltage back to a DC voltage, typically with a different voltage level than the voltage level of the DC source V6, using diodes Dl, D2. The converted DC voltage is provided as an output voltage after being filtered by the filter defined by output inductor LI and output capacitor CI. The converter shown in Fig. 4 preferably includes diode rectifiers Dl, D2. However, for low voltage applications, the diode rectifiers Dl, D2 can be replaced with synchronous rectifiers, which are usually MOSFETs, for example.

[0030] The active flux balance circuit includes two current-sensing resistors R7, Rll in the legs of the full-bridge circuit. The resistors R7, Rll measure the current during the conduction period of the primary switches Ql, Q2 and the prima ry switches Q3, Q4, respectively. The current measurement signals are averaged and filtered by low-pass RC filters, which include the combination of resistor R18 and capacitor C2 and the com bination of resistor R19 and capacitor C3, respectively. The averaged and filtered current measurement signals are connected to the input pins of the error amplifiers XI, X2 and the compensation network including capacitors C4, C5. The output pin of the error amplifiers XI is connected to transistor Q5 through resistor R21. The transistor Q5 is connected to the pin DT_L of half-bridge driver DRV1 and is used to change (decrease) the resistance value of pin DT_L, which changes the dead time of the primary switch Q4. Similarly, the output pin of the error amplifiers X2 is connected to transistor Q6 through resistor Rl. The transistor Q6 is connected to the pin DT_L of half-bridge driver DRV2 and is used to change (decrease) the resistance value of pin DT_L, which changes the dead time of the primary switch Q2.

[0031] The error amplifiers XI, X2 control the dead time of the half-bridge drivers DRV1, DRV2 to ensure that equal or substantially equal currents flow in both legs of the full-bridge circuit. The component tolerance and operational amplifier offset will create errors in measurement of the current. These current measurement errors create flux unbalances and DC current in the transformer Tl. The current measurement error is preferably kept as small as possible. For example, the difference in the currents is preferably less than about 5% of the minimum magnetizing current. The gap in transformer Tl, the transformer Tl, and the resistance of the primary switches Ql, Q2, Q3, Q4 will keep the transformer Tl from reaching saturation.

[0032] Conventional control methods attempt to ensure equal currents by controlling the pulse width of the drive signal applied to the primary switches using a PWM controller. This drive signal is passed through the drivers connected to the primary switches. In conventional control methods, the drivers can create unbalanced drive signals that must be balanced by changing the pulse width of drive signal from the PWM controller. That is, conventional control methods require modification of the PWM controller, which is very often impossible because the PWM controller's internal architecture cannot be changed. The preferred embodiments of the present invention can use standard PWM controllers with fixed symmetrical pulses. The incoming drive signals from the PWM controller are preferably equal but the duration of the pulses that drive the primary switches Ql, Q2, Q3, Q4 are modified by changing the driver DRV1, DRV2 dead time. The dead time is controlled by delaying the turn-on time of the primary switches Ql, Q2, Q3, Q4. For example, if the current measured at resistor R7 is higher than the current measured at resistor Rll, then the error amplifiers XI, X2 increase the dead time for the diagonal primary switches Ql, Q2 and decrease the dead time for the diagonal primary switches Q3, Q4. This increases the current at resistor Rll and makes both currents equal.

[0033] The control circuit controls the dead time to achieve flux balance while the PWM signals are symmetrical and controlled by the PWM controller. Because most standard PWM controllers use symmetrical PWM signals, this allows preferred embodiments of the present invention to use standard PWM controllers without having to modify the PWM controllers.

[0034] The PWM controller shown in Fig. 4 is located on the secondary side. The PWM controller can be a digital controller with a Power Management Bus or PM Bus communication interface. Typically, the communication interface is located on the secondary side. The communication interface is typically used to adjust and read operating parameters, such as output voltage, current limit, etc.

The PWM controller can also be located on the primary side, in which case, an isolated PMBUS interface is preferably used. The other portions of the control circuit will remain the same regardless of the location of the PWM controller. Because the PWM controller is located on the secondary side, a small signal isolator circuit is used to transfer the PWM signals from the secondary side to the primary side. The control circuit works with both analog and digital control and with fixed duty cycle control.

[0035] Changing the dead time by delaying the on-time (i.e., leading-edge pulse control) of the primary switches Ql, Q2, Q3, Q4 is an effective way to balance the flux in conventional hard-switching full-bridge topologies because drivers with turn-on delay dead time control are widely available. If the control circuit of a preferred embodiment of the present invention is used to balance the flux of the soft-switched zero-voltage-transition (ZVT) phase-shifted bridge, then drivers with turn-off time control (i.e., trailing-edge pulse control) are used. Accordingly, the drivers are provided with dead-time control of both the leading and trailing edges. Leading- edge dead time can be used to ensure ZVT conditions, and trailing-edge dead time control can be used for transformer flux balance control.

[0036] The current-sensing circuit can be implemented with a single current-sensing resistor or a current-sensing transformer, for example. A converter with a control circuit with one current-sensing resistor R7 according to a second preferred embodiment of the present invention is shown in Fig. 5. The principle of operation of the converter shown in Fig. 5 is similar to that of the converter shown in Fig. 4. The differences are in the current-sensing circuit. The primary current is sensed preferably with one resistor R7, for example.

[0037] Because only one current-sensing resistor R7 is preferably used, an additional unfolding circuit is provided to determine which leg of the full-bridge circuit is conducting. An example of such a circuit including switches Q7, Q8, and resistors R6, R8 is shown in Fig. 5. To reconstruct the same current measurement waveforms as in the converter in Fig. 4, two switches Q7, Q8 and two resistors R6, R8 are added. When the PWM_A signal is high, the switches Ql, Q2 conduct and switch Q8 shunts the current measurement signal going to the low-pass filter defined by the resistor R18 and the capacitor C2. When the PWM_B signal is high, the opposite full-bridge leg conducts (i.e., switches Q3, Q4). The switch Q7 is on and shunts the current signal going to the low-pass filter defined by the resistor R19 and the capacitor C3. The resistors R6, R8 are added to prevent filter capacitors C2, C3 from discharging when the switches Q7, Q8 are on. One implementation of the unfolding circuits defined by switches Q7, Q8, and resistors R6, R8 is shown in Fig. 5. Instead of being connected to shunt the signals, the two switches Q7, Q8 can be connected in series with the resistor R7.

[0038] A converter with a control circuit including a current-sensing transformer TXl according to a third preferred embodiment of the present invention is shown in Fig. 6. The principle of operation for the converter shown in Fig. 6 is similar to the converter shown in Fig. 5. The current-sensing transformer TXl is connected in series with DC source V6 such that current-sensing transformer TXl measures the current flowing into the full-bridge circuit defined by primary switches Ql, Q2, Q3, Q4.

[0039] The current-sensing resistor R7 is preferably replaced by a current-sensing transformer diode D3 and current-sensing resistor R7. The current-sensing transformer TXl can be arranged in series with primary winding as shown in Fig. 7. The primary current is measured at resistor R7. The unfolding circuit defined by switches Q7, Q8 and resistors R6, R8 is used to determine positive and negative current amplitude. The control circuit in Fig. 6 measures the current flowing into the full-bridge circuit defined by primary switches Ql, Q2, Q3, Q4. This current is unidirectional. The disadvantage of the control circuit in Fig. 6 is it is limited to the maximum duty cycle because of the required time to reset the current sense transformer TXl. The current-sense transformer TXl in Fig. 7 is located in series with the primary winding of the transformer Tl, measuring the current in the primary winding of the transformer Tl. The diodes D3, D4, D5, D6 define a rectifier bridge, which is required because the current in the primary winding of the transformer Tl is bi-directional. The control circuit in Fig. 7 does not have a current reset problem. The voltage at resistor R7 in Fig. 7 represents the power transformer current. The additional switches Q7, Q8 are added to unfold current signal CS to two current measurements (positive and negative cycles).

[0040] Fig. 8 shows another preferred embodiment of the present invention using current- sensing transformer TXl connected in series with the primary winding of the transformer Tl. In this preferred embodiment the primary current is sensed at resistors R7, R8. By using two current-sensing resistors R7, R8, it is not necessary to use the unfolding circuit defined by switches Q7, Q8 and resistors R6, R8. As with the preferred embodiment shown in Fig. 4, using two resistors R7, R8 eliminates the need to use expensive and space-consuming switches Q7, Q8, which are typically MOSFETs. It is also possible to use two current-sensing transformers instead of only using current-sensing transformer TXl; however, using two current-sensing transformers is more expensive and requires more space.

[0041] Preferred embodiments of the present invention can be used for a push-pull converter, for example. In push-pull converters, there are no high-side primary switches Ql,

Q3. The current will be measured in the same way as in the embodiments discussed above and dead time control will be applied on the low-side primary switches Q2 and Q4.

[0042] The diode rectifiers Dl, D2 in Figs. 4-6 can be replaced with synchronous rectifiers such as metal-oxide-semiconductor field-effect transistors (MOSFETs).

[0043] The preferred embodiments of the present invention work with any controller, regardless of the control method used, including, for example, voltage mode, peak current mode, average current mode, or unregulated bus converter. The preferred embodiments of the present invention can be applied with either digital or analog control. The preferred embodiments of the present invention do not need to use a DC blocking capacitor inserted in series with the primary winding.

[0044] The flux balance circuit of the preferred embodiments of the present invention is preferably located on the primary side. The PWM controller can be on the primary side or the secondary side. The PWM controller can control both the primary switches and the secondary- side rectifiers, if necessary, and can also communicate with a master controller or other power supplies. The PWM controller can communicate with a master controller that is a power management unit that can adjust output voltages, protection, etc.

[0045] The preferable location for the PWM controller is on the secondary side. The preferred embodiments of the present invention do not need any control or measurement signal to communicate with the PWM controller, which eliminates the need for crossing the safety isolation barrier.

[0046] It should be understood that the foregoing description is only illustrative of the present invention. Various alternatives and modifications can be devised by those skilled in the art without departing from the present invention. Accordingly, the present invention is intended to embrace all such alternatives, modifications, and variances that fall within the scope of the appended claims.

Claims

WHAT IS CLAIMED IS:

1. A converter comprising:

a DC voltage source;

a transformer;

a bridge circuit including first and fourth transistors in a first leg and second and third transistors in a second leg and controlled such that, when current flows in the first leg, current does not flow in the second leg and such that, when current flows in the second leg, current does not flow in the first leg; and

a flux balance circuit arranged to measure the currents in the first and second legs and to control a dead time of the first, second, third, and fourth transistors based on the

measurement of the currents in the first and second legs so that the currents flowing in the first and second legs are equal or substantially equal.

2. The converter of claim 1, wherein the dead times of the first, second, third, and fourth transistors are controlled based on a comparison of the currents in the first and second legs.

3. The converter of claim 2, further comprising error amplifiers that perform the comparison of the currents in the first and second legs.

4. The converter of claim 1, further comprising a PWM controller arranged to provide a pulse width modulation signal to the first, second, third, and fourth transistors.

5. The converter of claim 4, wherein the PWM controller is located on a secondary side of the transformer.

6. The converter of claim 1, wherein the flux balance circuit includes:

a first current-sensing resistor connected to the first leg and arranged to measure the current flowing in the first leg; and a second current-sensing resistor connected to the second leg and arranged to measure the current flowing in the second leg.

7. The converter of claim 1, wherein the flux balance circuit includes a current-sensing resistor connected to the first and second legs and arranged to measure the currents flowing in the first and second legs.

8. The converter of claim 7, further comprising an unfolding circuit connected to the first and second legs and controlled such that the current-sensing resistor measures at least one of the current flowing in the first leg and the current flowing in the second leg.

9. The converter of claim 1, wherein the flux balance circuit includes a current-sensing transformer.

10. The converter of claim 9, further comprising an unfolding circuit connected to the first and second legs and controlled such that the current-sensing transformer measures at least one of the current flowing in the first leg and the current flowing in the second leg.

11. The converter of claim 10, wherein the current-sensing transformer is connected in series with the DC voltage source.

12. The converter of claim 10, wherein the current-sensing transformer is connected in series with a primary winding of the transformer.

13. The converter of claim 9, further comprising a current-sensing resistor connected to the current-sensing transformer.

14. The converter of claim 9, further comprising first and second current-sensing resistors connected to the current-sensing transformer.

15. The converter of claim 1, wherein no DC blocking capacitors are included.

16. The converter of claim 1, wherein:

the transformer defines an isolation barrier; and

no control or measurement signal crosses the isolation barrier.

17. The converter of claim 1, further comprising a rectification circuit connected to the nsformer.

18. The converter of claim 17, wherein the rectification circuit includes diodes.

19. The converter of claim 17, wherein the rectification circuit includes MOSFETs.

20. The converter of claim 1, further comprising:

a first half-bridge driver connected to the first and fourth transistors; and

a second half-bridge driver connected to the second and third transistors.