TWI441490B - Digital video broadcast service discovery - Google Patents

Description

Digital Video Broadcasting Service Exploration Technology Field of invention

Embodiments are generally related to communication networks. More particularly, embodiments relate to digital video broadcast service discovery techniques.

Background of the invention

The digital broadband broadcast network enables end users to receive digital content including video, audio, data, and the like. With a mobile terminal, a user can receive digital content through a wireless digital broadcast network. Digital content can be transmitted in a nest in a network. In a communication network, a honeycomb cell can represent a geographic area that can be covered by a transmitter. A network can have multiple honeycomb cells, and the honeycomb cells can be adjacent to other honeycomb cells.

A receiving device, such as a mobile terminal, can receive a program or service in a data or transport stream. The transport stream carries individual elements of the program or service, such as audio, video and data components of a program or service. Typically, the receiving device locates different components of a particular program or service in a data stream via program specific information (PSI) or service information (SI) embedded in the data stream. However, PSI or SI signaling may not be sufficient in some wireless communication systems such as Handheld Digital Video Broadcasting (DVB-H) systems. The use of PSI or SI signaling in this system may result in a sub-optimal end-user experience, as PSI and SI tables carried in PSI and SI information may have long repetition periods. In addition, PSI or SI signaling requires a large amount of bandwidth, which is costly and also reduces the efficiency of the system.

Summary of invention

In order to provide a basic understanding of some aspects of the invention, a brief summary is presented below. The present invention has not been extensively summarized in this summary. It is not intended to identify key or critical elements of the invention, and is not intended to limit the scope of the invention. The following summary merely presents some concepts of the invention in a simplified form as a

Embodiments are directed to modulating binary phase shift keying of a first pilot symbol according to a reference sequence, and modulating differential binary phase shift keying of a second pilot symbol. The original reference sequence is then combined with the delayed differential modulation sequence prior to performing an inverse fast Fourier transform and inserting a guard interval. The receiver operation is a reversal of the transmitter operation just discussed. The receiver does not have to know the reference sequence. Embodiments are directed to specifying a plurality of seed sources, each of which has a bit pattern of r bits, not all of which have a zero value; by applying to each source A recursive formula expands the sources into their respective sequences; and uses one of the sequences as a comb sequence and uses the sequences other than the comb sequence as a binary phase shift keying pattern.

Simple illustration

A more complete understanding of the present invention and its advantages can be obtained by the description of the accompanying drawings in which <RTIgt;

Figure 1 illustrates a suitable digital broadband broadcast system in which one or more illustrative embodiments of the present invention may be implemented.

Figure 2 illustrates an example of a mobile device in accordance with one aspect of the present invention.

Figure 3 schematically illustrates an example of a honeycomb lattice in accordance with one aspect of the present invention, each of which may be covered by a different emitter.

Figure 4 shows a frame (a symbol for channel search and service discovery) and a frame and a frame of information according to one aspect of the present invention.

Figure 5 shows how a signal center frequency can coincide with a channel center frequency or with respect to a channel center frequency.

Figure 6 is a flow chart showing the steps performed by a receiver in accordance with at least one embodiment.

Figure 7 shows an example of the magnitude of a pilot signal bandwidth relative to a signal bandwidth and a channel raster bandwidth in accordance with one aspect of the present invention.

Figure 8 illustrates a sparse lead spacing of a pilot sequence of a pilot symbol in accordance with one aspect of the present invention.

Figure 9 is a flow chart showing the steps performed by a receiver to perform correlation in the frequency domain to detect the coarse offset being used.

Figure 10 is a flow diagram showing the steps associated with performing a service discovery in the time domain, in accordance with an embodiment.

Figure 11 shows an example of a sequence of pilot/telephone symbols in accordance with one aspect of the present invention.

Figure 12 is a flow diagram showing the steps of a method performed by a transmitter in accordance with at least one level of the present invention.

Figure 13 illustrates a packet structure in accordance with one aspect of the present invention.

Figure 14 illustrates a transmit window offset in accordance with one aspect of the present invention.

Figure 15 illustrates a current sub-letter of a frame in accordance with one aspect of the present invention. An offset between the number and the next sub-signal.

Figure 16 illustrates an additional packet structure that can be used to carry signaling information in accordance with one aspect of the present invention.

Figure 17 illustrates an exemplary flow chart for use in service exploration in accordance with one aspect of the present invention.

Figures 18 and 19 depict the relationship between P1, P2 and DATA symbols in accordance with one aspect of the present invention.

Figure 20 shows an exemplary frame and a time slot structure comprising OFDM symbols and a honeycomb cell in accordance with one aspect of the present invention.

Figure 21 illustrates the coherence bandwidth and differential modulation within a pilot symbol.

Figure 22 depicts the differential modulation between two P1 symbols in accordance with one aspect of the present invention.

Figure 23 shows two 1K symbols with a 1/1 guard interval and differential modulation between the symbols, according to an embodiment.

Figure 24 shows the calculation of the sum of energy received from one or more pilot symbols in accordance with one or more embodiments.

Figure 25 shows a transmitter in accordance with one or more embodiments.

Figure 26 shows a receiver in accordance with one or more embodiments.

Figure 27 is a flow diagram showing the steps that can be performed by a receiver in accordance with one or more embodiments.

Figure 28 is a diagram of the auto/cross correlation between the pilot sequences and their frequency offset versions in accordance with one or more embodiments.

Figure 29 is an enlarged version of the figure of Figure 28 showing the low cross-correlation range of the frequency offset.

Figure 30 is a diagram showing the envelope amplitude of a first pilot symbol signal in accordance with at least one embodiment.

Figure 31 is an enlarged version of the figure of Figure 30.

Detailed description of the preferred embodiment

In the following description of the various embodiments, reference is made to the accompanying drawings, in which FIG. It will be appreciated that other embodiments may be utilized and structural and functional modifications may be made without departing from the scope and spirit of the invention.

Embodiments are directed to service discovery and channel discovery in digital broadcast networks. From a user's perspective, relatively fast service discovery techniques are desirable. Of course, when the receiving device is used for the first time, a blind service discovery/channel search is performed. Moreover, when a terminal is turned off and moved to a different location, a new blind search is performed again. Typically, an action TV application sometimes performs a background channel search in order to detect possible new services. This blind service exploration should take only a few seconds to enable frequent rescans without bothering the end user.

Challenges associated with the discovery of conventional digital video broadcast services include the following. The DVB-H standard provides a lot of flexibility regarding signal bandwidth, FFT size, guard spacing, internal modulation, and more. The operator can use an offset for the DVB-H signal, ie the signal is not offset by a certain amount at the nominal center frequency of a channel. Different countries use different channel rasters and signal bandwidths. TPS (transmitter parameter signaling) is included in the standard to aid the receiver Sync and channel search. Unfortunately, the receiver needs to know several parameters before it can decode the TPS information. The signal bandwidth, frequency offset, FFT size, and guard interval need to be known before the TPS can be decoded. Most channels in the UHF band do not contain DVB-H services. Non-DVB-H channels use a trial error method to detect (using all modes to try to reach the lock), and this takes a lot of time. In fact, the time to detect non-DVB-H services primarily sets the speed at which channel searches can be achieved, since typically most of these channels are empty or contain non-DVB-H services.

An example of blind service exploration is calculated as follows: number of channels in UHF 35, (channel 21-55, 470-750 MHz); number of frequency offsets 7 (-3/6, -2/6, -1/) 6,0,+1/6,+2/6,+3/6 MHz); number of signal bandwidths 3 (6 MHz, 7 MHz, 8 MHz. 5 MHz is an individual instance for USA receiver only); FFT size The number of 3 (2K, 4K, 8K); the number of guard intervals 4 (1/32, 1/16, 1/8 and 1/4); and the average time to decode TPS for a mode is 120 ms (2K) 50 ms, 4K 100 ms, 8K 200 ms). These figures are exemplary.

In this example, the resulting time for blind service exploration would be: 35*7*3*3*4*120 ms=1058.4 seconds=17.64 minutes.

According to an embodiment, various methods can be used to reduce the time it takes to perform channel search/service exploration. The basic concept of these various methods introduces a portion of a signal that has known characteristics and maintains the same portion (e.g., initialization/synchronization symbol(s)) as the different digital video broadcast modes of operation. Thus, the known portion of the signal can be decoded without having to rely on the trial error method. The known portion of the signal contains the reference to the remainder of the signal The number; therefore, after the known portion is decoded, the remainder of the signal can be decoded without attempting an erroneous method. This known portion of the signal contains a subset of the available subcarriers and their modulation. The combination of pre-defined subcarriers (number of subcarriers) and their modulation is chosen such that the combination is unique for each offset-FFT size pair (or only unique for the different FFT sizes), And this combination can be used to identify the signal as a desired signal for one of the digital video broadcasts. Moreover, with the known portion of the signal, the channels containing the digital video broadcast service can be efficiently detected. If the fixed known portion is not found from the checked signal, the signal will be treated as a non-digital video broadcast signal or an empty channel, and the receiver can quickly proceed to the next channel/frequency. In this way, detecting non-digital video broadcasts and empty channels becomes relatively fast.

FIG. 1 illustrates a suitable digital broadband broadcast system 102 in which one or more illustrative embodiments may be implemented. A system such as the system described herein may utilize a digital broadband broadcast technology such as Handheld Digital Video Broadcasting (DVB-H) or a next generation DVB-H network. Examples of other digital broadcasting standards available to digital broadband broadcasting system 102 include terrestrial digital video broadcasting (DVB-T), terrestrial integrated services digital broadcasting (ISDB-T), advanced television system committee (ATSC) data broadcasting standard, terrestrial digital multimedia Broadcast (DMB-T), terrestrial digital multimedia broadcasting (T-DMB), satellite digital multimedia broadcasting (S-DMB), forward link only (FLO), digital audio broadcasting (DAB), and digital radio amplitude modulation alliance (DRM) . Other digital broadcast standards and technologies now known or later developed may also be used. The aspects of the present invention are also applicable to other such as, for example, T-DAB, T/S-DMB, ISDB-T, and ATSC. Multi-carrier digital broadcast systems, proprietary systems such as Qualcomm MediaFLO/FLO, and non-legacy systems such as 3GPP MBMS (Multimedia Broadcast/Multicast Service) and 3GPP2 BCMCS (Broadcast/Multicast Service).

Digital content may be created and/or provided by digital content source 104 and may include video signals, audio signals, data, and the like. The digital content source 104 can provide content to the digital broadcast transmitter 103 in the form of a digital packet, such as an Internet Protocol (IP) packet. A group of related IP packets that share a unique IP address or other source identifier is sometimes described as an IP stream. The digital broadcast transmitter 103 can receive, process, and forward transmissions of a plurality of digital content streams from a plurality of digital content sources 104. In various embodiments, the digital data streams can be IP streams. The processed digital content can then be passed to a digital broadcast tower 105 (or other physical transport component) for wireless transmission. Finally, the mobile terminal or device 112 can selectively receive and consume digital content originating from the digital content source 104.

As shown in FIG. 2, the mobile device 112 can include a processor 128, a memory 134, and/or other storage connected to the user interface 130, and can be used to display video content and services to a mobile device user. A display 136 that guides information and the like. The mobile device 112 can also include a battery 150, a speaker 152, and an antenna 154. The user interface 130 can further include a keyboard, a touch screen, a voice interface, one or more arrow keys, a rocker, a data glove, a mouse, a roller ball, and the like.

Computer executable instructions and materials used by processor 128 and other components within mobile device 112 may be stored in a computer readable memory 134. The memory can optionally include read-only electrical and non-electrical memory Any combination of memory modules or random access memory modules is implemented. The software 140 can be stored in the memory 134 and/or storage to provide instructions to the processor 128 to enable the mobile device 112 to perform various functions. Alternatively, computer executable instructions for some or all of the mobile devices 112 may be implemented in hardware or firmware (not shown).

Mobile device 112 can be configured to receive, decode, and process digital wideband broadcast transmissions, for example, based on the Digital Video Broadcasting (DVB) standard, such as DVB-H or DVB-T, through a particular DVB receiver 141. The mobile device can also be provided with other types of receivers for digital broadband broadcast transmissions. In addition, receiver device 112 can also be configured to receive, decode, and process transmissions through FM/AM radio receiver 142, WLAN transceiver 143, and telecommunications transceiver 144. In one aspect of the invention, mobile device 112 can receive radio data stream (RDS) messages.

In an example of the DVB standard, a DVB 10 Mbit/s transmission may have 200 50 kbit/s audio program channels or 50 200 kbit/s video (TV) program channels. The mobile device 112 can be configured to receive, decode, and process based on the Handheld Digital Video Broadcasting (DVB-H) standard or such as DVB-MHP, Satellite DVB (DVB-S), or terrestrial DVB (DVB-T) The transmission of other DVB standards. Similarly, other digital transmission formats can be selectively used to deliver such as ATSC (Advanced Television Systems Committee), NTSC (National Television System Committee), ISDB-T (Ground Integrated Services Digital Broadcasting), DAB (Digital Audio Broadcasting), Availability of content and information on additional services such as DMB (Digital Multimedia Broadcasting), FLO (forward link only) or DIRECTV. Furthermore, the digital transmission can be time divided, such as in DVB-H technology. time Segmentation reduces the average power consumption of a mobile terminal and enables smooth and seamless handover. In contrast to the bit rate required to transmit data using a conventional streaming mechanism, time division must use a higher instantaneous bit rate to transmit data in bursts. In this example, the mobile device 112 can have one or more buffer memories in preparation for storing the decoded time division transmission prior to presentation.

In addition, an electronic service guide (ESG) can be used to provide program or service related information. In general, an electronic service guide (ESG) enables a terminal to communicate what services are available to the end user and how the services can be accessed. The ESG includes multiple ESG split blocks that exist independently. Traditionally, ESG partitioning blocks include XML and/or binary files, but more recently, they have included a large number of items such as, for example, an SDP (Dialog Description Protocol) description, a text file, or an image. The ESG partitioning blocks describe one or more layers of currently available (or future) services or broadcast programs. These levels may include, for example, free text descriptions, schedules, geographic availability, prices, purchase methods, types, and other ancillary information such as preview images or clips. According to many different protocols, audio, video, and other types of data including the ESG partition can be transmitted over various types of networks. For example, using protocols of the Internet Protocol Suite such as the Internet Protocol (IP) and the User Data Element Agreement (UDP), data can be transmitted through a collection of networks commonly referred to as "Internet." . Data is often transmitted over the Internet and addressed to a single user. However, it can be addressed to a group of users, often referred to as multicast. In the case where the material is addressed to all users, this is called broadcasting.

One way to broadcast material is to utilize an IP data broadcast (IPDC) network. IPDC is a combination of digital broadcast and internet protocols. Through such an IP-based broadcast network, one or more service providers can supply different types of IP services including online newspapers, radios, and televisions. These IP services are organized into one or more media streams in the form of audio, video, and/or other types of data. To determine when and where these streams occur, the user refers to an Electronic Service Guide (ESG). One type of DVB is Handheld Digital Video Broadcasting (DVB-H). The DVB-H is designed to deliver 10 Mbps of data to a battery powered terminal device.

The DVB transport stream transmits compressed audio and video and data to a user via a third party delivery network. By means of a technique such as the Motion Picture Experts Group (MPEG), encoded video, audio, and data in a single program are multiplexed into a transport stream (TS) along with other programs. The TS is a packetized data stream having a fixed length packet including a header. Individual elements of a program (intelligence and video) are carried in a packet with a unique packet identification (PID). In order for a receiving device to locate different elements of a particular program within the TS, program specific information (PSI) embedded in the TS is provided. In addition, additional service information (SI) (a set of tables attached to the MPEG private section syntax) is incorporated into the TS. This enables a receiver device to properly process the data contained within the TS.

As explained above, the ESG partitioning blocks can be transmitted by the IPDC to the destination device through a network such as, for example, DVB-H. The DVB-H may include, for example, separate audio, video, and data streams. Then, the destination device must once again determine the order of the ESG partitions and combine them into useful ones. News.

In a typical communication system, a honeycomb frame can define a geographic area that can be covered by a transmitter. The honeycomb cell can be of any size and can have adjacent honeycomb cells. Figure 3 schematically illustrates an example of a honeycomb grid in which each of the honeycomb cells can be covered by a different emitter. In this example, the cellular grid 1 represents a geographic area covered by a transmitter for a communication network. The honeycomb grid 2 is adjacent to the honeycomb grid 1 and represents a second geographic area that can be covered by a different emitter. The hive cell 2 can be, for example, a different hive cell within the same network as the hive cell 1. Alternatively, the honeycomb grid 2 can be in a network different from the honeycomb grid 1. In this example, the honeycomb cells 1, 3, 4, and 5 are adjacent honeycomb cells of the honeycomb cell 2.

In accordance with one or more embodiments, data for channel search and service discovery is signaled using symbols at least at the beginning of a data frame carrying multimedia and other materials for service. In other embodiments, one or more of the symbols may also be inserted into the data frames. Moreover, one or more of the symbols may be inserted into and/or within a hyperframe consisting of two or more data frames.

In an embodiment, the symbols include a first symbol that can be used to identify the type of signal that is desired. Moreover, the first symbol can be used to detect an offset from the center frequency of the radio channel. The signals may include a second symbol that carries data relating to modulation parameters used in subsequent data symbols. In another embodiment, the symbols include a third symbol that can be used for channel estimation.

Figure 4 shows symbols in accordance with one aspect of the invention (for channel searching) And the synchronization symbol for service discovery, S1-S3) and the frame and super frame of data D.

In various digital broadcast networks, a multi-carrier signal can be set relative to the channel raster such that the signal center frequency (SCF) coincides with the channel center frequency (CCF), or it can be offset from the channel center frequency. The signal center frequency can be offset due to spectrum usage reasons (eg, interference from an adjacent channel). For this first symbol, not all available subcarriers are used. In various embodiments, the subcarriers selected for the first symbol can be evenly spaced and the location can be symmetric about the channel center frequency or the offset signal frequency.

Figure 5 shows how a signal center frequency can be consistent with a channel center frequency (CCF) or offset relative to a channel center frequency. In Figure 5, SCF A is consistent with its corresponding CCF, and SCF B and SCF C are offset with respect to the corresponding CCFs. The rectangle in Figure 5 illustrates the subcarriers selected by the first symbol from among the available subcarriers. For SCF A, SCF B, and SCF C, the selected subcarriers are centered on their respective SCFs. However, for SCF D, the selected subcarriers are centered on the CCF opposite the SCF.

For the first symbol for channel search and service discovery, the subcarriers can be selected such that they can be found regardless of the offset. In the first symbol, a fixed fast Fourier transform (FFT) can be utilized. The fixed FFT can be selected from available FFT sizes including 2K, 4K, 8K in current digital video broadcasting systems, but can also include 1K at the lower end and 16K at the higher end. In an embodiment, the lowest available FFT is used. Moreover, the first symbol can use one A fixed guard interval (GI) that can be selected from the GIs of the symbols used to carry the material. In an embodiment, the first symbol may have no guard interval.

The number of subcarriers used for the first symbol may be less than half of the available subcarriers.

When the first symbol is used for channel offset signaling, the carriers may be modulated using binary phase shift keying (BPSK) or quadrature phase shift keying (QPSK). The selected steering pattern may be different for different channel offset values, and in one embodiment, the guiding pattern and subcarrier modulation may be selected such that different guiding patterns are orthogonal to one another and The differences between each other are maximized to provide robustness in detection. In an embodiment, the different steering patterns may only signal the FFT size and the frequency offset is found by detecting a shift from the nominal center frequency.

For this second (and third, if any) symbol, the entire signal bandwidth (essentially all available carriers) can be used. In an embodiment, the second (and third) symbol may use the same FFT size and guard interval as the first symbol. In some embodiments, not all available subcarriers are used for the second (and third) symbol. In an embodiment, the second and third symbols may have the same subcarriers as the pilot subcarriers, and in a further embodiment, have additional subcarriers for use as bootstrap. In an embodiment, the second symbol further carries a signaling material and, further, carries forward error correction data (FEC) for the signaling material.

According to an embodiment, a signal has a known characteristic and maintains the same portion (eg, (multiple) initialization) as a different digital video broadcast mode of operation / sync symbol) was introduced. The known portion of the signal contains parameters for the remainder of the signal; therefore, after the known portion is decoded, the remainder of the signal can be decoded without using the trial error method. Moreover, with the known portion of the signal, the channels containing the digital video broadcast service can be efficiently detected. If the fixed known portion is not found from the checked signal, the signal will be treated as a non-digital video broadcast signal or a null channel, and the receiver can quickly proceed to the next channel/frequency.

Figure 6 is a flow chart showing the steps performed by a receiver in accordance with at least one embodiment. Based on the channel raster, a frequency synthesizer in the receiver is programmed to the nominal center frequency of the channel to receive a signal on the channel, as shown in step 602. By comparing the received signal with a set of stored known signals, an attempt is made to determine if the received signal is of a desired type and if an offset is used, as shown in step 604. If a match is found, the signal is determined to be the desired type, and the offset and FFT size for the signal can be determined. A determination as to whether a match is detected is made as shown in step 606. If a match is not detected, then the "NO" branch of step 606 is considered, the channel is considered to contain a non-digital video broadcast signal or the received signal is not of the desired type, and processing continues to the next channel. , as shown in step 608.

Otherwise, if a match is detected, then the "YES" branch of step 606 is used, and the determined frequency offset is used to re-plan the frequency synthesizer, as shown in step 610. The next sync symbol is demodulated to detect the modulation parameters for the data symbols, as shown in step 612. Finally, channel estimation And correction and data demodulation are then performed as shown in step 614.

If the resynchronization of the frequency synthesizer takes a considerable amount of time, the receiver can wait for the next set of initialization/synchronization symbols and demodulate the modulation parameters from the group.

Figure 7 shows an example of the magnitude of a pilot signal bandwidth relative to a signal bandwidth and a channel raster bandwidth in accordance with one aspect of the present invention. In an embodiment, the first symbol is a pilot symbol for coarse frequency and timing synchronization. The pilot symbol has a bandwidth that is less than the actual data symbol, such as in the example of an 8 MHz data symbol, which may be 7 MHz wide. The pilot symbol center frequency can be the same as the frequency used for the data symbols, i.e., if an offset is used for the data symbol, the offset can also be used for the pilot symbol. Where a smaller bandwidth of the pilot symbol is used, the RF portion of the receiver can be scheduled to the nominal channel center frequency during the initial synchronization phase and still set to receive the entire bandwidth of the pilot symbol . In the case where the smaller bandwidth of the pilot symbol is not used, the RF channel selection filter of the receiver will filter out portions of the pilot symbol.

In an embodiment, the pilot symbols can be selected using known (fixed) FFT and guard interval. Moreover, the number of boots utilized may be different from that used for data symbols, ie, portions of such guidance may disappear, for example 256 boots may be used. These boots can be modulated with a known sequence.

Figure 8 illustrates a sparse lead spacing for a pilot sequence of a pilot symbol in accordance with one aspect of the present invention. The modulation sequence "fingerprint" for this guided pattern can be known by the receiver. In addition to modulation, the subcarriers in the pilot symbols may also have different surge levels, as illustrated in FIG.

Figure 9 is a flow chart showing the steps performed by a receiver to perform correlation in the frequency domain to detect the coarse offset being used. A radio frequency portion (frequency synthesizer) of the receiver is programmed to the nominal center frequency of the channel (according to the channel raster) as shown in step 902.

An FFT is calculated using a pre-determined FFT size, as shown in step 904. The width of the pilot signal is less than the channel bandwidth. Therefore, the FFT can acquire the pilot symbol even when an initial setting to the frequency synthesizer is erroneous due to the offset.

The frequency offset is detected based on the offset of the pilot sync symbol in the frequency domain, as shown in step 906. If no correlation is found in the frequency domain, the signal is not a digital video broadcast signal and the channel search can continue to the next channel.

The offset is compensated by re-planning the frequency synthesizer of the receiver, as shown in step 908. The next sync symbol is demodulated to detect the modulation parameters for the data symbols, as shown in step 910. Based on the channel estimate symbols, channel estimation and correction are performed, as shown in step 912, and then the data is demodulated, as shown in step 914. In an embodiment, the receiver can wait for a sync symbol in the next set of sync symbols, thereby allowing the frequency synthesizer to be re-planned to the signal center frequency.

Different boot sequences (fingerprints) can be used depending on the offset in use. For example, if 7 offsets are possible (±3/6 MHz, ±2/6 MHz, ±1/6 MHz, 0), then 7 different boot sequences can be introduced. Several methods can be used to establish the bootstrap sequence, including but not limited to: a pseudo-random sequence, reverse per second, augmenting the center carrier, and the like. According to an embodiment, the receiver A correlation is performed in the time domain to detect the boot sequence used and thereby detect the offset used. Such fingerprints may be used in accordance with one or more embodiments for performing a time domain correlation. However, in a frequency domain embodiment, the offset can be detected in the frequency domain by a sliding correlator, that is, a single fingerprint can be used. Moreover, for frequency domain embodiments, for example, if different fingerprints are used for different FFT sizes, information like FFT size can be encoded. Then, a frequency domain correlation can be performed using multiple fingerprints. In an embodiment, if there are multiple fingerprints in use, the received fingerprints can be compared to multiple stored fingerprints at the same time. A received pilot sequence can be progressively translated in the frequency domain by the channel bandwidth, wherein a high correlation signal is generated when the pilot sequences are identical.

Figure 10 is a flow diagram showing the steps associated with performing a service discovery in the time domain, in accordance with an embodiment. A radio frequency portion (frequency synthesizer) of the receiver is programmed to the nominal center frequency of the channel (according to the channel raster) as shown in step 1002.

In an embodiment, a correlation of the received pilot sequence with a known pilot sequence is performed in the time domain to detect the offset used, as shown in step 1004. For example, if 7 offsets are in use, then 7 different boot sequences (fingerprints) are defined. Each coarse offset corresponds to a particular boot sequence fingerprint. Based on this correlation, the fingerprint used (i.e., the offset used) can be detected. The pilot sequence will be at the nominal center frequency of the channel (according to the channel raster). In an embodiment, a set of pilot symbols are defined such that each of them corresponds to a frequency offset-FFT size pair, wherein the offset and FFT size can be detected based on the detected correlation .

Based on the identified boot sequence fingerprint, the frequency offset is detected, as shown in step 1006. If none of the boot sequences has a display correlation, then the signal is not a desired digital video broadcast signal and the search can continue to the next channel.

The offset is compensated by re-planning the frequency synthesizer of the receiver, as shown in step 1008. The next sync symbol is demodulated to detect the modulation parameters for the data symbols, as shown in step 1010. Channel estimation and correction based on the channel estimate symbols are performed, as shown in step 1012, and then the data is demodulated as shown in step 1014. In an embodiment, the receiver can wait for the next set of sync symbols, thus allowing the frequency synthesizer to be re-planned.

After the offset has been found and the frequency synthesizer is re-planned, the second symbol (ie, the symbol following the pilot symbol) can be selected using a fixed FFT and guard interval, but the entire signal frequency is used. width. The second symbol can then contain specific information about the modulation parameters for the data symbols that follow. In another embodiment, the second symbol can use the FFT signaled in the first symbol.

A third symbol that can be traded can be inserted before the data symbols to facilitate channel estimation.

Figure 11 shows an example of a sequence of pilot/telephone symbols in accordance with one aspect of the present invention. The pilot symbols 1102 and the transmit symbols 1104 and 1106 can be repeated frequently enough in transmission, for example every 50 ms, to enable signal detection and synchronization to be as fast as desired. The first pilot symbol 1102 is used for coarse frequency and time synchronization, and in addition, it can also carry about The information of the FFT size of the face symbol. For the first symbol, the FFT, guard interval, and modulation are fixed. In an embodiment, the second symbol 1104 includes the same pilot subcarrier as the first symbol, but may additionally have more subcarriers used as pilot subcarriers. The second signaling symbol also carries signaling data including FFT size, guard interval, and modulation parameters. The third signaling symbol contains more bootstraps and is used for channel estimation and fine timing.

This modulation parameter for the data symbols (like the QPSK vs. 16QAM vs. 64QAM star map) can be changed frequently because the repeated transmit symbols carry information about the selected parameters.

Figure 12 is a flow diagram showing the steps of a method performed by a transmitter in accordance with at least one level of the present invention. A sequence of symbols is included that includes a pilot symbol as a first symbol followed by a next signaling symbol as a second symbol followed by a plurality of data symbols, wherein the pilot symbols are combined to convey a coarse frequency And timing synchronization information, the next signaling symbol is configured to transmit modulation parameters, as shown in step 1202. In an embodiment, a third signaling symbol may follow the second signaling symbol. Then, the symbol sequence is transmitted on a broadcast channel having a pilot signal bandwidth, the pilot signal bandwidth may be narrower than a data signal bandwidth, and the data signal bandwidth may further be narrower than a channel raster frequency of the broadcast channel. As shown in step 1204.

According to one or more embodiments, the channel seek time is typically quite low, such as a few seconds. If the pilot-symbol repetition rate is 50 ms, the average time for 3 bandwidths (6, 7 and 8 MHz) is approximately 35*3*50 ms = 5.25 s. Because the center frequencies of the channel rasters are different, the different bandwidths are individually searched.

In still another embodiment of the present invention, two pilot symbols P1 and P2 are respectively defined to enable fast channel search and service discovery within the frame. In addition, a P2-1 packet structure is defined for the transmission of OSI layer 1 (physical layer (L1)) and frame specific information within the P2 symbol. In addition to the L1 and frame specific information, the P2-1 packet can also carry OSI layer 2 (data link layer (L2)) signaling information (such as program specific information / service information (PSI / SI)) or actual Service information.

In one aspect of the invention, the pilot symbol P1 can enable a fast initial scan of the signal. In this initial signal scan, the pilot symbol P1 can also be used to signal the FFT-size and frequency offset to a receiver. In addition, pilot symbol P1 can be used to assist in coarse frequency and coarse timing synchronization of the receiver.

In yet another aspect of the invention, in addition to the initial synchronization achieved with the pilot symbol P1, the pilot symbol P2 can also be used for coarse and fine frequency synchronization and timing synchronization. Moreover, the pilot symbol P2 may also carry a physical layer (L1) signaling information, and the entity layer (L1) signaling information may describe the physical parameters of the transmission and the configuration of the TFS-frame. In addition, pilot symbol P2 may provide an initial channel estimate that may be needed to decode information in the P2 symbol, and the initial channel estimate, along with the scattered bootstrap, may decode information in the first few data symbols in the frame. Finally, the pilot symbol P2 can be provided to carry one of the layer 2 (L2) signaling information.

In an embodiment of the invention, two P2 packet structures can be implemented Carry signal information. The first one of the packets P2-1 can carry the primary signaling information needed in time-frequency segmentation (TFS). The P2-1 packet structure can also carry L2 signaling information and/or data. In another embodiment of the invention, a second packet structure P2-n can be used to provide sufficient spacing to encapsulate all of the required L2 signaling information. The P2-n packets can be carried as data content in the data symbols. The P2-n packets may follow the P2-1 packet.

Fig. 13 illustrates the structure of the P2-1 packet structure 1301. In P2-1, exemplary fields and lengths are defined for various embodiments as follows:

T (Type) This 8-bit field 1302 may indicate the type of the associated P2 symbol. This field provides flexibility to the network to deliver different P2 symbols. Based on this type of value, certain rules and semantics apply to the P2 symbol structure and usage. An example of the latter is, for example, the impact of different output stream types supported by the system (i.e., different combinations of TS and Generic Stream Encapsulation (GSE)). Some examples of this type of value can be found in Table 1 as explained below.

Those of ordinary skill in the art will appreciate that the notation DVB-T2 or T2 and DVB-H2 or H2 shown in Table 1 can be used for content intended for ground (fixed) reception, respectively, for utilizing the present invention. The various embodiments of the mobile handheld reception.

L (Length) This field 1304 may indicate the length of the P2-1 packet, counting all bits immediately following the field. According to this definition, the length can be expressed as a number of bits or a group of bytes.

E (end) This field 1306 contains a one-bit flag indicating whether there are other P2-n packets after this. When set to the value '1', then there are no P2-n packets after the packet. If the field is set to a value of '0', then there is one or more P2-n packets following the field after this field.

N (Notification) This 4-bit field 1308 may indicate whether a notification is carried in the current sub-signal. The detailed notification of the notification can be completed within the L2 signaling structure.

Res This 4-bit field 1310 can be reserved for future use.

Honeycomb ID This 8-bit field 1312 can indicate the cell _id of the signal carrying the current frame. The mapping between the cell _id and other network parameters is done within the L2 call. It should be noted that in order to reduce the additional burden, this field can be smaller than that used in conventional DVB systems.

Network ID This 8-bit field 1314 can indicate the network _id to which the signal carrying the current frame belongs. The mapping between network _id and other network parameters is done within the L2 call. It should be noted that in order to reduce With a small additional burden, this field can be smaller than that used in traditional DVB systems.

Frequency Index This field 1316 may indicate the frequency index of the current sub-signal. The frequency index can be mapped to the actual frequency, for example in L2 signaling (eg in PSI/SI). Table 2 lists an example of this latter mapping. In this example, four frequencies are utilized, but in other embodiments, the number can be smaller or larger.

The GI field 1317 may indicate a guard interval.

Frame Number This 8-bit field 1318 indicates the number of the current frame in a frame.

The transmit window offset This 8-bit field 1320 can indicate the start of the (time slot) transmission provided within the P2 symbol. The offset from the beginning of the frame within the current sub-signal is indicated by the number of OFDM cells. The total length of the time slot covered by the signaling window is equal to the length of the time slot in the current frame and sub-signals. For example, Figure 14 illustrates the concept of a transmission window offset. In Fig. 14, a messaging window 1402 may have a length of approximately one frame, but may not begin with the first time slot of the frame. Offset 1404 can define the starting point of the transmission carried within the TFS-frame. If the offset is zero, The window can then point directly to the current frame and signal all services within the frame. If the offset is a frame, the window can point to the next frame. If the offset is less than a frame, the signaling may begin with a service pointed to by the signaling offset and may signal some services that are approximately corresponding in length to a frame.

Transmitting Time Slot ID This field 1322 identifies the time slot carrying the P1 and P2 signaling data. It should be noted that the same time slot ID can also carry other information, such as L2 signaling or data containing actual services.

Number of Time Slots This 8-bit field 1324 may indicate the number of time slots included in the messaging window of the current sub-signal.

The field 1326, which may be part of the time slot circuit field 1325, may identify a time slot within the messaging window of the current sub-signal. The one-time slot identified by the identifier can carry the data containing the actual service or the L2 signaling data.

This field 1328, which may be part of the time slot loop field 1325, may indicate the modulation type of the associated time slot.

This field 1330, which may be part of the time slot loop field 1325, may indicate the code rate of the associated time slot.

This field 1332, which may be part of the time slot circuit field 1325, may indicate the length of the associated time slot. By definition, this length can be represented as the number of bits or bytes.

OFDM Fill Bits This 8-bit field 1334 may indicate the number of padding bits within the last OFDM cell of the frame.

Offset from the next T2 sub This 4-bit field 1336 may indicate the offset between the current and next sub-signals of the frame. For example, Figure 15 illustrates an offset 1502 from the next sub-signal field.

L2 Sending or Data This field 1338 can be reserved for carrying L2 messages or data. This type of field in P2-1 indicates the information carried in this field.

In an embodiment of the invention, a single P2-1 packet may not be large enough to carry all L2 signaling information. Therefore, additional P2-n packets may be required to carry and encapsulate L2 signaling. Figure 16 illustrates the structure of a P2-n packet 1602 that can be used to carry L2 signaling information such as PS/SI. The definition of the fields in P2-n is as follows, where for various embodiments, the field length is exemplary:

T (Type) This 8-bit field 1604 may indicate the type of the message carried within the payload of the packet. Based on this type of value, a receiver can correctly decode the carried message. Examples of the type of signaling may include, for example, only PSI/SI, PSI/SI, and signaling intended for mobile handheld services, in accordance with embodiments of the present invention.

L (Length) This field 1606 may indicate the length of the P2-n packets, counting all the bits immediately following the field. By definition, this length can be represented as the number of bits or bytes.

E (end) This field 1608 may include a one-bit flag indicating whether the current packet is the last P2-n packet or whether there are other P2-n packets following the packet. .

L2 sends this field 1610 to be reserved for carrying L2 signaling. This type field 1604 can indicate the information carried within the field.

A plurality of P2-n packets can be used depending on the number of L2 signaling materials.

Figure 17 depicts a flow chart illustrating service discovery in accordance with one aspect of the present invention. In Figure 17, the L2 signaling is carried in two packets (P2-1 and P2-n). Other changes to P2-n may include any combination of data carried in any one or two packet types with L2 signaling information.

In Fig. 17, in accordance with various embodiments of the present invention, a receiver can find a signal from a frequency band carrying a signal, as shown in FIG. With the preamble provided by P1, an appropriate frequency can be detected.

Based on the information carried in P1, a receiver may be able to decode the P2-1 packet 1702 and the p2-n packet 1704 carried in the following symbols, as shown in 2. In one aspect of the invention, four fields can be included in the P2 packet header. The "T" field 1706 can indicate the type of current signal. The "L" field 1708 may indicate the length of the P2. If the 'E' field 1710 is set to '1', then the current P2-1 packet is the 'last', ie there is no subsequent P2-n packet following. Finally, the 'N' field 1712 can indicate whether the current signal carries notification information.

According to the P2 packets (ie, P2-1 and P2-n), the receiver may be able to receive the L2 signaling data 1714, and the L2 signaling data 1714 may be carried in the P2-1 and P2-n packets. Within the load, as shown in 3.

Then, the L2 signaling material 1714, that is, the specific PSI/SI used for the type of transmission in the case of carrying only the type of transmission, may be mapped using the network and the cellular information 1716, as shown in FIG. . Network information Table (NIT) 1720, information about adjacent honeycomb cells (including the geographic location of each of the honeycomb cells 1718) can be provided.

Moreover, time frequency partitioning (TFS) specific information may be partially carried within the PSI/SI. An NIT can map each frequency portion of the same TFS frame, as shown in Figure 5. The NIT can map each frequency portion of the same TFS frame. Finally, the NIT maps the transport stream to a different frequency and, in addition, maps to a different TFS frame, as shown in 6.

In a further aspect of the invention, the transport stream can be mapped to a service within a Service Description Table (SDT) 1722, as indicated by 7, by following the semantics of conventional PSI/SI. As shown in Figure 8, by utilizing PAT and PMT, services can be further mapped to the PIDs of each transport stream, similar to in conventional DVB systems.

By adding additional descriptors, the mapping of each service in combination with time slot _id 1724 and frame_number 1726 can be done within the service loop of the SDT, as shown at 9.

Finally, as shown at 10, the receiver may continue the service discovery procedure within the P2-1 packet by checking which time slots may need to be received in order to receive the desired service advertised within the SDT. The tables NIT, SDT, PAT, and PMT are used as examples corresponding to current (legacy) DVB tables.

In a further aspect of the invention, the L1 information signaled within the P2-1 packet may be related to the particular messaging window. The start position of the transmission window can be indicated by 'Signal Window Offset - Field'. The total number of time slots located within the given messaging window can be indicated in the &apos;Time slot number field&apos;. For the purpose of P1 and P2-1 packets, a specific time slot ID can be signaled. The P2-n packets can be carried in the 'general time slot' and thus can also contain actual Content material. The time slot loop indicates the modulation, code rate, and length for each time slot advertised within the loop. In addition, the OFDM padding bit field can be used to indicate possible padding in the end of the frame. Finally, the 'Offset with Next T2 Sub' field may indicate the offset between the current and next sub-signals of the associated frame.

By way of example, Figures 18 and 19 depict the relationship between P1, P2, and DATA symbols (i.e., OFDM symbols). It can be seen from Figures 18 and 19 how the data has been segmented during the duration of P2 and data symbols. The data packets may be placed directly after the last P2-n and both are carried in the &apos;DATA symbols&apos;.

Figure 20 shows an exemplary frame and time slot structure in accordance with at least one level of the present invention. In FIG. 20, a frame 2002 may be comprised of one or more time slots 2004. For example, frame 2002 includes time slot 1 2006 to time slot 4 2012. Each time slot 2006-2012 may include multiple OFDM (Orthogonal Frequency Division Multiplexing) symbols, typically from a few symbols to tens of symbols. These services are assigned to these time slots such that one or more time slots are used for a service. For example, time slot 1 2006 may include some OFDM symbols 2014 through 2024. Furthermore, each OFDM symbol can include a number of OFDM cells. For example, OFDM symbol 2014 includes OFDM cells 2026 through 2034.

The embodiment relates to initial service exploration in the next generation terrestrial digital video broadcasting (DVB-T2) system. The DVB-T2 system can include a preamble that is intended to effectively identify available T2 signals. This preamble should not consume too much capacity, but it should be compatible with different Fast Fourier Transform (FFT) sizes (2k, 4k, 6k, 8k, and 32k). Minimize the extra burden already A 2k symbol (P1) is used for each FFT-size and the actual FFT-size is signaled within the symbol by modulating the carriers with different pseudo-random binary sequences (PRBS). To find the FFT-size of the following symbol, the receiver detects the modulated PRBS. The PRBS also indicates an integer frequency shift (the DVB-T2 signal can be shifted by +/- 1/6, +/- 2/6, +/- 3/6 MHz) compared to the nominal center frequency. In summary, the P1 symbols are used in the initial scan to: (1) detect the presence of the T2 signal; (2) estimate the frequency offset; and (3) detect the FFT size used.

After the initial scan, the P1 symbol may not be used during normal data reception or handover because the parameters carried by P1 (ie, FFT size and frequency offset) remain unchanged. Regarding handover, these parameters are the same between radio frequency (RF) channels or they are signaled prior to handover (eg, program specific information/service information according to ETSI EN 300 468 Digital Video Broadcasting (DVB) ( PSI/SI); Specification of Service Information (SI) in DVB systems). However, P1 can be used during normal data reception, for example, to detect the start of the frame or to improve synchronization and channel estimation algorithms. The P2 symbol(s) are one or more signaling and channel estimation symbols located after the P1.

The detection of P1 and thus the detection of the DVB-T2 signal is based on a guard interval correlation (GIC). In GIC, this guard interval is associated with the end of the symbol. A peak in the GIC indicates a possible DVB-T2 signal from which the possible DVB-T2 signal can be verified. The first problem is that to provide robust detection, the guard interval should be long (ie, a long guard interval provides a higher signal-to-noise ratio). However, a longer guard interval and thus a longer P1 reduces the data capacity.

Since P1 is the first symbol to be received, there is usually no prior knowledge of the channel situation. Therefore, the P1 symbol should include some method to overcome channel distortion. In practice, this would mean using, for example, an additional pilot carrier for channel estimation or differential modulation between the subcarriers.

Due to the lower FFT-size, the carrier spacing of the P1 symbols may not be as dense as in the following data symbols (eg, 2K for P1 and 32K for data). For a successful PRBS detection in P1, the coherence bandwidth of the channel should be less than the subcarrier spacing of a 2K symbol. However, the network can be designed for 32K mode, and long single frequency network (SFN) delays can produce much higher frequency selectivity.

The received complex-valued signal at carrier index k can be represented as r _k =h _k s _k +n _k , where s _k is the transmitted data symbol (eg, using quadrature phase shift keying (QPSK)), h _k is the channel response at carrier index k, and n _k is the noise term.

At the same tune in the mediation, by the guide, h _k is estimated first, and then, for example, by dividing the estimated r _k h _k, the effect of the channel is equalized.

If we consider DVB-T2 and P1 symbols, there is no guidance to estimate h _k . Therefore, in the absence of channel estimation, non-coherent demodulation will typically be used. This can be done by using differential modulation, such as Differential Binary Phase Shift Keying (DBPSK), where the information is encoded into a phase difference between two adjacent carriers. These two adjacent carriers can be expressed as r _k =h _k s _k +n _k and r _k+1 =h _k+1 s _k+1 +n _k+1 . The transmitted symbol can be decoded from the phase difference between the two received carriers: r _k+1 -r _k ==h _k+1 s _k+1 -h _k s _k +n.

Figure 21 illustrates the coherent bandwidth and differential modulation within a pilot (P1) OFDM symbol. It is assumed that the phase of the channel response h _k and h _k+1 is approximately the same as that shown in the upper graph of Fig. 21. However, in a highly frequency selective channel (e.g., the lower graph of Figure 21), the correlation between adjacent channel responses is quite low. This makes it impractical to use differential modulation between carriers.

The coherence bandwidth (ie, the bandwidth at which the channel response is highly correlated) is available To approximate, here τ _d is the delay spread of the channel. In order to use DBPSK between carriers, the coherence bandwidth of the channel should be lower than the carrier spacing. The FFT size of P1 is 2K and the carrier spacing in the 8 MHz channel is 4.46 kHz. From these carriers, every third or ninth carrier is used. Therefore, the actual carrier spacing can even be 40.1 kHz. Conversely, the delay spread in a large SFN network can be 448 μs (16k mode with 1/4 guard interval), resulting in a 2.2 kHz co-modulation bandwidth.

According to one aspect of the invention, two P1 symbols are used, for example, a 1k symbol with GI = 1/1. These two symbols are used separately in the GIC. When GI = 1/1, the entire symbol duration can be utilized in the GIC.

According to one aspect of the invention, differential modulation is applied between two P1 symbols, as shown in FIG. Since the differential modulation is currently performed on a subcarrier-by-subcarrier basis, the homology bandwidth is not required. (Optionally, the first P1 symbol can be used for channel estimation, which will allow for homology demodulation of the second P1 symbol.)

The time interval of the two P1 symbols is relatively short, whereby the channel does not change from the first symbol to the second symbol. Thus, in accordance with one or more embodiments, in the time domain, the differential modulation can be done between carriers having the same carrier number.

Embodiments also support action reception. According to an embodiment, the coherence time of the channel is longer than the duration of the two P1 symbols. This makes the correlation between r _k (1) and r _k (2) high. The channel's coherence time is available To approximate, here, F _d is the Doppler extension of the channel, and it consists of Given, here v is the speed of the receiver, c is the speed of light (3*10^8 m/s), and F _c is the carrier frequency. If v = 120 km/h and F _c = 666 MHz, F _d = 74 Hz and τ _coh = 13.5 ms, which is significantly longer than the duration of a P1 symbol (eg 280 μs).

According to one or more embodiments, the symbol synchronization of P1 can be improved. The P1 symbols can have a 1/1 guard interval that improves symbol synchronization and maximizes the guard interval correlation length with respect to additional burden. These P1 symbols can use a 1k FFT, which reduces the additional burden compared to the two 2k symbols.

Guard Interval Correlation (GIC) is a basic method for synchronization in Orthogonal Frequency Division Multiplexing (OFDM) symbols. Since the GI is a cyclical copy of the last portion of the actual OFDM symbol, the receiver can find the beginning of an OFDM symbol by detecting the correlation. In effect, the receiver continuously correlates the two blocks of the received signal, the two blocks being separated by N samples ( N is the FFT size and is also the number of data samples). A correlation peak is detected at the correct location.

Figure 23 shows two 1k symbols with a 1/1 guard interval and differential modulation between the symbols. As can be seen, the 1/1 guard interval means that the GI and the data portion have the same length and the samples are also equal. Equivalent to the 1/1 guard interval without the guard interval It is considered to have two equal symbols.

Due to the differential modulation, the consecutive symbols P1 and P1' are different, which means that a regular GIC should be applied within each P1 symbol. However, the correlation length is twice as large as a 2k 1/4 GI symbol (1/4*2048=512), and the correlations from the two symbols can be combined for further improvement. The 1k 1/1 GI symbol is still satisfactory because the guard interval correlation does not currently match the data patterns (2k, 4k, etc.).

Another embodiment speeds up the initial scan. It is desirable to quickly detect a non-T2 signal to enable the receiver to tune to the next frequency. This can be done by detecting the zero carrier in the P1 symbol by: (1) calculating the three sums of the received energy of the carriers belonging to the subsets r _3k , r _3k-1 , and r _3k+1 (see 24th) Figure 24 illustrates the calculation of the sum of the energy received from P1 according to one or more embodiments, where r is the kth carrier of the P1 symbol and k = 1, 2, 3 ... and (2) detecting the presence of the T2 signal by comparing the received energy of the three subsets; and (3) setting an energy threshold (eg, 5 dB below the maximum energy); and (4) As soon as one sum exceeds the threshold, a possible T2 signal is detected.

Figure 25 shows a transmitter in accordance with one or more embodiments. According to a reference sequence, the first P1 is modulated by BPSK, and the second P1 is modulated as follows: if PRBS _k =0→b _{k, 2} = b _k,1 ; if PRBS _k =1→b _{k, 2} = -b _k,1 (or vice versa), where PRBS _k is the kth element of the PRBS and b _k,m is the transmitted symbol on the kth carrier at the mth P1 symbol. The transmitter then combines the original reference sequence and the delayed differentially modulated sequence prior to inverse fast Fourier transform (IFFT) and guard interval insertion. N refers to the FFT size.

Figure 26 shows a receiver in accordance with one or more embodiments. The receiver performs the inverse of the transmitter operation discussed above in relation to Figure 25. That is, the receiver removes the guard interval from the P1 symbols (first and second pilot symbols), performs a fast Fourier transform on the P1 symbols, and then micro-decomposes the P1 symbol to obtain the transmitted An estimate of a pseudo-random binary sequence. The receiver does not have to know the reference sequence.

Figure 27 is a flow chart showing the steps that can be performed by a receiver in accordance with one or more embodiments. In this initial scan, the receiver can be tuned to the nominal center frequency of the channel and can begin looking for the P1 symbol. Then, the following procedure can be repeated at the selected channel (and bandwidth) - but it is not necessary to repeat at each frequency offset because the P1 symbol can be detected at the nominal center frequency regardless of the Offset.

The first task after bandwidth and nominal center frequency selection is to find the presence of a T2 signal. The P1 symbol can be found, for example, by a guard interval correlation that immunizes the frequency offset. The use of guard interval correlation also contributes to T2-signal detection because the absence of a 2k symbol implies a non-T2 channel.

The guard interval correlation is intended for the case where the delay spread of the channel remains within the guard interval, and this may not be an example of a P1 symbol in a large-scale SFN (e.g., in 32k mode). In this example, a delay that is longer than the guard interval, particularly a delay that is several times the duration of the useful symbol, produces an erroneous correlation.

However, it should be noted that the symbol timing in the presence of a powerful SFN echo is not just a specific P1 problem, since the receiver needs to be able to synchronize to the correct path anyway. The difference is due to the shorter GIC window, P1 related Has a high noise level.

The coarse time and fractional frequency synchronization are obtained from the guard interval correlation. These are rough estimates for the P1 symbols themselves, and they can be improved with the latter symbols. The FFT-size is found assuming these estimates are accurate enough to detect one of the five PRBS patterns.

For a fast initial scan, those channels that do not contain a T2-signal should be discarded fairly quickly. The foregoing structure according to an embodiment supports a step-by-step detection in which the non-T2 channels can be discarded relatively quickly, and by reading the L1 static transmission, the detection of a T2-signal can be confirmed.

With this guard interval correlation, the first cancellation can be completed. The P1 signal can be repeated every frame (approximately 200 ms) and is quite robust in terms of SNR requirements, so testing two consecutive P1 positions can be sufficiently reliable to detect the T2 signal. Each RF-channel will take approximately 500 ms. A receiver can then determine if a possible P1 symbol has been found. If done through 39 UHF-channels and even with 3 channel bandwidths, the total time for the scan is approximately 58 seconds. It is important to note that attempting to scan different bandwidths at the same time is actually not helpful because the channel rasters are different.

Once a possible P1 symbol has been found, the receiver can perform coarse synchronization and FFT. The receiver can then use a sparse carrier raster to distinguish between T2 and other 2k signals. Thus, the non-T2 signals will most likely be detected from the first received P1 symbol.

The detection of the frequency offset is based on finding the shifted guide pattern. The correct offset is first found by using the power on the assumed pilot carrier, and thereafter, the correlation is calculated for the five PRBSs, the frequency offset and the FFT- The detection of the size can be separated. On the other hand, the PRBSs can already be used when the frequency offset is found. This sparse carrier raster reduces the complexity of the search algorithm.

After the frequency offset has been detected, the receiver can be tuned to receive the data symbols. Another task is to find the guard interval used to decode the P2 symbol. Since the P1 symbol does not carry any signaling information of the GI, the receiver can detect the GI by utilizing the regular OFDM symbols during the frame. The P2 symbol immediately after the detected P1 cannot be decoded. However, there is enough time to detect the GI before the next frame because the entire 200 ms frame duration can be used. This adds an additional 200 ms to the signal acquisition time, but this is likely to occur only for the discovered T2-signals, not for each tested channel. Since the maximum number of parallel multiplexes is typically 7 to 8, the total time added to the scan sequence is less than 2 s.

If the frame duration is configurable, the frame synchronization can be obtained by recognizing the next P1 symbol. The detected parameters of the L1 static transmission from the P2 symbols are then confirmed.

In an embodiment, the first P1 is used for a channel estimate, and then the channel estimate is used to equalize the second P1. This reuses a basic concept of various embodiments, although the implementation is different. N refers to the FFT size.

According to the DVB-T2 standard, the P1 and P2 symbols are proposed as a solution to the transmission of initial scanning and signaling. According to an embodiment, the differential modulation between two P1 symbols can have the advantage of a highly frequency selective channel.

As discussed above, the P1 symbols are used in the initial scan to: (1) detect the presence of the T2 signal; (2) estimate the frequency offset; and (3) detect the FFT size used. One possible way to estimate the frequency offset (and to some extent detect the presence of the T2 signal) is to use a frequency domain 'comb', i.e., use a subset of the available subcarriers in the OFDM symbol. Suppose there are a total of L available subcarriers (= minus the FFT-size of the guard band). Moreover, assuming that every third subcarrier is available for this boot/synchronization purpose, this will have L'= /3 +1 effective subcarriers are used for the synchronization signal. Mathematically, the comb is available one yuan sequence P (0), P (1 ), ..., P (L '-1) is represented. Here, the bit P(k) informs whether the ( lowest ) + 3* k subcarrier contains a binary phase shift keying (BPSK) signal: '0' indicates a subcarrier that does not contain power, and '1' indicates a subcarrier containing a BPSK-modulated signal. The concept is that when the operator uses a channel frequency offset, the comb is shifted accordingly. Therefore, after timing synchronization and fractional frequency synchronization are achieved, the receiver can perform FFT and search for the integer frequency offset. Here, the receiver can use the received power of the assumed pilot carrier (ie, the comb) and find the frequency offset without demodulating the pseudo-random binary sequence. Then, in the presence of a substantially sufficient match with the shifted comb and the measured subcarrier signal power, the correct integer frequency offset (= an integer multiple of the subcarrier spacing) can be detected . Then, the FFT-size (for example, selected from 5 choices) is based on 5 BPSK-types S _m (0), S _m (1)... S _m ( L '-1) (for m A choice of =1, 2, 3, 4 or 5) is indicated.

This frequency offset (after adjusting its fractional portion) is equivalent to adding a constant offset n to the subscripts. And The number of collisions between the comb and its shifted version is calculated, and S (0) = N is equal to the number N of subcarriers in the comb. In order to validate the detection of the integer frequency offset, the total number of collisions S ( n ), n ≠0 should be small compared to the correct match N.

Ideally, the structure of the P1 signals should be such that it also supports other methods of detection, thus providing the hardware designer with freedom of choice. Another method to detect the problem of the presence of a P1-signal is based on time domain correlation. In order to also support this alternative method, these actual signals There should be good cross-correlation properties - not only for different values of m but also for different values of (m, n) pairs, ie for different values of (FFT-size, frequency offset) combination.

Other characteristics required for this set of signals are reasonable time domain autocorrelation properties and reasonable peak-to-average power ratio (PAPR) characteristics. Ideally, it should also be possible to reproduce the comb and the BPSK-sequences quickly and efficiently without relying on a large lookup table.

Embodiments are directed to: 1) a comb limited to every third subcarrier, and 2) a comb containing approximately one-half of the remaining subcarriers, so the number N of effective subcarriers should be approximately L/6. Due to these appropriate assumptions, the length is L '= /3 A shorter comb pattern/sequence of +1 is of interest.

According to an embodiment, a binary m-sequence of a suitable length is used to generate the comb, and the selected cyclic shift of the same m-sequence (currently interpreted as +1/-1 with respect to 0/1) is used Five BPSK-types were produced.

Six bit patterns (each consisting of r bits that are not all zero, which will later be referred to as seed sources) are specified. Then, by applying a recursive formula determined by a primitive polynomial of a first order r, the sources are expanded into a sequence of length 2 ^r -1. It is noted that this same recursive formula is applied to form each of the six sequences. One of the sequences is picked to determine the comb, and the other five determine the BPSK-type, by reinterpreting '0' as +1 and '1' reinterpreting -1 . Ideally, then L '= 2 ^r -1. Different usage examples and an alternative method for constructing the comb can also be used.

In the specific usage example of DVB-T2, L = 1531 subcarriers, so L '= 511 = 2 ⁹ -1, r = 9, and the primitive feedback polynomial 1 + x ⁴ + x ⁹ can be used. An exemplary set of provenances consists of 100 000 000 for the comb and 000 110 101, 110 001 100, 101 111 101, 101 101 111, 111 100 111 for the 5 BPSK-types (all Interpreted as +/-1) composition. By repeatedly applying the recursive formulas P ( k )= P ( k -4)+ P ( k -9)(mod2) and S _m ( k )= S _m ( k -4)* S _m ( k - 9) (for k = 9,10, ..., 510) , these sequences are expanded into a sequence of P and S _m (for m = 1,2,3,4, and 5).

One design criterion for selecting such sources is that when the generated sequences are cyclic shifts of the other, the amount shifted from one to the other should be relatively large. Likewise, the sources can be designed such that one of them cannot be mutually exclusive or exclusive from the comb sequence and another sequence obtained by cyclically shifting by a short (eg, less than 45 positions) ( XOR) is produced.

If the number of available carriers L ' is not in the form of 2 ^r -1, but is also relatively close to such a number, then the comb and the sequences can be truncated from the end of the m-sequence by a small segmentation It is shortened, or the pattern can be expanded by iteratively repeating it for a relatively short period of time. In the above example, the number of subcarriers can be reduced from 1531 to 1507 by cyclically shifting the comb pattern and one position of the BPSK-sequences. To achieve this, the 9-bit source can be extended to 10 bits by applying the recursive relationship once. After this, the first bit can be omitted, thus producing a 9-bit source. Therefore, the source 000 000 001 for the comb and the sources 001 100 010, 100 011 000, 011 111 010, 011 011 110, 111 001 111 for the BPSK-sequences can be replaced by the above proposals. use. Then the comb will start with 8 zeros, ie 24 empty subcarriers, and the P1-signal narrows down to as low as 1507 consecutive carriers. It is observed that the role played by the available bandwidth is less important because in a narrower frequency band (eg 5 MHz) application, the spacing between subcarriers is also narrower and there is still room for space About the same number of subcarriers.

An alternative method of generating a frequency domain comb is to use a squared residual sequence (=QR-sequence) known in the art. The resulting comb shares collision statistics between multiple shifted versions of the comb based on the m-sequence. This alternative method has the advantage that the length of a QR-sequence is a prime number p of modulo 4 and 3. Therefore, when the QP-sequence is used, the length available for the group is more flexible. The cyclically shifted version of the same sequence can also be used herein to construct the BPSK-sequences. However, generating a relatively long QP-sequence in real time is computationally cumbersome, and in fact a relatively large lookup table may have to be used.

According to at least one embodiment, the proposed five P1-signals are For m=1, 2, 3, 4 and 5. Here, n represents the integer part of the frequency offset. It is counted at a multiple of the subcarrier spacing, so in the proposed usage example, n = ±37, ±75, ±112 corresponds to a frequency offset of ±1/6, ±1/3, ±1/2 MHz (note that the fraction of subcarrier spacing is processed earlier, regardless of whether they are a result of a rounding error or a result of a clock difference between the receiver and the transmitter). But the proposed structure actually allows n to any integer value up to 134. Here, P and S _m (for m = 1, 2, 3, 4 and 5) are the sequences of length 511 discussed above. These signals occupy 256 subcarriers within a range of 1531 consecutive subcarriers.

There are various other options that are equally effective for these sources. For example, each of the 6 m-sequences can be cyclically shifted by the same amount without changing the relevant characteristics. These example values of the seed source work well when the integer portion n of the frequency offset is less than 3*45=135. Within this range, the cross-correlation in the offset versions of the sequences remains low. A computer search has revealed other provenance groups with equally good performance. The possibility of a slightly wider range of low correlation is not completely ruled out, but it is known that if n can be as large as 3*51=153, with this method, such a low correlation range cannot be achieved, regardless of the How to choose the source carefully.

A spacing of multiples of 3 allows the integer portion of the frequency offset to be detected relatively quickly because there is no collision between the actual comb and the tested version unless the tested is offset from the actual integer The difference is a multiple of 3. If that condition is met, then when we have the correct offset, the number of collisions is 256, otherwise the number of collisions is in the range 119...128, ie near the best pseudo-random intermediate point 128. For having a similar structure (= limited by every third subcarrier) and density (= one of a total of 6 subcarriers in total) a randomly generated comb of 'effective'), the expected range of the number of collisions (deviation from the expected value +/- 2 standard deviations) is from 104 to 144, so by making the change into a narrower range, This number is improved using m-sequences.

The basic algebraic structure of the m-sequences helps to ensure that almost all of the sequences produced in this manner have fairly good PAPR-characteristics (with the exception that the same provenance is used for one of the combs and one of the sequences) ) and fairly good time domain autocorrelation. Careful selection of such sources further helps to ensure good cross-correlation properties in the offset versions of the various sequences. Indeed, these non-ordinary correlations are very close to zero, as opposed to random upswing to +/- 2 SD levels of 32.

Figure 28 is a diagram of the auto/cross correlation between the pilot sequences and their frequency offset versions in accordance with one or more embodiments.

Figure 29 is an enlarged version of the figure of Figure 28 showing the low cross-correlation range of the frequency offset.

Figure 30 is a diagram showing the envelope amplitude of a first P1-signal (calculated with a center frequency of 666 MHz and a carrier spacing of 4464 Hz, sampled at 25 MHz to produce a single symbol of the graphs) in accordance with at least one embodiment. a graphic. This ratio is chosen such that the mean square amplitude is equal to one.

Figure 31 is an enlarged version of the figure of Figure 30. Figures 30 and 31 together show the reasonable PAPR-characteristics of this group.

In the following discussion of the BPSK and P1 sequences, F = GF (512) will represent a finite field of 512 elements, and g will be a primitive element of F satisfying Equation 1 + g ⁵ + g ⁹ =0 , so the power g ^{i is} the non-zero element of F, when the index i takes the equivalent value i=0, 1, ..., 510. We should further note that g ^-1 will be the one of the earlier feedback equations 1 + x ⁴ + x ⁹ =0. Let tr : F → GF (2) be the tracking function. The previous 0/1-valued m-sequence and all its cyclic shifts are obtained, such as the sequence m _α ( i )= tr ( αg ^{i -1} ), for i =1, 2,..., 511 and α ≠ 0. We write as e ( x )=(-1) ^{tr ( x )} , and ω = e ^{2 πj /511} . Therefore, we can choose element α F and β _j F , j =1, 2, 3, 4, 5, such that the comb of 0 and 1 is obtained, such as P ( i )= tr ( αg ^{i -1} )=(1- e ( αg ^{i -1} ))/ 2, and the BPSK-sequences are obtained, such as S _j ( i ) = e ( β _j g ^{i -1} ). The P1-sequences are thus given by the formula P 1 _j ( i )=(1- e ( αg ^{i -1} )) e ( β _jg ^{i -1} )/2.

We get the equation e ( x ± y ) = e ( x ). e ( y ) and (hereinafter referred to as equation (1) or (1)), whenever γ is non-zero, and the so-called Gauss sums (hereinafter referred to as equation (2) or (2)), wherein when both γ and k are non-zero, the Gauss sum Absolute absolute value And less than this value when one of them, instead of two zeros.

In this regard, we record that the proposed comb corresponds to the choice α =1.

Let us consider the pattern The number of collisions with its shifted version P ( k + n ), where n indicates the amount of shift (maximum 12/3 =37). If we continue to continue the pattern of the comb in cycles of 511, the number of collisions can be calculated. Represents the variable x = g ^k and takes the usual convention: F ^* is a collection of non-zero elements in field F. Then the number of 'collisions modulo 511' is (so, k+n is calculated with 511 as the modulus)

Here, the first sum is 511. Here, the first sum is 511. Since t < 511, the coefficients α , αg ", α (1 + g ⁿ ) are non-zero, and equation (1) tells us that the remaining sums are all equal to -1 (according to the fact that the items are missing in the sum e (0) = 1). In summary, we get that the shifted comb has 512/4 = 128 collisions with the looped comb. When we consider the tail effect due to the sum of k and n overflows > 511 , we see an expected drop in the number of collisions. At n = 1, 2, 3, 4, 6, 7, 8, there are 128 collisions, and the number is approximately linear with increasing n When n reaches the maximum value of 37, the number of collisions is 125. With the offset n=36, the lowest value of 119 collisions is reached. Therefore, due to the comb, the number of collisions between the two offset combs will be close. This ideal intermediate point 128.

We can calculate the cross-correlation between two P1-sequences (in the frequency domain, according to Parseval's theorem, whether this is done in the frequency or time domain is irrelevant), such as , so, the sum (1) tells us that the cross-correlation is equal to zero, but with the proviso that β _j - β _j is non-zero (in other words, the two sequences are different), and α + β _j - β _j is non- Zero (in other words, the two sequences are not bit-wise complementary to each other). An executable test for this is that for the two sequences of the type to be orthogonal, their initial partitions are different from each other, and the bitwise XOR of their initial partition is different from the initial of the comb P. Split the block.

As in the calculation of the number of collisions, we first cyclically extend the sequence in the frequency domain, calculate the cross-correlation between such extended pairs of signals, and more or less ignore the short 'tail' The short 'tail' is the sum of a small number of pseudo-random terms and will not do much. Therefore, the (frequency domain) cross-correlation between a P1-signal and another P1-signal shifted from the P1-signal by t positions is = (hereinafter referred to as equation (3)).

It is observed that here the indices j and j' can be equal, ie we are also interested in the correlation between a sequence and its offset version. From equation (1), we see that this main term is zero unless one of the coefficients in square brackets is zero. Since n takes values in a range about zero, what is left to us is to select the targets of the coefficients β ₁ ,..., β ₅ in such a way that the coefficients themselves are also the sum of α + These bases g discrete logarithm of β ₁ ,..., α + β ₅ are as far apart as possible from each other (cyclic model 511). Since there are a total of 10 field elements here, the minimum interval between the discrete logarithms cannot be higher than 11/10 =51. Due to the choice of the sample structure α = 1 = g ⁰ , a small heuristic search gives the set used in the above discussion: β ₁ = g ³³ , α + β ₁ = g ¹⁸¹ , β ₂ = g ¹³⁵ , α + β ₂ = g ⁴⁹⁹ , β ₃ = g ²⁴⁵ , α + β ₃ = g ³⁹⁸ , β ₄ = g ³⁴⁹ , α + β ₄ = g ⁸⁵ , β ₅ = g ⁴⁴⁵ , α + β ₅ = g ²⁹⁶ . Here, the discrete logarithms form a list {33, 135, 245, 349, 445, 181, 499, 398, 85, 296} - the first five discrete logarithms specify the elements β ₁ ,... , β ₅ , and the last five lists the discrete logarithm of the elements α + β ₁ ,..., α + β ₅ . Here, the minimum cycle interval 45 is between 499 and 33 because 33-499+511=45. Another discrete logarithmic sequence that also has a minimum cycle interval of 45 is {33, 135, 233, 339, 447, 181, 499, 388, 286, 80}. What is unknown is whether there is a choice that leads to a larger loop interval. Since 3*45=135 (subcarrier spacing) is greater than 112, this satisfies our purpose.

These figures explain the gap in Figure 28. Since there is no match with an offset of up to 44 in either direction, the width of the near zero region in Fig. 28 is 2*44 + 1 = 89 carriers. It is to be noted that this interval 45 corresponds to an item with a negative sign in equation (3). The minimum loop interval corresponding to the item with a positive sign is 96, and it appears between the pairs (45, 349) and (181, 85). This explains why the most recent side lobes are negative and also explains the wider gap of 2*96+1=193 carriers above the x-axis.

Here we show the boundaries It is meant how the autocorrelation of the proposed signals is maintained at a low level for at least a certain discrete set time shift. The time domain version of the proposed P1-signal is Wherein for convenience, we can include a frequency offset into f, and f △ so that the spacing between the two possible carrier signal P1- (= 3 times the carrier spacing 2k OFDM- symbol). Suppose we have a time error Δ t that is less than the guard interval. Time domain correlator view = (hereinafter referred to as equation (4)).

Here, the coefficients K and K' are for normalization and contain the power of the ascending power and the constant from the DFT and the integral. Therefore, the absolute value of the item depends only on the sum (based on the ratio). △ t is assumed that such a size with respect to the product of some integer n △ f △ t = n / 511 in. That is, the time error is an integer multiple of 1/511 of the common period of the subcarriers. So we can write e ^{2 πjk (△f.△ t )} = ω ^nk . Considering the fact that P 1 _j ( k +1)=(1- e ( αg ^k ))/2 depends only on the type of the comb (and does not depend on the BPSK-modulation at all), we see that At these values of the time error, the cross correlation is equal to (hereinafter referred to as equation (5)).

The sums in equations (1) and (2) tell us (ignoring the multiplier K) - its absolute value is independent of n) when n = 0 (ie when there is no timing error), the sum has a value of 256 And otherwise have absolute values . In summary: due to our signal, this has a discrete set of relatively dense time errors, and the discrete set of relatively dense time errors will cause the autocorrelation value to be approximately 10 dB below the synchronization value. Although this is not decisive, it is highly indicative that the autocorrelation properties of our proposed signals are relatively good.

In addition, the sums (1) and (2) are the main ones for our estimation. When we compare two different P1-signals P1 _j and P1 _j ', the calculations for the above equations (4) and (5) are derived.

Remember, we operate under the assumption of β _j - β _j ≠ α . If n = 0 here, the sum is estimated to be 0 by the formula (1), and otherwise, we have two Gauss sums here, so by trigonometric inequality, we can estimate . In other words, at the discrete set of time errors, the cross-correlations are less than a perfect match of 256*K" of at least 7 db.

Furthermore, the sum (2) allows us to make a relatively sharp estimate of the envelope power at the sampling instants Δ t = n / (511 ^{Δ f} ) (for all n = 0, 1, ..., 510) . We have .

Since α ≠ β _j , we get zero at n=0, and because of the result of equation (2) about Gauss sum, the upper limit of the sum in the absolute value sign is . In summary, the envelope power sampled is thus at most one. Here, the total signal energy is 256, so the average power is . Therefore, at this (Nyquist) sampling rate, the maximum ratio to the average envelope power is . There is a big bound here, and the general bounds tell us that in the worst case, the ratio of the continuous peak to the average envelope power is (And in practice, it is very likely to be better).

One or more layers of the present invention can be implemented in computer executable instructions, such as one or more program modules executed by one or more computers or other devices. Generally, a program module includes routines, programs, objects, components, data structures, etc. that perform particular tasks or implement particular abstract data types when executed by a processor in a computer or other device. The computer executable instructions can be stored on a computer readable medium such as a hard disk, optical disk, removable storage medium, solid state memory, RAM, or the like. Will be in the art It will be understood by those of ordinary skill that, in various embodiments, the functions of the program modules can be combined or separated as desired. Moreover, the functionality may be implemented in whole or in part in firmware or hardware equivalents such as integrated circuits, field programmable gate arrays (FPGAs), application specific integrated circuits (ASICs), and the like.

Embodiments include any novel feature or combination of features disclosed herein, either explicitly or in any generalized form. While the embodiments have been described in terms of specific examples of the presently preferred embodiments of the present invention, it will be understood by those of ordinary skill in the art that there are many variations and modifications of the systems and techniques described above. Therefore, the spirit and scope of the invention should be construed broadly as set forth in the appended claims.

102‧‧‧Digital Broadband Broadcasting System

103‧‧‧Digital Broadcast Transmitter

104‧‧‧Digital content sources

105‧‧‧Digital Broadcasting Tower

112‧‧‧Mobile terminal or device, mobile device, receiving device

128‧‧‧ processor

130‧‧‧User interface

134‧‧‧ computer readable memory

136‧‧‧ display

140‧‧‧Software

141‧‧‧DVB Receiver

142‧‧‧FM/AM radio receiver

143‧‧‧WLAN transceiver

144‧‧‧Telecom transceiver

150‧‧‧Battery

152‧‧‧Speaker

154‧‧‧Antenna

602-614‧‧‧Steps

902-914‧‧‧Steps

1002-1014‧‧‧Steps

1102‧‧‧Guide symbol/first boot symbol

1104-1106‧‧‧Signal symbol

1202-1204‧‧‧Steps

1301‧‧‧P2-1 packet structure

1302-1324‧‧‧ Field

1325‧‧‧ time slot circuit field

1326-1338‧‧‧ Field

1402‧‧‧Send window

1404‧‧‧Offset

1502‧‧‧Offset

1602‧‧‧P2-n packet

1604-1610‧‧‧ Field

1702‧‧‧P2-1 packet

1704‧‧‧P2-n packet

1706‧‧‧‘T’ column

1708‧‧‧‘L’ column

1710‧‧‧‘E’ column

1712‧‧‧‘N’ column

1714‧‧‧L2 mailing information

1716‧‧‧Network and Honeycomb Information

1718‧‧‧Location

1720‧‧‧Network Information Sheet (NIT)

1722‧‧‧Service Description Table (SDT)

1724‧‧‧time slot_id

1726‧‧‧ Frame - Number

2002‧‧‧ frame

2004-2012‧‧ ‧ slot

2014-2024‧‧ OFDM symbol

2026-2034‧‧‧OFDM cells

Figure 1 illustrates a suitable digital broadband broadcast system in which one or more illustrative embodiments of the present invention may be implemented.

Figure 2 illustrates an example of a mobile device in accordance with one aspect of the present invention.

Figure 3 schematically illustrates an example of a honeycomb lattice in accordance with one aspect of the present invention, each of which may be covered by a different emitter.

Figure 4 shows a frame (a symbol for channel search and service discovery) and a frame and a frame of information according to one aspect of the present invention.

Figure 5 shows how a signal center frequency can coincide with a channel center frequency or with respect to a channel center frequency.

Figure 6 is a flow chart showing the steps performed by a receiver in accordance with at least one embodiment.

Figure 7 shows an example of the magnitude of a pilot signal bandwidth relative to a signal bandwidth and a channel raster bandwidth in accordance with one aspect of the present invention.

Figure 8 illustrates a sparse lead spacing of a pilot sequence of a pilot symbol in accordance with one aspect of the present invention.

Figure 9 is a flow chart showing the steps performed by a receiver to perform correlation in the frequency domain to detect the coarse offset being used.

Figure 10 is a flow diagram showing the steps associated with performing a service discovery in the time domain, in accordance with an embodiment.

Figure 11 shows an example of a sequence of pilot/telephone symbols in accordance with one aspect of the present invention.

Figure 12 is a flow diagram showing the steps of a method performed by a transmitter in accordance with at least one level of the present invention.

Figure 13 illustrates a packet structure in accordance with one aspect of the present invention.

Figure 14 illustrates a transmit window offset in accordance with one aspect of the present invention.

Figure 15 illustrates an offset between a current sub-signal and a next sub-signal of a frame in accordance with one aspect of the present invention.

Figure 16 illustrates an additional packet structure that can be used to carry signaling information in accordance with one aspect of the present invention.

Figure 17 illustrates an exemplary flow chart for use in service exploration in accordance with one aspect of the present invention.

Figures 18 and 19 depict the relationship between P1, P2 and DATA symbols in accordance with one aspect of the present invention.

Figure 20 shows an exemplary frame and a time slot structure comprising OFDM symbols and a honeycomb cell in accordance with one aspect of the present invention.

Figure 21 illustrates the coherence bandwidth and differential modulation within a pilot symbol.

Figure 22 depicts a microscopic relationship between two P1 symbols in accordance with one aspect of the present invention. The pitch changes.

Figure 23 shows two 1K symbols with a 1/1 guard interval and differential modulation between the symbols, according to an embodiment.

Figure 24 shows the calculation of the sum of energy received from one or more pilot symbols in accordance with one or more embodiments.

Figure 25 shows a transmitter in accordance with one or more embodiments.

Figure 26 shows a receiver in accordance with one or more embodiments.

Figure 27 is a flow diagram showing the steps that can be performed by a receiver in accordance with one or more embodiments.

Figure 28 is a diagram of the auto/cross correlation between the pilot sequences and their frequency offset versions in accordance with one or more embodiments.

Figure 29 is an enlarged version of the figure of Figure 28 showing the low cross-correlation range of the frequency offset.

Figure 30 is a diagram showing the envelope amplitude of a first pilot symbol signal in accordance with at least one embodiment.

Figure 31 is an enlarged version of the figure of Figure 30.

1702‧‧‧P2-1 packet

1704‧‧‧P2-n packet

1706‧‧‧‘T’ column

1708‧‧‧‘L’ column

1710‧‧‧‘E’ column

1712‧‧‧‘N’ column

1714‧‧‧L2 mailing information

1716‧‧‧Network and Honeycomb Information

1718‧‧‧Location

1720‧‧‧Network Information Sheet (NIT)

1722‧‧‧Service Description Table (SDT)

1724‧‧‧time slot_id

1726‧‧‧ Frame - Number

Claims (29)

A communication method, comprising the steps of: transmitting a first pilot Orthogonal Frequency Division Multiplexing (OFDM) symbol, the first pilot OFDM symbol comprising a plurality of sparsely distributed secondary carriers; and selecting a set of pseudo-random transmission sequences And the set of pseudo-random transmission sequences are grouped with signaling information provided in the plurality of sparsely distributed secondary carriers; and the differential binary phase shift keying is used by the group of pseudo-random transmission sequences The plurality of sparsely distributed secondary carriers are modulated such that the signaling information is provided in the plurality of sparsely distributed secondary carriers. The method of claim 1, wherein the plurality of sparsely distributed subcarriers are selected from a middle segment of a frequency band. The method of claim 1, wherein the sparse distribution of the plurality of sparsely distributed subcarriers forms an identifiable pattern, the identifiable pattern indicating a frequency shift. The method of claim 1, wherein the sequence of the pseudo-random transmission sequence is orthogonal to each other. The method of claim 1, wherein the sending information is a frequency offset information. A communication device comprising: a processor; and a memory storing executable instructions executable by the processor configurable to cause the device to perform at least the following actions: causing one of the first boots to be positive Cross frequency division multiplexing (OFDM) a number is transmitted, the first pilot OFDM symbol comprises a plurality of sparsely distributed secondary carriers; a set of pseudo-random transmission sequences are selected, the set of pseudo-random transmission sequences being combined to provide the plurality of sparse distributions Transmitting information in the activated subcarrier; and using the differential binary phase shift key to modulate the plurality of sparsely distributed secondary carriers by the set of pseudorandom transmission sequences, so that the signaling information is A plurality of sparsely distributed subcarriers are provided. The apparatus of claim 6, wherein the plurality of sparsely distributed subcarriers are selected from a middle of a frequency band. The apparatus of claim 6, wherein the sparse distribution of the plurality of sparsely distributed subcarriers forms an identifiable pattern, the identifiable pattern indicating a frequency shift. The apparatus of claim 6, wherein the sequence of the pseudo-random transmission sequence is orthogonal to each other. The apparatus of claim 6, wherein the first pilot OFDM symbol uses a 1k fast Fourier transform. The apparatus of claim 6, wherein the first pilot OFDM symbol substantially occupies half of the secondary carrier, and the half of the secondary carrier is derived from a middle of a frequency band. The device of claim 6, wherein the sending information is a frequency offset information. A communication method comprising the following steps: Receiving a first pilot Orthogonal Frequency Division Multiplexing (OFDM) symbol; performing a fast Fourier transform on the first pilot OFDM symbol; detecting a frequency offset at least in part by finding a shifted subcarrier pattern; And modulating the first pilot OFDM symbol by a processor to generate an estimated one of a pseudo-random binary sequence. The method of claim 13, wherein the step of receiving the first pilot OFDM symbol indicates the occurrence of a desired signal. The method of claim 13, wherein the power of the secondary carrier is used to detect the frequency offset. The method of claim 13, wherein the transmission sequence is decoded from a phase difference between adjacent subcarriers. A communication device comprising: a processor; and a memory storing executable instructions configurable by the processor to cause the device to perform at least the following actions: receiving a first boot positive An alternating frequency division multiplexing (OFDM) symbol; performing a fast Fourier transform on the first pilot OFDM symbol; detecting a frequency offset at least in part by finding a shifted subcarrier pattern; and microdecomposing the first A pilot OFDM symbol to produce an estimated A pseudo-random binary sequence of the calculation. The apparatus of claim 17, wherein the memory further stores executable instructions executable by the processor configurable to cause the apparatus to use the first pilot OFDM symbol as an occurrence of a desired signal. An indicator. The device of claim 18, wherein the memory further stores executable instructions executable by the processor configurable to cause the device to utilize the power of the secondary carrier to detect the frequency offset. The apparatus of claim 17, wherein the transmission sequence is decoded from a phase difference between adjacent subcarriers. An one or more computer readable medium containing computer executable instructions configured to cause an apparatus to perform at least the following steps: causing a first guided orthogonal frequency division multiplexing (OFDM) when the computer executable instructions are executed a symbol is transmitted, the first pilot OFDM symbol includes a plurality of sparsely distributed secondary carriers; and a set of pseudo-random transmission sequences are selected, the set of pseudo-random transmission sequences being combined to provide the plurality of sparse distributions Transmitting information in the activated subcarrier; and using the differential binary phase shift key to modulate the plurality of sparsely distributed secondary carriers by the set of pseudorandom transmission sequences, so that the signaling information is A plurality of sparsely distributed subcarriers are provided. The computer readable medium of claim 21, wherein the memory further stores executable instructions configured by the processor to cause the loading The plurality of subcarriers that are sparsely distributed are selected from a middle of a frequency band. The computer readable medium of claim 21, wherein the sparse distribution of the plurality of sparsely distributed subcarriers forms an identifiable pattern, the identifiable pattern indicating a frequency shift. The computer readable medium of claim 21, wherein the pseudo-random sequence of the transmission sequences is orthogonal to each other. The computer readable medium of claim 21, wherein the transmission information is a frequency offset information. An one or more computer readable medium containing computer executable instructions configured to cause an apparatus to perform at least the following steps: receiving a first pilot orthogonal frequency division multiplexing (OFDM) when the computer executable instructions are executed a symbol; performing a fast Fourier transform on the first pilot OFDM symbol; detecting a frequency offset at least in part by finding a shifted subcarrier pattern; and micro-decomposing the first pilot OFDM symbol to produce an estimate A pseudo-random binary sequence. The computer readable medium of claim 26, wherein the step of receiving the first pilot OFDM symbol indicates the occurrence of a desired signal. The computer readable medium of claim 26, wherein the transmission sequence is decoded from a phase difference between adjacent subcarriers. The computer readable medium of claim 26, wherein the secondary carrier The power is used to detect the frequency offset.