RU2435171C1 - Phase direction finding method and phase direction finder for implementing said method - Google Patents

Abstract

FIELD: physics. ^ SUBSTANCE: phase direction finder which realises the disclosed phase direction finding method has receiving antennae 1, 2.i (i=1, 2,Ç, n), two receivers, a reference generator, a pulse generator, an electronic switch, a 90 phase changer, three phase detectors, an indicator, a heterodyne, a mixer, an intermediate frequency amplifier, two multipliers, two band-pass filters, a delay line, a differentiator, a correlator, a control pulse generator, two switches, two phase doublers, two phase halvers, two narrow band-pass filters, a phase metre, a computing unit and a recording unit. ^ EFFECT: broader functionalities of the method and device through determination of the distance to a phase-shift keyed composite signal source. ^ 2 cl, 3 dwg

Description

The proposed method and device relates to the field of electronics and can be used to determine the location of radiation sources of complex signals.

Known phase methods of direction finding and phase direction finders (RF patents Nos. 2003131, 2006872, 2010258, 2012010, 2134429, 2155352, 2175770, 2290658; I.E. Kinkulkin et al. “Phase method for determining coordinates”. M .: Sov. Radio, 1979; “Space Radio Engineering Complexes. Edited by S. I. Bychkov. M.: Sov. Radio, 1967, pp. 13-134, Fig. 2.3.9 and others).

Of the known technical solutions closest to the proposed are "Phase direction finding method and phase direction finder for its implementation" (RF patent No. 2290658, G01S 3/46, 2005), which are selected as prototypes.

The known method and device provide accurate and unambiguous determination of the angular coordinate α of the radiation source of complex signals in the azimuthal plane, but do not allow to estimate the distance R to the radiation source of complex signals.

An object of the invention is to expand the functionality of technical solutions by determining the range to the radiation source of complex signals with phase shift keying.

The problem is solved in that the phase direction finding method, based, in accordance with the closest analogue, on receiving signals, amplifying and limiting them in amplitude, comparing signals transmitted through two channels in phase, while the signal of one of the channels is pre-shifted in phase by 90 °, set in the azimuthal plane of n receiving antennas around a circle of radius d with the possibility of their electronic rotation with an angular velocity Ω around a receiving antenna located in the center of the circle, receiving antennas are placed along of a circle alternating with a frequency Ω, the signal received by an antenna located in the center of the circle is frequency-converted, an intermediate frequency voltage is extracted, it is multiplied with signals alternately received by n receiving antennas located around a circle, a phase-modulated voltage is extracted, it is multiplied with a local oscillator voltage , isolate a low-frequency voltage with a frequency of Ω and compare it in phase with the reference voltage, forming an accurate, but ambiguous direction finding scale of the signal source, simultaneously Phase-modulated voltage is subjected to autocorrelation processing, a low-frequency voltage with a frequency Ω is extracted, it is compared in phase with a reference voltage, forming a rough but unambiguous direction finding scale for the signal radiation source, differs from the closest analogue in that they determine the derivative of the correlation function of signals received by antennas located on the same line of sight with the radiation source of the signal, for which the signal received by the antenna located in the center of the circle is previously differentiated comfort in time, they form a control impulse at the moment the derivative of the correlation function passes through zero, which is used to resolve the further processing of the received signals, during which the received signals are multiplied and phase divided by two, the harmonic oscillations are extracted, the phase difference between them is measured and the range is determined to the signal source.

The problem is solved in that a phase direction finder, comprising, in accordance with the closest analogue, a first receiving antenna, a first receiver, a mixer, a second input of which is connected to the local oscillator output, and an intermediate frequency amplifier, a reference oscillator, a pulse generator, electronic, connected in series a switch, n inputs of which are connected to the outputs of n receiving antennas arranged around a circle of radius d with the possibility of their electronic rotation around the first receiving antenna, in the center of the circle, and the second receiver, the first multiplier, the second input of which is connected to the output of the intermediate frequency amplifier, the first bandpass filter, the delay line, the second phase detector, the second input of which is connected to the output of the first bandpass filter, 90 ° phase shifter, the first phase a detector, the second input of which is connected to the second output of the reference oscillator, and an indicator, connected in series to the output of the first bandpass filter, a second multiplier, the second input of which is connected to the output of the local oscillator, W A second bandpass filter and a third phase detector, the second input of which is connected to the third output of the reference generator, and the output is connected to the second input of the indicator, differs from the closest analogue in that it is equipped with a differentiator, a correlator, a control pulse shaper, two keys, two phase doublers, two phase dividers into two, two narrow-band filters, a phase meter, a computing unit, and a recording unit, and a differentiator, a correlator, and a second the path of which is connected to the output of the second receiver, the driver pulse shaper, the first key, the second input of which is connected to the output of the first receiver, the first phase doubler, the first phase divider into two, the first narrow-band filter, phase meter, computer registration unit and registration unit, to the output of the driver a second pulse, the second input of which is connected to the output of the second receiver, the second phase doubler, the second phase divider into two and the second narrow-band filter, the output of which connected to the second input of the phase meter.

The structural diagram of the phase direction finder that implements the proposed phase direction finding method, is presented in figure 1. The relative position of the receiving antennas 1, 2.i and the source of radio emissions (IRI) on the same line of sight is shown in figure 2. The correlation function R (τ) and its derivative d ∗ R (τ) / d ∗ τ are shown in FIG. 3.

The phase direction finder comprises in series a first receiving antenna 1, a first receiver 3, a mixer 12, a second input of which is connected to the output of the local oscillator 11, an intermediate frequency amplifier 13, a first multiplier 14, a first band-pass filter 15, a delay line 16, a second phase detector 17, and a second the input of which is connected to the output of the first band-pass filter 15, the phase shifter 8 by 90 °, the first phase detector 9, the second input of which is connected to the second output of the reference generator 5, and the indicator 10. To the first output of the reference generator 5 after A pulse generator 6, an electronic switch 7, the n inputs of which are connected to the outputs of n receiving antennas 2.i (i = 1, 2, ..., n) placed on a circle of radius d with the possibility of their electronic rotation around the first receiving antenna 1, are carefully connected placed in the center of the circle, and a second receiver 4, the output of which is connected to the second input of the first multiplier 14. To the output of the first bandpass filter 15, a second multiplier 18 is connected in series, the second input of which is connected to the output of the local oscillator 11, the second bandpass filter 19 and three the second phase detector 20, the second input of which is connected to the third output of the reference generator 5, and the output is connected to the second input of the indicator 10. To the output of the first receiver 3, a differentiator 21, a correlator 22, the second input of which is connected to the output of the second receiver 4, are connected in series 23 a control pulse, the first key 24, the second input of which is connected to the output of the first receiver 3, the first phase doubler 26, the first phase divider 28 into two, the first narrow-band filter 30, phase meter 32, the computing unit 33 and the recording unit 34. A second key 25, the second input of which is connected to the output of the second receiver 4, the second phase doubler 27, the second phase divider 29 into two and the second narrow-band filter 31, the output of which is connected to the second input of the phase meter 32, is sequentially connected to the output of the control pulse generator 23.

The proposed method is implemented as follows.

Received complex signals, for example with phase shift keying (PSK):

U ₁ (t) = υ _s ∗ Cos [(ω _s ± Δω) t + φ _k (t) + φ _s ],

U ₂ (t) = υ _с ∗ Cos [(ω _c ± Δω) t + φ _k (t) + φc + 2π ∗ d / λ ∗ Cos (Ωt-α)], 0≤t≤T _C ,

where υ _с , ω _c , φ _c , T _C - amplitude, carrier frequency, initial phase and signal duration;

± Δω instability of the carrier frequency of the signal due to various destabilizing factors, including the Doppler effect;

φ _k (t) = {0, π} is the manipulated phase component that displays the law of phase manipulation in accordance with the modulating code, and φ _k (t) = const for k ∗ τ _e <t <(k + 1) ∗ τ _e and can change abruptly at t = k ∗ τ _e , i.e. at the borders between elementary premises (k = 1, 2, ..., N-1);

τ _e , N - the duration and number of chips that make up the signal of duration T _C (T _C = N ∗ τ _e );

d is the radius of the circle on which the receiving antennas are located 2.i (i = 1, 2, ..., n), (measuring base);

Ω is the speed of electronic rotation of the receiving antennas 2.i (i = 1, 2, ..., n) around the receiving antenna 1;

α - bearing (azimuth) to the signal source,

from the outputs of the receiving antennas 1, 2.i (i = 1, 2, ..., n) directly and through the electronic switch 7 are fed to the inputs of the receivers 3 and 4, and then to the first inputs of the mixer 12, multiplier 14, differentiator 21 and correlator 22 respectively. The second input of the mixer 12 from the output of the local oscillator 1 i receives voltage

U _g (t) = υ _g ∗ Cos (ω _g t + φ _g ),

where υ _g , ω _g , φ _g - amplitude, frequency and initial phase of the local oscillator voltage.

At the output of the mixer 12, voltages of combination frequencies are generated. The amplifier 13 is allocated the voltage of the intermediate (differential) frequency

U _pr (t) = υ _pr ∗ Cos [((ω _pr ± Δω) t + φ _k (t) + φ _pr ], 0≤t≤T _C ,

where υ _pr = 1/2 ∗ υ _s ∗ υ _g ;

ω _CR = ω _with -ω _g - intermediate (difference) frequency;

φ _CR = φ _s -φ _g ,

which is fed to the second input of the multiplier 14. At the output of the multiplier 14 is formed phase-modulated (FM) oscillation at a frequency ω _{g of the} local oscillator 11

U ₃ (t) = υ ₃ ∗ Cos [ω _g t + φ _g + 2π ∗ d / λ ∗ Cos (Ωt-α)], 0≤t≤T _C ,

where υ ₃ = 1/2 ∗ υ _с ∗ υ _pr ,

which is allocated by the band-pass filter 15 and supplied to the first inputs of the phase detector 17, delay line 16 and multiplier 18. The second input of the latter is supplied with the voltage U _g (t) of the local oscillator 11. A harmonic voltage is generated at the output of the multiplier 18

U ₄ (t) = υ ₄ ∗ Cos [2π ∗ d / λ ∗ Cos (Ωt-α], 0≤t≤T _C ,

where υ ₄ = 1/2 ∗ υ ₃ ∗ υ _g ,

which is allocated by the band-pass filter 19 and supplied to the first input of the phase detector 20. The reference voltage is applied to the second input of the phase detector 20 from the third output of the reference generator 5

U ₀ (t) = υ ₀ ∗ CosΩt.

A constant voltage is generated at the output of the phase detector 20

U _н1 (α) = υ _н1 ∗ Cosα,

where υ _н1 = 1/2 ∗ υ ₄ ∗ υ ₀ ,

which is fixed by indicator 10. Thus, a direction finding scale is formed, which is an accurate but ambiguous scale.

At the same time, the phase-modulated oscillation U ₃ (t) is subjected to autocorrelation processing using an autocorrelator consisting of a delay line 16 and a phase detector 17.

In phase-modulated voltage U ₃ (t) the value

mφ = 2π ∗ d / λ,

called the phase modulation index, characterizes the maximum value of the phase deviation from the zero value that occurs during the electronic rotation of the receiving antennas 2.i (i = 1, 2, ..., n) around the receiving antenna 1 (Fig.2).

Receiving antennas 2.i (i = 1, 2, ..., n) are switched alternately with a frequency Ω using an electronic switch 7 controlled by an n-phase pulse generator 6. The control pulses are generated by the 6 pulse generator from the harmonic voltage generated by the reference generator 5

U ₀ (t) = υ ₀ ∗ CosΩt.

However, for d / λ <1/2, ambiguity of the reference angle α occurs. The elimination of this ambiguity by reducing the d / λ ratio usually does not justify itself, since the main advantage of the wide-base direction finder is lost. In addition, in the range of meter and especially decimeter waves, it is often not possible to take small values of d / λ due to design considerations.

In connection with the stated reason, the problem arises of decreasing the phase modulation index without decreasing the relative size of the measuring base d / λ. This is achieved by autocorrelation processing of the phase-modulated voltage U ₃ (t) using the delay line 16 and the phase detector 17. Moreover, the delay time t of the delay line 16 is chosen so as to reduce the phase modulation index to a value

m _φ1 = 2π ∗ d ₁ / λ,

where d ₁ <d,

where the inequality holds

d ₁ / λ <1/2,

providing unambiguous direction finding of the signal source.

The output of the phase detector 17 produces a harmonic voltage

U ₅ (t) = υ ₅ ∗ Cos (Ωt-α), 0≤t≤T _C ,

where υ ₅ = 1/2 ∗ υ ₃ ² ,

which through the phase shifter 8 through 90 ° enters the first input of the phase detector 9, the second input of which from the third output of the reference generator 5 is supplied with a reference voltage U ₀ (t). The output of the phase detector 9 produces a constant voltage

U _n2 (α) = υ _n2 ∗ Sinα,

where υ _Н2 = 1/2 ∗ υ ₅ ∗ υ ₀ ,

which is fixed by indicator 10. So the direction finding scale is formed, which is a rough, but unambiguous scale.

To assess the distance R to the radiation source of the QPSK signals, it is necessary to fix the line of sight on which the stationary receiving antenna 1, one of the mobile antennas 2.i and the radio emission source (IRI) are located (Fig. 2), and also select from the QPSK signals received by the indicated antennas, harmonic oscillations. This can be done by removing phase manipulation from the received PSK signals. Having received two harmonic oscillations and evaluating their phase relationships, we can measure the distance R to the IRI.

The line of sight can be fixed using the correlator 22, to the two inputs of which signals U ₁ (t) and U ₂ (t) are supplied from the outputs of the first 3 and second 4 receivers. At the output of the correlator 22, a correlation function R (τ) is formed (Fig. 3, a). However, in the region of the maximum, the correlation function R (τ) has a small slope and changes insignificantly with changes in τ. Much more favorable for finding the maximum is the form of the derivative of the correlation function d ∗ R (τ) / dτ (Fig. 3, b). At the point τ = 0, the derivative has a significant steepness and, in addition, changes sign depending on the position relative to the point τ = 0.

Thus, finding the maximum of the correlation function (the maximum principle is an extreme problem) is engaged in the minimum principle - stabilization of the zero value of the controlled variable. The minimum method of the derivative of the correlation function (passing through zero), along with high accuracy and sensitivity, has another very significant advantage of the zero method, namely: the amplitude of the input signals and its fluctuations do not affect the measurement result.

The derivative d ∗ R (τ) / dτ is formed due to the inclusion of a differentiator 21 in the channel of the first signal. At the time the derivative passes through zero, the driver 23 generates a control pulse, which is supplied to the control inputs of the keys 24 and 25, opening them. In the initial state, keys 24 and 25 are always closed.

In this case, the QPSK signals from the outputs of receivers 3 and 4 through the public keys 24 and 25 are fed to the inputs of the doublers 26 and 27 of the phase, respectively, at the output of which harmonic voltages are generated:

U ₅ (t) = υ ₅ ∗ Cos [2 (ω _c ± Δω) t + 2φ _c ],

U ₆ (t) = υ ₅ ∗ Cos [2 (ω _c ± Δω) t + 2φ _∑ ], 0≤t≤T _C ,

where υ ₅ = 1/2 ∗ υ _c ² ;

2φ∑ = φ _c + 2π ∗ d / λ ∗ Cos (Ωt-α)];

2φ _k (1) = {0.2π}.

The indicated voltages arrive at the inputs of the phase dividers 28 and 29 into two, but the output of which produces harmonic voltages, respectively:

U ₇ (t) = υ ₇ ∗ Cos [(ω _c ± Δω) t + φ _c ],

U ₈ (t) = υ ₇ ∗ Cos [(ω _c ± Δω) t + φ _∑ ], 0≤t≤T _C.

These voltages are allocated by narrow-band filters 30, 31 and are fed to two inputs of the phasemeter 32, which measures the phase difference

Δφ = φ ₁ -φ ₂ = m ₁ Rm ₂ Rm ₂ d = R ∗ (m ₁ -m ₂ ) -m ₂ d,

where φ ₁ = m ₁ R is the phase shift of the electromagnetic wave during the propagation of the signal from the IRI to the antenna 2.i;

φ ₂ = m ₂ (R + d) - phase incursion at antenna 1;

m ₁ , m ₂ - wave numbers (m ₁ = 2π / λ ₁ , m ₂ = 2π / λ ₂ ).

In the computing unit 33 determines the range of R to Iran

R = Δφ + m ₂ d / m ₁ -m ₂ ,

which is fixed by the registration unit 34.

Thus, the proposed method in comparison with the prototype provides the fundamental possibility of determining the range to Iran. Knowing the range R and azimuth α, we can determine the location of the radiation source of the PSK signals. At the same time, the formation of the line of sight, on which the stationary receiving antenna 1, the mobile antenna 2.i, and the IRI radiation source are simultaneously located, is carried out using the derivative of the correlation function d ∗ R (τ) / dτ, which, along with high accuracy and sensitivity, has Another very significant advantage of the zero method, namely: the amplitude of the input signals and its fluctuations do not affect the measurement result.

Thus, the functionality of known technical solutions is expanded.

Claims (2)

1. The phase direction finding method, based on receiving signals, amplifying and limiting them in amplitude, comparing signals transmitted through two channels in phase, while the signal of one of the channels is pre-phase shifted by 90 ° in phase, set n receiving antennas in the azimuth plane circles of radius d with the possibility of their electronic rotation with an angular velocity Ω around a receiving antenna located in the center of the circle, receive antennas are arranged around the circle, alternately with frequency Ω, the signal received by the antenna is placed at the center of the circle, it is converted in frequency, the intermediate frequency voltage is isolated, it is multiplied with signals alternately received by n receiving antennas located around the circle, the phase-modulated voltage is isolated, it is multiplied with the local oscillator voltage, the low-frequency voltage is isolated with the frequency Ω and the phase is compared in phase with a reference voltage, forming an accurate but ambiguous direction finding scale of the signal radiation source, simultaneously phase-modulated voltage is subjected to autocorrelation processing They isolate a low-frequency voltage with a frequency Ω, compare it in phase with the reference voltage, forming a rough but unambiguous direction finding scale for the signal radiation source, characterized in that they determine the derivative of the correlation function of the signals received by antennas located on the same line of sight with the signal radiation source why the signal received by the antenna located in the center of the circle is previously differentiated by time, a control pulse is formed at the moment of passage of the derivative th function through zero, which is used to permit further processing of the received signals, in which process the received signals are multiplied and divided in phase two, isolated harmonic oscillations measured phase difference Δφ between them is determined and the range R to the signal radiation source by the formula:
R = Δφ + m ₂ d / m ₁ -m ₂ ,
m ₁ = 2π / λ ₁ , m ₂ = 2π / λ ₂ - wave numbers. 2. A phase direction finder comprising a first receiving antenna, a first receiver, a mixer, the second input of which is connected to the local oscillator output, and an intermediate frequency amplifier, a reference generator, a pulse generator, an electronic switch, n inputs of which are connected to the outputs of n receiving antennas, in series placed around a circle of radius d with the possibility of their electronic rotation around the first receiving antenna located in the center of the circle, the second receiver, the first multiplier, the second input which is connected to the output of the intermediate frequency amplifier, a first bandpass filter, a delay line, a second phase detector, a second input of which is connected to the output of the first bandpass filter, a 90 ° phase shifter, a first phase detector, the second input of which is connected to the second output of the reference oscillator, and an indicator connected in series to the output of the first bandpass filter, a second multiplier, the second input of which is connected to the local oscillator output, a second bandpass filter and a third phase detector, the second input of which is connected nen with the third output of the reference generator, and the output is connected to the second input of the indicator, characterized in that it is equipped with a differentiator, a correlator, a control pulse shaper, two keys, two phase doublers, two phase dividers into two, two narrow-band filters, a phase meter, and a computing unit and a registration unit, and a differentiator, a correlator, the second input of which is connected to the output of the second receiver, a driver of the control pulse, are first connected to the output of the first receiver the key, the second input of which is connected to the output of the first receiver, the first phase doubler, the first phase divider into two, the first narrow-band filter, the phase meter, the computing unit and the registration unit, the second key is connected in series to the output of the control pulse former, the second input of which is connected to the output of the second a receiver, a second phase doubler, a second phase divider into two and a second narrow-band filter, the output of which is connected to the second input of the phase meter, while the computing unit determines the distance R to and source of signal emission from the formula R = Δφ + m ₂ d / m ₁ -m ₂
where Δφ = φ ₁ -φ ₂ is the phase difference,
φ ₁ = m ₁ R is the phase shift of the electromagnetic wave during the propagation of the signal from the radio source to the antenna 2.i (i = 1, 2, ..., n),
φ ₂ = m ₂ (R + d) - phase incursion to antenna 1,
m ₁ = 2π / λ ₁ , m ₂ = 2π / λ ₂ - wave numbers.

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