RU2338219C1 - Method of target tracking and design of giant-pulse radiolocation station for method implementation - Google Patents

Abstract

FIELD: radiolocating equipment.

SUBSTANCE: after quadrature-phase detection of summary and difference signals, suppression of video frequencies is carried out, where target signal shifted by Doppler frequency is a priori unavailable (mirror video frequencies). Suppression is carried out by passage of quadratures of inlet video signal through appropriate phase-shifting circuits with singular amplification, phase-frequency characteristics of which in working range are shifted in respect to each other by 90 degrees, and summation of received signals.

EFFECT: provides the possibility for giant-pulse radiolocating station to track both target signal and noise source with simultaneous increase of communication potential by target signal.

5 cl, 8 dwg

Description

The present invention relates to radar technology and can be used in coherent pulsed radars using complex signals and a single-pulse direction-finding principle, designed to measure both the coordinates of the reflecting target and the angular position of noise interference sources.

Currently, the monopulse principle, complex signals, and Doppler filtering are widely used to track targets in range and angle.

A device is known for generating an angular error in a coherent pulse-Doppler monopulse radar when receiving a noise signal [1]. In this device, the sum and difference signals after the monopulse antenna are transferred to the intermediate frequency and after amplification at the intermediate frequency using the reference signal generator and two phase detectors, they are transferred to the video frequency to obtain real sum and difference signals. The received signals (u _∑ and u _Δ ) are digitized, using the discrete Fourier transform, the M-point spectrum of the complex signal x = u _∑ + ju _{Δ is} calculated, the real (Re) and imaginary (Im) parts of which are taken the real digitized total u _∑ and difference and u _Δ signals, respectively. From the spectral components of the found spectrum X, the spectra of the analytical total S _∑ and difference S _Δ signals in the frequency range i = 0 ... M / 2 are restored. Spectra S _∑ and S _Δ are used to calculate the angular error of the equal signal direction (RSN) relative to the direction of the noise source as the quotient of the mathematical expectation of the scalar products of the signals S _∑ and S _Δ for all spectral components for i = 0 ... M / 2 the mathematical expectation of power S _∑ at the same frequencies (the quadrature components are used as the coordinates of the vectors S _∑ and S _Δ ). To compensate for the non-identity of the amplitude-frequency characteristics (AFC) of the total and difference channels in the test test mode, a test signal is simulated by which the relative phase shift between the channels is estimated, which is taken into account when forming the final estimate of the angular error.

The disadvantage of this device is the ability to work only on a noise signal and the complexity of the calculations for evaluating the error signal, because the estimation is performed according to the results of calculating the mathematical expectations of the scalar products S _∑i and S _Δi and powers S _ii for all i-th spectral components from i = 0 to i = M / 2.

There is a method of target tracking, described in [2] and taken as a prototype, which allows to increase the noise immunity of the tracking of a reflecting target with respect to passive and active noise interference while increasing tracking accuracy. According to the method, after setting the axis of equal signal direction (RSN) in the direction of the intended target, according to the previous review, the reflected signal is received with the formation of the sum and difference signals at the output of the monopulse antenna, the received signals are amplified by low-noise amplifiers at the carrier frequency, converted to an intermediate frequency, amplified, transferred to the video frequency using quadrature-phase detection to obtain the quadrature components of each signal, in a given range At a range of distances, consistent filtering is performed. For each element of the range, multichannel Doppler filtering of the total signal is carried out, the power of the signals at the outputs of the Doppler filters is determined, threshold detection of signals, determination of the spectral width of the detected signals and comparison with the threshold for selection of the true signal from the interference reflected by a false target. After selecting a signal for auto tracking, the target coordinates (range and Doppler frequency) are determined and the target is tracked by Doppler frequency and range, the target is mapped angularly relative to the RSN as the sum of the products of the same quadrature components of the total and difference signals at the monitored Doppler frequency. The received error signal is used to control the position of the PCN of a single-pulse antenna (close the tracking system in angle, gated, as usual, with a range gate).

The disadvantage of this method, the prototype is the lack of suppression of specular video frequencies of a broadband noise signal after quadrature-phase detection. In view of this, the components of the input spectrum of the broadband noise signal at the positive and negative (mirror) video frequencies overlap each other, respectively, it is impossible to correctly evaluate both the quadrature components of the total and difference signals, and the signal of the angular mismatch of the direction of the RSN to the noise interference source. In addition, due to the absence of suppression of specular video frequencies, the noise band of the receiver is 2 times wider than the optimum, respectively, the communication potential when receiving target signals is less than the calculated by 3 dB.

The aim of the invention is to enable tracking of both the target signal and the noise source of interference while increasing the communication potential of the target signal by suppressing the mirror noise band at the video frequency.

To solve the problem in the target tracking method [2], including:

setting the RSN axis of the monopulse antenna in the direction of the intended target according to the previous review;

emission of pulsed coherent signals in a given direction;

receiving high-frequency signals in the interval between probing;

sum-difference conversion of received signals, their superheterodyne conversion to an intermediate frequency;

amplification of the total and difference signals at an intermediate frequency, the conversion of signal spectra into the region of video frequencies by means of quadrature-phase detection using reference oscillations f _o , the frequency of which is equal to the intermediate frequency;

, в диапазоне дальностей,

,coordinated filtering of total and difference signals with obtaining M-point time realizations of analytical total v _∑ij and difference v _Δij signals at each j- _th repetition period,

, in the range of ranges,

,

multichannel Doppler filtering of the complex envelope of the pulse sequence of the total signal for each range element in a given interval in the Doppler frequency range from minus 1 / 2T _p to 1 / 2T _p with a band ΔF and the number of channels N = 1 / T _p ΔF, where T _p is the period repetition of probe pulses, ΔF is the width of the spectrum of inter-period fluctuations of signals from true targets

Z _∑in = F (v _∑ij ),

where i is the number of the range element,

j is the number of the repetition period,

,n is the number of the Doppler filter,

,

F is the Fourier transform operator;

calculation of the amplitude of the spectral components of the total signal

,

,

, М_A≤М;detection of signals for which P _∑in exceeds the detection threshold in the a priori range range,

, M _A ≤ M;

search for the target signal with the spectrum width ΔF _D less than the threshold Δ _threshold , including the calculation of the spectral width ΔF _{D of the} total signal in decreasing order of their amplitudes in range, comparing the current calculated value of the spectral width ΔF _D with a given threshold value Δ _threshold , when it is exceeded, a decision is made about detection of a false target, cyclic repetition of the calculation of the spectrum width of the signal following the amplitude, until the target signal with the spectrum width is less than the threshold;

determination of the number of the Doppler filter n _c with the maximum amplitude of the target spectrum;

determining a frequency mismatch signal, closing the frequency tracking loop, adjusting the frequency mismatch signal to position n _{c the} tracking strobe in frequency;

narrow-band filtering with a bandwidth ΔF = 1 / NT _{n of the} complex envelope of the pulse sequences of signals in the total and difference channels at the tracked n _C Doppler frequency to obtain complex spectral components of the total Z _∑ (i, n _c ) and difference Z _Δ (i, n _c ) signals;

selection of the amplitude envelope of the signals in the total channel

;

;

the selection of the error signal in range in the total channel, the closure of the range tracking loop and this signal adjusts the position (i _C ) of the tracking range gate;

calculating the angular error γ signal (i _c, n _c) PCH relative to the direction to the target based on the results of calculating the spectral components of the sum Z _Σ (i _c, n _c) and the difference Z _Δ (i _u, n _i) signals at the tracking point of the scene distance - Doppler frequency (i _C , n _C );

closing the tracking contour in the angle according to the signal of the angular mismatch γ (i _c , n _c ) with adjusting the position of the RSN of the single-pulse antenna;

characterized in that after quadrature-phase detection, the band of specular video frequencies is suppressed, where a priori there is no target signal shifted by the Doppler frequency, the received actual sum-difference signals are digitized with a sampling frequency F _in exceeding the frequency band occupied by the modulating signal by more than two times;

before starting the search for the target signal after the threshold detection, the number of detected signals w is determined; if during the cyclic search of the target signal for w cycles no target signal is detected, the following is performed:

, где нет отраженных сигналов с получением комплексных спектров суммарных сигналов Z_∑in и амплитуд спектральных составляющих суммарного сигнала

;multichannel Doppler filtering of the total signals in the frequency range from zero to 1 / 2T _p at ranges

where there are no reflected signals to obtain complex spectra of the total signals Z _∑in and the amplitudes of the spectral components of the total signal

;

threshold detection of noise interference source signals by amplitudes P _{∑in of} spectral components that exceed the detection threshold of the noise source;

finding the extreme points N _e (i _e, n _e) from the maximum amplitude spectral component P _Σin, with calculation for their spectral components of the difference signal _Δ Z (i, n);

calculation of the RSN mismatch signal relative to the direction of the noise interference source γ _un as the average value of the angular mismatches γ _e for N _e extreme points (i _e , n _e ) from the quadrature spectral components of the total Z _∑ (i _e , n _e ) and difference Z _Δ ( i _e, n _e) signals;

closing the tracking system in an angle with pointing the axis of the antenna according to the signal of the angular mismatch of the RSN antenna to the source of noise interference;

suppression of the band of mirrored video frequencies is performed by converting a pair of quadrature video signals into a new pair of signals, further shifted 90 ° relative to each other in the frequency band occupied by the modulation of the probing signal ΔF _c , and their subsequent addition;

, m-номер частоты спектральной составляющей, восстановление спектров аналитических суммарных S_∑mj и разностных S_Δmj сигналов по спектрам Х_mj, перемножение спектров аналитических суммарных и разностных сигналов с комплексно сопряженным спектром зондирующего сигнала S*_m с получением спектров сжатых сигналов V_∑mj=S_∑mjS*_m и V_Δmj=S_ΔmjS*_m, обратное М-точечное преобразование Фурье с получением для каждого j-того периода повторения М-точечных аналитических временных реализаций сжатых суммарных v_∑ij и разностных v_Δij сигналов.matched filtering in each repetition period is performed by a sequence of operations including the M-point Fourier transform of the sequences of complex numbers x _ij = u _∑ji + ju _Δij , the real part of which is equal to the digitized samples of the total signal u _∑ij , and the imaginary - to the samples of the difference signal u _Δij , obtaining M-point spectra X _mj = F (x _ij ),

, the m-number of the frequency of the spectral component, the restoration of the spectra of the analytical total S _∑mj and the difference S _Δmj signals from the spectra X _mj , the multiplication of the spectra of the analytical total and difference signals with the complex conjugate spectrum of the probe signal S * _m to obtain the spectra of the compressed signals V _∑mj = S _∑mj S * _m and V _Δmj = S _Δmj S * _m , the inverse M-point Fourier transform with obtaining for each j- _th repetition period M-point analytical time realizations of the compressed total v _∑ij and difference v _Δij signals.

в диапазоне дальностей

, включающим участок априорного нахождения отраженного сигнала и участок, где нет отраженного сигнала.According to the proposed method, monopulse total-differential reception of reflected signals is performed in the direction set according to the data of the previous review. After amplification and conversion of the received sum-difference signals to an intermediate frequency, they are transferred to the video frequency by quadrature-phase detection using reference oscillations whose frequency is equal to the intermediate frequency. The band of specular video frequencies is suppressed, where a priori there is no target signal shifted by the Doppler frequency by converting quadrature video signals into a new pair of signals, further shifted 90 ° relative to each other in the frequency band occupied by the modulation of the probe signal ΔF _c , and their subsequent addition. The obtained actual sum-difference signals are sampled with _a sampling frequency F> 2ΔF _c and further subjected to matched filtering to obtain an M-point time realizations analytical summary v _Σij v _Δij and difference signals at each j-the period of repetition,

in range

including a portion of the a priori location of the reflected signal and a section where there is no reflected signal.

, m-номер частоты спектральной составляющей, восстановление спектров аналитических суммарных S_∑mj и разностных S_Δmj сигналов по спектрам X_mj [3]:Consistent filtering includes the M-point Fourier transform of the sequences of complex numbers x _ij = u _∑ij + ju _Δij , the real part of which is equal to the digitized samples of the total signal u _∑ij , and the imaginary is equal to the samples of the difference signal u _Δij , with obtaining M-point spectra X _mj = F (x _ij ),

, m-number of the frequency of the spectral component, restoration of the spectra of the analytical total S _∑mj and difference S _Δmj signals from the spectra X _mj [3]:

,Re (S _∑mj ) = Re [X _mj + X _{Mm, j} ] / 2, for frequencies

,

Im (S _∑mj ) = Im [X _mj -X _{Mm, j} ] / 2, for frequencies

,Re (S _Δ mj) = Im [X _mj + X _{Mm, j} ] / 2, for frequencies

,

,Im (S _Δmj ) = - Re [X _mj -X _{Mm, j} ] / 2, for frequencies

,

Re (S _{∑ 0j} ) = Re (X _0j ), for the frequency m = 0

Im (S _∑0j ) = Re (S _{∑m / 2, j} ), for the frequency m = 0

Re (S _Δ0j ) = Im (X _0j ), for the frequency m = 0

Im (S _Δ0j ) = Re (S _{ΔM / 2j} ), for the frequency m = 0

S _∑mj = S _Δmj = 0, for frequencies m≥M / 2,

the multiplication of the spectra of analytical total and difference signals with a complex conjugate spectrum of the probe signal S * _m to obtain the spectra of compressed signals:

V _∑mj = S _∑mj S * _m and V _Δmj = S _Δmj S * _m ,

the inverse M-point Fourier transform with obtaining for each j-th repetition period M-point analytical time realizations of the compressed total v _∑ij and difference v _Δij signals,

, М_A≤М, осуществляют многоканальную доплеровскую фильтрацию комплексной огибающей импульсной последовательности суммарного сигнала в диапазоне частот Доплера от минус 1/2 Т_п до 1/2 Т_п с полосой ΔF и числом каналов N=1/Т_пΔF, где Т_п - период повторения зондирующих импульсов, ΔF - ширина спектра межпериодных флюктуаций сигналов от истинных целейfor each ith range element in the a priori range range

, M _A ≤M, carry out multichannel Doppler filtering of the complex envelope of the pulse sequence of the total signal in the Doppler frequency range from minus 1/2 T _p to 1/2 T _p with a band ΔF and the number of channels N = 1 / T _p ΔF, where T _p is the probe pulse repetition period, ΔF is the width of the spectrum of inter-period fluctuations of signals from true targets

Z _∑in = F (v _∑ij ),

where i is the number of the range cell,

n is the number of the Doppler filter,

j is the number of the repetition period,

F is the Fourier transform operator,

the amplitudes of the spectral components of the total signal are calculated

,

,

form a detection threshold at each point of the operating range, the range is the Doppler frequency according to the average level of interference in a moving window in range and Doppler frequency [4, p. 107], the center of which is the analyzed point (i, n)

where 2k _i is the size of the sliding window in range,

k ₀ is a factor depending on the probability of correct detection and the probability of false alarm,

perform the detection of signals in the working range of the amplitudes of the total spectral component P _∑in exceeding the detection threshold P _in , find the number of ranges w at which the detection threshold is exceeded, calculate the width of the Doppler spectra ΔF _{D of the} detected total signals in decreasing order of their amplitudes in range compare the current calculated value of the width of the spectrum ΔF _D with a predetermined threshold value Δ _threshold , when exceeded which they decide to detect a false target, calculate the number o for the analysis of the signals w: = w-1 and, for w> 0, the calculation of the spectrum width of the next signal with the amplitude amplitude until the target signal with the spectrum width is smaller than the threshold Δ _threshold is cyclically determined, the number of the target Doppler filter n _c is determined by the position of the maximum of the target signal spectrum, determining the frequency mismatch signal, closing the frequency tracking loop and adjusting the frequency mismatch signal to the position n _{c of the} tracking gate in frequency; filtering performed narrow- banded with bandwidth ΔF _f = 1 / NT _n complex envelope of pulse sequences of signals in sum and difference channels n to _n -th tracking Doppler frequency, amplitude envelope signal calculated in the total channel _Σints P, is calculated by the error signal range in total channel close the range tracking loop and adjust the position (i _c ) of the range tracking gate with this signal, calculate the angular mismatch signal γ (i _c , n _c ) of the RSN relative to the direction to the target based on the result tatam calculation of the spectral components of the total and difference signals at the tracked point range - frequency (i _C , n _C )

,

,

Short circuit tracking the angle of the angular error signal γ _u adjusted position PCH monopulse antenna.

за w циклов не обнаружен сигнал цели, производится:If during a cyclic search for target coordinates in the a priori range range

for w cycles, no target signal was detected;

, с получением комплексных спектров суммарных сигналов Z_Σin и амплитуд спектральных составляющих суммарного сигнала

;multichannel Doppler filtering of the total signals in the frequency range from zero to 1 / 2T _p at ranges where there are no reflected signals,

, with obtaining the complex spectra of the total signals Z _Σin and the amplitudes of the spectral components of the total signal

;

threshold detection of signals from the noise source by exceeding the amplitude of the spectral component P _{∑in the} noise detection threshold;

finding among signals that exceeded the detection noise threshold, N _e extreme signals, with a maximum spectrum amplitude P _∑in and their coordinates (i _e , n _e ),

calculating for N _{e the} found points of the spectral components of the difference signal Z _Δ (i _e , n _e ),

, по квадратурным спектральным составляющим суммарного Z_∑k и разностного Z_Δk сигналовcalculation of signals of the angular mismatch of RSN γ _k for N _e extreme points,

, by quadrature spectral components of the total Z _∑k and difference Z _Δk signals

,

,

calculation of the angular mismatch of RSN to the noise signal source as the average value of the angular mismatch of extreme points

,

,

the tracking system closes in angle according to the signal of the angular mismatch γ _IP with the correction of the position of the monopulse antenna to the source of noise interference.

The invention is illustrated by the further description and drawings of a monopulse radar that implements this method: The prototype for it, as well as for the inventive method, is the radar shown in [2].

Figure 1 - structural diagram of a monopulse radar;

Figure 2 - structural diagram of the antenna system 1;

Figure 3 is a structural diagram of a transmitter 3;

Figure 4 - structural diagram of the driver of the reference signal 7;

5 is a structural diagram of a high-frequency receiver 9 (10);

6 is a structural diagram of a block suppression mirror channel 13 (14);

7 - the algorithm of the computer 17 for the formation of compressed signals;

Fig. 8 is a flow chart of a digital computer 17 for selecting a target for escort, transition to escort and operations during escort.

Figure 1 presents the structural diagram of a monopulse radar, where the following notation:

1 - monopulse antenna system (AC), a variant of which is shown in figure 2;

2 - antenna switch (AP), can be made in the form of a three-arm ferrite Y-circulator;

3 - transmitter (PRD), a variant of the transmitter shown in figure 3;

4 - pathogen (B), a set of quartz oscillator with frequency multipliers, performed according to the scheme of frequency multipliers, including using loops of digital phase locked loop (PLL) [15, p. 35, Fig. 11; 16, p. 63, fig. 3.2];

5 - synchronizer (CHX) represents a combination of code - period converters, code - duration and digital-to-analog converters;

6 - modulator of the modulated signal (FMS), as the former can be a generator controlled by voltage frequency;

7 - driver signal reference (FOS), a structural diagram of which is shown in figure 4;

8 - switch (P), a single-pole switch for two positions [9] can be used as a switch, a matched load is connected to the free output of the switch, which is not shown;

9 - high-frequency receiver of the total signal (RFL-∑), the structure of the receiver is shown in figure 5;

10 - high-frequency receiver of the differential signal (Vpr-Δ), similar to the high-frequency receiver of the total channel 9;

11 - quadrature-phase detector of the total signal (KVFd-∑), an example may be a quadrature mixer [7];

12 - quadrature-phase detector of the difference signal (KVFd-Δ), similar to the quadrature-phase detector 11;

13 - block suppression of the mirror channel of the total channel (BPZK-∑), the block diagram of the block is shown in Fig.6;

14 - block suppression of the mirror channel of the differential channel (BPC-Δ), similar to block 13;

15 - video amplifier total channel (VU-∑);

16 - video amplifier differential channel (VU-Δ);

17 - an on-board computer (BTsVM), as an option may be a BTsVM [12].

In the diagram of figure 1, the radar station contains a series-connected pathogen 4, a transmitter 3, an antenna switch 2, a monopulse antenna system 1, a second output of the pathogen 4 through a series-connected synchronizer 5, a modulated signal shaper 6 and a reference signal shaper 7 connected to the third input of the onboard computer machines (BTsVM) 17, the third output of the antenna switch 2 through a series-connected high-frequency receiver of the total signal 9, a quadrature phase detector sum the signal 11, the suppression unit of the mirror channel of the sum channel 13 and the video amplifier of the sum signal 15 are connected to the fourth input of the digital computer 17, the output of the switch 8 through the series-connected high-frequency receiver of the difference signal 10, the quadrature phase detector of the difference signal 12, the block suppression of the mirror channel of the difference channel 14 and the video amplifier of the differential signal 16 is connected to the fifth input of the BTsVM 17, the first input-output of the BTsVM 17 is connected to the third inputs of the high-frequency receivers total 9 and differential about 10 channels, the third input of switch 8, the fourth input-output of the monopulse antenna system 1, the fourth input of the transmitter 3, the input of the exciter 4, and the first input of the synchronizer 5, the second and third outputs of the monopulse antenna system 1 are connected to the first and second inputs of the switch 8, the first output of the exciter 4 is connected to the second inputs of the high-frequency receivers of the total 9 and differential 10 signals, the output of the modulator of the signal 6 is connected to the third input of the transmitter 3, the second output of the synchronizer 5 is connected with the inputs of the modulator 6 and transmitter 3 of the same name, the third output of the pathogen 4 is connected to the first input of the driver of the reference signal 7, the first inputs of the quadrature phase detectors 11 and 12, the third output of the synchronizer 5 is connected to the second inputs of the video amplifiers 15 and 16, the second outputs of the quadrature phase detectors 11 and 12 are connected to the same inputs of the suppression units of the mirror channel 13 and 14, respectively, the fourth output of the synchronizer 5 is connected to the sixth input of the computer 17, the second input is the output of which Xia input - output of the radar.

Figure 2 presents a variant of a monopulse antenna system 2 based on a two-mirror antenna [10], where the following notation:

18 - parabolic mirror;

19 - twist-reflector;

20 - sum-difference converter (SRP), which can be built on the basis of waveguide T-bridges;

21 ₁ ... 21 ₄ - irradiators;

22 - gyrostabilizer (GS), an example of which can be a gyrostabilizer [11].

Monopulse antenna system 1, shown in figure 2, contains a parabolic mirror 18, a movable mirror (twist reflector) 19, mechanically connected to the platform of the gyrostabilizer 22, four irradiators 21 ₁ ... 21 _{4 of the} monopulse antenna system 1 are connected to the same inputs and outputs in total -differential converter 20, the fifth input-output of which is the first total input-output of a monopulse antenna system 1, the sixth and seventh outputs of the total-differential converter 20 are the second (azimuthal), third (angle natural) difference outputs of the monopulse antenna system 1, respectively, the information input-output of the gyrostabilizer 22 is the fourth input-output of the monopulse antenna system 1.

Figure 3 shows the structural diagram of the transmitter 3, which adopted the following notation:

23 - the first mixer (CM1) can be performed as a balanced mixer according to the scheme [13, p.144];

24 - the first band-pass filter (PF1);

25 - power amplifier (UM) can be implemented depending on the required power and the band of amplified frequencies based on an amplitron, a traveling wave lamp or a semiconductor device [13, pp. 19-20, 25, 42-46].

In the transmitter 3 circuit shown in FIG. 3, a first mixer 23, a first bandpass filter 24 and a power amplifier 25 are connected in series, the output of which is the output of the transmitter 3, the first input of the first mixer 23 is the third input of the transmitter 3, the second input of the power amplifier 25 is the fourth input of the transmitter 3, the second input of the transmitter 3 is connected to the second input of the first band-pass filter 24 and the third input of the power amplifier 25, the first input of the transmitter 3 is the second input of the first mixer 23.

Figure 4 shows the structural diagram of the driver of the reference signal 8, which adopted the following notation:

26 - second mixer (CM2);

27 - second band-pass filter (PF2);

28 - the first video amplifier (VU1).

In the diagram of the driver of the reference signal 7, shown in Fig. 4, the output of the second mixer 26 is connected through a series-connected second band-pass filter 27 and the first video amplifier 28 to the output of the driver of the reference signal 7, the first and second inputs of the second mixer 26 are the same inputs of the driver of the reference signal 7 .

Figure 5 shows the structural diagram of high-frequency receivers (RF) 9 and 10, where the following notation:

29 - limiter (Ogre);

30 - high frequency amplifier (UHF);

31 - mixer with suppression of the mirror channel (SMPZK), an example of the mixer is the mixer shown in [13, Fig. 36 p.144];

32 - a filter on surface acoustic waves (SAW-F), an example of which may be the filter shown in [8];

33 - intermediate frequency amplifier (IFA).

An example of a module in which a limiter 29, a high-frequency amplifier 30, a mixer with suppression of the mirror channel 31 are combined is a module [14].

In the diagram of the high-frequency receiver 9, shown in Fig. 5, a limiter 29, a high-frequency amplifier 30, a mixer with suppression of the mirror channel 31, a SAW filter 32 and an intermediate-frequency amplifier (UPCH) 33 are connected in series, the output of which is the output of the high-frequency receiver 9, the input of the limiter 29 is the first input of the high-frequency receiver 9, the second input of the mixer with suppression of the mirror channel 31 is the second input of the high-frequency receiver 9, the second input of the amplifier 33 is the third input of the high-frequency capacity 9.

Figure 6 shows the structural diagram of the suppression blocks of the mirror channel 13, 14, which adopted the following notation:

34 and 35 - the first and second phase-shifting circuit (FSC 1 and FSC 2),

36 - adder (Sum).

In the diagram of the suppression units of the mirror channel 13 (14) shown in Fig. 6, the outputs of the first 34 and second 35 phase-shifting circuits are connected to the first and second inputs of the adder 36, respectively, the output of which is the output of the suppression unit of the mirror channel 13, the first and second inputs of the block suppression of the mirror channel 13 are the inputs of the first 34 and second 35 phase-shifting circuits, respectively.

In accordance with the diagrams of FIGS. 1-8, a single-pulse radar that implements the proposed method works as follows.

In the initial state, the radar is in standby mode, while the power is supplied to the minimum required number of radar units, including the BCVM 17. The radar is activated by commands from an external control system by the inclusion commands with the issuance of the corresponding flight missions arriving at the second input of the BCVM 17. V in both cases, the digital computer 17 includes power to all radar units except the output power amplifier 25 in the transmitter 3 (figure 3). To this end, the corresponding command comes to the fourth input of the transmitter 3 from the digital computer 17. Next, the digital computer 17 performs calculations of the radiation parameters (repetition period, duration of the probe pulse, the law of intrapulse modulation, the initial values of the signal amplification level in high-frequency receivers (9 and 10) and in video amplifiers (15 and 16) depending on the a priori distance to a given target and its reflecting characteristics), after that the BCVM 17 through the control bus from the first input - output, enters the calculated settings into the radar units, including:

setting the initial signal amplification in high-frequency receivers of total 9 and differential 10 signals (RF AGC signal),

connecting the azimuthal difference signal from the second output of the monopulse antenna system 1 through the switch 8 to the first input of the high-frequency receiver of the difference signal 10,

introduction to the monopulse antenna system 1 (hereinafter referred to as antenna system 1) for the fourth input of the codes of the speeds of the axis of sight in azimuth and elevation. These codes are transmitted to the gyrostabilizer 22 (figure 2), with which the axis of sight of the antenna system is sent to the desired start of the scanning sector. The current coordinates of the axis of the antenna system in azimuth and elevation angle (ε, β) are issued from the gyrostabilizer 22 to the BCM 17 at its request from the fourth input - the output of the antenna system 1. When the specified start of the scanning sector (ε ₀ , β ₀ ) of the BCM 17 is reached gyro stabilizer 22 stops the axis of the antenna system 1,

the introduction into the pathogen of 4 codes for setting the local oscillator frequency f _g at the first output and the reference frequency f _o at the third. The frequency f _o is equal to the intermediate f _pr and is used in quadrature-phase detectors 11 and 12 when transferring the received total and difference signals from the intermediate frequency to the video frequency,

introducing into the synchronizer 5 parameters of the pulse of the amplitude modulation of the probe pulses (repetition period T _p and duration τ at the second output of the synchronizer 5), a random code of intrapulse frequency manipulation of the probe signal transmitted through a digital-to-analog converter in the form of a U _FM voltage to the first input of the modulated signal generator 6, the gain control code of the signal in video amplifiers 15 and 16, converted to an analog (AGC VU signal), issued from the third output of the synchronizer 5.

After preparatory operations, the pathogen 4 generates a reference frequency f _o equal to the intermediate frequency supplied to the quadrature-phase detectors 11, 12, the local oscillator frequency f _g supplied to the transmitter 3 and high-frequency receivers of the total 9 and differential 10 signal, quartz frequency f _square on the second the output fed to the synchronizer 5. The frequency f _{kv is} used in the synchronizer 5 in the formation of pulses of amplitude modulation U _AM digitally. Pulses of amplitude modulation from the second output of the synchronizer 5 are fed to the second inputs of the shaper of the modulated signal 6 and the transmitter 3 to modulate the duration of the signals generated by them. At the fourth output of the synchronizer 5, a quartz sampling frequency f _{in is formed} , which is used in the digital computer 17 to synchronize the sampling times received and converted to the video frequency of the signals from the outputs of the video amplifiers 15 and 16. If there are pulses of amplitude and frequency modulation at the first and second outputs of the synchronizer 5, the modulator signal 6 generates a frequency-manipulated signal at an intermediate frequency f _pr receiver arriving at the third input of the transmitter 3.

The subsequent work of the radar includes reconnaissance, target selection, capture and tracking of the selected target. The computer 17 according to the laid program includes the scanning mode of the antenna system 1 in azimuth in a given angular sector. For this purpose, the BCMC 17 introduces into the gyrostabilizer 22 of the antenna system 1 control signals in sign and scanning speed, turns on the transmitter 3, by giving a turn-on command to its fourth input. From this moment, a powerful signal is generated at the transmitter output (Fig. 3) at a carrier frequency obtained by mixing the local oscillator frequency arriving at the first input of the first mixer 23 with a frequency-manipulated pulse coming from the modulated signal generator 6 to the first input of the first mixer 23 at intermediate frequency. The tuning frequency of the first band-pass filter 24 (its attenuation at the carrier frequency) is controlled by an amplitude modulation pulse coming from the synchronizer 5 to the second input of the first band-pass filter 24, while in the pauses between the probe pulses, its attenuation is maximized, suppressing a continuous stray signal at the output of the power amplifier 25 to an acceptable level. The output signal of the transmitter 3 through the antenna switch 2 enters the antenna system 1 and is emitted in the scanned direction. Antenna system 1 receives a reflected signal and forms a sum and two difference signals (azimuthal at the second output and angular at the third output). Depending on the measured angular coordinate (azimuthal or elevation), according to the BCVM command 17, which is input to switch 8 through the third input, either the azimuthal or angular difference signal arrives at the high-frequency receiver of the difference channel 10 with the antenna of system 1. The total signal from the first output of the antenna system 1 passes through the antenna switch 2 to the high-frequency receiver of the total signal 9. Further processing of the total and difference signals before they arrive on the digital computer 17 is carried out by similar channels consisting of high-frequency receivers 9 and 10, quadrature-phase detectors 11 and 12, the suppression units of the mirror channel 13 and 14 and the video amplifiers 15 and 16. In this case, in the high-frequency receivers 9 and 10 (Fig. 5), the received signals are converted to an intermediate frequency and gain, in quadrature-phase detectors 11 and 12 receive quadrature sum and difference signals at the video frequency, in the block suppression of the mirror channel 13 and 14 (Fig.6) receive real single-cavity signals, which are further amplified in video amplifiers 15 and 16. The frequency of the local oscillator f _g to the second the inputs of high-frequency receivers 9 and 10 comes from the first output of the pathogen 4. To obtain the quadrature of the received signal at the video frequency, the reference frequency f _o comes to the first inputs of the quadrature-phase detectors 11 and 12 from the pathogen 4. The amplification level of single-cavity signals in video amplifiers 15 and 16, as well as in high-frequency receivers 9 and 10, is regulated by the BCM 17 based on the analysis of the level of the total video signal at the output of the video amplifier 15. In this case, the adjustment code of the video amplifiers 15 and 16 is fed to the BCM 17 to the synchronizer 5, which converts it to the analog voltage of the AGC VU supplied to the second inputs of the video amplifiers 15 and 16. The output total and differential video signals from the outputs of the video amplifiers of the total 15 and 16 differential channels are fed to the fourth and fifth analogs stems inputs onboard computer 17, which is digitized, stored and processed.

The processing of signals received during scanning of the axis of the antenna system in the working sector of the angles includes for the current position of the axis of the antenna system a consistent filtering of signals at each repetition period and the subsequent multi-channel Doppler for each resolved range element in a given a priori range range. The algorithm for coordinated filtering of the total and differential signals is shown in Fig.7. A feature of intra-period matched filtering is that in each period, as a reference u _op (i, j), the signal from the output of the signal modulator 6 is used, which passed through the reference signal generator 7 to the third analog input of the digital computer 17, digitized and stored. The spectrum of the reference analytical signal S (m, j) is obtained using the M point fast Fourier transform (FFT) from u _op (i, j) in accordance with the expression:

,

,

where F is the Fourier transform operator.

и углов сканирования β с получением комплексных спектров Z_∑in(β) и амплитуд Р_∑in(β). Далее для каждого положения β вычисляется порог обнаружения П_inβ (поз.44 фиг.8) по среднему уровню помех в скользящем окне по дальности, центром которого является анализируемая точка (i,n,β) [4, с.107]The algorithm for the subsequent search for the signal, the selection of the target signal, its capture and tracking is shown in Fig. 8. First (pos. 43 of Fig. 8), Doppler filtering of the compressed total signals v _∑ij (β) is _performed in the working range

and scanning angles β to obtain complex spectra Z _∑in (β) and amplitudes P _∑in (β). Next, for each position β, the detection threshold P _inβ (pos. 44 of Fig. 8) is calculated by the average noise level in a sliding window in range, the center of which is the analyzed point (i, n, β) [4, p. 107]

,

,

where 2k _i is the size of the sliding window,

k ₀ is a factor depending on the probability of correct detection and the probability of false alarm.

Using a threshold to determine _inβ n w the number of signals that exceed the detection threshold. For w signals exceeding the detection threshold, the spectrum width ΔF _D (i, β) of the total signal is calculated in the order of decreasing range amplitude (pos. 46, 47, 51, 52 of Fig. 8) until a signal with a bandwidth ΔF _{d is} smaller than the threshold Δ _threshold (pos. 47 of Fig. 8), after which the radar operates along the branch from pos. 48 to pos. 50 and 56, corresponding to the detection of a target with signal capture by Doppler frequency, range and angle. If from w signals exceeding the detection threshold P _inβ , the target signal is not found (ΔF _d <Δ _threshold ), the radar is tuned to detect, capture and track a noise source of interference (pos. 53-55 of Fig. 8). In this case, sequentially at ranges where there is no reflected signal, the power scan of the noise signal is calculated along the azimuth angle P (β):

,

,

where M _w is the number of ranges over which the noise signal is analyzed, the angular position β _{w of the} maximum power of the noise signal is found, and P (β _w ) is compared with the detection threshold. If a threshold is exceeded, are N _e extreme points (i _e, j _e) for the azimuth β _br, spectra total Z _Σ (i _e, n _e, β _w) and difference signal Z _Δ (i _e, n _e, β _w) , then γ _{k is} calculated.

The information obtained at pos. 50 or 55 of Fig. 8 is used in pos. 56 of Fig. 8 to calculate the error signal by the angle (either γ _C or γ _IP ) and to close the tracking system according to the angle from the angular error signal.

Switching the differential signals in azimuth and elevation through switch 8 at the input of the high-frequency receiver of the differential signal 10 according to the BCMC 17 commands makes it possible to simultaneously monitor both angular coordinates of the target (ε _c , β _c ) and their angular velocities (ω _εc , ω _βc ). Tracking the range and Doppler frequency of the target signal provides a measurement of both the current range to the target and the radial velocity of the target (D _c , V _Dc ). For tracking the noise signal source, the BCVM 17 measures both the angular coordinates (ε _IP , β _IP ) and the angular velocities (ω _{ε IP} , ω _{β IP} ) for tracking.

The logic of the radar in combat mode can be changed to purely passive according to the source of noise interference or active only for the reflective target, including with the tuning of the carrier frequency from pack to pack, which is provided by the possibility of synthesizing signal frequencies in the exciter 4 on a modern element base.

The measurement results of the BCMC 17 are provided to the control system at its request via a separate information bus connected to its second output.

Figure 2 shows a variant of the antenna system 1 on the basis of two Cassegrain mirror antennas, including a parabolic mirror 18, four horn irradiators 21 ₁ ... 21 ₄ , a twist-reflector 19, mechanically connected to the gyrostabilizer 22. Supply of the probe signal of the transmitter to the horn irradiators 21 ₁ ... 21 ₄ and conversion of the signal received by the four beams of the two mirror antennas into sum and difference signals is performed by the sum-difference converter 20. In this case, the difference signals in azimuth and elevation are generated at hundredth and seventh outputs of the total-differential converter 20, respectively, the total - at the fifth. The direction of radiation-reception is determined by the tilt of the twist-reflector 19, mechanically set by the gyrostabilizer 22.

The structural diagram of the driver of the reference signal 6 is shown in figure 4. Here, the frequency-manipulated intermediate frequency signal is fed to the second input of the second mixer 26, where it is transferred to the video frequency using the intermediate frequency signal f _pr from the fourth output of the exciter 4, then it is sequentially filtered by the second band-pass filter 27 and amplified by the first video amplifier 28, which is the output device of the reference driver channel 6.

An option for high-frequency receivers 9 and 10 is the receiver depicted in figure 5. Here, through a series-connected limiter 29 and high-frequency signal amplifier 30, the received signal is supplied to the mixer with suppression of the mirror channel 31. The heterodyne frequency comes to the second input of the mixer 31 from the exciter 4. The signal converted to the intermediate frequency through the SAW filter and the intermediate frequency amplifier 33 is fed to high-frequency receiver output 9 (10).

A variant of the suppression unit of the mirror channel 13 (14) at the video frequency is shown in Fig.6. Here, the first and second phase-shifting circuits 34 and 35 can be made on the basis of circuits with a uniform amplitude-frequency characteristic, the phase-frequency characteristics of which in the entire frequency range have a relative shift of 90 degrees. [6, p. 51, fig. 2].

The advantage of the proposed method over the prototype is: providing radar tracking both for reflective targets and noise sources, this allows, depending on the jamming situation and flight mission, to direct the aircraft to either the target or the noise source, and accordingly increase combat capabilities LA

increasing the communication potential by 3 dB due to suppression of the mirror video channel, respectively, reducing the noise band,

the compression schemes of the sum and difference channels are performed using the digitized real signal (one quadrature component), and not the quadrature one, consisting of two components, while the requirements for the sampling frequency are minimal (F _in ≥2ΔF _c ), the requirements for the identity of the amplitude-frequency and phase-frequency the characteristics of the video amplifiers standing after the quadrature-phase detection in front of the ADC are recorded (a quadrature-phase detector and a second video amplifier are not required), the orthogonality of the squared compressed signals ensures INDICATES purely digital processing without increasing the requirements for the number of operations and speed ADC,

reduction in the number of operations during compression of the sum and difference signals due to the convolution of the received signals with the reference signal in the frequency domain, where the spectrum of the received sum and difference signals is calculated by one FFT, instead of two, while the corresponding spectra of analytical signals are restored.

The proposed device monopulse radar that implements the specified method, in addition to what the method allows, provides:

increasing the communication potential by 3 dB due to suppression of the mirror channels when transferring the received total and differential signals to an intermediate frequency;

protection of the receiver from overload by both the probing signal and powerful organized interference due to the limitation of the received signal at the carrier frequency;

Reducing the requirements for the coherence of sounding signals by transferring each sounding signal to a video frequency, digitizing and using it as a reference.

Using the information presented in the application materials, the proposed radar can be manufactured in production using known materials, elements, units and technology, used to detect, track and measure coordinates of both real reflecting targets and sources of noise interference, which proves the industrial applicability of the invention.

In accordance with the application materials, a prototype of the device was manufactured, the tests of which confirmed the achievement of the indicated technical effect.

LITERATURE

1. Patent of Russia No. 225349 dated 02.17.2004, cl. G01S 3/14. Error generating device for receiving a noise signal

2. Patent of Russia №2117960 from 08.20.1998, cl. G01S 13/34. A method of tracking a target with a monopulse radar station.

3. Introduction to Digital Filtering, ed. R. Bogner and A. Konstanidis. - M.: World 1976 (p. -128-131).

4. Protection of radar systems from interference. Status and development trends, ed. A.I. Kanaschenkov and V.I. Merkulov. - M .: Radio engineering, 2003 (p. 107).

5. Resonant microwave power limiter on semiconductor diodes. M .: Radio engineering 1987, No. 10.

6. Quadrature scheme on operational amplifiers. - M .: Electronics, 1975, No. 17 (p. 50, 51).

7. Quadrature mixer LT 5516 company Linear Technologis.

8. SAE filters, type AE 455 N 804. Prospectus of the company AEC Design, St. Petersburg.

9. One-way switch to 2 positions SW-338 from MA-COM.

10. M.S. Zhuk, Yu.B. Molochkov. Design of lens, scanning broadband antennas and feeder devices. - M.: Energy, 1973 (p. 82, Fig. 2-7).

11. Gyrostabilized drive of the antenna system GS-26, developed by OAO Temp-Avia.

12. On-board computer VB-480-01. Operation manual NKShR.466535.020 RE.

13. Guide to radar ed. M .: Skolnik, v. 3. - M .: Soviet Radio, 1979 (p. 19 ... 52, 103 ... 107, 144, Fig. 36).

14. The receiving module of the microwave “Opara”, developed by ZAO “Salyut 25”, Nizhny Novgorod.

15. L.A. Belov. Synthesizers of frequencies and signals (ser. "Lecture notes on radio engineering disciplines, issue 9". - Sainz-Press, 2002 (p. 35, Fig. 11).

16. V. Ryzhkov, Popov V.N. Frequency synthesizers in radio technology. - M.: Radio and Communications, 1991 (p. 63, Fig. 3.2).

Claims (5)

A method of tracking a target in a monopulse radar station (RLS), including setting the axis of the equal signal direction (RSN) of the monopulse antenna in the direction of the intended target according to a previous review; emission of pulsed coherent signals in a given direction; receiving high-frequency signals in the interval between probing; sum-difference conversion of received signals, their superheterodyne conversion to an intermediate frequency; amplification of the sum and difference signals at an intermediate frequency; the conversion of signal spectra into the region of video frequencies by means of quadrature-phase detection using reference oscillations f _o , the frequency of which is equal to the intermediate frequency; matched filtering the sum and difference signals to obtain the M-point time realizations analytical summary v _Σij v _Δij and difference signals at each j-th repetition period,

, in the range of ranges,

multichannel Doppler filtering of the complex envelope of the pulse sequence of the total signal for each range element in a given interval in the Doppler frequency range from minus 1 / 2T _p to 1 / 2T _p with a band ΔF and the number of channels N = 1 / T _p ΔF, where T _p - probe pulse repetition period, ΔF is the width of the spectrum of inter-period fluctuations of signals from true targets Z _Σin = F (v _Σij ), where i is the number of the range element, j is the number of the repetition period, n is the number of the Doppler filter,

, F is the Fourier transform operator; calculation of the amplitude of the spectral components of the total signal

detection of signals for which P _Σin exceeds the detection threshold in the a priori range range,

, M _A ≤ M; search for a target signal with a spectrum width ΔF _d less than a threshold Δ _threshold , including calculating the spectral width ΔF _{d of the} total signal in descending order of their amplitudes in range, comparing the current calculated value of the spectrum width ΔF _d with a given threshold value Δ _threshold , when it is exceeded, a decision is made about detection of a false target, cyclic repetition of the calculation of the spectrum width of the signal following the amplitude, until the target signal with a spectrum width less than the threshold is found; determination of the number of the Doppler filter n _c with the maximum amplitude of the target spectrum; determining a frequency mismatch signal, closing the frequency tracking loop, adjusting the frequency mismatch signal to position n _{c the} tracking strobe in frequency; narrow-band filtering with a bandwidth ΔF = 1 / NT _{n of the} complex envelope of the pulse sequences of signals in the total and difference channels at the tracked n _C Doppler frequency to obtain complex spectral components of the total Z _Σ (i, n _c ) and difference Z _Δ (i, n _c ) signals, the selection of the amplitude envelope of the signals in the total channel

, isolating the error signal by range in the total channel, closing the range tracking loop and adjusting the position (i _c ) of the range tracking gate with this signal, calculating the angular mismatch signal γ (i _c , n _c ) of the RSN relative to the direction to the target according to the results of calculating the spectral components total Z _Σ (i _c , n _c ) and differential Z _Δ (i _c , n _c ) signals at the tracked point in the scene, the range is the Doppler frequency (i _c , n _c ), the angle tracking circuit is closed by the angle mismatch signal γ (i _i, n _i) from the correctness Rovkov position PCH monopulse antenna, characterized in that after the quadrature-phase detection suppress strip mirror video frequencies where priori offline shifted by the Doppler frequency signal objectives obtained actual sum-difference signals is digitized with sampling frequency F _B, exceeding a frequency band occupied by the modulating signal ΔF _c more than twice; before starting the search for the target signal after the threshold detection, the number of detected signals w is determined; if during the cyclic search of the target signal for w cycles the target signal is not detected, the following is performed: multi-channel Doppler filtering of the total signals in the frequency range from zero to 1/2 T _p at ranges

where there are no reflected signals to obtain complex spectra of the total signals Z _Σin and the amplitudes of the spectral components of the total signal

, threshold detection of noise interference source signals by the amplitudes P _{Σin of} spectral components that exceed the detection threshold of the noise source, finding N _e extreme points (i _e , n _e ) with a maximum amplitude of the spectral component P _Σin , with the calculation of the spectral components of the difference signal Z _Δ (i, n), the calculation error signal RDA to the direction to the source of noise interference γ _u as the average angular mismatches γ _e n _e for the extreme points (i _e, n _e) of the quadrature spectrum cial integral total Z _Σ (i _e, n _e) and the difference Z _Δ (i _e, n _e) signals; short circuit of the tracking system in the angle with pointing the axis of the antenna according to the signal of the angular mismatch of the RSN antenna to the source of noise interference. 2. The method according to claim 1, characterized in that the suppression of the mirror video frequencies of the input signal is performed by converting the quadrature components of the input video signal into a new pair of signals, further shifted by 90 ° relative to each other in the frequency band occupied by the modulation of the probe signal ΔF _c , and the subsequent their addition. 3. The method according to claim 1, characterized in that the matched filtering in each repetition period is performed by a sequence of operations including the M-point Fourier transform of the sequences of complex numbers x _ij = u _Σij + ju _Δij , the real part of which is equal to the digitized samples of the total signal u _Σij , and imaginary - samples of the difference signal u _Δij , with the receipt of M-point spectra X _mj = F (x _ij ),

, m is the frequency number of the spectral component, reconstructing the spectra of the analytical total S _Σmj and difference S _Δmj signals from the spectra of X _mj , multiplying the spectra of the analytical total and difference signals with the complex conjugate spectrum of the probe signal S * _m to obtain the spectra of the compressed signals V _∑mj = S _∑mj S * _m and V _Δmj = S _Δmj S * _m , the inverse M-point Fourier transform with obtaining for each j-th repetition period M-point analytical time realizations of the compressed total v _Σij , and difference v _Δij signals. 4. Monopulse radar station (RLS) that implements the method according to claim 1, comprising a monopulse antenna system, the first input-output of which is connected to the second input-output of the antenna switch, an exciter, quadrature-phase detectors of the sum and difference channels, a synchronizer, characterized in that a transmitter is inserted into it, connected in series with the antenna switch, connected in series with a modulated signal driver, a reference signal generator and an on-board computer (BTsV M), intended for the issuance of flight tasks, calculation of radiation parameters, calculation of the initial values of the signal gain level when receiving, depending on the a priori distance to a given target and its reflecting characteristics, input of calculated tuning values into radar units, high-frequency receivers of total and difference signals, first the inputs of which are connected to the third output of the antenna switch and the output of the switch, respectively, the block suppression of the mirror channel in the total channel, connected through a video amplifier the sum channel with the fourth input of the digital computer, the block suppressing the mirror channel in the difference channel, connected through the video amplifier of the difference channel to the fifth input of the digital computer, while the high-frequency receiver of the total signal through the quadrature-phase detector of the total signal is connected to the first input of the block suppressing the mirror channel in the total channel, the high-frequency receiver of the difference signal through the quadrature-phase detector of the difference signal is connected to the first input of the block suppression of the mirror channel in the difference m channel, the first output of the exciter is connected to the second inputs of the high-frequency receivers of the total and differential signals, the first input of the transmitter, the third output of the exciter is connected to the first inputs of the driver of the reference signal, quadrature-phase detectors of the total and difference signals, the second outputs of which are connected to the same inputs of the suppression units mirror channels of the total and differential channels, respectively, the first input-output of the computer is connected to the third inputs of the high-frequency receivers of the total and p input channel, the third input of the switch, the fourth input-output of the monopulse antenna system, the fourth input of the transmitter, the exciter input and the first input of the synchronizer, the second and third outputs of the monopulse antenna system are connected to the first and second inputs of the switch, the output of the modulated signal former is connected to the third input of the transmitter , the second synchronizer output is connected to the inputs of the modulator and transmitter of the same name, the third synchronizer output is connected to the second E inputs of sum and difference video amplifier channels coupled to the second output of the exciter with the same input synchronizer, a first output of which is connected with the same input of the modulated signal, the fourth output of the synchronizer is connected to a sixth input digital computer, a second input-output of which is input-output of the radar. 5. Monopulse radar according to claim 4, characterized in that the mirror channel suppression unit contains two phase-shifting chains and an adder, the inputs of which are connected to the outputs of phase-shifting chains, the inputs of phase-shifting chains are quadrature inputs of the mirror-channel suppression unit, the adder output is the output of the mirror-suppression unit signal.

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