RU2141719C1 - Method and electric drive for vector control of permanent-magnet synchronous motor - Google Patents

Abstract

FIELD: speed, torque and angular position control for various machines and mechanisms dispensing with transducers. SUBSTANCE: method for vector control of equiphase and orthophase stator currents involves measuring two-phase stator voltage and varying phases of equiphase and orthophase stator currents and rotor speed according to measured two-phase voltages and stator currents. Electric drive implementing this method has two-phase voltage sensor whose outputs are connected through integrators to sine and cosine inputs of direct and inverse Cartesian coordinate converters and to inputs of rotor speed and angular position computing unit. EFFECT: improved stability, enlarged range and improved precision of torque and speed control. 5 cl, 3 dwg

Description

 The invention relates to electrical engineering, in particular to controlled AC electric drives and can be used to control the moment, speed and angular movement of the working bodies of machines and mechanisms, including for deep submersible electric pumps in oil production, in underwater vehicles, in nuclear reactors, electric vehicles, machine spindles and special robots, on-board automation and special tracking systems.

 The purpose of the invention is to increase the stability, range and accuracy of regulation of the moment and speed of the electric motor without sensors structurally associated with the electric motor.

 There is a method of vector control of an electric motor, in which the electric motor is supplied with an m-phase current and from an m-phase inverter, the actual phase currents are measured, the phase, frequency and amplitude of the phase currents are regulated by switching the power switches of the m-phase inverter, depending on the mismatch between the set and measured currents, while the given phase currents are formed in the form of two orthogonal components of the current vector relative to the scan signal characterizing the given synchronization phase, and the phases of the phase currents are shifted from ositelno synchronization phase by an angle proportional to the ratio of said orthogonal components of the current vector [1].

 The electric drive for implementing this method comprises a m-phase inverter, an m-phase current sensor and an m-phase electric motor in series, while the control inputs of the m-phase inverter are through key management devices, an m-phase pulse-width modulator and an m-phase current controller connected to the outputs of the Cartesian coordinate transducer [1].

 The disadvantages of this technical solution are the instability of the rotor movement, the small range and low accuracy of regulating the moment and speed of the electric motor without a sensor structurally associated with the electric motor, which is caused by the uncertainty of the orientation of the coordinates of the current vector and the mismatch of the synchronization phase of the temporary current vector specified at the input of the Cartesian coordinate converter to the initial and the current angle of the spatial vector of the flux linkage of the rotor of the electric motor relative to the stationary geometrical stator axis.

Closest to the proposed method is a vector control of a synchronous permanent magnet motor on the rotor [2], according to which the stator windings of the electric motor are supplied with phase currents equal to the difference of two separately regulated periodic stator currents - cosine (in-phase) and sinusoidal (orthophase) stator currents, variable a clock phase φ _s function characterizing the spatial angle of the current angular position of the longitudinal magnetic axis of the permanent magnets on the rotor, of in relative axis α of the stator, which is measured two-phase stator current i _sα, i _sβ, according to the values which according to the sine and cosine functions of the clock phase φ _s is calculated measured values of the amplitudes ortofaznogo and phase stator currents, depending on which amplitude ortofaznogo current is adjusted proportionally a predetermined value of the electric motor moment, and the common-mode current amplitude is regulated at a zero level in the main speed control zone to the boundary speed and increases with increasing speed above the boundary velocity grow [2].

 The electric drive for implementing the vector control method for a synchronous permanent magnet motor on the rotor comprises an inverter through a phase current sensor connected to the stator windings of the synchronous permanent magnet motor on the rotor, the control inputs of the inverter through a pulse generation unit and a block of PWM phase current regulators are connected to the outputs of the direct converter of two-phase-three-phase coordinates, the inputs of which are connected to the outputs of the direct converter of Cartesian coordinates, connected by the first two inputs to the outputs of the orthophase and common mode current regulators, the feedback inputs of which are connected to the outputs of the inverse Cartesian coordinate converter, the two first inputs connected through the inverse of the two-phase-three-phase coordinates to phase current sensors, the outputs of which are also connected to the inputs of the PWM controller block phase currents, the other two inputs of the forward and reverse transducers of the Cartesian coordinates, respectively, are combined and form the sine and cosine inputs of the transform cartesian coordinates, while the control inputs of the in-phase and orthophase current regulators are connected through the current vector control unit to the output of the speed controller [2].

 The disadvantage of this technical solution is the instability of the rotor movement, the small range and low accuracy of regulating the moment and speed of the electric motor without sensors structurally associated with the electric motor, which is caused by the use of the method of orientation of the Cartesian coordinates of current regulation by the angular orientation of zero marks and sensor signals on the stator and rotor of the electric motor.

 This drawback significantly reduces the efficiency and limits the scope of application of permanent magnet synchronous motors on the rotor, since in chemically active, liquid radioactive media and in mobile units it is often impossible or inefficient to use sensors structurally coupled to an electric motor, including deep-well submersible electric pumps for oil production, in underwater vehicles, in nuclear reactors, in electric vehicles, machine spindles and special robots, in large Automation and special tracking systems. The exclusion of the rotor angular position sensor from the electric drive in such machines and mechanisms leads to uncontrollability by the mutual orientation of the current and magnetic flux vectors, loss of synchronism, motor shutdowns, torque and speed fluctuations, and a decrease in the control range, especially with wide load changes.

 The aim of the invention is to eliminate this drawback, namely, increasing the stability, range and accuracy of regulation of the moment and speed of the electric motor without sensors structurally associated with the electric motor.

To do this, in the vector control method of a permanent magnet synchronous electric motor on the rotor, according to which the stator windings of the electric motor are supplied with phase currents equal to the difference of two separately adjustable periodic stator currents - cosine (in-phase) and sinusoidal (orthophase) stator currents, changed as a function of the synchronization phase φ _s characterizing the spatial angle of the current angular position of the longitudinal magnetic axis of the permanent magnets on the rotor relative to the axis α of the stator, for which measure the two-phase stator current i _sα , i _sβ , according to the values of which, depending on the sine and cosine functions of the synchronization phase φ _{s, the} measured values of the amplitudes of the orthophase and in-phase stator currents are calculated, depending on which the amplitude of the orthophase current is regulated in proportion to the set value of the motor moment, and the amplitude common-mode current is regulated at zero level in the main zone for controlling the speed of the boundary speed and increase with increasing speed above the boundary speed, the phases are additionally regulated stator current equal to the synchronization phase φ _s, depending on the stator voltage magnitude, which is measured two-phase stator voltage u _sα, u _sβ, u _sβ then the measured values of the two-phase stator voltage u _sα, u _sβ, u _sβ and the two-phase stator current i _sα , i _sβ change the phase of the orthophase current of the stator according to the law
sinφ _s = ∫ (u _sβ -R _s i _sβ ) dt-L _s i _sβ ,
and the phase of the common-mode current of the stator is changed according to the law
cosφ _s = ∫ (u _sα -R _s i _sα ) dt-L _s i _sα ,
where R _s is the active resistance of the stator phase winding, L _s is the stator inductance of the electric motor, and the current value of the rotor speed is calculated from the measured values of the biphasic stator voltage and biphasic stator current as a derivative of the synchronization phase φ _s and the rotor speed is regulated depending on the calculated current value speed by changing the set value of the moment of the electric motor in proportion to the mismatch of the calculated current value of speed with a given value of speed, which can measure nyatsya based on the error value proportional to the phase synchronization and a predetermined value of the angular position of the rotor.

 In an electric drive for implementing a vector control method comprising an inverter, connected to the stator windings of a synchronous electric motor with permanent magnets on the rotor through phase current sensors, the inverter control inputs are connected to the outputs of the direct two-phase three-phase coordinate converter via the pulse generation unit and the PWM phase current regulator unit , the inputs of which are connected to the outputs of the direct Cartesian coordinate converter connected by the two first inputs to the outputs of the orthophase regulators phase and common-mode currents, the feedback inputs of which are connected to the outputs of the inverse Cartesian coordinate converter, the first two inputs connected through the inverse of the two-phase-three-phase coordinates to the phase current sensors, the outputs of which are also connected to the inputs of the block of PWM phase current regulators, two other direct inputs and the inverse of the Cartesian coordinates transducers are respectively combined and form the sine and cosine inputs of the Cartesian coordinates transducers, while the control inputs of the reg Common-mode and orthophase current generators are connected through the current vector control unit to the speed controller output, in addition, a two-phase voltage sensor is connected to the inverter outputs, two outputs of which are connected through the first pair of adders to the inputs of two integrators, the outputs of which are connected through the second pair of adders to the cosine and sine inputs Cartesian coordinates transducers, the second inputs of each of the two pairs of adders are connected to the outputs of the inverse transducer of two-phase-three-phase coordinates, and the sine The input and output inputs of the Cartesian coordinates converters are connected to two inputs of the angle and speed calculation unit, the output of the speed calculation channel of which is connected to the feedback input of the speed controller.

 In addition, a position controller is additionally introduced into the electric drive, the feedback input of which is connected to the output of the angle channel of the angle and speed calculation unit, and the control input forms the input of the rotor angular position setting, while the output of the position controller is connected to the control input of the speed controller.

 Additionally, in the electric drive, the angle and velocity calculation unit contains a division node at two inputs, the output of which is connected through the arctangent function calculation node to the sign calculation node, in addition, the angle calculator connected via the derivative calculation node to the multiplication node, the second input of which is connected to the output of the sign calculation node, and the output of the multiplication node forms the output of the speed calculation channel, the output of the angle calculation channel is formed by the output of the rotor angular position calculator, the first input of which connected to the output of the angle calculator, and the second input is connected to the output of the sign calculation node.

 In FIG. 1 is a functional diagram of an electric drive with a synchronous electric motor; FIG. 2 is a diagram of spatial state vectors of a motor; FIG. 3 is a diagram of time vectors explaining a vector control method.

 An electric drive with a synchronous electric motor with permanent magnets on the rotor (Fig. 1) contains a power source 1 at the power input, made in the form of a rectifier 1-a or a battery 1-b. The outputs of the power source 1 through the capacitor 2 are connected to the power inputs of the inverter 3.

The outputs a, b, c of the inverter 3 through the sensors 4, 5 of the phase current are connected to the stator windings of the synchronous electric motor 6 with permanent magnets 7 on the rotor. The control inputs of the inverter 3 are connected to the outputs of the pulse shaping unit 8, the inputs of which are connected to the outputs of the block 9 of the PWM current regulators. The inputs of block 9 of the PWM current controllers are connected to the outputs i _sa ^* , i _sb ^* , i _sc ^{* of the} direct transducer 10 of two-phase-three-phase coordinates, the inputs of which are connected to the outputs i $\genfrac{}{}{0ex}{}{\text{*}}{\text{sα}}$ , i $\genfrac{}{}{0ex}{}{\text{*}}{\text{sβ}}$ direct converter 11 Cartesian coordinates. The first two inputs of the direct converter 11 of the Cartesian coordinates are connected to the outputs of the regulators 12, 13 of the orthophase and common mode currents, the feedback inputs of which I _ort , I _{syn are} connected to the outputs of the inverse converter of the 14 Cartesian coordinates. The first two inputs i _sα , i _{sβ of the} inverse transducer 14 of the Cartesian coordinates are connected through the inverse transducer 17 of the two-phase-three-phase coordinates to the outputs of the phase current sensors 4, 5, which are also connected to the inputs of the block 9 of the PWM phase current regulators.

 The other two inputs of the direct and inverse converters 11, 14 of the Cartesian coordinates are respectively combined and form the sine and cosine inputs sin, cos of the converters 11, 14 of the Cartesian coordinates.

The control inputs I _ort ^* , I _syn ^{* of the} regulators 12, 13 in-phase and orthophase currents are connected via the current vector control unit 15 to the output of the speed controller 16.

Power sensor 20 includes a two-phase voltage u _sα, u _sβ, connected at the outputs of the inverter 3. The sensor 20 comprises a two-phase voltages at the input resistors 21, 22, 23, included in a pattern corresponding circuit of the stator phase windings of the synchronous motor 6 compounds, for example according to the scheme " star "shown in FIG. 1. The outputs of the two resistors 21, 22 are connected to the inputs of the galvanic isolation elements 24, 25, the outputs of which are connected to the inputs of the adder 26. The output of the first galvanic isolation element 24, which is connected to the phase resistor 21 of the “a” inverter 3, forms the first output u _sα sensor 20 two-phase voltage. The output of the adder 26 forms a second output u _sβ of the two-phase voltage sensor 20.

Two outputs u _sα , u _sβ of the two-phase voltage sensor 20 are connected through the first pair of adders 27, 28 to the inputs of two integrators 29, 30, the outputs of which through the second pair of adders are connected to the cosine and sine inputs cosφ _s , sinφ _{s of the} converters 11, 14 Cartesian coordinates . The second inputs of each of the pairs of adders 27, 28 and 31, 32 are connected to the outputs of the inverter 17 of the two-phase-three-phase coordinates. The sine and cosine inputs of the forward and reverse converters 11, 14 of the Cartesian coordinates are connected to two inputs of the angle and velocity calculation unit 33, which contains the output of the channel for calculating the speed ω and the output of the channel for calculating the angle θ. The output of the channel for calculating the speed ω is connected to the feedback input of the speed controller 16.

подключенный ко входу узла 38 вычисления производной, выход которого соединен со входом узла 39 умножения. Выход вычислителя 37 угла может быть подключен ко входу вычислителя 40 углового положения ротора, второй вход которого соединен с выходом узла 36 вычисления знака. Выход узла 36 вычислителя знака подключен ко второму входу узла 39 умножения.The angle and velocity calculation unit 33 comprises, at two inputs cosφ _s , sinφ _s, a division unit 34, the output of which is connected through the arc tangent function calculation unit 35 to the sign calculation unit 36. In addition, the block 33 calculates the angle and speed contains the calculator 37 angle output

connected to the input of the derivative calculation node 38, the output of which is connected to the input of the multiplication node 39. The output of the angle calculator 37 can be connected to the input of the rotor angular position calculator 40, the second input of which is connected to the output of the sign calculation unit 36. The output of the node 36 calculator character is connected to the second input of the node 39 multiplication.

 The output of the multiplication unit 39 forms the output of the speed calculation channel ω of the block 33, and the output of the rotor angular position calculator 40 forms the output of the angle calculation channel θ of the angle and speed calculation unit 33.

 The electric drive may contain at the input a position controller 41, the output of which is connected to the control input of the speed controller 16.

 The power source 1 can be made in the form of a three-phase diode rectifier connected to the industrial network in option 1-a, or, in the case of autonomous, mobile and on-board machines and mechanisms, in the form of a battery in option 1-b. Inverter 3 can be made in the form of a three-phase transistor inverter, or in the form of a solid-state IGBT module with six keys on insulated gate bipolar transistors (IGBTs), as shown in FIG. 1.

 Block 8 pulse formation can be made in the form of a special integrated circuit six-channel driver, and the combination of blocks 9 PWM phase current controllers, converters 10, 11, 14, 17 coordinates can be made in the form of a signal microprocessor, which can be integrated analog-digital signal converters of current sensors 4, 5 and two-phase voltage sensor 20.

 The device unit 15 control the current vector is made in accordance with [2].

 Block 33 and controllers 12, 13, 16, 41 can be made in microprocessor design, for example, on a digital signal processor.

 The electric drive operates as follows.

From the output of the power supply 1, the constant voltage smoothed by the capacitor 1 is supplied through the transistor switches of the inverter 3 and the phase sensors 4, 5 to the stator windings of the synchronous electric motor 6 with permanent magnets on the rotor 7. The output pulses of the pulse generating unit 8 control the open and closed state of the six transistor keys inverter 3 by the method of pulse-width modulation of the error signals between the given phase currents i _sa ^* , i _sb ^* , i _sc ^* and the changed phase currents i _sa . i _sb , i _sc using block 9 PWM phase current regulators.

Двухфазная система заданных фазных токов i $\genfrac{}{}{0ex}{}{\text{*}}{\text{sα}}$ ,i $\genfrac{}{}{0ex}{}{\text{*}}{\text{sβ}}$ образуется на выходе прямого преобразователя 11 декартовых координат, в котором заданный вектор тока статора

из вращающихся декартовых координат d, g приводится к неподвижным декартовым координатам α,β (фиг. 2). Угловая скорость вращения декартовых координат и угол θ вращающейся продольной оси d относительно неподвижной оси α статора электродвигателя 6 (фиг. 2) задаются сигналами cos, sin на входах прямого преобразователя 11 декартовых координат (фиг. 1), на два других входа которого поступают сигналы задания проекций i_sd ^*, i_sg ^* заданного вектора тока статора в координатах d, g согласно выражениям:

где φ_s - фаза синхронизации, характеризующая временной процесс управления вектором тока

синхронно с изменением временных декартовых координат x, y (фиг. 3);
θ(t) - пространственный угол продольной оси d относительно пространственной неподвижной оси α (фиг. 2);
Z_p - число пар полюсов электродвигателя;
θ_м(t) - угол положения ротора электродвигателя.At the outputs of the direct transducer 10 of two-phase-three-phase coordinates, a three-phase system of predetermined phase currents is formed depending on the input two-phase system of predetermined currents i $\genfrac{}{}{0ex}{}{\text{*}}{\text{sα}}$ , i $\genfrac{}{}{0ex}{}{\text{*}}{\text{sβ}}$ according to the formulas:

Two-phase system of preset phase currents i $\genfrac{}{}{0ex}{}{\text{*}}{\text{sα}}$ , i $\genfrac{}{}{0ex}{}{\text{*}}{\text{sβ}}$ is formed at the output of the direct transformer 11 Cartesian coordinates, in which a given stator current vector

from the rotating Cartesian coordinates d, g is reduced to the fixed Cartesian coordinates α, β (Fig. 2). The angular velocity of rotation of the Cartesian coordinates and the angle θ of the rotating longitudinal axis d relative to the fixed axis α of the stator of the electric motor 6 (Fig. 2) are set by the signals cos, sin at the inputs of the direct transformer 11 of the Cartesian coordinates (Fig. 1), to which the other two inputs receive reference signals projections i _sd ^* , i _sg ^{* of a} given stator current vector in coordinates d, g according to the expressions:

where φ _s is the synchronization phase characterizing the time process of controlling the current vector

synchronously with the change in temporal Cartesian coordinates x, y (Fig. 3);
θ (t) is the spatial angle of the longitudinal axis d relative to the spatial stationary axis α (Fig. 2);
Z _p - the number of pairs of poles of the motor;
θ _m (t) is the angle of the rotor position of the electric motor.

Синхронизация процесса управления вектором тока

во времени t (фиг. 3) с процессом пространственного углового перемещения пространственных векторов (фиг. 2) производится с помощью замкнутых контуров регулирования синфазного и ортофазного токов

образованных регуляторами 12, 13 ортофазного и синфазного токов, обратного преобразователя 14 декартовых координат, блока 15 управления вектором тока и регулятора 16 скорости и обратного преобразователя 17 двухфазно-трехфазных координат с сумматором 18.In a direct transducer of 11 Cartesian coordinates, the two-phase current i $\genfrac{}{}{0ex}{}{\text{*}}{\text{sα}}$ , i $\genfrac{}{}{0ex}{}{\text{*}}{\text{sβ}}$ by vector transformation formulas:

Synchronization of the current vector control process

in time t (Fig. 3) with the process of spatial angular displacement of spatial vectors (Fig. 2) is performed using closed-loop control circuits in-phase and orthophase currents

formed by the regulators 12, 13 of the orthophase and common mode currents, the inverse transducer 14 of the Cartesian coordinates, the current vector control unit 15 and the speed controller 16 and the inverse transducer 17 of the two-phase-three-phase coordinates with the adder 18.

где T₁, T₂ - постоянные времени,
p - оператор,

Измеренные величины амплитуд I_ort, I_syn ортофазного и синфазного токов, образуются на двух выходах обратного преобразователя 14 декартовых координат в соответствии с уравнениями обратного преобразования декартовых координат из неподвижной декартовой системы α,β во вращающуюся декартовую систему x, y:

На входы обратного преобразователя 14 декартовых координат поступают сигналы измеренного двухфазного тока i_sα,i_sβ, преобразованного с помощью обратного преобразователя 17 двухфазно-трехфазных координат и сумматора 18 согласно формулам

Измеренные амплитуды I_ort, I_syn в виде выходных сигналов обратного преобразователя 14 декартовых координат сравниваются на входах регуляторов 13, 14 с заданными амплитудами I_ort ^*, I_syn ^*, которые вычисляются в блоке 15 управления вектором тока статора, предложенном в прототипе [2]. На два входа блока 15 управления вектором тока статора подаются сигналы, пропорциональные заданному моменту M^* и граничной скорости ω_гр, который ограничивает диапазон регулирования скорости в режиме нулевого синфазного тока I_syn ^*=I_syn = 0. В этой основной зоне регулирования скорости угол εd пространственного вектора тока статора

(фиг. 2) относительно продольной оси d постоянных магнитов составляет 90^o, а амплитуда ортофазного тока I_ort ^*=I_ort пропорциональна заданному моменту M^*.Using the regulators 12, 13 of the orthophase and common mode currents, the amplitudes of the orthophase current I _ort and the common mode current I _{syn are} regulated according to the proportional-integral law, depending on the mismatches ΔI _ort and ΔI _syn in accordance with the equations:

where T ₁ , T ₂ - time constants,
p is the operator,

The measured amplitudes I _ort , I _{syn of the} orthophase and in-phase currents are formed at the two outputs of the inverse transducer of 14 Cartesian coordinates in accordance with the equations of the inverse transformation of the Cartesian coordinates from a fixed Cartesian system α, β to a rotating Cartesian system x, y:

The inputs of the inverse transducer 14 of the Cartesian coordinates receive the signals of the measured two-phase current i _sα , i _sβ , transformed using the inverse transducer 17 of the two-phase-three-phase coordinates and the adder 18 according to the formulas

The measured amplitudes I _ort , I _syn in the form of the output signals of the inverse transducer 14 Cartesian coordinates are compared at the inputs of the regulators 13, 14 with the given amplitudes I _ort ^* , I _syn ^* , which are calculated in the block 15 of the stator current vector control proposed in the prototype [2] . The signals of the stator current vector control unit 15 are supplied to the two inputs proportional to the given moment M ^* and the boundary velocity ω _gr , which limits the speed control range in the regime of zero common mode current I _syn ^* = I _syn = 0. In this main speed control zone, the angle εd spatial stator current vector

(Fig. 2) relative to the longitudinal axis d of the permanent magnets is 90 ^o , and the amplitude of the orthophase current I _ort ^* = I _{ort is} proportional to the given moment M ^* .

When the speed increases above the boundary speed ω> ω _gr at the output of the current vector control unit 15 (Fig. 1), the signal for setting the common-mode current amplitude I _syn > 0 increases and the angle εd decreases from the conditions for maintaining the output power with a limited voltage of the power source 1 and a variable coefficient communication I _ort ^* and M ^* according to the technical solution [2].

где Δω = ω^*-ω - рассогласование скорости,

коэффициент усиления пропорциональной части регулятора скорости.The speed controller 16 generates a signal for setting the motor moment M ^* depending on the input signal of the speed mismatch Δω in accordance with:

where Δω = ω ^* -ω is the speed mismatch,

gain of the proportional part of the speed controller.

The value of the given moment M ^* changes in proportion to the mismatch of the velocity Δω with the proportionality coefficient K _pc and additionally varies according to the integral law with the integration time constant T ₄ .

с вектором потокосцепления магнитов ψ_м согласно выражению:

где I_s - амплитуда фазного тока статора, регулируемая с помощью блока 9 ШИМ-регуляторов фазного тока и сумматора 19 с использованием обратных связей по фазным токам i_sa, i_sb, i_sc.The actual moment of the electric motor M is formed due to the interaction of the stator current vector

with the flux linkage vector of the magnets ψ _m according to the expression:

where I _s is the amplitude of the phase current of the stator, controlled by block 9 of PWM phase current controllers and the adder 19 using feedbacks on phase currents i _sa , i _sb , i _sc .

In order to continuously control the moment and speed without sensors structurally connected with the electric motor and to ensure the synchronization condition (2), the synchronization signals cos, sin at the inputs of the forward and reverse transducers 11, 14 Cartesian coordinates are generated as a function of the measured two-phase stator voltage u _sα , u _sβ , the values of which are measured using the sensor 20 of the two-phase voltage at the output of the inverter 3.

Опорная фаза "a" для тока i_sa, выраженного уравнениями (1), (7) и для напряжения U_sa, выраженного уравнением (10) реализована в электроприводе по схеме на фиг. 1 непосредственной связью соответствующих входов и выходов каналов "a" и каналов "α", что обеспечивает условия относительности и синхронизации управления и измерения всех векторов относительно неподвижной ориентирующей оси α (фиг. 2) и совпадение осей α и a (фиг. 3).On the resistors 21, 22, 23, connected to the output of the inverter 3 according to the scheme corresponding to the connection diagram of the phase windings of the stator of the electric motor 6, voltages are generated proportional to the phase voltages U _sa , U _sb , U _{sc of the} stator of the electric motor. Through two galvanic isolation elements, two voltage signals U _sa , U _sb are supplied to the inputs of the adder 26, as a result of which the signals of the measured two-phase voltage of the stator are formed at the two outputs of the two-phase voltage sensor 20 in accordance with the expressions:

The reference phase "a" for the current i _sa expressed by equations (1), (7) and for the voltage U _sa expressed by equation (10) is implemented in the electric drive according to the circuit in FIG. 1 by a direct connection of the corresponding inputs and outputs of channels "a" and channels "α", which provides the conditions for relativity and synchronization of control and measurement of all vectors relative to a fixed orienting axis α (Fig. 2) and the coincidence of the axes α and a (Fig. 3).

где ψ_sα,ψ_sβ - потокосцепления статора, которые вычисляются с помощью интеграторов 29, 30 путем интегрирования указанной в (11) разности напряжения и падения напряжения по каждой из осей α,β.
На выходах интеграторов 29, 30 образуются сигналы косвенно измеренных
потокосцеплений статора ψ_sα,ψ_sβ, которые изменяются в функции измеренных величин двухфазного напряжения u_sα,u_sβ и двухфазного тока i_sα,i_sβ. Сигналы косвенно измеренных потокосцеплений статора ψ_sα,ψ_sβ с выходов интеграторов 29, 30 поступают на первые входы сумматоров 31, 32, на вторые входы которых подаются сигналы i_sα,i_sβ с выходов обратного преобразователя 17 двухфазно-трехфазных координат.From the outputs of the sensor 20 of the two-phase voltage u _sα , u _sβ . the signals are fed to the inputs of the adders 27, 28, the other inputs of which are fed the signals of the measured two-phase current i _sα , i _sβ . The outputs of the adders 27, 28 produce signals proportional to the derivatives of the stator flux linkages according to the equations:

where ψ _sα , ψ _sβ are stator flux _linkages , which are calculated using integrators 29, 30 by integrating the voltage difference and voltage drop indicated in (11) along each of the axes α, β.
The outputs of the integrators 29, 30 form signals of indirectly measured
stator flux _linkages ψ _sα , ψ _sβ , which vary as a function of the measured values of the two-phase voltage u _sα , u _sβ and the two-phase current i _sα , i _sβ . The signals of indirectly measured stator flux _linkages ψ _sα , ψ _sβ from the outputs of integrators 29, 30 are fed to the first inputs of adders 31, 32, the second inputs of which are fed signals i _sα , i _sβ from the outputs of the inverter 17 of two-phase and three-phase coordinates.

Проекции ψ_мα,ψ_мβ на оси α,β, получаемые в виде выходных сигналов сумматоров 31, 32, выражаются согласно векторной диаграмме фиг. 2 следующими функциями пространственного угла θ продольной оси d постоянных магнитов относительно неподвижной ориентирующей оси α статора:

где ψ_м - постоянная амплитуда потокосцепления постоянных магнитов на роторе.Using these adders 31, 32, the measured quantities ψ _sα , ψ _sβ , i _sα , i _{sβ are} converted into indirectly measured values of the flux linkage of the magnets ψ _mα , ψ _mβ in accordance with the vector diagram shown in FIG. 2 and expressions:

The projections ψ _mα , ψ _mβ on the α, β axis, obtained as the output signals of adders 31, 32, are expressed according to the vector diagram of FIG. 2 by the following functions of the spatial angle θ of the longitudinal axis d of the permanent magnets with respect to the fixed orienting axis α of the stator:

where ψ _m is the constant amplitude of flux linkage of permanent magnets on the rotor.

The adders 31, 32 have a gain inverse to the constant value ψ _m , and therefore the output signals of the adders 31, 32 have a normalized unit amplitude and are periodic functions of the spatial angle θ of the d axis relative to the α axis.
The normalized periodic signals cosθ (t), sinθ (t) come from the outputs of the adders 31, 32 to the cosine and sine inputs cos, sin of the direct and inverse converters 11, 14 of the Cartesian coordinates and to the inputs of the angle and velocity calculation unit 33.

на выходе узла 38 вычисления производной получается сигнал модуля скорости

Сигнал знака поступает на вход узла 39 умножения и одновременно может подаваться на вход вычислителя 40 положения ротора, в котором с учетом числа пар полюсов электродвигателя Z_p производится вычисление угла положения ротора.The division unit 34 serves to calculate the tangent tanθ (t), the site 35 for calculating the arctangent function, and the unit 36 for calculating the sign perform the operation of determining the sign of the angle θ and the direction of rotation of the electric motor rotor. At the output of the angle calculator 37, an angle modulus signal is generated

at the output of the derivative calculation node 38, a speed module signal is obtained

The sign signal is fed to the input of the multiplication unit 39 and at the same time can be fed to the input of the rotor position calculator 40, in which the rotor position angle is calculated taking into account the number of pole pairs of the electric motor Z _p .

 From the output of the multiplication unit 39, the speed signal ± ω, the magnitude and sign of which depends on the ratio of the measured values of the phase voltages and phase currents at the output of the inverter 3 and characterizes the actual speed ω of the rotor of the electric motor, is fed to the feedback input of the speed controller 16.

The speed feedback ω also arrives at the input of the current vector control unit 15, which compares it with the boundary speed signal ω _gr and determines the ratio of the given amplitudes I _ort ^* , I _syn ^* depending on three values: M ^* , ω, ω _gr .
A signal of a predetermined speed ω ^* is supplied to the input of the speed controller 16, which can be set from the stand-alone electric drive control panel, or from the output of the position controller 41, to the feedback input of which a signal θ _m is supplied from the output of the rotor angular position calculator 40.

From an autonomous drive control panel, one of three possible control modes can be set by an external task: 1) a given moment M ^* ; 2) a given speed ω ^* ; 3) a given position θ $\genfrac{}{}{0ex}{}{\text{*}}{\text{m}}$ . The other two parameters automatically change depending on a given control action, thereby achieving the goal of vector motor control.

управляют в зависимости от вектора напряжения

вызвавшего этот ток, причем таким образом, что фаза тока статора, равная фазе синхронизации φ_s, изменяется в функции величин напряжения, а синфазный и ортофазный ток, образующие ток статора электродвигателя, регулируют в координатах фазы синхронизации φ_s, синхронизированной во время с пространственным углом θ текущего углового положения продольной оси d постоянных магнитов на роторе за счет изменения фазы ортофазного тока статора по закону:
sinφ_s= ∫(u_sβ-R_si_sβ)dt-L_si_sβ (15)
и фазы синфазного тока статора по закону:
cosφ_s= ∫(u_sα-R_si_sα)dt-L_si_sα. (16)
Регулирование момента M электродвигателя осуществляется за счет векторных обратных связей путем измерения амплитуд ортофазного и синфазного токов I_ort, I_syn относительно фазы синхронизации φ_s, задаваемой измеренным двухфазным напряжением статора по законам (15), (16).The essence of the vector control method is that the current vector

control depending on the voltage vector

which caused this current, so that the phase of the stator current equal to the phase of synchronization φ _s varies as a function of voltage values, and the common-mode and orthophase current, forming the stator current of the electric motor, are regulated in the coordinates of the phase of synchronization φ _s , synchronized during time with a spatial angle θ of the current angular position of the longitudinal axis d of permanent magnets on the rotor due to a change in the phase of the stator orthophase current according to the law:
sinφ _s = ∫ (u _sβ -R _s i _sβ ) dt-L _s i _sβ (15)
and the phase of the common-mode current of the stator according to the law:
cosφ _s = ∫ (u _sα -R _s i _sα ) dt-L _s i _sα . (16)
The torque M of the electric motor is controlled by vector feedbacks by measuring the amplitudes of the orthophase and in-phase currents I _ort , I _syn with respect to the synchronization phase φ _s specified by the measured two-phase stator voltage according to the laws (15), (16).

 The motor speed ω is controlled by vector feedbacks by measuring the two-phase current and two-phase voltage and calculating the current speed as a function of the synchronization phase determined by the laws (15), (16).

Regulation of the position θ _m and the angular displacement θ _m (t) of the electric motor rotor is carried out by indirect feedback on the angular position θ formed by vector feedbacks on the two-phase voltage and two-phase current at the inverter output.

The specified vector control method with forced orientation of the current vector along the d axis and automatic generation of a sine and cosine control action sinθ, cosθ directly at the inverter output without sensors structurally coupled to the electric motor ensures stability of maintaining the given values of the moment M ^* , speed ω ^* and angular position θ ^* in a wide range of load and rotation speed of a synchronous permanent magnet motor on the rotor.

2) автоматической компенсацией отклонения угла фазового сдвига ε_d вектора тока

(фиг. 2) от заданного оптимального значения за счет регулирования синфазного и ортофазного токов в декартовых координатах x, y, вращаемых в функции измеренных векторов

и образуемой по формулам фазы φ_s; 3) автоматической компенсацией отклонения момента M и скорости ω от заданных величин M^*, ω^* за счет пропорционально-интегрального регулирования скорости и момента в функции текущей скорости ω, вычисленной по измеренным векторам U_s, i_s.Industrial applicability and efficiency of using the invention for machines and mechanisms where the use of sensors on the engine is difficult under operating conditions is due to the increased stability, accuracy and range of regulation of the moment, speed and angular position of the rotor, which are achieved by a combination of three main results in the vector control method and electric drive: 1) automatic compensation of the deviation of the synchronization phase φ _s (Fig. 3) from the current spatial angle θ (Fig. 2) and forced orientation vector transformations about the axis of the magnets, performed as a function of the stator voltage vector

2) automatic compensation of the deviation of the phase shift angle ε _d current vector

(Fig. 2) from a given optimal value due to the regulation of in-phase and orthophase currents in Cartesian coordinates x, y, rotated as functions of the measured vectors

and formed by the formulas of the phase φ _s ; 3) automatic compensation of the deviation of the moment M and speed ω from the given values M ^* , ω ^* due to the proportional-integral control of the speed and moment as a function of the current speed ω calculated from the measured vectors U _s , i _s .

Technical and economic efficiency is achieved by the fact that almost absolute rigidity and linearity of mechanical characteristics are obtained over a wide range of load changes M _s and in a wide range of smooth speed control from zero speed ω = 0 (in stop mode) to a double speed from the boundary value ω≤2ω _gr without a speed sensor and without any other sensors structurally associated with the electric motor. This new property makes it possible to use in a new way the most compact and energy-saving synchronous electric motor with excitation from permanent magnets on the rotor in chemically active, liquid, radioactive environments, in remotely controlled and mobile units where installation of any sensors on the electric motor is impossible.

Application for a new purpose of a synchronous electric motor with permanent magnets on the rotor is achieved thanks to the control method and electric drive proposed in the invention, while meeting the requirements for stability, accuracy and range of torque and speed regulation with the greatest economic effect in the following directions:
- deep submersible electric pumps in oil wells for oil production, where the use of half-shortened permanent magnet motors instead of permanent magnet induction motors instead of asynchronous electric motors, primarily for directional wells, provides an economic effect for oil producers of more than $ 10 million a year ;
- underwater vehicles with a liquid-filled electric motor when powered by rechargeable batteries and requiring a maximum resource before recharging;
- servo drives in nuclear reactors for regulating the position of graphite rods, for crane mechanisms and manipulators, where the use of photoelectric sensors on an electric motor is impossible due to the radioactivity of the medium;
- electric vehicles for which the requirement of minimum energy consumption and minimum mass and dimensions of the electric motor are combined with the requirements of expanding the speed control range with a constant power above the main speed in the absence of sensors on the electric motor;
- machine spindles, where the dimensions of the sensor on the electric motor limit the speed and working area of processing, and the exclusion of the sensor on the electric motor gives an extension of the processing capabilities;
- special robots, on-board automation systems, special tracking systems that require precise testing of a given movement of the working body of the mechanism in the absence of sensors on the shaft, especially in cases of using microelectric motors, the mass and dimensions of which are comparable or less in comparison with standard sensors of speed and angular position .

Sources of information
1. Patent of the Russian Federation N 1458951, cl. 4 H 02 M 7/40 Mishchenko V.A. , Mishchenko N.I. A method of controlling a multiphase inverter and a device for its implementation. Priority 03/26/1984 Registered in the State Register 11/15/1988, 02/15/1989 Issue. N 6. Valid in place of the copyright certificate since July 1, 1991. Patent holder: Mishchenko V.A.

 2. Copyright certificate N 1681371, cl. 5 H 02 P 5/40 Mishchenko V.A., Mishchenko N.I. Vector control method for a permanent magnet synchronous electric motor on a rotor. Priority 03/31/1987. Registered on 06/01/1991, publ. 09/30/91 Bull. N 36.

The list of figures for the application for the invention "Method of vector control of a synchronous electric motor with permanent magnets on the rotor and an electric drive to implement this method"
FIG. 1 Functional diagram of an electric drive with a synchronous electric motor.

 FIG. 2. The diagram of the spatial state vectors of the motor.

 FIG. 3. A diagram of time vectors explaining the vector control method.

The notation in FIG. 1 "A vector control method for a permanent magnet synchronous electric motor on the rotor and an electric drive for implementing this method"
1 - power source (rectifier, battery);
2 - filter capacitor;
3 - inverter (transistor on IGBT modules or IGBT modules);
4,5 - phase current sensor;
6 - synchronous electric motor;
7 - permanent magnets on the rotor;
8 - pulse shaping unit (driver of IGBT modules);
9 - block PWM current controllers;
10 - direct converter of two-phase-three-phase coordinates;
11 - direct converter of Cartesian coordinates;
12 - orthophase current regulator;
13 - common mode current regulator;
14 - inverse transducer of Cartesian coordinates;
15 - current vector control unit;
16 - speed controller;
17 - inverse transducer of two-phase-three-phase coordinates;
18 - adder;
19 - adder;
20 - two-phase voltage sensor;
21, 22, 23 - resistors;
24, 25 - element of galvanic isolation;
26 - adder;
27, 28 - adder;
29, 30 - integrator;
31, 32 - adder;
33 - unit for calculating the angle and speed;
34 - division unit;
35 - node calculation arctangent function;
36 - node calculation sign;
37 - angle calculator;
38 - node calculation of the derivative;
39 - multiplication node;
40 - calculator of the angular position of the rotor;
41 - position adjuster.

Claims (5)

1. The method of vector control of a synchronous permanent magnet motor on the rotor, which consists in the fact that the stator windings are supplied with phase currents equal to the difference of two separately adjustable periodic currents, in-phase and orthophase, varying in the function of the synchronization phase characterizing the spatial angle of the current angular position of the longitudinal magnetic axis of the permanent magnets on the rotor relative to the stator axis, for which the measured stator current in the phases, it is converted into a two-phase system of currents i _sα, i _sβ, on ve for the reasons of which, depending on the sine and cosine functions of the synchronization phase, the amplitudes of the in-phase and orthophase currents are calculated, which are used to control the amplitudes of the in-phase and orthophase currents depending on the mismatch between the calculated and given amplitudes of the in-phase and orthophase currents, while the predetermined amplitude of the orthophase current is changed proportionally to the value of a given moment of the electric motor, and a given amplitude of the common-mode current in the main zone of speed regulation to the boundary speed live at zero level and increase with increasing speed above the boundary speed, characterized in that the stator current phase is regulated equal to the synchronization phase φ _s , for which the stator voltage U _sβ , U _{sα are measured} , then the function of the synchronization phase is calculated in accordance with the formulas

where R _s is the active resistance of the stator phase winding;
L _s is the inductance of the phase stator winding,
calculate the current value of the rotor speed as a derivative of the synchronization phase and change the value of the given moment in proportion to the mismatch of the calculated current speed and the set.  2. The method according to claim 1, characterized in that the predetermined speed is changed depending on the mismatch of the value proportional to the phase of synchronization, and the specified value of the angular position of the rotor.  3. A vector control device for a permanent magnet synchronous electric motor on the rotor, comprising an inverter, connected to the stator windings of the electric motor through phase current sensors, and the control inputs of the inverter through the pulse generation block of the control connected to the outputs of the phase current regulators with pulse-width modulation (PWM controllers) , a direct Cartesian coordinate converter connected by two inputs to the outputs of the orthophase and common mode current regulators, the feedback inputs of which are connected the outputs of the inverse transducers of the Cartesian coordinates, the phase current sensors, the outputs of which are connected to the inputs of the block of PWM phase current regulators, the other two inputs of the forward and reverse transducers of the Cartesian coordinates are respectively combined and form the sine and cosine inputs of the transducers of the Cartesian coordinates, while the control inputs of the in-phase and orthophase currents are connected via the current vector control unit to the output of the speed controller, characterized in that the inputs of the direct converter of two azno-three-phase coordinates are connected to the outputs of the direct Cartesian coordinates converter, and the outputs are connected to the inputs of the PWM phase current regulator unit, the inverse Cartesian coordinates converter is connected to the phase sensors by two inputs to the phase current sensors, a voltage sensor is connected to the outputs of the current inverter whose two outputs are connected through the first pair of adders to the inputs of two integrators, the outputs of which through the second pair of adders are connected to the cosine and sine inputs Cartesian coordinate transformers, the second inputs of each of the two pairs of adders are connected to the outputs of the inverse transducer of two-phase-three-phase coordinates, and the sine and cosine inputs of the direct and inverse Cartesian transducers are connected to two inputs of the block for calculating the rotor rotation angle and rotor speed, the output of the speed calculation channel which is connected to the feedback input of the speed controller.  4. The device according to claim 3, characterized in that it further comprises a position controller, the feedback input of which is connected to the output of the channel for calculating the angle of the block for calculating the rotor rotation angle and rotor speed, the control input forms an input for setting the rotor angle, and the output is connected to control input of the speed controller.  5. The device according to claim 3 or 4, characterized in that the unit for calculating the angle of rotation of the rotor and rotational speed of the rotor contains a division node at two inputs, the output of which is connected through the node for calculating the arc tangent function of the angle of rotation of the rotor to the node for calculating the sign of the angle of rotation and direction of rotation a rotor with an output connected to the input of the node for calculating the derivative of the angle of rotation of the rotor, the output of which is connected to the first input of the node of the multiplication, the second input of which is connected to the output of the node for calculating the sign of the angle of rotation of the rotor, and the output the multiplication node forms the output of the speed calculation channel, the output of the angle calculation channel is formed by the output of the rotor angular position calculator, the first input of which is connected to the output of the rotor angle calculator, and the second input is connected to the output of the rotor angle sign calculation node.

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