KR960007997B1 - Converter using zero voltage switching - Google Patents

Abstract

The circuit is for reducing the switching loss and for stabilizing control signals and for giving better power consumption characteristics. The converter comprises; a source voltage stabilizing circuit(100) operating only at an initial start-up time or under an arbitrary voltage level of Vcc and being turned off after the normal system operation; a main power switching circuit(300) to induce an resonant inductor current into the anode through the cathode of the diode until a gate-source voltage reaches to Vgs(on); a main power circuit(400); and an control circuit to provide the power switching circuit(500) with control signals.

Description

Zero voltage switching resonant converter

1 is a circuit diagram of the present invention.

2 is a power supply voltage stabilization circuit diagram of FIG.

3 is a detailed view of the switching part of FIG.

4 is a waveform diagram of each part of FIG.

5 is a block diagram of a control circuit unit.

6 is a sawtooth wave generating circuit diagram of FIG.

7 is a square wave generating circuit diagram of FIG.

8 is a circuit diagram illustrating a driving current blocking circuit of FIG.

9 is a waveform diagram of each part of FIG.

10 is a conventional circuit diagram.

11 is an operation waveform diagram of a conventional circuit.

The present invention relates to an electronic ballast suitable for operating a discharge load such as a fluorescent lamp.

Ballasts are essential devices for the purpose of limiting the current after discharge in all discharge lamps including fluorescent lamps. As the demand for electronic ballasts has recently been rapidly spreading and spreading, there has been a growing interest in ballasts and many technical job problems have been discovered. Because the ballast is conventionally used mainly choke coil (choke coil) has a disadvantage such as heavy and bulky, audible noise generation and low system efficiency. The discovery of this problem and the solution to it are an electronic ballast using a high frequency switching DC / AC converter.

FIG. 10 shows a resonant converter circuit of a conventional switching method, which is supplemented. A square wave pulse is formed by alternating the zero voltage section (+) and (-) by the control circuit unit and the transformer 200 from the DC voltage obtained by full-wave rectification and planarization of the commercial AC power supply voltage. 340 and 380 are switched alternately. In addition, the resonance current i _L according to the operation of the switching elements 340 and 380 is supplied to the load to drive the discharge lamp.

In the conventional resonant converter for load driving, four driving transistors T ₁ to T ₄ outputting a V _t signal applied to the voltage transformer 200 as shown in FIG. 10 in operating a main power switching device. In the switching operation, each transistor has a period in which transistors connected to each other in series for a very short time, for example, tens of ns to ns, that is, a section in which T ₁ , T ₂ , T ₃ , and T ₄ of FIG. 10 are simultaneously turned on. As shown in FIG. 11, the short-circuit current i _p appears every time the on / off signal is applied to the transistor base.

This short-circuit current not only becomes a switching loss of the output transistor, but also acts as a noise source for the circuits generating the X and Y signals, thereby causing the X and Y control signals to be unstable, thereby causing malfunction.

It is an object of the present invention not only to solve this problem but also to provide a zero voltage resonant converter suitable for discharge load driving which is advantageous in terms of power consumption.

The apparatus according to the first aspect of the present invention for achieving the above object is a power supply voltage stabilization circuit that receives a DC voltage supplied through the first and second power supply line, and than the first square wave falling edge in accordance with the voltage stabilization circuit A driving signal generating circuit for generating a second square wave which is expressed before a predetermined section, first and second switching means receiving the first and second square waves, and blocking a driving current connected between the first switching means and the power supply voltage stabilization circuit. A transformer having a circuit, a first winding receiving the output of the first and first switching means, a second winding, and a third winding for alternately outputting a switching signal and a fourth winding for outputting the regenerated current to the first winding. A load driving switching means connected to an output of the transformer, an inductor for transmitting a high frequency switching signal to a load according to the switched signal, a second power supply line, and the first power supply line; And first and second diodes connected in parallel to a second switching means, respectively, and third and fourth diodes connecting the diode and the first winding and connecting the connection point and the power supply voltage stabilization circuit. .

In addition, the apparatus according to the second aspect of the present invention is a power supply voltage stabilization circuit for receiving a DC voltage supplied through the first and second power supply line, and a positive and negative switching having a zero level interval in accordance with the voltage stabilization circuit A transformer having a control circuit for generating a driving pulse, a first winding for receiving the output of the control circuit, a second winding for alternately outputting a switching signal, and a fourth winding for outputting the sacrificed current to the first winding; Formed with a form, load drive switching means connected to the output of the transformer, and an inductor for transmitting a high frequency switching signal to the load in accordance with the switched signal, the control circuit is a sawtooth wave generator circuit and the sawtooth wave by the same four-phase Square wave generating circuit for generating a square wave having a width, and the square wave is alternately displayed at the time of sawtooth wave receiving the square wave, and the upper section is longer than the sawtooth period. A drive signal generating circuit for outputting first and third extended square waves, first and second switching means receiving first and second driving signals, and driving connected between the first switching means and the power supply voltage stabilization circuit. A current interrupting circuit, first and second diodes connected in parallel to the second power supply line and the first and second switching means, respectively, and the diode and the first winding. It is characterized by consisting of the third and fourth diodes for connecting.

As can be seen from the above configuration, in the present invention, by using the energy sacrificed from the main power circuit as the power source of the control circuit, it is more economical than the method used as the power source of the control circuit through the resistor and capacitor from the system power as in the prior art. It is. That is, in the present invention, when the system is in normal operation, no other power from the control circuit portion for the on / off switching action of the main power switching is required. In the conventional method, power for driving the main power switch is supplied from the control circuit unit at every switching moment, but the present invention does not require a separate power since the main power switch is turned on and off using only energy sacrificed from the main power circuit unit. Do not.

In the present invention, the reverse recovery time of the body diode configured in the main power switch is actively used for the switching operation until the gate-source voltage of the main power switch reaches V _{g3 (on)} . Has a switching mechanism that causes the resonant inductor current i _L to flow through the diode's cathode toward the anode.

In the present invention, the turn-off time point of the output transistor of the control circuit part is not dependent on the base signal but is a time point at which the conduction of the diode connected in parallel with the transistor ends, that is, a time point at which the secondary current i _s is zero-crossing. The rising slope of the collector-emitter voltage of the output transistor is dependent on the resonant inductor current i _L and the secondary current i _s which is a function of the inductance of the primary control winding.

In addition, the present invention has a power supply voltage stabilization circuit of the control circuit portion, and operates only at the start-up moment when the system power is applied or when the power supply of the control circuit portion is below an arbitrary voltage level, and after the system starts normal operation, the normal turn -Have operation staying in off mode.

With reference to the accompanying drawings showing an embodiment of the present invention for the above configuration and operation will be described in more detail.

FIG. 1 shows the overall circuit configuration including the apparatus of the present invention, wherein the Ed DC power source 1 indicated by the DC power source in the circuit shows a DC power source obtained by rectifying and smoothing a commercial AC power source. The power supply is supplied to the circuit block of various functions through the first power supply line 1A and the second power supply line 1B, which is also referred to as ground (G). That is, power is supplied to the power supply voltage stabilization circuit unit 100, the main power switch unit 300, and the main power circuit unit 400 according to the configuration of the present invention. The control circuit 500 receives power from the power supply voltage stabilization circuit 100 and outputs a control signal to the main power switching unit 300. First, except for the internal configuration of the control circuit 500, the circuit configuration and operation of FIG. 1 will be described next.

FIG. 2 is a circuit diagram showing a detailed circuit configuration of the power supply voltage stabilization circuit unit 100 of FIG. As one of the features of the present invention, the power supply voltage stabilization circuit 100 of the control circuit unit 500 is a system power supply, the initial operation start (start-up) time when Ed is applied or will be described later, but the power supply of the control circuit unit 400 In the normal operation mode, the control circuit operates only with power sacrificed from the main power circuit by operating only under the condition that the (Vcc) level is below an arbitrary voltage level, and in the turn-off mode after the system starts normal operation. It is to use power efficiently.

Next, the power supply voltage stabilization circuit 100 shown in FIG. 2 will be described.

When the DC power source Ed is initially applied through the first power supply line 1A and the second power supply line 1B, the DC power Ed is distributed at a ratio as transmitted by the resistors 140, 145, and 150, Therefore, the DC voltage distributed by the resistors 145 and 150 is clamped by the voltage of the zener diode 155 to be applied to the gate of the MOS transistor 110 which receives the voltage, thereby making it switched on. Then, as the MOS transistor 110 is turned on, power is applied to the resistor 105 and the capacitor 120, so the capacitor starts to accumulate charge.

The voltage induced in the capacitor 120 is supplied to the control circuit 500 through the signal line 3. That is, the voltage at the capacitor is designed to reach a voltage level Vcc sufficient to operate the control circuit.

When the initial power is turned on, since the capacitor has no charge, when the charge gradually accumulates and reaches Vcc, the control circuit 500 operates normally and the entire system of FIG. 1 operates normally, but will be described later, but the current sacrificed from the main power circuit unit 400. The capacitor 120 is charged by the components of this signal line along the A and B lines from the driving transformer 200 partially shown in FIG.

The capacitor is charged not only by the power supply Ed but also by the sacrificial current so that the capacitor voltage continues to rise, but is limited by the control diode 115 connected in parallel with the capacitor, and at this point, a series resistor connected in parallel to the capacitor. The switching transistor 135 connected to this resistor by 125 and 130 is turned on and the resistor 150 connected between the collector of this transistor 135 and ground 1B is disconnected, thus resulting in a MOS transistor. 110 is off.

The series of operations described above occurs when the initial power is turned on. In the subsequent operation state, the MOS transistor 110 is turned off so that the DC power supply Ed is no longer supplied to the circuit. Therefore, the control circuit 500 is operated by the sacrificed current from the main power circuit portion, not the power supply voltage stabilization circuit 100.

In the process, the control circuit 500 operates to generate a 'Vt' voltage as shown in FIG. However, the control circuit will be described later.

In the control circuit 500, the switching control signal Vt is an alternating square wave pulse having a zero voltage section and is applied to the primary coil of the transformer 200 as shown in FIG. As shown in (b) and (c), 'Vgs ₁ ' and 'Vgs ₂ ' signals are generated.

3 is again challenged to explain the transformer 200 of FIG. 1 and the main power switching unit 300 connected to the output of the transformer 200, which will be described with reference to the waveform diagram of FIG. 4.

As mentioned, the outputs of the second and third windings 240 and 260 in which the primary side of the transformer 200 is the secondary side by the voltage of the first winding 220 are the same as those of FIGS. 4B and 4C. The state is reversed after the first output (Vgs ₁ ) is in the positive level and the second output (Vgs ₂ ) is in the off period of the negative level. The switching means 320 and 370 are never turned on at the same time.

The two alternating pulses appearing on the secondary side of the transformer 200 are adapted to alternately turn on / off the main power switching means 320, 370. First, when the MOS transistor 320 used as the switching means is in the on state at the time t ₁ to t ₂ , the resonance current i _L flowing in the coil 410 is shown in FIG. 4 (j). As described above, the main power circuit 400 continues to increase according to the resonance characteristics of the main power circuit unit 400.

When the transformer secondary winding output Vgs ₁ becomes zero level as shown in FIG. 4 (b) through the control circuit unit 400 at time t ₂ , the drain current i _p that flows to the drain of the first MOS transistor 340. ) Flows continuously through the capacitor 330 connected between the drain and the source of the first MOS transistor 340, as shown in FIG. 4 (h), and thus the capacitor voltage Vs ₁ appears in FIG. Rises together. While the rising Vs ₁ voltage becomes equal to the DC voltage Ed level, the voltage Vs ₂ appearing between the drain and the source of the second MOS transistor 380 is the same as that of the transistor third winding output Vgs ₂ . has a zero interval in the time interval (t ₂ ~ t ₃₎ it decreases to 4 degrees claim (e) time as shown in (t ₂ ~ t ₃₎ agent at the regional level. At this time, since the resonant current i _L is free-wheeled by the drain-source connected diode 390 of the second MOS transistor, the resonant current continuously decreases. The switching signal voltages Vgs ₁ and Vgs ₂ until the current i _L is zero-crossed, i.e., to time t ₄ , remain at zero voltage and the resonant current i _L From the moment zero-crossing), the diode 390 enters reverse recovery mode. Therefore, the current i _L resonates in the opposite direction and increases in level. In this current flow path, the reverse current flows from the cathode of the diode to the anode during the reverse recovery time (trr) of the diode 390. This operation is achieved by this. However, since the voltage Vt of the first control winding 220 is increased in the negative direction as shown in FIG. 4 (a) at the time when the resonance current i _L crosses zero, the transformer third order When the winding output Vgs ₂ reaches the Vgs (on) level at time t ₅ as shown in FIG. 4 (4), the reverse recovery operation of the diode 390 ends at this time, and the second MOS transistor 370 The resonant current i _L flows in the reverse direction to the drain of the gate and continues to flow.

The second switching device 370 is the time interval (t ₃ ~ t ₆₎ that is on at the time, from t ₆ again, the second turn-MOS transistor when turned off, this parallel-connected capacitor 395 is charged The operation proceeds in the form as described above.

In this operation, the voltage Vt applied to the primary winding of the transformer is an alternating square wave pulse having a zero level interval as shown in FIG. 4 (a) or (i). It is provided from the control circuit 500 being provided. However, as described above, the control circuit 500 is supplied with the main power only at the initial start and operated using the sacrificed power of the main power circuit unit 400 during the normal operation.

The main power circuit unit 400 includes an inductor 410, a discharge-loaded fluorescent lamp 420, and capacitors 440, 450, diodes 460, and 470 that accumulate and discharge charges.

5 is a block diagram of the control circuit, and FIG. 6 is a detailed view of the sawtooth wave generating circuit 600 of the control circuit. FIG. 7 is a square wave generating circuit 650 and a driving signal generating circuit 700 of the control circuit. 8 is a detailed circuit diagram of the driving current blocking circuit 750 of the control circuit. 9 is an operation waveform diagram related to the control circuit. Next, the configuration and operation of the control circuit 500 will be described.

By the initial power-on, the capacitor 120 voltage of FIG. 2 is applied to each circuit block of the control circuit through the output line 3 to perform the initial operation. The circuit connected to the line 3 for power supply includes a reference voltage generator 550, a square wave generator 650, a drive signal generator 700 and a drive current blocking circuit 750, as shown in FIG. to be.

The reference voltage generator 550 applies a predetermined voltage to the sawtooth wave generator circuit 600 by the supplied power. A detailed view of the sawtooth wave generating circuit 600 receiving the reference voltage is shown in FIG. The sawtooth wave generating circuit 600 may adopt a well-known circuit or use the circuit used in the present embodiment.

This circuit is a constant current source 605 generated by the source E, the voltage of the capacitor 610 and the reference voltage ( 620 and a switching means 615 which is switched on at the time of rise of the comparator output and discharges the capacitor charge, and the capacitor voltage is output as a sawtooth wave.

That is, the voltage of the capacitor 610 increases linearly from the constant current source 605 so that the reference voltage ( ), The output of comparator 625 appears to be at a high level and thus this signal is applied to the base of transistor 615 to turn on the transistor. As a result, the voltage of the capacitor 610 is represented by a sawtooth wave as shown by V ₆₁₀ of FIG. This sawtooth voltage is applied to the square wave generating circuit 650 to generate a four-phase square wave.

7 shows a square wave generating circuit generated by receiving sawtooth waves at four different time points. The square wave generation circuit 650 clocks the comparator 655 which compares the input sawtooth wave with the reference voltage 660, the second D flip-flop 670 having the comparator output as a clock, and an inverted signal of the comparator output. A first D-type flip-flop 665, and the inverted output of the first D-type flip-flop 665 is input to the second D-type flip-flop 670 and the second D-type flip-flop 670 Is connected to the input of the first D flip-flop.

The comparator 655 compares the reference voltage Vr level of Fig. 9 (a) with the sawtooth level and outputs a square wave waveform as shown in Fig. 9 (b). Then, FIG. 9 (b) the output _{(Q 1), (Q 1} ) of the output and the comparator output is the D-type flip-flop of claim 1 D-type flip-flop serves as a clock inverted by the inverter 675 of the ninth (C) and (e), and the outputs Q ₂ and Q ₁ of the second D-type flip-flop are shown in FIGS. 9 (d) and (f), and thus a, b, and c. The output d is a square wave of the same width which is expressed at four different time points. Four square waves are applied to the driving signal generation circuit 700 to output two square waves as shown in FIGS. 9 (g) and 9 (h).

The driving signal generation signal 700 is composed of two NAND gates 705 and 710, and the outputs a and b of the first and second D-type flip-flops are the first NAND gate 705. ), And the inverting outputs (c) and (d) of the first and second D-type flip-flops are input to the second NAND gate 710. The two square waves of the drive signal generation circuit 700 overlap each other in part and this part corresponds to the zero-level interval.

Two square waves (X) and (Y), which are alternately generated at each time of sawtooth wave generation and have a pulse width longer than the sawtooth period by a phase interval, are input to the driving unit as shown in FIG. 8 to output transistor 815, 835 is turned on to output an alternating square wave Vt having a zero-level interval as shown in FIG. 9 (i) and input it to the primary winding of the transformer 200.

Next, when the Vcc voltage is applied to the drive current blocking circuit 750 through the output line 3 in FIG. 8 at the beginning of power-on, the PNP transistor 760 is in an on state, and thus the base current of the transistor 760 is turned on. As a result, the voltage of the capacitor 765 gradually rises and becomes equal to Vcc so that the PNP transistor 760 is turned off. Therefore, the voltage Vcc supplied from the power supply voltage stabilization circuit unit 100 is initially blocked.

However, the driving power is required for the on / off operation of the main power switch. As described above, when the power supply is cut off, the driving power cannot be operated. However, the required power does not consume energy sacrificed from the main power circuit unit 400 to the control circuit 500. By using it, it becomes possible to drive. This operation is described next.

First, the output stage will be described in detail.

In the waveform diagram of FIG. 9 (j), a process in which the current i _s in the time t ₂ to t ₃ section is shown as shown is as follows.

In FIG. 8, a diode is connected to the collector of the driving transistor 835, which receives the Y output of the driving signal generation circuit 700 through the resistor 810, and the anode is connected to ground. The diode 820 is connected to the cathode of the first control winding 220 and the collector of the driving transistor 815 which receives the X output of the driving signal generating circuit 700 through the resistor 805 and the anode is connected to ground. The current path (Current Regeneration) is made to the power supply voltage stabilization circuit 100 by forming a flow path leading to). At this time, when the X signal is applied to the switching driving transistor 815 at the time t ₃ , the energy of the primary control winding 220 suddenly becomes a peak current as the transistor 815 is turned on and the sacrificial current i _s Denotes a short-circuit current as shown in FIG. As a result, the energy in the transformer 200 rapidly becomes zero.

The transistor 815 then remains on until the resonant inductor current i _L is zero-crossed. At this time, the current (i _s ) is induced in the first winding by the third winding of the transformer, and the flow diagram leading to the diode 840, the primary control winding 220, and the transistor 815 constitutes a loop. When the resonant current (i _L ) is zero-crossed and the direction of the current is changed, the collector-emitter voltage V _CE of the transistor 835, which is already in the off state as the diode 840 is turned off, gradually increases. . This voltage increase rate is expressed as a function of the resonance current i _L and the inductance value of the control winding 220.

The sacrificial current i _s in a section in which the collector-emitter voltage V _CE of the transistor 835 is smaller than the power supply voltage Vcc becomes a zero value and the time t ₅ to t in FIG. _It is indicated in section ₆ . When the voltage V _CE becomes equal to the power supply voltage Vcc, the diode 830 connected between the collector of the transistor 835 and the output line 'C' is turned on and the diode 820 is simultaneously turned on. In this case, the current flow chart is a flow chart of the diode 820, the control winding 220, the diode 830, and the power supply voltage supply line. In this way, current sacrificiality is reestablished as shown in time 't6'. As can be seen in the above operation process, the turn-off of the transistors 815 and 835 becomes a zero-crossing point of the resonant current i _L , and the turn-on depends on the control signals X and Y. It can be seen that it is determined according to the control signal rising time of FIG. 9 (g) and (h).

In the switching drive of the juice switching means according to the present invention, there is an effect of removing the short-circuit current, and the energy consumed from the main power unit is utilized in the control circuit unit, which is advantageous in power consumption.

Claims (10)

A power supply voltage stabilization circuit that receives a DC voltage supplied through the first and second power supply lines, and a drive signal generation circuit that generates a second square wave that is expressed a predetermined period ahead of the first square wave falling edge according to the voltage stabilization circuit; First and second switching means for receiving the first and second square waves, a driving current blocking circuit connected between the first switching means and the power supply voltage stabilization circuit, and a first receiving means for receiving the output of the first and second switching means. A transformer having a first winding, a second winding for alternately outputting a switching signal, and a fourth winding for outputting the sacrificed current to the first winding; a load driving switching means for connecting to the output of the transformer, respectively; An inductor for transmitting a high frequency switching means to a load according to the switched signal, and a first diode and a second diode connected in parallel to the second power supply line and the first and second switching means, respectively. , Third and fourth diodes zero voltage resonant converter of a switching system, characterized in that consists of connecting the diode and the first winding and connected to the connection point and the power supply voltage stabilizing circuit. The zero-voltage switching type resonant converter according to claim 1, wherein the power supply voltage stabilization circuit comprises a capacitor that accumulates electric charges upon initial power-up, and a switching means that cuts off the power supply at a constant voltage of the capacitor. The zero voltage resonance type converter of claim 2, wherein the sacrificial current of the transformer is supplied to a capacitor of the power supply voltage stabilization circuit through the third and fourth diodes. According to claim 1, wherein the drive current blocking circuit has a capacitor connected to the second power supply line or ground, one side of the emitter receiving the power supply stabilization circuit, the base connected to the capacitor and the first drive A zero voltage resonant converter comprising a PNP type transistor having a collector connected to a first switching means for receiving a current. A power supply voltage stabilization circuit for receiving a DC voltage supplied through the first and second power supply lines, a control circuit for generating a switching drive pulse for alternating a zero level interval positively and negatively according to the voltage stabilization circuit, and the control circuit A transformer having a first winding receiving an output of a winding, a second winding for alternately outputting switching means, and a fourth winding for outputting a sacrificed current to the first winding; and a load drive connected to an output of the transformer, respectively. A switching means and an inductor for transmitting a high frequency switching signal to a load according to the switched signal, wherein the control circuit comprises a sawtooth wave generating circuit and a square wave generating circuit for generating a square wave having the same width as a four-phase by the sawtooth wave; A sphere that receives a square wave and alternately appears at each time of sawtooth wave generation, and outputs first and second square waves extending longer than the sawtooth period. A signal generating circuit, first and second switching means for receiving first and second drive signals, a drive current blocking circuit connected between the first switching means and the power supply voltage stabilization circuit, a second power supply line and the first And first and second diodes connected in parallel to the first and second switching means, respectively, and third and fourth diodes connecting the diode and the first winding and connecting the connection point and the power supply voltage stabilization circuit. Zero voltage switching type resonant converter. 6. The zero voltage type resonant converter according to claim 5, wherein the power supply voltage stabilization circuit comprises a capacitor that accumulates electric charge when the initial power is turned on, and a switching means that cuts off the power supply at a constant voltage of the capacitor. The zero voltage resonance type converter of claim 6, wherein the sacrificial current of the transformer is supplied to the capacitor of the power supply voltage stabilization circuit through the third and fourth diodes. 6. The driving circuit of claim 5, wherein the driving current blocking circuit comprises: a capacitor having one side connected to a second power supply line or ground; an emitter receiving power from the power supply stabilization circuit; a base connected to the capacitor and the first driving; A zero voltage resonant converter comprising a PNP type transistor having a collector connected to a first switching means for receiving a current. 6. The sawtooth wave generating circuit according to claim 5, wherein the sawtooth wave generating circuit includes a constant current source for receiving a supply power and outputting a constant current, a capacitor for accumulating the current of the constant current source, a comparator for comparing the capacitor voltage with a reference voltage, and an output of the comparator. And a switching means configured to switch to discharge the capacitor and generate a sawtooth output. 6. The circuit according to claim 5, wherein the square wave generator circuit comprises: a comparator for comparing the input sawtooth wave with a reference voltage, a second D flip-flop having the comparator output as a clock, and a first D having an inverted signal of the comparator output as a clock; A flip-flop, the inverted output of the first D-type flip-flop is connected to the input of the second D-type flip-flop, and the output of the second D-type flip-flop is connected to the input of the first D-type flip-flop, and the drive The signal generation circuit includes a first NAND gate to which the outputs of the first and second D flip-flops are input, and a second NAND gate to receive the inverted outputs of the first and second D flip-flops. Zero-voltage resonant converter.