KR102705539B1 - Interleaved Converters Using The Secondary LLC Resonant Circuit, And Power Converting Apparatus Using The Same - Google Patents

Abstract

The present invention can control the output voltage (V _o ) from 0 voltage to the gain control range (G v = 0~2) and transmit power through the interleaved phase shift control (D _P : Interleaved Modulation, IM) of individual primary full-bridge circuits (FB1, FB2) at a constant switching frequency (f _s ) around the resonant frequency (f _r ) during forward operation through the proposed interleaved control secondary _LLC resonant circuit, and in the case of reverse operation, the LLC resonant gain characteristics of the magnetizing inductance (L _m1 , L _m1 ') of the transformer (T ₁ , T ₂ ) and the primary and secondary leakage inductances (L _l1 , L _l1 ', L _l2 , L _l2 ') and the secondary resonant capacitor (C _s ) are utilized to control the output stage energy through variable frequency control (FM) of the secondary full-bridge (FB3) switching element. A novel secondary LLC resonant circuit-applied interleaved converter is provided, which is a main circuit type capable of transmitting power to an input terminal.

Description

Interleaved converter using the secondary LLC resonant circuit and power converting apparatus using the same {Interleaved Converters Using The Secondary LLC Resonant Circuit, And Power Converting Apparatus Using The Same}

The present invention relates to an interleaved converter applying an LLC resonant circuit, and more specifically, to an interleaved converter applying a secondary-side LLC resonant circuit by changing two existing primary-side LLC resonant circuits into one secondary-side resonant circuit.

Recently, a DC-DC converter (Bi-directional DC-DC Converter) power conversion device with a wide input/output voltage control range is required in application fields such as microgrid systems, energy storage systems (ESS), electric vehicle OBC (One Board Charger) and battery charging/discharging systems, and fuel cell power conversion devices.

These DC-DC converters (Bi-directional DC-DC Converters) include uni-directional DC-DC converters (Uni-directional DC-DC Converters) and bi-directional DC-DC converters (Bi-directional DC-DC Converters). First, the uni-directional DC-DC converter (Uni-directional DC-DC Converter) will be explained.

Recently, insulated boost DC-DC converters are being applied and reviewed in PCS (Power Conditioning Systems) and ESS (Energy Storage Systems) that require a high voltage step-up ratio from a low input power source such as a battery or hydrogen fuel cell. For this purpose, as a DC-DC converter, a boost converter that can control gain through variable switching frequency control by considering transformer leakage inductance (resonant inductor can be applied in series) and magnetizing inductance as resonant elements as shown in Fig. 1 and that applies a primary LLC (P-LLC) resonant converter that can perform zero voltage switching operation (ZVS) under wide input/output voltage conditions is being reviewed (First Prior Art). However, there is no change in gain above the resonant frequency (f _s /f _r = 1), and especially in the high switching frequency operating range as shown in Fig. 2, the gain increases according to the parasitic capacitance (C _p ) present in the output rectifier diode and the secondary winding of the transformer as shown in Fig. 1, so there is a limitation in wide output voltage control through variable switching frequency control (FM, Frequency Modulation).

And, in order to improve this, an LLC-LC resonant converter with a notch filter, which is an LC parallel resonant circuit, applied to the primary side or the secondary side of the resonant circuit shown in FIGS. 3 and 4 was applied, so that a wide output voltage control range could be achieved in a narrow switching frequency control range (Second Prior Art). However, it has a disadvantage that a notch filter, which is a primary side L _P C _P (or secondary side L _f C _f ) parallel resonant circuit, must be applied separately, and a large parallel resonant current flows in the parallel resonant circuit (Notch Filter) when the output voltage (V _o ) is reduced to 0 and the output load is short-circuited.

Meanwhile, to supplement this, as shown in FIGS. 5a and 5b, a “hybrid rectification method interleaved LLC resonant converter” has been disclosed, which can control the output by operating as a hybrid rectifier as shown in FIGS. 5a and 5b depending on the output voltage control range, in which two primary full-bridge (or half-bridge) LLC resonant converters are connected in parallel on the primary side, and the secondary side operates as a hybrid rectifier as shown in FIGS. 5a and 5b (Third Prior Art). The hybrid rectification method interleaved LLC resonant converter controls the output voltage through two controls, as shown in FIGS. 5a, 5b, and 6. First, in the case of interleaved phase shift control (φ, PM : Phase -shifted Modulation ), as shown in FIGS. 5a and 6, the two primary full-bridge LLC resonant converters operate with an output voltage control range (G _{v = 1~2) through interleaved phase shift control (PM) at a fixed constant switching frequency (f s )} near _the resonant frequency (f r ). Second, in the case of variable switching frequency control ( FM : Frequency Modulation ), as shown in FIGS. 5b and 6, the interleaved phase shift control (φ = π) is in-phase, and the primary parallel full-bridge LLC resonant converters are switched in synchronization in a frequency range lower than the resonant frequency (f _r ), thereby controlling the output with a higher gain control range through the variable switching frequency ( FM, f _s ). The primary switching element of the interleaved LLC resonant converter having the hybrid rectifier diode of Fig. 5a can perform zero voltage switching (ZVS) and respond to input/output voltage control in a wide range. However, when the interleaved full-bridge LLC resonant converters having the hybrid rectifier diodes as shown in Fig. 5a operate under the interleaved phase shift control ( PM ), even if the phase shift control (φ) is reduced to 0 (Out-Phase), the output voltage cannot be reduced to 0 due to the limitation of the gain control range (G _v = 1 to 2), making it difficult to operate with a wider output voltage control range. In particular, since the output voltage cannot be reduced to 0, excessive current may flow under overload and short-circuit loads.

On the other hand, an interleaved LLC resonant converter with a half-bridge coupled inductor is applied to reduce current imbalance and achieve zero voltage switching, but it is difficult to respond to a short-circuit load because two converters, not an interleaved phase shift control circuit, perform variable frequency control switching in phase and the output voltage cannot be reduced to zero.

On the other hand, in order to widen the output voltage control range (G _v = 0 to 2), an interleaved full-bridge LLC resonant converter main circuit method is being considered, which adopts a bridge rectifier method instead of a hybrid rectifier, as shown in Fig. 7, and ensures that the secondary windings of the transformer are always connected in series (4th Prior Art). However, there is a major disadvantage of primary resonance current imbalance (I _Lr1 , I _Lr2 ) depending on the interleaved phase shift control (φ). In addition, since the interleaved phase shift control (φ) becomes minimum (0, Out-Phase), the output voltage (V _o ) can be reduced to 0, so that operation can be performed in a wide output voltage control range. However, when the interleaved phase shift control (φ) is controlled to a minimum, the output voltage (V _o ) becomes too low, so that the voltage induced to the primary side of the transformer is small, and the primary magnetizing current (I _Lr1 = I _Lm1 , I _Lr2 = I _Lm2 ) of the transformer required for zero voltage switching (ZVS) does not flow, causing a problem of hard switching of the primary side full-bridge LLC resonant converter switching elements.

Finally, as shown in Fig. 8, in order to have a wide input/output voltage control range, a bidirectional converter that adopts a bridge rectifier method rather than a hybrid rectifier method is being reviewed for application of an interleaved phase shift control full-bridge LLC resonant converter main circuit method so that the secondary winding of the transformer is always operated in series connection during forward operation (5th prior art). However, as shown in FIG. 7 above, although the output voltage (V _o ) can be reduced to 0 when the interleaved phase control (φ) is minimum (0), thereby providing a wide output voltage control range, when the interleaved phase shift control (φ) is performed, an imbalance in the primary resonant current (I _Lr1 = I _P1 , I _Lr2 = I _P2 ) may occur, and in particular, when the interleaved phase control (φ) is controlled to a minimum as described above, the output voltage (V _o ) becomes too low and the voltage induced to the primary side becomes small, so that the primary magnetizing current (I _Lr1 = I _P1 = I _m1 , I _Lr2 = I _P2 = I _m1 ') of the transformer does not flow, and as shown in FIG. 9, a problem occurs in which the primary full-bridge LLC resonant converter switching elements hard-switch. In addition, as shown in Fig. 8, in the case of reverse operation, since it has the SRC (Series Resonant Converter) gain characteristic instead of the LLC resonant gain characteristic that can be boosted and lowered, voltage control is not possible, and therefore a separate non-isolated buck-boost converter that can be boosted and lowered is required on the secondary side, which has many disadvantages in that the power conversion stage is large.

The present invention is to solve the above problems, and its purpose is to control the output voltage (V _o ) from 0 voltage to the gain control range (G v = 0~2) and transmit power through the interleaved phase shift control ( _{D P} : Interleaved Modulation, IM) of individual primary full-bridge circuits (FB1, FB2) at a constant switching frequency (f s ) around the resonant frequency (f _r ) during forward operation through the proposed interleaved _control secondary LLC resonant circuit (see Figs. 12a to 12c), and in the case of reverse operation, by utilizing the LLC resonant gain characteristics of the magnetizing inductance (L _m1 , L _m1 ') of the transformer (T ₁ , T ₂ ) and the leakage inductances (L _l1 , L _l1 ', L _l2 , L _l2 ') of the primary and secondary sides and the secondary resonant capacitor (C _s ). The present invention provides a novel secondary LLC resonant circuit-applied interleaved converter (see Figs. 13a to 13c) capable of transferring output end energy to input end power through variable frequency control (FM) of secondary end full-bridge (FB3) switching elements.

The above objects and other additional objects will be apparent to those skilled in the art without departing from the technical spirit of the present invention by the appended claims.

According to the first aspect of the present invention for solving the above problem, an interleaved converter applying a secondary LLC resonant circuit comprises: n (n≥2) transformers; n full-bridge circuits respectively connected to the primary sides of the n transformers; and an interleaved secondary LLC (S-LLC) resonant circuit which is connected in series with each other to the secondary sides of the n transformers and applies one resonant circuit to the secondary sides; Including the n full-bridge circuits on the primary side, the n transformers' respective primary-side resonance current (I _P1 , I _P2 ) imbalances induced on the primary side are minimized while allowing the n full-bridge circuits to operate in parallel on the primary side, and by applying one resonant circuit on the secondary side, the component elements are minimized while reducing the phase shift control (D _P ) to 0 (Out-Phase) even under overload and short-circuit load, thereby reducing the output voltage (V _o ) to 0 voltage, thereby limiting the overload or short-circuit current.

At this time, it is possible to additionally include an auxiliary parallel inductor (L _A ) on the secondary side.

Meanwhile, according to the second aspect of the present invention for solving the above problem, an interleaved converter applying a secondary LLC resonant circuit comprises: n (n≥2) transformers; n half-bridge circuits respectively connected to the primary sides of the n transformers; and an interleaved secondary LLC (S-LLC) resonant circuit which is connected in series with each other to the secondary sides of the n transformers and applies one resonant circuit to the secondary side; Including the n half-bridge circuits on the primary side, the n transformers' respective primary resonance current (I _P1 , I _P2 ) imbalances induced on the primary side are minimized while allowing the n half-bridge circuits to operate in parallel on the primary side, and by applying one resonant circuit on the secondary side, the component elements are minimized while reducing the phase shift control (D _P ) to 0 (Out-Phase) even under overload and short-circuit load, thereby reducing the output voltage (V _o ) to 0 voltage, thereby limiting the overload or short-circuit current.

Preferably, DC blocking capacitors (C _B1 , C B2 ) are connected in series between one of the primary side terminals ( _ab , cd) of each of the n full-bridge circuits and one of the primary side terminals composed of the primary side leakage inductance (L l1 , L _l1 ') and the magnetizing inductance (L _m1 , L _{m1 '} ) of the corresponding transformer, and the other of the primary side full-bridge terminals (ab, cd) of the full-bridge circuit and the other of the primary side terminals of the transformer are connected.

Also preferably, DC blocking capacitors (C _B1 , C _{B2 ) are connected in series between one of the primary half-bridge terminals (ab, cd) of each of the n half-bridge circuits and one of the primary terminals composed of the primary leakage inductance (L l1 , L l1 '} ) and the magnetizing inductance (L _m1 , L _{m1 '} ) of the corresponding transformer, and the other of the primary half-bridge terminals (ab, cd) of the half-bridge circuit and the other of the primary terminals of the transformer are connected.

In addition, preferably, the secondary windings of the transformers including the secondary leakage inductances (L _l2 , L _l2 ') of each of the n transformers are connected in series, and one of the series-connected terminals and one of the terminals (ef) of the secondary bridge rectifier diode section consisting of switching elements (S ₁ -S ₂ , S ₃ -S ₄ ) are connected in series with a secondary resonant capacitor (C _s ) to form a secondary LLC resonant circuit.

Also preferably, the interleaved converter is characterized in that it is a bidirectional converter.

Also preferably, the primary side of the transformer (T ₁ , T ₂ ) is characterized by additionally including a separate resonant inductor (L _rP1 , L _rP2 ) instead of a leakage inductance.

Also preferably, the secondary side of the transformer (T ₁ , T ₂ ) is characterized by additionally including a separate resonant inductor (L _rS ) instead of a leakage inductance.

Most preferably, it is characterized by n=2.

On the other hand, the interleaved converter with the secondary LLC resonant circuit according to the third aspect of the present invention for solving the above problem is a converter having an interleaved phase shift controlled secondary LLC resonant circuit, which can control the output voltage (V _o ) from 0 voltage to the gain control range (Gv) and transmit power through the interleaved phase shift control between individual primary full-bridge circuits at a constant switching frequency (f _s ) around the resonant frequency (f _r ) during forward operation, and in the case of reverse operation, the magnetizing inductance (L _m1 , L _m1 ') of the transformer (T ₁ , T ₂ ) and the leakage inductance (L _l1 , L _l1 ', The LLC resonance gain characteristic with the secondary resonant capacitor (C _s ) and the secondary resonant capacitor (L _l2 , L _{l2 '} ) is used to transfer the energy of the output (V _o ) terminal to the input terminal (V _in ) through variable frequency control, and in the forward operation, the primary and secondary leakage inductances of the transformer (T ₁ , T _{2 )} are used. The forward power is transmitted through interleaved phase shift control (D _P , PM) by utilizing the resonance between the secondary resonant capacitor (C _s ) and the secondary resonant capacitor (L _l2 , L _l2 '), _and when operating in the reverse direction _{, the magnetizing inductance (L m1 , L m1 ' )} and leakage inductance (L _l1 , L _l1 ', It is characterized by having resonance gain characteristics in both directions by transmitting power in the reverse direction through variable switching frequency control (FM) according to LLC resonance gain characteristics and _a secondary resonance capacitor (C _s ) composed of an equivalent leakage inductance (L _l2 , L l2 ').

On the other hand, according to the fourth aspect of the present invention for solving the above problem, the interleaved converter having the secondary LLC resonant circuit is a converter having the secondary LLC resonant circuit with interleaved phase shift control, which can control the output voltage (V _o ) from 0 voltage to the gain control range (Gv) and transmit power through the interleaved phase shift control between individual primary half-bridge circuits at a constant switching frequency (f _s ) around the resonant frequency (f _r ) during forward operation, and in the case of reverse operation, the magnetizing inductance (L _m1 , L _m1 ') of the transformer (T ₁ , T ₂ ) and the leakage inductance (L _l1 , L _l1 ', The LLC resonance gain characteristic with the secondary resonant capacitor (C _s ) and the secondary resonant capacitor (L _l2 , L _{l2 '} ) is used to transfer the energy of the output (V _o ) terminal to the input terminal (V _in ) through variable frequency control, and in the forward operation, the primary and secondary leakage inductances of the transformer (T ₁ , T _{2 )} are used. The forward power is transmitted through interleaved phase shift control (D _P , PM) by utilizing the resonance between the secondary resonant capacitor (C _s ) and the secondary resonant capacitor (L _l2 , L _l2 '), _and when operating in the reverse direction _{, the magnetizing inductance (L m1 , L m1 ' )} and leakage inductance (L _l1 , L _l1 ', It is characterized by having resonance gain characteristics in both directions by transmitting power in the reverse direction through variable switching frequency control (FM) according to LLC resonance gain characteristics and _a secondary resonance capacitor (C _s ) composed of an equivalent leakage inductance (L _l2 , L l2 ').

Preferably, the gain control range (Gv) is characterized as 'G _v = 0 to 2'.

Also preferably, the interleaved phase shift control (D _P ) is characterized by 'D _P =0~0.5'.

Also preferably, the forward operation of the main circuit has six operation modes according to phase control (D _P , PM), and during the forward operation, the switching operation frequency (f _s ) performs the switching operation in a frequency range that is slightly lower or higher than the resonant frequency (f _r ) or the resonant frequency (f _r ), and the secondary bridge switching elements (S ₁ , S ₂ , S ₃ , S ₄ ) can be applied as synchronous rectifiers, but are turned off and perform a rectification operation through an anti-parallel diode.

Also preferably, the reverse operation of the main circuit has four operation modes according to variable switching frequency control (FM), and, in the reverse operation, the switching operation frequency (f _s ) is characterized by performing the switching operation in a frequency range (f _min ~f _r ) in which a discontinuous resonant current flows below the resonant frequency (f _r ).

Additionally, according to the fifth aspect of the present invention, a power conversion device employing the secondary-side LLC resonant circuit applied interleaved converter is provided.

Preferably, the power conversion device is characterized in that it is used in any one of a microgrid system, an energy storage system (ESS), an electric vehicle OBC (One Board Charger), a battery charging/discharging system, and a fuel cell power conversion device.

According to the interleaved converter applying the secondary-side LLC resonant circuit according to the present invention, by changing the circuit of the existing two primary-side LLC resonant circuits into an interleaved secondary-side LLC (S-LLC) resonant circuit applying a single resonant circuit as a means for reducing current imbalance and achieving zero voltage switching, even if the primary-side full-bridge or half-bridge converter operates in parallel and interleaved phase shift control (D _P ) is operated, the individual primary-side resonant current (I _P1 , I _P2 ) imbalance induced in the primary side is minimized, and component elements can be minimized by applying a single resonant circuit instead of two resonant circuits. In addition, it is an invention of a main circuit capable of limiting overload or short-circuit current by reducing the phase shift control (D _P ) to 0 (Out-Phase) for overload and short-circuit load and reducing the output voltage (V _o ) to 0 voltage.

Meanwhile, additional features and advantages of the present invention will become more apparent through the following description.

FIGS. 1 to 9 illustrate main circuits, gain characteristics, and operating waveforms according to the prior art, and FIGS. 10 to 18 illustrate main circuits, gain characteristics, and operating waveforms for interleaved phase shift control (D _P ) converters with a secondary LLC resonant circuit applied according to the present invention.
Figure 1 is a primary-side LLC (P-LLC) DC-DC converter.
Figure 2 shows the gain characteristics of Figure 1.
Figure 3 shows an LLC-LC DC-DC converter with a primary or secondary notch filter applied.
Fig. 4 shows the gain characteristics of the circuits of Fig. 3 with the notch filter applied.
Figures 5a and 5b illustrate an interleaved phase shift control (φ) LLC DC-DC converter with a hybrid rectifier.
Figure 6 shows the gain characteristics of Figures 5a and 5b.
Fig. 7 is an interleaved phase shift control (φ) LLC DC-DC converter.
Fig. 8 is an interleaved phase shift control bidirectional converter.
Fig. 9 is an operation waveform of each part of the interleaved phase shift control LLC DC-DC converter of Figs. 7 and 8.
Fig. 10 is a full-bridge interleaved phase shift controlled secondary LLC DC-DC resonant converter according to the first embodiment of the present invention.
Fig. 11 is a half-bridge interleaved phase shift controlled secondary LLC DC-DC resonant converter according to the second embodiment of the present invention.
Figures 12a to 12c are the forward operation of the interleaved phase shift control (D _P , IM) of Figure 10.
Fig. 12a is a forward operation equivalent circuit, Fig. 12b is a forward operation gain characteristic, and Fig. 12c shows a forward operation waveform.
Figures 13a to 13c are reverse operation of the variable switching frequency control (FM) of Figure 10.
Fig. 13a is an equivalent circuit for reverse operation, Fig. 13b shows the gain characteristics for reverse operation, and Fig. 13c shows a waveform for reverse operation.
Figures 14a to 14f are operating modes during forward operation of interleaved phase shift control (D _P , IM), respectively.
FIGS. 15a to 15d, respectively, are operation modes during reverse operation of variable switching frequency control (FM).
Fig. 16 is a secondary LLC DC-DC resonant converter with interleaved phase shift control using an auxiliary inductor (L _A ) according to a third embodiment of the present invention.
Figures 17a and 17b show the resonance gain characteristics of Figure 16.
Fig. 18 shows the forward operation of the interleaved phase shift control (D _P , IM) under short-circuit load.
Fig. 19 is a full-bridge interleaved phase shift controlled secondary LLC DC-DC resonant converter according to a modified example of the first embodiment of the present invention.
FIG. 20a and FIG. 20b are operation modes in the forward operation of the interleaved phase shift control (D _P , IM) in the open state of the relay contact (S _AW ) of the circuit of FIG. 19, corresponding to FIG. 14a and FIG. 14f, respectively.
FIG. 20c and FIG. 20d are the operation modes of the interleaved phase shift control (D _P , IM) forward operation in the relay contact (S _AW ) operation state of the circuit of FIG. 19, corresponding to FIG. 14a and FIG. 14f, respectively.
FIG. 21a and FIG. 21b are operation modes of the variable switching frequency control (FM) reverse operation in the open state of the relay contact (S _AW ) of the circuit of FIG. 19, corresponding to FIG. 15a and FIG. 15d, respectively.
FIG. 21c and FIG. 21d are operation modes of the variable switching frequency control (FM) reverse operation in the relay contact (S _AW ) operation state of the circuit of FIG. 19, corresponding to FIG. 15a and FIG. 15d, respectively.
Fig. 22 is a half-bridge interleaved phase shift controlled secondary LLC DC-DC resonant converter according to a modified example of the second embodiment of the present invention.
Table 1 shows the main ratings and parameters of the bidirectional secondary LLC resonant converter applied in the PSIM simulation.

A preferred embodiment of the present invention will be described with reference to the attached drawings.

However, the embodiments of the present invention may be modified in various forms, and the scope of the present invention should not be construed as being limited to the embodiments described in detail below, but are provided to more completely explain to those skilled in the art.

Furthermore, in this specification, the examples and embodiments described below should be considered as illustrative and not restrictive, and the present invention is not limited to the details given herein, but may be modified into substituted and equivalent other embodiments within the scope and equivalents of the appended claims.

In the present invention, the hybrid interleaved LLC resonant converter method of the third conventional technique (wherein two full-bridge converters on the primary side, as shown in FIGS. 5a and 5b, among the conventional insulated DC-DC converters operate in parallel and perform phase shift control and the secondary side applies a hybrid rectification method to operate with a wide output voltage control range) is improved. In the case of the third conventional technique, even if the interleaved phase control (φ) becomes a minimum (0), since it is a hybrid rectification method, it is only reduced to half of the maximum output voltage (V _o ), and the output voltage (V _o ) does not become 0, so it does not reduce to a low voltage in the event of an overload or short-circuit load, so there was a problem that the overload current flowed and it was difficult to respond.

In addition, in the present invention, the third and fourth conventional techniques of the bridge rectification method shown in FIGS. 7 and 8 are also improved. In the case of the third and fourth conventional techniques, when the interleaved phase control (φ) is minimum (0, Out-Phase), the output gain control range can be reduced from (G _v = 1 to 2) to 0, so that operation can be performed in a wide output voltage control range. However, when the interleaved phase shift control (φ) is controlled to a minimum, the output voltage (V _o ) becomes too low, so that the voltage induced to the primary side of the transformer is small, and thus the magnetizing current on the primary side of the transformer (I _Lr1 = I _P1 = I _m1 , I _Lr2 = I _P2 = I _m1 ') required for zero voltage switching (ZVS) does not flow, so that a problem occurs in which the switching elements of the full-bridge LLC resonant converters connected in parallel on the primary side are hard-switched. Furthermore, even though the secondary sides of the transformer are connected in series, resonant current imbalance (I _Lr1 = I _P1 , I _Lr2 = I _P2 ) occurs due to differences in the gain characteristics of the individual LLC resonant circuits on the primary side, so the resonant current (I _P , I _P2 ) flows ahead of the terminal voltage (V _P1 , V _P2 ) of the primary full-bridge LLC resonant converter where a lot of resonant current flows, and there are limitations in the operating characteristics such as reduced efficiency due to hard switching.

Therefore, instead of applying the existing two primary-side LLC resonant circuits as shown in FIGS. 5, 7, and 8 as a means for reducing the imbalance of the resonant current (I _Lr1 = I _P1 , I _Lr2 = I _P2 ) and for zero-voltage switching, in the present invention, as shown in FIGS. 10 and 11, the circuit is changed to a secondary-side LLC resonant circuit applying one resonant circuit, so that even if the primary-side full-bridge or half-bridge converter operates in parallel and interleaved phase shift control (= D _P ), the resonant current (I _P1 , I _P2 ) imbalance induced in the primary side is minimized, and component elements can be reduced by applying one resonant circuit instead of two resonant circuits. In addition, it is an invention of a main circuit capable of limiting overload or short-circuit current by reducing the voltage to 0 by controlling the interleaved phase displacement (=D _P ) to 0 (Out-phase) even in overload and short-circuit load.

(Examples 1 and 2)

Hereinafter, the interleaved control unidirectional and bidirectional resonant converters applied with the secondary LLC resonant circuit according to the first and second embodiments of the present invention will be described in detail with reference to FIGS. 10 to 15 and FIG. 18.

FIG. 10 is a full-bridge interleaved phase shift control secondary-side LLC DC-DC resonant converter according to the first embodiment of the present invention, and FIG. 11 is a half-bridge interleaved phase shift control secondary-side LLC DC-DC resonant converter according to the second embodiment of the present invention, and FIGS. 12a to 12c are diagrams showing the interleaved phase shift control (D _P , IM) forward operation of FIG. 10, wherein FIG. 12a is a forward operation equivalent circuit, FIG. 12b is a forward operation gain characteristic, and FIG. 12c shows a forward operation waveform. FIGS. 13a to 13c are diagrams showing the variable switching frequency control (FM) reverse operation of FIG. 10. Fig. 13a is an equivalent circuit for reverse operation, Fig. 13b shows the gain characteristics for reverse operation, and Fig. 13c shows a waveform for reverse operation.

FIGS. 14a to 14f are operation modes during forward operation of interleaved phase shift control (D _P , IM), respectively, FIGS. 15a to 15d are operation modes during reverse operation of variable switching frequency control (FM), respectively, and FIG. 18 is forward operation of interleaved phase shift control (D _P , IM) under short-circuit load.

In the present invention, as a means for reducing the imbalance of resonant current (I _P1 , I _P2 ) that may occur in the primary-side individual LLC resonant circuits shown in FIGS. 5, ₇ , and 8 and improving zero-voltage switching, instead of applying a resonant circuit to each terminal of a primary-side full-bridge ( _FB1 , _FB2 of FIG. ₁₀ ) (or a half-bridge of FIG. 11) for interleaved phase shift control (D _P , PM) composed of primary-side switching elements (Q ₁ -Q ₂ /Q 3 -Q 4 , Q ₁ '-Q ₂ '/Q 3 '-Q 4 ') as shown in FIGS. 10 and 11, one of the individual primary-side full-bridge terminals (ab, cd) and the transformer primary leakage inductance (L _l1 , L _l1 ') are connected to block the DC bias. A DC blocking capacitor (C _B1 , C _B2 ) is connected in series between one of the primary terminals of a transformer (T ₁ , T ₂ ) consisting of a magnetizing inductance (L _m1 , L _m1 '), and the other of the primary terminals (ab, cd) of the full-bridge is connected to the other of the primary terminals of the transformer (T ₁ , T ₂ ). Also, the secondary windings of the transformers (T ₁ , T ₂ ) including the secondary leakage inductance (L _l2 , L _l2 ') of the transformers are connected in series, and one of the terminals connected in series and one of the terminals (ef) of the secondary bridge rectifier diode section consisting of switching elements (S ₁ -S ₂ , S ₃ -S ₄ ) is connected in series with a secondary resonant capacitor (C _s ) to form a secondary LLC resonant circuit, and the interleaved phase shift control secondary-side LLC resonant converters and secondary-side LLC bidirectional resonant converters are proposed. (Here, instead of the primary and secondary leakage inductances of the transformers (T ₁ , T ₂ ), a separate resonant inductor (L _rP1 , L _rP2 or L _rS ) can be applied.)

As shown in the gain characteristics of Figs. 12a and 12b and the operating waveform of Fig. 12c, the proposed interleaved phase shift controlled secondary LLC resonant circuit can control the output voltage (V _o ) from 0 voltage to the gain control range (G _v = 0 to 2) and transmit power through the interleaved phase shift control (D _P : 0 to 0.5, Phase-shifted Modulation, PM) between individual primary full-bridge circuits at a constant switching frequency (f _s ) near the resonant frequency (f _r ) in the forward operation, and in the case of the reverse operation shown in the gain characteristics of Figs. 13a and 13b and the operating waveform of Fig. 13c, the magnetizing inductance (L _m1 , L _m1 ') of the transformer (T ₁ , T ₂ ) and the leakage inductance (L _l1 , L _l1 ', The invention relates to a main circuit capable of transferring energy _from the output (V _o ) terminal to the input terminal (V _in ) terminal through variable frequency control (FM, Frequency Modulation) by utilizing LLC resonant gain characteristics with _a secondary resonant capacitor (C _s ) and a secondary resonant capacitor (L _l2 , L _l2 '). In particular, the reason why both directions have resonant gain characteristics is that when operating in the forward direction, the primary and secondary leakage inductances of the transformer (T ₁ , T ₂ ) The forward power is transmitted through interleaved phase shift control (D _P , PM) by utilizing the resonance between the secondary resonant capacitor (C _s ) and the secondary resonant capacitor (L _l2 , L _l2 '), _and when operating in the reverse direction _{, the magnetizing inductance (L m1 , L m1 ' )} and leakage inductance (L _l1 , L _l1 ', The reverse power transfer operation is performed through variable switching frequency control (FM) according to the LLC resonance gain characteristics and the equivalent leakage inductance (L _l2 , L _l2 ') and the secondary resonance capacitor (C _s ).

The forward motion gain formula of Fig. 10 and Fig. 12a is as follows [Mathematical Formula 1].

이고, Here,

And,

The gain formula according to the change in phase control (D _P ) is as follows.

,

,

,

Here,

,

,

,

The reverse operation gain formula of Fig. 10 and Fig. 13a is as follows [Mathematical Formula 2].

Here,

,

,

,

,

,

,

,

,

Here, in FIGS. 10 and 11, a DC blocking capacitor (C _B1 , C _B2 ) is inserted between one of the full-bridge primary terminals (ab, cd) and one of the transformer primary winding terminals to block the DC bias current flowing in each of the primary transformer (T ₁ , T ₂ ) windings according to the interleaved phase shift control (D _P , PM) during forward operation. In addition, it has the advantage of a gain reduction characteristic above the resonant frequency during reverse operation under variable switching frequency (FM) control.

Therefore, as shown in Figs. 10 and 11, by changing the circuit of the existing two primary-side LLC resonant circuits into secondary-side LLC resonant circuits as a means for reducing the imbalance of resonant currents (I _P1 , I _P2 ) and for zero-voltage switching, even if the primary-side full-bridge or half-bridge converters operate in parallel with interleaved phase shift control (D _P , PM), the imbalance of resonant currents (I _P1 , I _P2 ) induced in the primary side can be minimized, and component elements can be minimized by applying one LLC resonant circuit instead of two resonant circuits. In addition, the invention is a main circuit capable of limiting the overload or short-circuit current by reducing it to 0 voltage through the interleaved phase shift control (D _P , PM) even under overload and short-circuit.

This is to explain based on the main circuit of the interleaved control secondary LLC bidirectional resonant converter capable of implementing bidirectional power transfer through interleaved phase shift control (D _P , PM) in forward operation of Fig. 12 and variable switching frequency control (f _s , FM) in reverse operation of Fig. 13.

The forward operation of the main circuit in Fig. 10 will be explained in terms of six operation modes according to phase control (D _P , PM), as shown in Figs. 12 and 14. At this time, it is assumed that the switching operation frequency (f _s ) performs the switching operation in a frequency range _{that is slightly higher than the resonant frequency (f r )} , and that the secondary bridge switching elements (S ₁ , S ₂ , S ₃ , S ₄ ) are turned off to perform a rectifying operation through the anti-parallel diode.

Mode 1 (t ₁ ~t ₂ ) : This is a section in which forward power is transferred according _to the interleaved phase shift control (D _P , PM) of the full-bridge switching elements (Q _{1 -Q 2} /Q ₃ -Q ₄ , Q ₁ '-Q ₂ '/Q ₃ '-Q ₄ ') connected in parallel on the primary side during the section t 1 ~t 2 . Since Q ₁ and Q ₄ and Q ₁ ' and Q ₄ ' are turned on, input voltage (V _in ) is applied to each of the primary full-bridge terminals (ab, cd), and the primary resonance current (I _P1 , I _P2 ) of the transformer (T ₁ , T ₂ ) starts to flow, and the sum voltage of the secondary windings of the series-connected transformer (T ₁ , T ₂ ) is applied to the secondary bridge switching elements (S ₁ , S ₄ ). This is the section where the secondary resonant current (I _S ) is rectified through the anti-parallel diode and power is transmitted to the output load. At this time, the secondary bridge switching rectifier elements (S ₁ -S ₂ , S ₃ -S ₄ ) are turned off, so they only act as rectifying bridge diodes and perform rectification operation.

Mode 2 (t ₂ ~t ₃ ) : At time t ₂ , Q ₁ and Q ₄ are continuously turned on according to the interleaved phase shift control (D _P , PM), and when Q ₁ ' and Q ₄ ' are turned off, the parasitic output capacitances (C _OSS ) of the switching elements Q ₁ ' and Q ₄ ' begin to charge due to the primary full-bridge terminal peak current (I _P2 ) and are charged to the input voltage (V _in ). At the same time, the voltage of the parasitic output capacitances (C _OSS ) of Q ₂ ' and Q ₃ ' decreases to 0, and this mode ends when current starts to flow through the anti-parallel diodes of Q ₂ ' and Q ₃ '. At this time, during the dead time section t ₂ to t ₃ , the lower stage primary side full-bridge switching elements (Q ₁ ', Q ₂ ', Q ₃ ', Q ₄ ') can easily perform zero voltage switching (ZVS) operation under all input/output voltage and load conditions by the terminal peak current (I _P2 ).

Mode 3 (t ₃ ~t ₄ ) : Before time t ₃ , Q ₂ ' and Q ₃ ' are turned on under zero voltage switching condition, and negative input voltage (-V _in ) is applied to the primary full bridge terminal (cd) and the resonant circuit. Therefore, the voltage polarities of the primary and secondary sides of the transformer (T ₂ ) are reversed, and the sum voltage of the secondary windings of the series-connected transformers (T ₁ , T ₂ ) becomes 0. However, this is the section where the secondary resonant current (I _S ) that was flowing continues to flow through the anti-parallel diode of the secondary bridge switching elements (S ₁ , S ₄ ) and is reset by the output voltage (V _o ).

Mode 4 (t ₄ ~t ₅ ) : Since the sum of the secondary windings of the series-connected transformers (T ₁ , T ₂ ) at time t ₄ is 0, the resonant current (I _S ) flowing through the secondary windings of the transformers is reduced to 0 by the reverse bias of the output voltage (V _o ), and the individual primary full-bridge stage resonant current (I _P1 =I _m1 , I _P2 =I _m1 ') flows only through the primary magnetizing inductance (L _m1 , L _m1 ') of the transformers (T ₁ , T ₂ ).

Mode 5 (t ₅ ~t ₆ ) : At time t ₅ , Q ₁ and Q ₄ are turned off, and the parasitic output capacitances (C _OSS ) of switching elements Q ₁ and Q ₄ begin to charge due to the resonant current (I _P1 = I _m1 ), which is the magnetizing current flowing through the primary magnetizing inductance (L _m1 ) of the transformer (T 1 ), and are charged to the input voltage (V _in ). At the same time, the parasitic output capacitance (C _OSS ) voltages of Q ₂ and Q ₃ decrease to 0, and when current begins to flow through the anti-parallel diodes of Q ₂ and Q _{3 ,} when Q ₂ and Q ₃ are turned on, zero voltage switching (ZVS) operation occurs. Therefore, since zero voltage switching (ZVS) operation is performed by the primary resonant magnetizing current (I _P1 =I _m1 ) of the primary full-bridge stage transformer (T ₁ ) during the dead time section of t ₅ to t _{6 ,} optimal design and application of the magnetizing inductance (L _m1 , L _m1 ') of the transformers (T ₁ , T ₂ ) for this purpose is required.

Mode 6 (t ₆ ~t ₇ ) : In the section t ₆ ~t ₇ , Q 2 and Q ₃ and Q ₂ '-Q ₃ ' are turned on according to the interleaved phase shift control (D _P , PM) of the primary full-bridge switching elements (Q _{1 -Q 2} /Q ₃ -Q ₄ , Q ₁ '-Q ₂ '/Q 3 '-Q ₄ ') _, so that a negative input voltage (-V _in ) is applied to each of the resonant circuits connected to the primary full-bridge terminals (ab, cd), and the primary resonant current (I _P1 , I _P2 ) of the transformer (T ₁ , T ₂ ) starts to flow, and the negative sum voltage of the secondary windings of the series-connected transformer (T ₁ , T ₂ ) is applied to the secondary bridge switching rectifier (S This is the section where the secondary resonant current (I _S ) is rectified through the anti-parallel diodes of _{S 2} , S ₃ ) and power is transmitted to the output load. At this time, the secondary bridge switching rectifier elements (S ₁ -S ₂ , S ₃ -S ₄ ) are turned off, so they only act as rectifying bridge diodes and perform rectification operation.

The subsequent semi-cycle movements are similar repetitive movements, so their description is omitted.

Meanwhile, the reverse operation of the circuit of Fig. 10 is The reverse power transfer operation is _{performed through variable switching frequency control (FM) according to the LLC resonance gain characteristics with the equivalent leakage inductance composed of the magnetizing inductance (L m1 , L m1 ') and leakage inductance (L l1 , L l1 ' , L l2 ,} L _l2 ') of the transformer (T ₁ , T 2 ) and the secondary resonant capacitor (C _s ). At this time, the individual primary full-bridge switching elements (Q ₁ , Q ₂ , Q ₃ , Q ₄ and Q ₁ ', Q ₂ ', Q ₃ ', Q ₄ ') are turned off, and the antiparallel diodes of the switching elements are applied as rectifier diodes.

In Figs. 13 and 15 As shown, the reverse operation is controlled by variable switching frequency (FM, fmax∼fmin). At this time, it is assumed that the switching operation frequency (fs) is switching in the frequency range (fmin∼fr) where the discontinuous resonant current flows below the resonant frequency (fr), and the four operation modes are explained.

Mode 1 (t ₁ ~t ₂ ) : During the t ₁ ~t ₂ section _, the secondary main switching elements S ₁ and S ₄ are turned on, so that the output voltage (V _o ) is applied to the LLC resonant circuit terminal (ef) composed of the magnetizing inductance (L _m2 , L _m2' ) and the leakage inductance (L _l1 /L _l1 ', L _l2 /L _l2 ') of the series-connected transformers (T 1 , T ₂ ) and the equivalent leakage inductance (L _eq ) and the secondary resonant capacitor (C _s ), and the secondary resonant current (I _S ) flows, and the primary resonant currents (I _P1 , I _P2 ) flow through the anti-parallel diodes of the individual primary full-bridge switching elements (Q ₁ , Q ₄ , and Q ₁ ', Q ₄ '). It is rectified and power is transmitted to the input power supply (V _in ).

Mode 2 (t ₂ ~t ₃ ): During the period t ₂ ~t ₃ , S ₁ and S ₄ are turned on, but the primary resonant current (I _P1 , I _P2 ) commutated through the anti-parallel diodes of the individual primary full-bridge switching elements (Q ₁ , Q ₄ , and Q ₁ ', Q ₄ ') becomes 0, so only the secondary magnetizing current (I S =I m2 =I m2 ') flowing through the secondary leakage inductance (L _l2 /L _l2 ') and magnetizing inductance (L _m2 , L _m2 ') of the transformer ( _{T 1 , T 2} ) and the secondary resonant capacitor (C _s ) flows.

Mode 3 (t ₃ ~t ₄ ): When S ₁ and S ₄ are turned off at time t ₃ , the parasitic output capacitance (C _OSS ) of S ₁ and S 4 starts to charge due to the secondary resonance current (I _S =I _m2 =I _m2 ') flowing through the secondary resonance capacitor (C _s ), secondary leakage inductance (L _l2 /L _l2 ') _of the transformer (T ₁ , T ₂ ), and magnetizing inductance (L _M1 , L _M1 '), and is charged to the output voltage (V _o ). At the same time, the voltage of the parasitic output capacitance (C _OSS ) of S ₂ and S ₃ decreases to 0, and this mode ends when current starts to flow through the anti-parallel diodes of S ₂ and S ₃ .

Therefore, since the zero voltage switching (ZVS) operation is achieved by the current (I _S =I _m2 =I _m2 ') flowing in the secondary magnetizing inductance (L _m2 , L _m2 ') of the transformer (T ₁ , T ₂ ), optimal design and application of the secondary magnetizing inductance (L _m2 , L _m2 ') of the transformer (T ₁ , T ₂ ) is required to obtain zero voltage switching (ZVS) and gain control range.

Mode ₄ (t ₄ ~t ₅ ): When current flows through the anti-parallel diodes of S ₂ and S ₃ at time t ₄ , zero-voltage switching (ZVS) is performed when S ₂ and S ₃ are turned on, and a negative output voltage (-V _o ) _is applied to the LLC resonant circuit terminal (ef) composed of the magnetizing inductance (L _m2 , L _m2 ') and leakage inductance (L _l1 /L _l1 ', L _l2 /L _l2 ') of the series-connected transformers (T _{1 ,} T 2 ) and the equivalent leakage inductance (L _eq ) and the secondary resonant capacitor (C _s ), and the resonant current (I _S ) flows, and the individual primary full-bridge switching elements (Q ₂ , Q ₃ , and Q ₂ ' , The primary resonant currents (I _P1 , I _P1 ') are rectified through the anti-parallel diodes of Q ₃ ' and power is transmitted to the input power supply (V _in ).

The subsequent semi-cycle movements are similar repetitive movements, so their description is omitted.

The present invention can be applied to unidirectional DC-DC converters and bidirectional DC-DC converters having a wide input/output voltage control range in application fields such as microgrid systems, energy storage systems (ESS), electric vehicle OBCs (One Board Chargers) or battery charging/discharging systems, and fuel cell power conversion devices.

(Example 3)

Now, the interleaved control unidirectional and bidirectional resonant converters with secondary LLC resonant circuit application according to the third embodiment of the present invention will be described in detail with reference to FIGS. 16 to 18.

Fig. 16 is a secondary LLC DC-DC resonant converter with interleaved phase shift control applied to an auxiliary inductor (L _A ) according to a third embodiment of the present invention, Figs. 17a and 17b are resonant gain characteristics of Fig. 16, and Fig. 18 is a forward operation of interleaved phase shift control (D _P , IM) under short-circuit load.

Another main circuit application circuit for unidirectional or forward operation, As shown in FIGS. 16 and 17, the interleaved phase shift control (PM) operation mode section in which the output voltage (V _o ) is controlled from 0 voltage to the gain control range (G _v = 0 to 2) through interleaved phase shift control (PM) at a constant switching frequency (f _s ) around the resonant frequency (f _r ) depending on the applied secondary auxiliary parallel inductor (L A ), and the variable frequency control (FM) operation mode section in which the output voltage (V _o ) is controlled from 0 voltage to the gain control range (G v = 0 to 2) through variable switching frequency control (Frequency Modulation, FM) operation from a switching frequency around the resonant frequency (f _r ) to _a low minimum frequency (f min ) can be divided into the operation mode section. Depending on the applied secondary-side auxiliary parallel inductor (L _A ), the gain characteristics can be further improved in the variable switching frequency control (FM) operating range from zero voltage switching in the primary-side switching element and the switching frequency near the resonant frequency (f _r ) to the low minimum frequency (f _min ).

In addition, in order to prevent current imbalance due to DC bias that may be generated by a very small duty difference of switching elements in the primary resonance currents (I _P1 , I _P2 ) of the full-bridge DC-DC converter, a DC blocking capacitor (C _B1 , C _B2 ) is applied between one of the individual primary full-bridge terminals (ab, cd) and the terminals of the primary windings of the transformer (T ₁ , T ₂ ). In addition, a resonant inductor (L _rP1 , L _rP2 or L _rS ) can be applied separately from the primary leakage inductance (L _l1 , L _l1 ', L _l2 , L _l2 ') of the transformer (T ₁ , T ₂ ).

(variant)

Additionally, as a modified example of the first and second embodiments of the present invention, a secondary LLC (S-LLC) resonant converter using a doubler element comprising a contact (S _AW ) and secondary voltage-dividing capacitors (Co1, Co2) is described with reference to FIGS. 19 to 22.

FIG. 19 is a full-bridge interleaved phase shift controlled secondary LLC DC-DC resonant converter according to a modified example of the first embodiment of the present invention, and FIG. 22 is a half-bridge interleaved phase shift controlled secondary LLC DC-DC resonant converter according to a modified example of the second embodiment of the present invention.

FIGS. 20a and 20b are operation modes in the forward operation of the interleaved phase shift control (D _P , IM) in the open state of the relay contact (S _AW ) of the circuit of FIG. 19, which correspond to FIGS. 14a and 14f , respectively, and FIGS. 20c and 20d are operation modes in the forward operation of the interleaved phase shift control (D _P , IM) in the operating state of the relay contact (S _AW ) of the circuit of FIG. 19, which correspond to FIGS. 14a and 14f , respectively, and FIGS. 21a and 21b are operation modes in the reverse operation of the variable switching frequency control (FM) in the open state of the relay contact (S _AW ) of the circuit of FIG. 19, which correspond to FIGS. 15a and 15d , respectively, and FIGS. 21c and 21d are operation modes in the reverse operation of the relay contact (S _AW ) of the circuit of FIG. 19, which correspond to FIGS. 15a and 15d , respectively. This is the operating mode when the variable switching frequency control (FM) is in reverse operation.

In application fields such as battery chargers and electric vehicle chargers, the battery voltage is being doubled, and in order to respond to various batteries, a main circuit method to respond to this is required. These modified examples are intended to respond to this.

To this end, in order to expand a wider input/output voltage control range in the forward and reverse operations of the full-bridge or half-bridge interleaved secondary-side LLC (S-LLC) resonant converter of FIGS. 10 and 11 described above, as shown in FIGS. 19 and 22, instead of the output-side capacitor (Co) of FIGS. 10 and 11, a first output-side capacitor (Co1) and a second output-side capacitor (Co2) are configured, and a relay contact (S _AW ) is applied between the contact (f) of the secondary-side switching elements (S3, S4) and the contact (g) of the first and second output-side capacitors (Co1, Co2), thereby enabling operation in a wider input/output voltage range in the forward and reverse operations.

In the case of the existing secondary side full-bridge rectifier method during forward operation, the gain range was Gv=0~2, but in the doubler element, the output voltage (V _o ) control range can be expanded to double (Gv=0~2~4) depending on the operation of the relay contact (S _AW ).

A simplified operation mode according to the relay contact (S _AW ) for this is shown in FIGS. 20a to 20d . When the relay contact (S _AW ) is not operated and is separated as in FIGS. 20a and 20b , a full-bridge rectification operation is performed as in the first and second embodiments, but when the relay contact (S _AW ) is operated and connected as in FIGS. 20c and 20d , the output voltage (V _o ) is expanded to double through a doubler operation and operated.

Even in reverse operation, power transmission to the input power supply (V _in ) can be controlled within the output voltage (V _o ) control range depending on the operation of the relay contact (S _AW ) in the doubler element.

A simplified operation mode according to the relay contact (S _AW ) for reverse operation is shown in FIGS. 21a to 21d. When the relay contact (S _AW ) is not operated and is separated as in FIGS. 21a and 21b , a full-bridge switching operation is performed, but when the relay contact (S _AW ) is operated and is connected as in FIGS. 21c and 21d , the input voltage (V _in ) is controlled in response to the output voltage (V _o ) divided through the half-bridge switching operation. At this time, the secondary switching elements S ₃ and S ₄ of one bridge are turned off, and the secondary switching elements S ₁ and S ₂ of one bridge are switched at a 50% duty ratio and the reverse power transfer is controlled through variable frequency control.

Finally, in order to show the optimal embodiment of the present invention, for a 1kW bidirectional secondary-side LLC (S-LLC) resonant converter using the circuit of FIG. 10, the main input/output specifications and parameters such as transformers (T ₁ , T ₂ ), primary and secondary leakage inductances (L _l1 , L _l1 ', N ² L _l2 , N ² L _l2 ') and magnetizing inductances (L _m1 , L _m1 ') as seen from the primary side, a secondary-side resonant inductor (L _SR ) and a secondary-side resonant capacitor (C _S ) are applied, and the resonance gain characteristics through [Mathematical Formula 1] and [Mathematical Formula 2] during forward and reverse operations and the simulation using the PSIM program are shown in the example. To this end, in the main circuit of Fig _. 10 having _the inherent gain characteristics of the secondary-side LLC (S-LLC) resonant converter in bidirectional operation, the output voltage (V _o ) is controlled through phase control (D P , _PM ) at the resonant frequency (f _r ) or a near constant switching frequency (f _s : 100 kHz) under the input voltage range (V _in : 50 V dc to 80 V _dc ) and output voltage (V _o : 700 V _dc ) conditions in the forward operation, and the resonant gain characteristics and PSIM simulation operation waveforms in which reverse power is transferred through variable switching frequency control (f _s : 47 kHz to 105.7 kHz, FM) for a wide input voltage range (V _in : 50 V _dc to 80 V _dc ) at a fixed output voltage (V o ) in the reverse operation are shown in Figs. 12b and 12c, and Figs. 13b and 13c.

[Table 1] shows the main ratings and parameters of the bidirectional secondary LLC resonant converter applied to the PSIM simulation.

In the forward operation of the main circuit specifications shown in Fig. 10 and [Table 1], when the interleaved phase shift control (D _P , PM) is performed and a short-circuit load condition occurs, the interleaved phase shift control (D _P , PM) becomes 0 (Out-Phase) as shown in Fig. 18, reducing the output voltage (V _o ) to 0, and showing the simulation results that stably respond to the short-circuit load.

According to the interleaved converter applying the secondary-side LLC resonant circuit of the present invention, by changing the circuit of the existing two primary-side LLC resonant circuits into an interleaved secondary-side LLC (S-LLC) resonant circuit applying a single resonant circuit as a means for reducing current imbalance and zero-voltage switching, even if the primary-side full-bridge or half-bridge converter operates in parallel and interleaved phase shift control (D _P ) is operated, the individual primary-side resonant current (I _P1 , I _P2 ) imbalance induced in the primary side is minimized, and by applying one resonant circuit instead of two resonant circuits, the component elements can be minimized, and in addition, by reducing the phase shift control (D _P ) to 0 (Out-Phase) even under overload and short-circuit, the output voltage (V _o ) can be reduced to 0 voltage, thereby limiting the overload or short-circuit current.

Input/output specifications (V _in / V _o / P _o ) 50V _dc ~80V _dc / 700V _dc / 1kW Seen directly from the first side
Transformer
Parameter Transformer
(T ₁ ) Primary leakage inductance (L _l1 ) 0.2691 uH Secondary leakage inductance (N ² L _l2 ) 0.9391 uH Magnetizing inductance (L _m ) 20 uH Transformer
(T ₂ ) Primary leakage inductance (L _l1 ') 0.2691 uH Secondary leakage inductance (N ² L _l2 ') 0.9391 uH Magnetizing inductance (L _m ') 20 uH Turning ratio (N _P1 /N _S1, N _P2 /N _S2 ) 0.15(3T/20T), 0.15(3T/20T) Forward
In action Application of PSIM simulation
Primary side main switching element (Q ₁ ~ Q ₄ ) IPP048N12N3 (120V,100A,
R _DS : 4.8mΩ, Power MOSFET) Primary side blocking capacitor (C _B1 , C _B2 ) 20 uF resonant frequency 100 kHz Switching frequency (f _r /f _s ) 100 kHz Reverse
In action Application of PSIM simulation
Secondary side main switching element (S ₁ ~ S ₄ ) UJ3C120040K3S (1200V,65A,
R _DS : 35mΩ, SiC MOSFET) Secondary resonant capacitor (C _s )/resonant inductor (L _SR ) 14 nF/80 uH Resonant frequency (f _r ) 100 kHz Switching frequency (f _s ) 47 kHz ~ 105.7 kHz

Although the present invention has been described above according to one embodiment of the present invention, it is obvious that changes and modifications made by a person having ordinary skill in the art to which the present invention pertains within a scope that does not depart from the technical idea of the present invention also belong to the present invention.

T ₁ , T ₂ : Transformer
FB1, FB2: Primary full-bridge circuit
FB3: Secondary full-bridge circuit
ab, cd: primary full-bridge terminals
ef: secondary bridge rectifier diode terminal
Q ₁ -Q ₂ /Q ₃ -Q ₄ , Q ₁ '-Q ₂ '/Q ₃ '-Q ₄ ' : Primary switching elements
S ₁ -S ₂ , S ₃ -S ₄ : Secondary switching elements
L _l1 , L _l1 ' : Primary leakage inductance
L _m1 , L _m1 ' : Primary magnetizing inductance
C _B1 , C _B2 : Primary side blocking capacitor
L _l2 , L _l2 ' : Secondary leakage inductance
L _m2 , L _m2 ' : Secondary magnetizing inductance
C _s : Secondary resonant capacitor
L _rS : resonant inductor
L _A : Auxiliary inductor

Claims (19)

In the interleaved converter applying LLC resonant circuit,
n (n≥2) transformers;
n full-bridge circuits each connected to the primary side of the n transformers; and
Interleaved secondary LLC (S-LLC) resonant circuit in which the secondary sides of the above n transformers are connected in series with each other and one resonant circuit is applied to the secondary side;
Including,
The n full-bridge circuits are operated in parallel on the primary side, and the primary resonance current (I _P1 , I _P2 ) imbalance of each of the n transformers induced on the primary side is minimized.
By applying a single resonant circuit on the secondary side, the component elements are minimized and the phase shift control (D _P ) is reduced to 0 (Out-Phase) even under overload and short-circuit load, thereby reducing the output voltage (V _o ) to 0 voltage, thereby limiting overload or short-circuit current.
An interleaved converter with a secondary LLC resonant circuit, characterized in that the output-side capacitor is composed of a first output-side capacitor (Co1) and a second output-side capacitor (Co2), and a doubler element that applies a relay contact (S _AW ) between the contact (f) of the secondary-side switching element (S3, S4) and the contact (g) of the first and second output-side capacitors (Co1, Co2), thereby enabling operation in a wider input/output voltage range during forward and reverse operations. In the interleaved converter applying LLC resonant circuit,
n (n≥2) transformers;
n half-bridge circuits each connected to the primary side of the n transformers; and
Interleaved secondary LLC (S-LLC) resonant circuit in which the secondary sides of the above n transformers are connected in series with each other and one resonant circuit is applied to the secondary side;
Including,
The n half-bridge circuits are operated in parallel on the primary side, and the primary resonance current (I _P1 , I _P2 ) imbalance of each of the n transformers induced on the primary side is minimized.
By applying a single resonant circuit on the secondary side, the component elements are minimized and the phase shift control (D _P ) is reduced to 0 (Out-Phase) even under overload and short-circuit load, thereby reducing the output voltage (V _o ) to 0 voltage, thereby limiting overload or short-circuit current.
An interleaved converter with a secondary LLC resonant circuit, characterized in that the output-side capacitor is composed of a first output-side capacitor (Co1) and a second output-side capacitor (Co2), and a doubler element that applies a relay contact (S _AW ) between the contact (f) of the secondary-side switching element (S3, S4) and the contact (g) of the first and second output-side capacitors (Co1, Co2), thereby enabling operation in a wider input/output voltage range during forward and reverse operations. In the interleaved converter applying LLC resonant circuit,
n (n≥2) transformers;
n full-bridge circuits each connected to the primary side of the n transformers; and
Interleaved secondary LLC (S-LLC) resonant circuit in which the secondary sides of the above n transformers are connected in series with each other and one resonant circuit is applied to the secondary side;
Including,
The n full-bridge circuits are operated in parallel on the primary side, and the primary resonance current (I _P1 , I _P2 ) imbalance of each of the n transformers induced on the primary side is minimized.
By applying a single resonant circuit on the secondary side, the component elements are minimized and the phase shift control (D _P ) is reduced to 0 (Out-Phase) even in overload and short-circuit loads, thereby reducing the output voltage (V _o ) to 0 voltage, thereby limiting overload or short-circuit current.
In addition, an auxiliary parallel inductor (L _A ) is included on the secondary side.
An interleaved converter with a secondary LLC resonant circuit, characterized in that the output-side capacitor is composed of a first output-side capacitor (Co1) and a second output-side capacitor (Co2), and a doubler element that applies a relay contact (S _AW ) between the contact (f) of the secondary-side switching element (S3, S4) and the contact (g) of the first and second output-side capacitors (Co1, Co2), thereby enabling operation in a wider input/output voltage range during forward and reverse operations. In the first paragraph,
An interleaved converter applying a secondary LLC resonant circuit, characterized in that DC blocking capacitors (C B1 , C B2 ) are connected in series between one of the primary side terminals consisting of the primary side leakage inductance (L _l1 , L _l1 ') and the magnetizing inductance ( _{L m1} , L _{m1 '} ) of the transformer corresponding to each of the n primary side full-bridge terminals (ab, cd) of the full-bridge circuit, and the other of the primary side full-bridge terminals (ab, cd) of the full-bridge circuit is connected to the other of the primary side terminals of the transformer. In the second paragraph,
An interleaved converter applying a secondary LLC resonant circuit, characterized in that DC blocking capacitors (C _B1 , C _B2 ) are connected in series between one of the primary half-bridge terminals (ab, cd) of each of the n half-bridge circuits and one of the primary terminals consisting of the primary leakage inductance (L _l1 , L _l1 ') and the magnetizing inductance (L _m1 , L _{m1 '} ) of the corresponding transformer, and the other of the primary half-bridge terminals (ab, cd) of the half-bridge circuit is connected to the other of the primary terminals of the transformer. In claim 1 or 2,
An interleaved converter applying a secondary LLC resonant circuit, characterized in that the secondary windings of the transformers including the secondary leakage inductances (L _l2 , _{L l2} ') of each of the n transformers are connected in series, and one of the series-connected terminals and one of the terminals (ef) of the secondary bridge rectifier diode section consisting of switching elements (S ₁ -S ₂ , S 3 -S ₄ ) are connected in series with a secondary resonant capacitor (C _s ) to form a secondary LLC resonant circuit. In claim 1 or 2,
The above interleaved converter is an interleaved converter with a secondary LLC resonant circuit, characterized in that it is a bidirectional converter. In any one of claims 1 to 3,
An interleaved converter with a secondary LLC resonant circuit, characterized in that the primary sides of the above n transformers (T ₁ , T ₂ ) additionally include separate resonant inductors (L _rP1 , L _rP2 ) instead of leakage inductance. In any one of claims 1 to 3,
An interleaved converter with a secondary LLC resonant circuit, characterized in that the secondary sides of the above n transformers (T ₁ , T ₂ ) additionally include a separate resonant inductor (L _rS ) instead of a leakage inductance. In any one of claims 1 to 3,
Interleaved converter with secondary LLC resonant circuit characterized by n=2. A converter having an interleaved phase shift control secondary LLC resonant circuit,
In forward operation, power can be transmitted by controlling the output voltage (V _{o ) from 0 voltage to the gain control range (Gv ) through interleaved phase shift control between individual primary full-bridge circuits at a constant switching frequency (f s )} near the resonant frequency (f _r ).
In the case of reverse operation, the magnetizing inductance (L _m1 , L _m1 ') of the transformer (T ₁ , T ₂ ) and the leakage inductances of the primary and secondary sides (L _l1 , L _l1 ', It includes a main circuit that can transfer power from the output (V _o ) end energy to the input end (V _in ) through variable frequency control by utilizing the LLC resonant gain characteristics with the secondary resonant capacitor (C _s ) and the _{secondary resonant capacitor} (L l2 , L l2 ').
When operating in the forward direction, the leakage inductance of the primary and secondary sides of the transformer (T ₁ , T ₂ ) (L _l1 , L _l1 ', The forward power is transmitted through interleaved phase shift control by utilizing the resonance between the secondary resonant capacitor (C _s ) and the secondary resonant capacitor (L _l2 , L _l2 '), _and when operating in the reverse direction _{, the magnetizing inductance (L m1 ,} L _m1 ') and leakage inductance (L _l1 , L _l1 ', By transmitting the reverse power through the equivalent leakage inductance (L _l2 , L _l2 ') and the secondary resonant capacitor (C _s ) and the variable switching frequency control (FM) according to the LLC resonant gain characteristics,
Interleaved converter with a secondary LLC resonant circuit characterized by having resonant gain characteristics in both directions. A converter having an interleaved phase shift control secondary LLC resonant circuit,
In forward operation, power can be transmitted by controlling the output voltage (V _{o ) from 0 voltage to the gain control range (Gv ) through interleaved phase shift control between individual primary half-bridge circuits at a constant switching frequency (f s )} near the resonant frequency (f _r ).
In the case of reverse operation, the magnetizing inductance (L _m1 , L _m1 ') of the transformer (T ₁ , T ₂ ) and the leakage inductances of the primary and secondary sides (L _l1 , L _l1 ', It includes a main circuit that can transfer power from the output (V _o ) end energy to the input end (V _in ) through variable frequency control by utilizing the LLC resonant gain characteristics with the secondary resonant capacitor (C _s ) and the _{secondary resonant capacitor} (L l2 , L l2 ').
When operating in the forward direction, the leakage inductance of the primary and secondary sides of the transformer (T ₁ , T ₂ ) (L _l1 , L _l1 ', The forward power is transmitted through interleaved phase shift control by utilizing the resonance between the secondary resonant capacitor (C _s ) and the secondary resonant capacitor (L _l2 , L _l2 '), _and when operating in the reverse direction _{, the magnetizing inductance (L m1 ,} L _m1 ') and leakage inductance (L _l1 , L _l1 ', By transmitting the reverse power through the equivalent leakage inductance (L _l2 , L _l2 ') and the secondary resonant capacitor (C _s ) and the variable switching frequency control (FM) according to the LLC resonant gain characteristics,
Interleaved converter with a secondary LLC resonant circuit characterized by having resonant gain characteristics in both directions. In clause 11 or 12,
An interleaved converter applied with a secondary LLC resonant circuit, characterized in that the above gain control range (Gv) is 'G _v = 0 to 2'. In clause 11 or 12,
The above interleaved phase shift control is an interleaved converter applied with a secondary LLC resonant circuit, characterized in that 'D _P = 0~0.5'. In clause 11 or 12,
An interleaved converter with a secondary LLC resonant circuit, characterized in that the forward operation of the main circuit has six operation modes according to phase control (D _P , PM), and during forward operation, the switching operation frequency (f _s ) performs switching operation in a frequency range lower or higher than the resonant frequency (f _r ) or within a preset range than the resonant frequency (fr), and the secondary bridge switching elements (S ₁ , S ₂ , S ₃ , S ₄ ) can be applied as synchronous rectifiers, but are turned off and perform rectification operation through anti-parallel diodes. In clause 11 or 12,
An interleaved converter with a secondary LLC resonant circuit, characterized in that the reverse operation of the main circuit operates according to variable switching frequency control (FM, fmax∼fmin), and at this time, when operating in the reverse direction, the switching operation frequency (fs) performs switching operation in the range from the maximum switching frequency (fmax) to the minimum switching frequency (fmin) in which a discontinuous resonant current flows below the resonant frequency (fr) based on the resonant frequency (fr). A power conversion device employing an interleaved converter applying a secondary LLC resonant circuit according to any one of claims 1 to 3, 11 and 12. In Article 17,
The above power conversion device is a power conversion device characterized in that it is used in any one of a microgrid system, an energy storage system (ESS), an electric vehicle OBC (One Board Charger), a battery charging/discharging system, and a fuel cell power conversion device. delete