KR102677033B1 - Systems and methods related to linear and efficient broadband power amplifiers - Google Patents

Abstract

Systems and methods related to linear efficient broadband power amplifiers are disclosed. A power amplifier (PA) system can include an input circuit configured to receive a radio-frequency (RF) signal and split the RF signal into a first portion and a second portion. The PA system may further include a Doherty amplifier circuit including a carrier amplification path coupled to the input circuit to receive the first portion and a peaking amplification path coupled to the input circuit to receive the second portion. there is. The PA system may further include an output circuit coupled to the Doherty amplifier circuit. The output circuit may include a balance to unbalance (BALUN) circuit configured to combine the outputs of the carrier amplification path and the peaking amplification path to produce an amplified RF signal.

Description

SYSTEMS AND METHODS RELATED TO LINEAR AND EFFICIENT BROADBAND POWER AMPLIFIERS}

Cross-reference to related application(s)

This application follows U.S. Provisional Application No. 61/992,842, filed May 13, 2014, under the title SYSTEMS AND METHODS RELATED TO LINEAR AND EFFICIENT BROADBAND POWER AMPLIFIERS, CIRCUITS, DEVICES AND METHODS RELATED TO COMBINERS FOR Claiming priority to U.S. Provisional Application No. 61/992,843, filed under the title DOHERTY POWER AMPLIFIERS, and U.S. Provisional Application No. 61/992,844, filed May 13, 2014, under the title SYSTEMS AND METHODS RELATED TO LINEAR LOAD MODULATED POWER AMPLIFIERS. , the disclosures of which are expressly incorporated herein by reference in their entirety.

Technology field

This disclosure relates generally to radio-frequency (RF) power amplifiers (PA).

Traditionally, it has been widely believed that Doherty PA is not suitable for linear PA applications in handsets due to its size, complexity and non-linear behavior. In practice, in base station applications, predistortion linearizers are commonly used with Doherty PAs to satisfy linearity requirements. As described herein, issues such as size, complexity and linearity associated with Doherty PAs can be appropriately addressed.

According to some implementations, the present disclosure includes: an input circuit configured to receive a radio-frequency (RF) signal and split the RF signal into a first portion and a second portion; A power amplifier (PA) system comprising a Doherty amplifier circuit including a coupled carrier amplification path and a peaking amplification path coupled to an input circuit for receiving a second portion, and an output circuit coupled to the Doherty amplifier circuit. It's about. The output circuit may include a balance to unbalance (BALUN) circuit configured to combine the outputs of the carrier amplification path and the peaking amplification path to produce an amplified RF signal.

In some embodiments, the PA system can further include a pre-driver amplifier configured to partially amplify the RF signal prior to reception by the input circuit. In some embodiments, at least one of the input circuit and the output circuit may be implemented as a lumped-element circuit.

In some embodiments, the carrier amplification path can include a carrier amplifier and the peaking amplification path can include a peaking amplifier, and the carrier amplifier and peaking amplifier each include a driver stage and an output stage. In some embodiments, the input circuit may include a modified Wilkinson power divider configured to provide DC power to each of the carrier amplifier and peaking amplifier. In some embodiments, DC power may be provided to the carrier amplifier and peaking amplifier through a choke inductance. In some embodiments, the carrier amplification path and the peaking amplification path each include a DC blocking capacitance. In some embodiments, a modified Wilkinson power divider can be further configured to provide impedance matching between the driver stages and the pre-driver amplifier. In some embodiments, the carrier amplification path and the peaking amplification path can each include an LC matching circuit with an inductive coupling to ground and a capacitance along the path.

In some embodiments, the modified Wilkinson power divider (c) may be configured to provide the required phase shifting to compensate or tune the AM-PM effect associated with the peaking amplifier. In some embodiments, the modified Wilkinson power divider may be further configured to provide the required attenuation adjustment at the input of the carrier amplifier or peaking amplifier to compensate or tune AM-AM effects associated with the carrier amplifier and peaking amplifier. there is. In some embodiments, the modified Wilkinson power divider includes a capacitance that couples the first node to ground along the carrier amplification path, and an impedance that couples the second node to ground along the peaking amplification path. In some embodiments, the modified Wilkinson power divider further includes an isolation resistor implemented between the first node and the second node, wherein the isolation resistor prevents source-pulling effects between the carrier amplification path and the peaking amplification path, or is selected to reduce

In some embodiments, the balun circuit may include an LC balun converter. In some embodiments, the peaking amplifier may be configured to behave as a short circuit or low impedance node when in the off state, and the carrier amplifier may be configured to behave as a single-section matching network with a series inductance and shunt capacitance when using an LC balun converter. It can be configured to behave as a single-ended amplifier. In some embodiments, the LC balun converter can be configured such that the impedance seen by the carrier amplifier increases when in a low power mode. In some embodiments, the impedance seen by the carrier amplifier can be approximately doubled in low power mode.

In some embodiments, the peaking amplifier may be further configured to operate in a similar manner to a push-pull amplifier where the RF current from the carrier amplifier is influenced by the RF current from the peaking amplifier. In some embodiments, push-pull operation can improve linearity by reducing even-harmonics.

In some embodiments, the LC balun converter can include a first path that couples the output of the carrier amplifier to the output node, and a second path that couples the output of the peaking amplifier to the output node. In some embodiments, each of the first path and the second path can be inductively coupled to a DC port to provide a DC feed to the output stage. In some embodiments, each of the first path and the second path can include a harmonic trap. In some embodiments, the harmonic trap may include a second harmonic trap having an LC shunt and a series inductance to ground. In some embodiments, the second path can include a shunt capacitance and a series capacitance configured to provide phase compensation to the output of the peaking amplifier. In some embodiments, at least one of the shunt capacitance and series capacitance may be a surface-mount technology (SMT) capacitor.

In some embodiments, the LC balun converter can be configured to provide reduced losses in the carrier amplification path to maintain high efficiency in back-off and in high power mode.

In some embodiments, load modulation of the peaking amplifier is such that the impedance loci for the peaking amplifier are optimal when the peaking amplifier contributes approximately the same power as the carrier amplifier from a circuit that is approximately short-circuited when the peaking amplifier is in the off state. It can be configured to proceed with the load impedance.

In some embodiments, the input circuit may be a wideband circuit, at least in part due to a lead-lag network configured to provide a wideband phase shift.

In some embodiments, the input circuit is configured to provide broadband performance, while providing reactive to real impedance matching, and isolation between the carrier amplifier and the peaking amplifier.

In some implementations, the disclosure relates to a method for amplifying a radio-frequency (RF) signal, comprising providing a Doherty amplifier circuit having a carrier amplification path and a peaking amplification path, receiving the RF signal. splitting the RF signal into a first portion and a second portion, the first portion being provided to a carrier amplification path and the second portion being provided to a peaking amplification path, and a balance to unbalance (BALUN) circuit. and calculating an amplified RF signal by combining the outputs of the carrier amplification path and the peaking amplification path.

In some implementations, the present disclosure relates to a power amplifier module. A power amplification module may include a packaging substrate configured to receive a plurality of components, and a power amplifier (PA) system implemented on the packaging substrate. The PA system may include an input circuit configured to receive an RF signal and split the RF signal into a first portion and a second portion. The PA system may further include a Doherty amplifier circuit having a carrier amplification path coupled to the input to receive the first portion and a peaking amplification path coupled to the input circuit to receive the second portion. The PA system may further include an output circuit coupled to the Doherty amplifier circuit. The output circuit may include a balance to unbalance (BALUN) circuit configured to combine the outputs of the carrier amplification path and the peaking amplification path to produce an amplified RF signal. The power amplification module may further include a plurality of connectors configured to provide electrical connections between the PA system and the packaging substrate.

In some implementations, the disclosure relates to a wireless device including a transceiver configured to generate a radio-frequency signal, a power amplification (PA) module in communication with the transceiver, and an antenna in communication with the PA module, the antenna It is configured to facilitate transmission of the amplified RF signal. The PA module may include an input circuit configured to receive an RF signal and split the RF signal into a first portion and a second portion. The PA module may further include a Doherty amplifier circuit having a carrier amplification path coupled to the input circuit to receive the first portion and a peaking amplification path coupled to the input circuit to receive the second portion. The PA module may further include an output circuit coupled to the Doherty amplifier circuit. The output circuit may include a balance to unbalance (BALUN) circuit configured to combine the outputs of the carrier amplification path and the peaking amplification path to produce an amplified RF signal. The transceiver communicates with the PA module and may further include an antenna configured to facilitate transmission of the amplified RF signal.

According to some implementations, the present disclosure relates to a signal combiner including a balun converter circuit having a first coil and a second coil. The first coil may be implemented between the first port and the second port. The second coil is implemented between the third and fourth ports. The first port and the third port are coupled by a first capacitance. The second port and the fourth port are coupled by a second capacitance. The first port is configured to receive a first signal. The fourth port is configured to receive a second signal. The second port is configured to produce a combination of the first signal and the second signal. The signal coupler further includes a termination circuit that couples the third port to ground.

In some embodiments, the first port can be configured to receive a carrier-amplified signal from a Doherty power amplifier (PA), and the fourth port can be configured to receive a peaking-amplified signal from the Doherty PA. In some embodiments, the termination circuit may include a capacitor. In some embodiments, the capacitor may have a capacitance approximately equal to the reciprocal of [2 x π x the operating frequency of the Doherty PA x the characteristic impedance of the load coupled to the Doherty PA].

In some embodiments, the first port can be configured to receive a peaking-amplified signal from a Doherty power amplifier (PA), and the fourth port can be configured to receive a carrier-amplified signal from the Doherty PA. In some embodiments, the termination circuit may include an inductor. In some embodiments, the inductor may have an inductance approximately equal to [characteristic impedance of the load coupled to the Doherty PA / (2 x π x the operating frequency of the Doherty PA)].

In some embodiments, the S-parameter between the first one of the ports and the second one of the ports may be approximately equal to (1+j)/2. In some embodiments, the S-parameter between the first one of the ports and the second one of the ports may be approximately equal to (1-j)/2. In some embodiments, the S-parameter matrix of S-parameters between ports may only include values approximately 0, (1+j)/2, and (1-j)/2.

In some embodiments, the balun converter circuit may be implemented as an integrated passive device. In some embodiments, the integrated passive device further implements an auto-converter based impedance matching circuit.

In some implementations, the present disclosure relates to a power amplifier module including a packaging substrate configured to receive a plurality of components and a signal combiner implemented on the packaging substrate. The signal combiner includes a balun converter circuit having a first coil and a second coil. The first coil is implemented between the first port and the second port. The second coil is implemented between the third and fourth ports. The first port and the third port are coupled by a first capacitance. The second port and the fourth port are coupled by a second capacitance. The first port is configured to receive a first signal. The fourth port is configured to receive a second signal. The second port is configured to produce a combination of the first signal and the second signal. The signal coupler further includes a termination circuit that couples the third port to ground.

In some embodiments, the balun converter circuit may be implemented as an integrated passive device. In some embodiments the integrated passive device may further implement an auto-converter based impedance matching circuit.

In some embodiments, the PA module may further include a Doherty PA implemented on a packaging substrate. The Doherty PA may have a carrier amplification path that produces a carrier-amplified signal and a peaking amplification path that produces a peaking-amplified signal. In some embodiments, the first port can be configured to receive a carrier-amplified signal and the fourth port can be configured to receive a peaking-amplified signal. In some embodiments, the termination circuit may include a capacitor having a capacitance approximately equal to the reciprocal of [2 x π x the operating frequency of the Doherty PA x the characteristic impedance of the load coupled to the Doherty PA]. In some embodiments, the first port can be configured to receive a peaking-amplified signal and the fourth port can be configured to receive a carrier-amplified signal. In some embodiments, the termination circuit may include an inductor having an inductance approximately equal to [characteristic impedance of the load coupled to the Doherty PA / (2 x π x the operating frequency of the Doherty PA)].

In some embodiments, the S-parameter matrix of S-parameters between ports includes only values approximately 0, (1+j)/2, and (1-j)/2.

In some implementations, the present disclosure relates to a wireless device that includes a transceiver configured to generate a radio-frequency (RF) signal. The wireless device further includes a power amplifier (PA) module in communication with the transceiver. The PA module includes an input circuit configured to receive an RF signal and split the RF signal into a first portion and a second portion. The PA module further includes a Doherty PA having a carrier amplification path coupled to the input circuit to receive the first portion and a peaking amplification path coupled to the input circuit to receive the second portion. The PA module further includes an output circuit coupled to the Doherty amplifier circuit. The output circuit includes a balun converter circuit having a first coil and a second coil. The first coil is implemented between the first port and the second port. The second coil is implemented between the third and fourth ports. The first port and the third port are coupled by a first capacitance. The second port and the fourth port are coupled by a second capacitance. The first port is configured to receive a first signal through a carrier amplification path. The fourth port is configured to receive the second signal through the peaking amplification path. The second port is configured to produce a combination of the first signal and the second signal as an amplified RF signal. The wireless device further includes an antenna in communication with the PA module. The antenna is configured to facilitate transmission of the amplified RF signal.

In some implementations, the present disclosure relates to a method for amplifying a radio-frequency (RF) signal. The method includes providing a Doherty amplifier circuit having a carrier amplification path and a peaking amplification path, receiving an RF signal, splitting the RF signal into a first portion and a second portion, the first portion providing a carrier amplification path. and the second part is provided to the peaking amplification path, and using a balun converter circuit to combine the output of the carrier amplification path and the output of the peaking amplification path to produce an amplified RF signal. The balun converter circuit includes a first coil and a second coil. The first coil is implemented between the first port and the second port. The second coil is implemented between the third and fourth ports. The first port and the third port are coupled by a first capacitance. The second port and the fourth port are coupled by a second capacitance. The first port is configured to receive the output of the carrier amplification path. The fourth port is configured to receive the output of the peaking amplification path. The second port is configured to produce an amplified RF signal.

According to some implementations, the present disclosure relates to a power amplifier (PA) system including an input circuit configured to receive a radio-frequency (RF) signal and split the RF signal into a first portion and a second portion. . The PA system further includes a Doherty amplifier circuit including a carrier amplifier coupled to the input circuit to receive the first portion and a peaking amplifier coupled to the input circuit to receive the second portion. The first part and the second part have different phases and different powers. The PA system further includes an output circuit coupled to the Doherty amplifier circuit. The output circuit is configured to combine the outputs of the carrier amplifier and the peaking amplifier to produce an amplified RF signal.

In some embodiments, the input circuit can include a phase shifter configured to cause the first portion and the second portion to have different phases. In some embodiments, the phase shifter and peaking amplifier may be implemented within the peaking amplification path. In some embodiments, the first portion and the second portion may be out-of-phase by 10 to 20 degrees. In some embodiments, different phases may reduce at least one of AM/AM distortion or AM/PM distortion compared to the same phases.

In some embodiments, the input circuit can include an attenuator configured to cause the first portion and the second portion to have different powers. In some embodiments, an attenuator and carrier amplifier may be implemented in the carrier amplification path. In some embodiments, different powers may reduce at least one of AM/AM distortion or AM/PM distortion compared to the same powers.

In some embodiments, the input circuit may include a pre-driver amplifier.

In some embodiments, the peaking amplifier includes a driver stage configured to operate in a first biasing mode and an output stage configured to operate in a first biasing mode. In some embodiments, the first biasing mode is a class B biasing mode. In some embodiments, Class B biasing mode increases the PAE of the peaking amplifier compared to Class AB biasing mode. In some embodiments, the carrier amplifier includes a driver stage configured to operate in a second biasing mode. In some embodiments, the second biasing mode is a class AB biasing mode. In some embodiments, the carrier amplifier further includes an output stage configured to operate in a first biasing mode. In some embodiments, the carrier amplifier further includes an output stage configured to operate in a second biasing mode.

In some implementations, the present disclosure relates to a power amplifier (PA) module. The PA module includes a packaging substrate configured to receive a plurality of components and a PA system implemented on the packaging substrate. The PA system includes an input circuit configured to receive a radio-frequency (RF) signal and split the RF signal into a first portion and a second portion. The PA system further includes a Doherty amplifier circuit including a carrier amplifier coupled to the input circuit to receive the first portion, and a peaking amplifier coupled to the input circuit to receive the second portion. The first part and the second part have different phases and different powers. The PA system further includes an output circuit coupled to the Doherty amplifier circuit. The output circuit is configured to combine the outputs of the carrier amplifier and the peaking amplifier to produce an amplified RF signal.

In some embodiments, at least one of the input circuitry or the output circuitry may be implemented as an integrated passive device. In some embodiments, at least one of the input circuitry or the output circuitry may be implemented on a single GaAs die.

In some implementations, the present disclosure relates to wireless devices. A wireless device includes a transceiver configured to generate a radio-frequency (RF) signal. The wireless device includes a power amplifier (PA) module in communication with a transceiver. The PA module includes an input circuit configured to receive an RF signal and split the RF signal into a first portion and a second portion. The PA module includes a Doherty amplifier circuit including a carrier amplifier coupled to the input circuit to receive the first portion and a peaking amplifier coupled to the input circuit to receive the second portion. The first part and the second part have different phases and different powers. The PA module includes an output circuit coupled to a Doherty amplifier circuit. The output circuit is configured to combine the outputs of the carrier amplifier and the peaking amplifier to produce an amplified RF signal. The wireless device further includes an antenna in communication with the PA module. The antenna is configured to facilitate transmission of the amplified RF signal.

In some implementations, the present disclosure relates to a method for amplifying a radio-frequency (RF) signal. The method includes providing a Doherty amplifier circuit having a carrier amplification path and a peaking amplification path, receiving an RF signal, splitting the RF signal into a first portion and a second portion, the first portion providing a carrier amplification path. , the second part is provided to the peaking amplification path, the first part and the second part have different phases and different powers, and an RF signal amplified by combining the output of the carrier amplification path and the output of the peaking amplification path. It includes calculating .

For the purpose of summarizing the disclosure, certain aspects, advantages, and novel features of the inventions are described herein. It should be understood that all of these advantages may not be achieved in any particular embodiment of the invention. Accordingly, the invention, as taught or suggested herein, may be implemented or practiced in a manner that achieves or optimizes one advantage or group of advantages as taught herein without necessarily achieving other advantages. there is.

1 shows that, in some embodiments, a power amplifier may be implemented as a linear, efficient, wideband power amplifier.
Figure 2 shows an example architecture of a power amplifier including a carrier amplification path and a peaking amplification path.
3 shows an exemplary configuration of a modified Wilkinson-type power divider.
4 shows an example configuration of a combiner that can provide balance to unbalance (BALUN) converter functionality.
5 shows first example load modulation profiles of a peaking amplifier and carrier amplifiers using a balun converter configuration.
6 shows second example load modulation profiles of carrier amplifiers and peaking amplifiers using a balun converter configuration.
7 shows an example configuration of a power amplifier including a modified Wilkinson-type power divider.
Figure 8 shows an exemplary wideband phase shift response.
9 shows example impedance responses including harmonic traps.
10 shows example adjacent channel leakage-to-power ratio (ACLR) curves and power-added efficiency (PAE) curves.
11 illustrates a wireless device having one or more features described herein.
Figure 12 shows an example combiner configuration with both the carrier amplifier and peaking amplifier in the on state.
Figure 13 shows an example combiner configuration with the carrier amplifier in the on state and the peaking amplifier in the off state.
Figure 14 shows an exemplary Doherty combiner comprising two or more quarter wave transmission lines.
Figure 15 shows an example Smith chart for the combiner of Figure 14.
Figure 16 shows an exemplary Doherty coupler including a 3 dB coupler.
Figure 17 shows an example Smith chart for the combiner of Figure 16.
Figure 18 shows an example hybrid circuit that can be used as a Doherty combiner.
Figure 19 shows another example hybrid circuit that can be used as a Doherty combiner.
Figure 20 shows an example S-parameter matrix for the combiner of Figure 16.
Figure 21 shows an example S-parameter matrix for the combiner of Figure 18.
FIG. 22 shows an exemplary Doherty combiner configuration using the hybrid circuit of FIG. 18.
Figure 23 shows impedance traces resulting from Doherty action in the coupler of Figure 22.
FIG. 24 shows another example Doherty combiner configuration using the hybrid circuit of FIG. 18.
Figure 25 shows an example of integration of hybrid circuit and auto-converter based impedance matching as an integrated passive device (IPD).
Figure 26 shows an example Smith chart of an inverted load-modulation trajectory.
Figure 27 shows another example of integration of a hybrid circuit as an IPD.
Figure 28 shows an example architecture of a power amplifier in which a Doherty combiner having one or more features as described herein may be implemented.
Figure 29 depicts a wireless device having one or more features disclosed herein.
30 illustrates an example architecture of a power amplifier (PA) having one or more features as described herein.
Figure 31 shows an example of a combiner circuit for a Doherty PA.
Figure 32 shows an example splitter circuit for a Doherty PA.
FIG. 33 shows an example of a power splitter that can be used as the splitter of FIG. 30.
Figure 34 shows another example of a power splitter that can be used as the splitter of Figure 30.
Figure 35 shows an example of a coupler that can be used as the coupler of Figure 30.
Figure 36 shows another example of a coupler that can be used as the coupler of Figure 30.
Figure 37 shows an example of a low headroom class AB bias circuit.
Figure 38 shows an example of a low headroom class B bias circuit.
Figure 39 shows an example of the beneficial effects of using class B biasing of the driver stage for a peaking amplifier.
Figure 40 shows another example of the beneficial effects of using class B biasing of the driver stage for a peaking amplifier.
Figure 41 shows an example of a linearization effect that can be achieved by introducing a phase shift between the RF signals associated with carrier amplification and peaking amplification.
Figure 42 shows an example of the linearization effect that can be achieved by introducing non-uniform power splitting between the RF signals associated with carrier amplification and peaking amplification.
Figure 43 shows an example of a combined linearization effect that can be achieved by combining phase shift and non-uniform power splitting.
44 shows example plots of power-added efficiency (PAE) and adjacent channel power (ACP) at various operating frequencies for a front-end module (FEM).
Figure 45 shows a wireless device having one or more features described herein.

The subheadings provided herein, if present, are for convenience only and do not necessarily affect the scope or meaning of the claimed invention. Systems, devices, circuits and methods related to radio-frequency (RF) power amplifiers (PA) are described herein.

Power amplification using balun converters

1 , in some embodiments, a PA 100 having one or more features as described herein may be configured to provide broadband capability with either or both desirable linearity and efficiency. PA 100 is shown to receive an RF signal (RF_IN) and generate an amplified signal (RF_OUT). Various examples related to such PAs are described in much more detail herein.

2 shows an example architecture of PA 100 having one or more features as described herein. The architecture shown is the Doherty PA architecture. Although the various examples are described in the context of this Doherty PA architecture, it will be understood that one or more features of the present disclosure may also be implemented in other types of PA systems.

The exemplary PA 100 is shown as including an input port (RF_IN) for receiving an RF signal to be amplified. This input RF signal may be partially amplified by pre-driver amplifier 102 before splitting into carrier amplification path 110 and peaking amplification path 130. This division may be accomplished by divider 104. Examples relating to splitter 104 are described in greater detail herein, including examples relating to Figures 3 and 7.

2, carrier amplification path 110 is shown as comprising amplification stages collectively indicated as attenuator 112 and 114. Amplification stages 114 are shown including a driver stage 116 and an output stage 120. Driver stage 116 is shown as biased by bias circuit 118 and output stage 120 is shown as biased by bias circuit 122. In some embodiments, there may be more or fewer amplification stages. In various examples described herein, amplification stages 114 are sometimes described as amplifiers, but it will be understood that such amplifiers may include one or more stages.

In FIG. 2 , peaking amplification path 130 is shown as including a phase shifting circuit 132 and amplification stages 134 . Amplification stages 134 are shown including a driver stage 136 and an output stage 140. Driver stage 136 is shown as biased by bias circuit 138 and output stage 140 is shown as biased by bias circuit 142. In some embodiments, there may be more or fewer amplification stages. In various examples described herein, amplification stages 134 are sometimes described as amplifiers, although it will be understood that such amplifiers may include one or more stages.

2 shows that the carrier amplification path 110 and the peaking amplification path 130 may be combined by a combiner 144 to produce an amplified RF signal at the output port (RF_OUT). Examples relating to coupler 144 are described in greater detail herein, including examples relating to Figures 4 and 7.

In some embodiments, splitter 104 of FIG. 2 may be implemented as a lumped-element power splitter. This power splitter may be implemented as a modified Wilkinson-type power splitter configured to provide DC power to each of the driver stages (e.g., 116, 136 in FIG. 2). FIG. 3 shows an example configuration of a modified Wilkinson-type power splitter 104 that may be implemented as the splitter 104 of FIG. 2 . Figure 7 shows an example of how a modified Wilkinson-type power divider 104 may be implemented in the circuit example of PA 100 of Figure 2.

3, the modified power divider 104 is shown including an input port 150 configured to receive an input RF signal. As shown in the example PA circuit 100 of FIG. 7, input port 150 may be coupled to the collector of transistor Q0 of pre-driver amplifier 102. Input port 150 is further shown coupled to splitter node 156 via node 152. Node 152 is shown coupled to DC supply port 154 via inductance L1 (eg, an inductor). DC power for each of the driver stages can be obtained through DC supply port 154. In Figure 3, L1 may be part of a modified Wilkinson-type splitter that matches the impedance looking into the splitter to the impedance presented to the pre-driver PA collector. At the same time, L1 can serve as a DC path to the pre-driver.

In Figure 3, the carrier amplification path (110 in Figure 2) is shown to include a path from splitter node 156 to node 160 through capacitance C1, node 158, and capacitance C3. Node 160 may or may not be connected to port 162 to facilitate coupling to a forward path carrier amplifier (e.g., 114 in FIG. 2). Node 158 is shown coupled to ground through capacitance C2. Node 160 is shown coupled to ground through inductance L2.

In Figure 3, the peaking amplification path (130 in Figure 2) is shown to include a path from splitter node 156 to node 166 through capacitance C4, node 164, and capacitance C5. Node 166 may or may not be connected to port 168 to facilitate coupling to a forward path peaking amplifier (e.g., 134 in FIG. 2). Node 164 is shown coupled to ground through inductance L3. Node 166 is shown coupled to ground through inductance L4.

In Figure 3, resistor R1 is shown coupling node 158 of the carrier amplification path and node 164 of the peaking amplification path. Resistor R1 may be selected to function as an isolation resistor to prevent or reduce source-pulling effect(s) from the carrier and/or peaking amplifiers.

3, capacitance C1 may be selected to provide DC blocking functionality for the carrier amplification path. Similarly, capacitance (C4) can be selected to provide DC blocking functionality for the peaking amplification path.

3, capacitance C3 and inductance L2 may be selected to provide impedance matching between the pre-driver amplifier (e.g., 102 in FIGS. 2 and 7) and the carrier amplifier 114. Similarly, C5 and inductance L4 may be selected to provide impedance matching between the pre-driver amplifier (e.g., 102 in FIGS. 2 and 7) and peaking amplifier 134.

3, the capacitance (C2) associated with the carrier amplification path and the inductance (L3) associated with the peaking amplification path can be selected to provide the desired phase shifting between the two paths. This phase shift may be selected to compensate and/or tune AM-PM phenomena associated with peaking amplifier 134, for example. In Figure 2, this phase-shifting functionality is shown as block 132 along peaking amplification path 130.

In some embodiments, and as shown in Figure 2, attenuator 112 may be connected to carrier amplification path 110 (e.g., before carrier amplifier 114) or peaking amplification path 130 (e.g., It may be provided along (in front of) the peaking amplifier 134. Such attenuators may be configured to provide a desired attenuation adjustment to compensate and/or tune AM-AM phenomena associated with either or both the carrier amplifier and the peaking amplifier. These attenuators can also promote uneven power splitting between the two amplification paths.

Note that prior corrections and/or tuning of AM-AM and/or AM-PM effects can result in PA 100 of FIGS. 2 and 7 being substantially linear. This linearity can be achieved without requiring digital pre-distortion, which typically reduces the efficiency of the PA system for portable wireless devices and the applicability of the PA system in amplifiers. Additionally, the linearity achieved by PA 100 of FIGS. 2 and 7 (without digital pre-distortion) may be similar to the linearity performance associated with a class AB single-ended amplifier.

In some embodiments, combiner 144 of FIG. 2 may be implemented as, or similar to, a lumped constant balun (balanced to unbalanced) converter. Figure 4 shows an example configuration of combiner 144 that can provide this balun converter functionality. Figure 7 shows an example of how combiner 144 may be implemented in the circuit example of PA 100 of Figure 2.

In FIG. 4 , combiner 144 is shown as comprising a portion of peaking amplification path 130 and a portion of a carrier amplification path (e.g., 110 in FIG. 2 ) connected at peaking combining node 186 . Coupling node 186 is shown coupled to output port 198 (RF_OUT in FIGS. 2 and 7).

In Figure 4, a portion of the carrier amplification path is shown coupling node 186 and node 182 through inductance L13. Node 182 may or may not be connected to port 180 to facilitate coupling to a forward path carrier amplifier (e.g., 114 in FIG. 2). Node 182 is shown coupled to ground through capacitance C11 and inductance L12. Node 182 is also shown coupled to port 184 via inductance L11.

4, a portion of the peaking amplification path is shown coupling nodes 186 and 192 through inductance L16, node 196, and capacitance C14. Node 192 may or may not be connected to port 190 to facilitate coupling to a forward path peaking amplifier (e.g., 134 in FIG. 2). Node 192 is shown coupled to ground through capacitance C12 and inductance L15. Node 192 is also shown coupled to port 194 via inductance L14. Node 196 is shown coupled to ground through capacitance C13.

In FIG. 4 , node 182 may be coupled to the collector of the output stage of carrier amplifier 114 (e.g., 120 in FIG. 2 ) via port 180 . Accordingly, a DC feed may be provided to the output stage 120 of the carrier amplifier 114 through port 184 and inductance L11. Similarly, node 192 may be coupled to the collector of the output stage of peaking amplifier 134 (e.g., 140 in FIG. 2) via port 190. Accordingly, a DC feed may be provided to the output stage 140 of peaking amplifier 134 through port 194 and inductance L14.

4, capacitance C11, inductance L12, and inductance L13 may be selected to function as a second harmonic trap for the output of carrier amplifier 114. Similarly, capacitance C12, inductance L15, and inductance L16 may be selected to function as a second harmonic trap for the output of peaking amplifier 134.

4, capacitance C13 and capacitance C14 may be selected to provide phase compensation for the output of peaking amplifier 134. In some embodiments, C13 and C14 may be implemented as surface-mount technology (SMT) capacitors. In these embodiments, combiner 144 can be implemented as a broadband power combiner, using as little as two SMT capacitors.

The example combiner 144 of FIG. 4 can provide desirable functionality for operations of Doherty PA architectures. For example, the peaking amplifier in a Doherty PA architecture is typically required to behave as a short circuit or very low impedance path when it is turned off, and the carrier amplifier is typically required to behave as a short circuit or a very low impedance path when using an LC balun configuration, and the carrier amplifier is typically required to behave as a short circuit or very low impedance path when it is turned off, and the carrier amplifier is typically required to behave as a short circuit or a very low impedance path when it is turned off, and the carrier amplifier is normally required to behave as a short circuit or very low impedance path when it is turned off, and the carrier amplifier is typically required to behave as a short circuit or very low impedance path when it is turned off. For example, it typically operates as a single-ended amplifier with an equivalent circuit similar or identical to series L and shunt C). In this condition, the impedance seen by the carrier amplifier can be doubled.

When the peaking amplifier is turned on, the PA system can operate in a manner similar to a “push-pull” amplifier. For example, RF current from a carrier amplifier can see current from a peaking amplifier. In this condition, linearity can be improved because the even harmonic content can be reduced.

As described herein, combiner 144 with an exemplary LC balun configuration can be implemented in a compact form, using as few as two SMT components (e.g., capacitors). This combiner may include RF chokes and harmonic traps, configured to provide impedance matching from, for example, a 50-ohm output to the transistor-collectors of the peaking amplifier and carrier amplifier.

As described herein, combiner 144 with an exemplary LC balun configuration can be implemented to reduce losses in the carrier amplifier path compared to other Doherty topologies. This feature in turn can facilitate maintaining high efficiency in back-off and high power modes. Additionally, the LC balun configuration can provide the required or desired impedance and phase adjustment for the carrier amplifier. This feature can be important when designing an asymmetrically loaded Doherty transmitter.

In some embodiments, the load modulation associated with a peaking amplifier as described herein is generally opposite to that in conventional Doherty transmitters. Figure 5 shows load modulation profiles for the carrier amplifier 200 and peaking amplifier 202 of a conventional Doherty transmitter using a balun converter configuration. Figure 6 shows load modulation profiles for the carrier amplifier 204 and peaking amplifier 206 of a Doherty transmitter using a balun converter configuration as described herein (e.g., Figure 7). For the peaking amplifiers of Figures 5 and 6, the impedance loci varies from their respective short circuit conditions (e.g., when the peaking amplifier is turned off) to their respective optimal load impedance conditions (e.g. For example, when the peaking amplifier contributes the same power as the carrier amplifier), it can be seen that it progresses in the opposite direction. For the conventional example of Figure 5, the impedance (loci) of the peaking amplifier progresses in the same direction as the impedance of the carrier amplifier as power increases. For the example of Figure 6, the impedance (loci) of the peaking amplifier goes in the opposite direction of the impedance of the carrier amplifier as power increases.

7 shows an example of a PA 100 having one or more features as described herein. The PA may include a pre-driver amplifier 102, such as a one-stage single-ended amplifier. The output of pre-driver amplifier 102 is shown as being provided to a divider 104, such as the example described with respect to FIG. 3. The divided outputs of splitter 104 are shown as being provided to carrier amplifier 114 and peaking amplifier 134. The outputs of carrier amplifier 114 and peaking amplifier 134 are shown coupled by combiner 144, such as the example described with respect to FIG. 4.

In the example PA 100 of FIG. 7, splitter 104 and combiner 144 may produce broadband combining. For example, divider 104 is wideband in nature, for example due to the lead-lag circuitry that provides wideband phase shift. An example of this phase shift response is shown as curve 250 in FIG. 8. The exemplary response curve 250 represents the typical phase difference between matching reactive base impedances and the driver amplifier collector. It is further noted that divider 104 provides advantageous features such as imaginary to real impedance matching, isolation between the carrier amplifier and peaking amplifier, and still yielding wideband performance.

In another example, combiner 144 with its own LC balun configuration may also contribute to the wideband performance of PA 100. As described herein, an LC balun may include harmonic traps configured to maintain the impedance trajectory within lower constant Q circles. Examples of these impedance responses are shown in Figure 9 as curves 260, 262, and 264. Example response curves 260, 262, 264 represent collector load impedance versus frequency for different ZP values. ZP1 represents the load impedance seen by the carrier PA collector when both the carrier PA and peaking PA are turned on (running), which is for example about 5.7+j0.119 ohms. ZP2 is the collector impedance at the peaking PA collector, similar to the previous case (i.e., the same impedance when both PAs are on). ZP4 is the impedance seen by the carrier PA collector when the peaking PA is off, which is effectively doubled to, for example, approximately 10.86+j0.058 ohms. This feature actually improves the PA architecture bandwidth because the impedances versus frequency do not spread along the Smith chart.

A PA architecture having one or more features as described herein, including the examples in FIGS. 1-4 and 7, can be configured to yield excellent linear and efficient wideband performance. For example, for a -37-dBc adjacent channel leakage-power ratio (ACLR) of 21% using an LTE signal (e.g., 10-MHz BW, QPSK, 12 RB), Relative bandwidth can be achieved. Figure 10 shows ACLR curves and power-added efficiency (PAE) curves for different samples. The top set of curves 270 and 292 are for power added efficiency (PAE) for 27.5 and 27 dBm output power levels, respectively. The middle set of curves 274, 276 are for ACLR1 for 27.5 and 27 dBm output power levels, respectively. The dashed curve 278 is for ACLR2 for 27.5 dBm output power. In the context of ACLR performance, the 37-dBc ACLR bandwidth at 27-dBm output power is approximately 525 MHz (e.g., between markers “m39” and “m38”), which is approximately 2,500 MHz (e.g., between markers “m39” and “m38”). It can be seen that it is approximately 21% of the center frequency of "m48"). Note that the bandwidth can be much wider if the ACLR level is allowed to increase.

In some implementations, a device and/or circuit having one or more features described herein may be included in an RF device, such as a wireless device. These devices and/or circuits may be implemented directly within a wireless device, in modular form as described herein, or some combination thereof. In some embodiments, such wireless devices may include, for example, cellular phones, smart-phones, hand-held wireless devices with or without phone functionality, wireless tablets, etc.

11 schematically depicts an example wireless device 400 having one or more advantageous features described herein. For example, one or more PAs 110, collectively referred to as PA architecture 100, may include one or more features as described herein. Such PAs may facilitate multi-band operation of wireless device 400, for example.

PAs 110 may receive their respective RF signals from transceiver 410, which may be configured and operable to generate RF signals to be amplified and transmitted, and to process the received signals. Transceiver 410 is shown interacting with a baseband subsystem 408 that is configured to provide switching between data and/or voice signals suitable for a user and RF signals suitable for transceiver 410. Transceiver 410 is also shown connected to a power management component 406 that is configured to manage power for operation of wireless device 400. This power management unit may also control the operations of the baseband subsystem 408 and PAs 110.

Baseband subsystem 408 is shown as connected to user interface 402 to facilitate various input and output of voice and/or data provided to and received from a user. Baseband subsystem 408 may also be connected to memory 404, which is configured to store data and/or instructions to facilitate operation of wireless device 400 and/or provide storage of information to a user. there is.

In the example wireless device 400, the outputs of the PAs 110 are matched (via matching circuits 420) and their respective duplexers 412a-412d and band-select switch 414. It is shown as being routed to antenna 416 through. Band-select switch 414 may be configured to allow selection of an operating band. In some embodiments, each duplexer 412 may be allowed to transmit and receive operations simultaneously using a common antenna (e.g., 416). In Figure 11, the received signals are shown as being routed to “Rx” paths (not shown), which may include, for example, a low noise amplifier (LNA).

A number of different wireless device configurations may utilize one or more features described herein. For example, a wireless device need not be a multi-band device. In another example, a wireless device may include additional antennas, such as a diversity antenna, and other connectivity features such as Wi-Fi, Bluetooth, and GPS.

Signal combining using coiled balun transducers

The combiner can be implemented as part of a Doherty PA and is commonly used to provide a number of functions. For example, the combiner can be configured to provide the same power coupling when the PA is operating at full power. Figure 12 shows an example of this configuration where the combiner can operate as a conventional power combiner. In Figure 12, the values of various performance and operating parameters are examples; It can be adjusted appropriately for different applications.

Accordingly, in Figure 12, the configuration 2100 is illustrated when both the carrier amplifier 2110 and the peaking amplifier 2112 are on. In some implementations, carrier amplifier 2110 and peaking amplifier are saturated and have a power-added efficiency (PAE) of greater than 50%. The outputs of the carrier amplifier 2110 and the peaking amplifier 2112 are supplied to the respective input ports 2131 and 2132 of the transmission line coupler 2120. At both the first input port 2131 and the second input port 2132, there may be an impedance of 50 ohms. Transmission line coupler 2120 has a 50-ohm transmission line 2121 coupled between the first input port 2131 and the second input port 2132, and the second input port 2132 and the output port 2133. Includes a coupled 35.5-ohm transmission line 2122. The input of 35.5-ohm transmission line 2122 may present an impedance of 25 ohms.

In another example, a coupler may be configured to provide impedance transfer between a PA and a load coupled to the PA. For example, a 2:1 impedance transfer can be implemented from the load to the output of the carrier amplifier when the peaking amplifier is idle. This transfer functionality is depicted in Figure 13. Again, the values of various performance and operating parameters are examples; It can be adjusted appropriately for different applications. The foregoing functionality may be desirable for as small a bandwidth as possible to achieve economical coverage of multiple operating frequencies using a single amplifier.

Accordingly, in Figure 13, configuration 2150 is illustrated with carrier amplifier 2110 in the on state and peaking amplifier 2112 in the off state. In some implementations, carrier amplifier 2110 is saturated and has a PAE of greater than 50%. In this configuration 2150, an impedance of 100 ohms may be present at the first input port 2131 and a very high impedance (nearly open) may be present at the second input port 2132.

Figure 14 shows an example of a common Doherty coupler 2120 comprising two or more quarter wave transmission lines 2121, 2122 configured in such a way that both coupling and impedance conversion functions are achieved. These implementations are typically relatively bulky, especially at low frequencies. Accordingly, this combiner 2120 may be used in devices such as radio-frequency integrated circuits (RFICs), monolithic microwave integrated circuits (MMICs), and other RF modules. It may not be particularly suitable for certain applications. The impedance spread versus frequency for the Doherty coupler 2120 of FIG. 14 is shown in the example Smith chart 2144 of FIG. 15.

Other types of Doherty combiners may be based on lumped elements. Most of these implementations are limited to relatively narrow operating bands.

16 shows another example of a Doherty coupler 2220 using a 3 dB coupler 2221 with a nearly open-ended impedance at the isolation port 2222. Although this implementation is more compact than the example coupler 2120 of FIG. 14, it is still typically too large for applications such as RFICs, MMICs, and other RF modules at low frequencies due to its quarter-wave wavelength. big. The coupler 2220 has a first port coupled to the first input port 2231 of the coupler 2220, a second port coupled to the second input port 2232 of the coupler 2220, and a second port coupled to the first input port 2231 of the coupler 2220. A 3 db coupler 2221 having a third port coupled to output port 2233, and a fourth port coupled to a substantially open-ended impedance (e.g., split port 2222). The impedance spread versus frequency for the Doherty coupler 2220 of FIG. 16 is shown in the example Smith chart 2244 of FIG. 17.

Figures 18 and 19 show examples of hybrid circuits that can be used as Doherty combiners. This hybrid circuit can be configured to be particularly suitable for applications such as RFIC, MMIC and other RF modules. Figure 18 shows a schematic representation of this hybrid circuit, and Figure 19 shows an example layout of this hybrid circuit.

The hybrid circuit of Figures 18 and 19 can be implemented as a semi-lumped 90 degree hybrid based on a balun. Due to the compact nature of the balun used, this design can be easily implemented on insulating/semi-insulating substrates such as silicon, GaAs and IPD (eg glass or silicon).

In the hybrid circuit of Figures 18 and 19, the values of various performance and operating parameters are examples; It can be adjusted appropriately for different applications.

Accordingly, in FIG. 18, the signal combiner 2320 is shown as including a first port 2331, a second port 2332, a third port 2333, and a fourth port 2334. The first capacitor 2322 couples the first port 2331 and the second port 2332. The second capacitor 2323 couples the third port 2333 and the fourth port 2334. Signal combiner 2320 also includes a converter 2321 having four ports each coupled to four ports 2331-2334 of signal combiner 2320. 19, a substantially similar signal coupler 2390 is illustrated comprising a balun converter 2391 including a first coil and a second coil.

Figure 20 shows an example S-parameter (variance parameter) matrix that may represent the example of Figures 16, and Figure 21 shows an example S-parameter matrix that may represent the example of Figures 18 and 19. It can be seen that the S-parameter matrix of FIG. 21 is significantly different from the S-parameter matrix of FIG. 20. In the example of Figure 16, an open termination at the isolation port may result in Doherty operation. 18 and 19, specific terminations may be provided at the isolation ports to achieve Doherty operation. Examples of terminations are described in much greater detail herein.

It can be seen that in some embodiments, this particular termination may be implemented as a capacitor (eg, a capacitor) whose reactance is equal to the magnitude of the characteristic impedance of the system. Accordingly, this capacitance can be expressed as C = 1/(2πfZ ₀ ), where f is the operating frequency of the Doherty PA and Z ₀ is the characteristic impedance of the load coupled to the Doherty PA.

Figure 22 shows an example of a Doherty combiner configuration 2400 using the hybrid circuit of Figures 18 and 19. Configuration 2400 includes a first input port 2431 that can be configured to receive a carrier-amplified signal of the Doherty PA, a second input port 2432 that can be configured to receive a peaking-amplified signal of the Doherty PA, and It includes an output port 2433 that outputs a combination of signals received from the first input port 2431 and the second input port 2432. Configuration 2400 includes a transducer (e.g., a balun transducer) having a first coil 2401 and a second coil 2402, wherein the first coil 2401 has a first port 2411 and a second port. (2412), and the second coil (2402) is implemented between the third port (2413) and the fourth port (2414). The first port 2411 and the third port 2413 are coupled by the first capacitor 2421, and the second port 2412 and the fourth port 2414 are coupled by the second capacitor 2422. do. Third port 2413 is coupled to ground, in FIG. 22, through a termination circuit including third capacitor 2423. In some implementations, the capacitance of first capacitor 2421 and second capacitor 2422 is the same. In some implementations, the capacitance of third capacitor 2423 is twice the capacitance of first capacitor 2421 and/or second capacitor 2422.

Impedance trajectories 2444 resulting from Doherty operation in combiner 2400 of FIG. 22 are shown in FIG. 23. The spread of the impedance trajectories is somewhat wider than the example in Figure 17, but is superior to the example in Figure 15. In the Doherty combiner of Figure 22, the values of various performance and operating parameters are examples; It can be adjusted appropriately for different applications.

It can be shown that alternative configurations with inductive terminations of L = Z ₀ /(2πf) can provide Doherty coupler functionality in a similar way. The port positions of the carrier amplifier and peaking amplifier can be swapped in this case. Figure 24 shows an example where this inductive termination is used.

The Doherty combiner configuration 2500 of FIG. 24 includes a first input port 2531 that can be configured to receive a carrier-amplified signal of a Doherty PA, a second input port 2531 that can be configured to receive a peaking-amplified signal of a Doherty PA. It includes an input port 2532, and an output port 2533 that outputs a combination of signals received from the first input port 2531 and the second input port 2532. Configuration 2500 includes a transducer (e.g., a balun transducer) having a first coil 2501 and a second coil 2502, wherein the first coil 2501 has a first port 2511 and a second port. (2512), and the second coil (2502) is implemented between the third port (2513) and the fourth port (2514). The first port 2511 and the third port 2513 are coupled by the first capacitor 2521, and the second port 2512 and the fourth port 2514 are coupled by the second capacitor 2522. do. Third port 2513, in FIG. 24, is coupled to ground through a termination circuit including inductor 2523. In the Doherty combiner 2500 of Figure 24, the values of various performance and operating parameters are examples; It can be adjusted appropriately for different applications.

In some embodiments, examples described with respect to FIGS. 18, 19, and 20-24 are RFIC, MMIC, and RF modules (e.g., hybrid modules) where impedance matching is achieved by the use of magnetic transducers or automatic transducers. It can be particularly useful for configurations. In some embodiments, the nearly-open output impedance of the peaking amplifier device is not inverted by the matching circuit and may therefore be present at the peaking amplifier port of the Doherty combiner.

Figure 25 shows an example of integration of auto-converter based impedance matching as a hybrid circuit and integrated passive device (IPD) having one or more features as described herein. Circuit 2600 includes an IPD 2602 that includes an impedance matching network 2610 that includes one or more automatic transducers. The IPD further includes a coupler 2620, e.g., as described above. Circuit 2600 further includes MMIC 2601 having a carrier amplifier 2611 and a peaking amplifier 2612.

If an impedance inverting matching circuit such as a Pi-network, T-network or quarter-wave converter is used, the peaking amplifier typically presents approximately a short-circuit impedance to the input of the Doherty combiner when it is idle. In these examples, an inverted load-modulation trajectory is typically required or desired from a Doherty combiner (e.g., from 0.5*Rload impedance to Rload impedance as shown in the example Smith chart 2744 in FIG. 26). In some embodiments, this functionality can be achieved by reversing the carrier amplifier input and peaking amplifier input. Figure 27 shows an example of this reversed configuration. Accordingly, in Figure 27, circuit 2700 includes IPD 2702 including coupler 2720, for example, as described above. Circuit 2700 further includes MMIC 2701 having a carrier amplifier 2711 and a peaking amplifier 2712. The circuit further includes an impedance inversion matching circuit 2710. Although not illustrated in FIG. 27, impedance inversion matching circuit 2710 may be implemented within IPD 2702.

25-27, the values of various performance and operating parameters are examples; It can be adjusted appropriately for different applications.

28, an example architecture of a PA 2800 is shown in which a Doherty combiner having one or more features as described herein may be implemented. The architecture shown is the Doherty PA architecture. Although the various examples are described in the context of this Doherty PA architecture, it will be understood that one or more features of the present disclosure may also be implemented in other types of PA systems.

The exemplary PA 2800 is shown as including an input port (RF_IN) for receiving an RF signal to be amplified. This input RF signal may be partially amplified by pre-driver amplifier 2802 before being split by carrier amplification path 2810 and peaking amplification path 2830. This division may be accomplished by splitter 2804. Examples related to splitter 2804 (also referred to herein as a splitter or power splitter) are described in greater detail herein.

28, carrier amplification path 2810 is shown as including amplification stages, collectively indicated as attenuator 2812 and 2814. Amplification stages 2814 are shown including a driver stage 2816 and an output stage 2820. Driver stage 2816 is shown as biased by bias circuit 2818, and output stage 2820 is shown as biased by bias circuit 2822. In some embodiments, there may be more or fewer amplification stages. In various examples described herein, amplification stages 2814 are sometimes described as amplifiers, although it will be understood that such amplifiers may include one or more stages.

28, peaking amplification path 2830 is shown as including a phase shifting circuit 2832 and amplification stages 2834. Amplification stages 2834 are shown including a driver stage 2836 and an output stage 2840. Driver stage 2836 is shown as biased by bias circuit 2838, and output stage 2840 is shown as biased by bias circuit 2842. In some embodiments, there may be more or fewer amplification stages. In various examples described herein, amplification stages 2834 are sometimes described as amplifiers, although it will be understood that such amplifiers may include one or more stages.

FIG. 28 further illustrates that carrier amplification path 2810 and peaking amplification path 2830 may be combined by combiner 2844 to produce an amplified RF signal at the output port (RF_OUT). Examples related to coupler 2844 are described in greater detail herein. For example, combiner 2844 may be implemented as one of the combiners of FIGS. 22 and 24 .

In some implementations, a device and/or circuit having one or more features described herein may be included in an RF device, such as a wireless device. These devices and/or circuits may be implemented directly within a wireless device, in modular form as described herein, or in some combination thereof. In some embodiments, such wireless devices may include, for example, cellular phones, smart-phones, hand-held wireless devices with or without phone functionality, wireless tablets, etc.

FIG. 29 schematically depicts an example wireless device 2900 having one or more advantageous features described herein. For example, one or more PAs 2910, collectively referred to as PA architecture 2101, may include one or more features as described herein. These PAs can facilitate multi-band operation of wireless device 2900, for example.

PAs 2110a-2110d may receive their respective RF signals from transceiver 2910, which may be configured and operable to generate RF signals to be amplified and transmitted, and to process the received signals. Transceiver 2910 is shown interacting with a baseband subsystem 2908 that is configured to provide conversion between data and/or voice signals suitable for a user and RF signals suitable for transceiver 2910. . Transceiver 2910 is also shown connected to a power management component 2906 that is configured to manage power for operation of wireless device 2900. This power management unit may control the operations of the baseband subsystem 2908 and PAs 2110a-2110d.

Baseband subsystem 2908 is shown connected to user interface 2902 to facilitate various input and output of voice and/or data provided to and received from a user. Baseband subsystem 2908 may also be connected to memory 2904 that is configured to store data and/or instructions to facilitate operation of wireless device 2900 and/or provide information to a user. You can.

In the example wireless device 2900, the outputs of the PAs 2110a-2110d are matched (via matching circuits 2920a-2920d) and their respective duplexers 2912a-2912d and band-select switches. It is shown routed via 2914 to antenna 2916. Band-select switch 2914 may be configured to allow selection of an operating band. In some embodiments, each duplexer 2912 may allow transmit and receive operations to be performed simultaneously using a common antenna (e.g., 916). 29, the received signals are shown as being routed to “Rx” paths (not shown), which may include, for example, a low noise amplifier (LNA).

A number of different wireless device configurations may utilize one or more features described herein. For example, a wireless device need not be a multi-band device. In another example, a wireless device may include additional antennas, such as a diversity antenna, and additional connectivity features such as Wi-Fi, Bluetooth, and GPS.

Power amplification with improved linearization

Various examples related to Doherty power amplifier (PA) applications are disclosed, such as for high peak-to-average power ratio (PAPR) 4G modulated signals used in 3G and 4G handset applications. In some embodiments, by using the Doherty method for other designs, up to 10% higher peak power added efficiency (PAE) levels can be achieved for the same adjacent power level ratio (ACLR) levels. These PAE performances can match those of envelope tracking (ET) PAs for much less overall system complexity.

Traditionally, it has been widely believed that Doherty PA is not suitable for linear PA applications in handsets due to its size, complexity, and non-linear behavior. In practice, in base station applications, predistortion linearizers are commonly used with Doherty PAs to satisfy linearity requirements. As described herein, issues such as size, complexity, and linearity associated with Doherty PAs can be appropriately addressed.

Figure 30 shows an example architecture of PA 3100 having one or more features as disclosed herein. The architecture shown is the Doherty PA architecture. Although the various examples are described in the context of a Doherty PA architecture, it will be understood that one or more features of the present disclosure may also be implemented in other types of PA systems.

The exemplary PA 3100 is shown as including an input port (RF_IN) for receiving an RF signal to be amplified. This input RF signal may be partially amplified by pre-driver amplifier 3102 before splitting into carrier amplification path 3110 and peaking amplification path 3130. This division may be accomplished by divider 3104. Examples related to splitter 3104 (also referred to herein as a splitter or power splitter) are described in greater detail herein.

30, carrier amplification path 3110 is shown as including an attenuator 3112 and amplification stages collectively indicated as 3114. Amplification stages 3114 are shown including a driver stage 3116 and an output stage 3120. Driver stage 3116 is shown as biased by bias circuit 3118, and output stage 3120 is shown as biased by bias circuit 3122. In some embodiments, there may be more or fewer amplification stages. In various examples described herein, amplification stages 3114 are sometimes described as amplifiers, although it will be understood that such amplifiers may include one or more stages.

30, peaking amplification path 3130 is shown as comprising amplification stages collectively indicated as phase shifting circuit 3132 and 3134. Amplification stages 3134 are shown to include a driver stage 3136 and an output stage 3140. Driver stage 3136 is shown as biased by bias circuit 3138, and output stage 3140 is shown as biased by bias circuit 3142. In some embodiments, there may be more or fewer amplification stages. In various examples described herein, amplification stages 3134 are sometimes described as amplifiers, although it will be understood that such amplifiers may include one or more stages.

30 further illustrates that carrier amplification path 3110 and peaking amplification path 3130 may be combined by combiner 3144 to produce an amplified RF signal at the output port (RF_OUT). Examples related to coupler 3144 are described in greater detail herein.

Figure 31 shows an example of a combiner circuit for a Doherty PA. These combiners can be configured to provide moderate bandwidth performance. In Figure 31, the peaking amplifier signal and the carrier amplifier signal are shown received from their respective collectors (not shown) and combined to produce an output that can be provided to a duplexer, for example. 31, it will be understood that the impedance values, as well as the values of various capacitance and inductance elements, are examples and other values may also be implemented.

Combiner 3200 has a first input port 3211 (capable of receiving a peaking amplifier signal), a second input port 3212 (capable of receiving a carrier amplifier signal), and a first input port 3211 and It includes an output port 3213 that provides a combination of signals received at a second input port 3212.

The first input port 3211 is coupled to the first node 3211. The first node 3221 is further coupled to ground (via the first capacitor 3241 and the third inductor 3233) and to the second node 3222 (via the first inductor 3231). Second node 3222 is coupled to ground (via second capacitor 3242) and to third node 3223 (via second inductor 3232).

The second input port 3212 is coupled to the fourth node 3224. The fourth node is further coupled to ground (via third capacitor 3243 and fifth inductor 3235) and to fifth node 3225 (via fourth inductor 3234). Fifth node 3225 is coupled to ground (via fourth capacitor 3244) and to third node 3223 (via fifth capacitor 3245).

Output port 3213 is coupled to the sixth node 3226. The sixth node 3226 is further coupled to ground (via a sixth inductor 3236) and to the third node 3223 (via a sixth capacitor 3246).

The first input port 3211, the second input port 3212, the first capacitor 3241, the third inductor 3233, the third capacitor 3243, and the fifth inductor 3235 are integrated passive devices (IPD). ) can be implemented as. In some embodiments, components may be implemented on a single GaAs die 3270.

The impedance present in the second node 3222 and the fifth node 3225 may each be 25 ohms. The impedance present at the third node 3223 may be 12.5 ohms.

Figure 32 shows an example power splitter circuit for a Doherty PA. Such a splitter may be used in conjunction with the example combiner of FIG. 31 and may be configured to provide adequate bandwidth performance. 32, an input radio-frequency (RF) signal is received at input 3311 and is shown splitting into two paths. The first path can be coupled to the peaking PA at first output 3312 and the second path can be coupled to the carrier PA at second output 3313. There is an inductor 3331 along the first path, and a capacitor 3341 along the second path. In Figure 32, the values of various capacitance and inductance elements are examples; It will be appreciated that other values may also be implemented.

FIG. 33 shows an example of a power splitter 3400 that can be used as splitter 3104 of FIG. 30. 33, power splitter 3400 includes a converter 3450 with two coils positioned relative to each other. The first coil may have interleaved windings coupled to each other, one winding coupled to input 3411 and the other winding coupled to first output 3414. The second coil may have interleaved windings coupled to each other, one winding coupled to the isolation port 3412 and the other winding coupled to the second output 3413.

The example of Figure 33 can be configured as an orthogonal splitter with wideband capability. This splitter can be configured as a semi-concentrated 90 degree power splitter that can be implemented as an IPD design for low frequencies and also as an integrated splitter on a GaAs die for higher frequencies.

The power splitter 3400 may further include capacitors 3441 and 3442 that couple the coils. In some embodiments, first capacitor 3441 is coupled between input 3411 and isolation port 3412 and second capacitor 3442 is coupled between first output 3413 and second output 3414. is coupled to

Using the previous configuration, the power of the RF signal received at the input port can be split into two output ports 3413 and 3414. These divided signals can be provided to the carrier amplifier and peaking amplifier of FIG. 30.

FIG. 34 shows an example of a power splitter 3500 that can be used as splitter 3104 of FIG. 1. Additional details regarding these power splitters are described above, including but not limited to the section entitled “Power Amplification Using Balun Converters.”

The example of Figure 34 can be configured as an orthogonal splitter with wideband capability. In some embodiments, this splitter is a lumped 90-degree power splitter that can be implemented as an SMT circuit for low frequencies, and also as an integrated (e.g., IPD) splitter on a GaAs die for higher frequencies. It can be composed as.

Figure 35 shows an example of a coupler 3600 that can be used as coupler 3144 of Figure 30. Additional details regarding these combiners are described above, including but not limited to the section entitled “Power Amplification Using Balun Converters.”

The example of Figure 35 can be implemented as an SMT circuit with wideband capability. In some embodiments, this combiner may include power combining and dynamic load pulling functionality implemented through the use of a lumped balun.

Figure 36 shows another example of a coupler 3700 that can be used as coupler 3144 of Figure 30. Additional details regarding these combiners are described above, including but not limited to the section entitled “Signal Combining Using Coiled Balun Converters.”

The example of Figure 36 can be implemented as an IPD with broadband capabilities. In some embodiments, this combiner may include power combining and dynamic load pulling functionality implemented through the use of a semi-centralized 90 degree hybrid configuration.

30, in some embodiments, each of the driver stage 3116 and output stage 3120 of the carrier amplifier 3114 may be configured to operate in class AB mode. Additionally, each of the driver stage 3136 and output stage 3140 of the peaking amplifier 3134 may be configured to operate in class B mode. For these configurations, bias circuits such as those shown in Figures 38 and 39 may be used to bias the stages of carrier amplifier 3114 and peaking amplifier 3134, respectively. Accordingly, carrier amplifier 3114 and peaking amplifier 3134 may operate in different biasing modes. Additionally, for each amplifier 3114, 3134, each stage 3116, 3120 and 3136, 3140 may operate in different biasing modes. Different biasing modes may include Class A, Class B, Class AB, Class C, Class D, Class F, Class G, Class I, Class S, Class T, or any other biasing mode.

37 shows an example of a low headroom class AB bias circuit that can be used to provide a bias voltage (VBIAS) to the stage (driver 3116 or output 3120) of the carrier amplifier 3114. Accordingly, a class AB bias circuit may provide the biasing functionality of bias circuit 3118 and/or bias circuit 3122 of FIG. 30. Appropriate selections of transistors, diodes, capacitances and resistors may be implemented to accommodate such driver and output stage functionality. In some embodiments, the example bias circuit of FIG. 37 may be particularly suitable for integration with external band gap references for CMOS or GaAs, where low voltage headroom makes the use of conventional 2xVbe bias circuits difficult. The bias circuit of FIG. 37 may include sufficient bandwidth at baseband frequencies to support wideband signals such as LTE.

38 shows an example of a low headroom Class B bias circuit that can be used to provide a bias voltage (VBIAS) to the stage (driver 3136 or output 3140) of peaking amplifier 3134. Accordingly, a Class B bias circuit may provide the biasing functionality of bias circuit 3138 and/or bias circuit 3142 of FIG. 30. Appropriate selections of transistors, diodes, capacitances and resistors may be implemented to accommodate such driver and output stage functionality.

Figure 39 shows an example of the beneficial effects of using class B biasing of the driver stage for a peaking amplifier (3134 in Figure 30). Graph 4000 of FIG. 39 includes plots of output stage current as a function of output power for different configurations. For the carrier amplifier, the solid line 4011 is for a configuration where the driver and output stages are each biased in class B mode, while the dashed line 4011 is for class AB biasing of the driver stage and class AB biasing of the output stage. This is for a configuration with B biasing. Similarly, for the peaking amplifier, the solid line 4021 is for a configuration in which the driver and output stages are each biased in class B mode, while the dashed line 4022 is for a configuration in which the driver stage is biased in class AB and the output stage is biased in class B mode. This is for a configuration with class B biasing. As shown in Figure 39, the use of class B biasing in the driver stage within the peaking amplifier greatly reduces the current consumption of the output stage. However, the use of class B biasing in the driver stage within the carrier amplifier slightly increases the current consumption of the output stage.

Figure 40 shows an example of the beneficial effects of using class B biasing of the driver stage for a peaking amplifier (3134 in Figure 30). Graph 4100 of FIG. 40 includes plots of power-added efficiency (PAE) as a function of output power for different configurations. The solid line 4101 represents a configuration in which each of the driver and output stages of the peaking amplifier is biased in class B mode. Dashed line 4102 is for a configuration where the driver stage is biased in class AB mode and the output stage is biased in class B mode. The dot-dotted line 4103 is for an equivalent non-Doherty amplifier biased in class AB mode. As shown in Figure 40, the use of class B biasing in the driver stage within the peaking amplifier significantly increases PAE performance.

Figure 41 shows an example of a linearization effect that can be achieved by introducing a phase shift between the RF signals associated with carrier amplification and peaking amplification. This phase shift may be introduced, for example, by phase shift component 3132 in FIG. 30. Graph 1200 of FIG. 41 includes plots of AM/AM (left vertical axis) and AM/PM (right vertical axis) as a function of output power. For AM/AM plots 4211, 4212, Figure 41 shows that the curve corresponding to the configuration with phase shift has less AM/AM distortion than the configuration without phase shift, especially at higher output powers. . Similarly, for the AM/PM plots 4221 and 4222, Figure 41 shows that the curve corresponding to the configuration with phase shift has less AM/PM distortion than the configuration without phase shift, especially at higher output powers. shows.

As described herein, the power split into the carrier amplification path and the peaking amplification path may be different. Figure 42 shows an example of the linearization effect that can be achieved by introducing this non-uniform power split between the RF signals associated with carrier amplification and peaking amplification. This uneven power split may be introduced or facilitated, for example, by attenuator component 3112 of FIG. 30. Graph 4300 in FIG. 42 includes plots of AM/AM (left vertical axis) and AM/PM (right vertical axis) as a function of output power. For AM/AM plots 4311, 4312, FIG. 42 shows that the curve corresponding to the configuration with uneven power splitting is smaller than the configuration with uniform power splitting configuration, especially at higher output powers. It shows that there is distortion. Similarly, for the AM/PM plots 1321 and 1322, FIG. 13 shows that the curve corresponding to the configuration with uneven power split is similar to that of the curve corresponding to the configuration with a uniform power split configuration, especially at medium to higher output powers. It is shown to have smaller AM/PM distortion than the configuration.

Figure 43 shows an example of a combined linearization effect that can be achieved by combining the previous phase shift and non-uniform power split features described with respect to Figures 41 and 42. Graph 4400 in FIG. 43 includes plots of gain (left vertical axis) and PAE (right vertical axis) as a function of output power. In particular, line 4411 shows the gain for a non-Doherty amplifier, line 4412 shows the gain for a Doherty amplifier without phase shift and uniform power splitting, and line 4413 shows the gain with phase shift and non-uniform power splitting. The gain for a Doherty amplifier with power splitting is shown. Similarly, line 4421 shows the PAE for a non-Doherty amplifier, line 4412 shows the PAE for a Doherty amplifier with no phase shift and no uniform power split, and line 4413 shows the PAE with and without phase shift. The PAE for a Doherty amplifier with non-uniform power splitting is shown.

Figure 43 shows that a linear load modulated amplifier (Doherty PA with phase shift and non-uniform power splitting) has a very similar gain compression curve to a non-Doherty PA (e.g., a class AB/F amplifier). It shows. Figure 43 shows that the PAE of a linear load modulated amplifier (Doherty PA with phase shift and non-uniform power splitting) is only slightly smaller (e.g., smaller than that of a typical non-linear Doherty amplifier (Doherty PA without linearization)) (about 3% smaller than the high output power) is also shown.

44 shows the PAE (left vertical axis) and adjacent channel at various operating frequencies for a front-end module (FEM) with a dual-band Doherty PA, and an FEM with an average power tracking (APT) PA configured for LTE operation. Power (ACP) (right vertical axis) is shown. Figure 44 shows that PAE is generally higher and the magnitude of ACP is generally lower for Doherty PA than for APT PA. In the example shown, the improvement is approximately 10%.

In some implementations, a device and/or circuit having one or more features described herein may be included in an RF device, such as a wireless device. These devices and/or circuits may be implemented directly within a wireless device, in modular form as described herein, or in some combination thereof. In some embodiments, such wireless devices may include, for example, cellular phones, smart-phones, hand-held wireless devices with or without phone functionality, wireless tablets, etc.

Figure 45 schematically depicts an example wireless device 3801 having one or more advantageous features described herein. For example, one or more PAs 3110a-3110d, collectively referred to as PA architecture 3101, may include one or more features as described herein. These PAs can facilitate multi-band operation of wireless device 3801, for example.

PAs 3110a-3110d may receive their respective RF signals from transceiver 3810, which may be configured and operable to generate RF signals to be amplified and transmitted, and to process the received signals. The transceiver 3810 is shown interacting with a baseband subsystem 3808 that is configured to provide switching between data and/or voice signals suitable for the user and RF signals suitable for the transceiver 3810. Transceiver 3810 is also shown as connected to a power management component 3806 that is configured to manage power for operation of wireless device 3801. This power management may also control the operations of baseband subsystem 3808 and PAs 3110a-3110d.

Baseband subsystem 3808 is shown connected to user interface 3802 to facilitate various input and output of voice and/or data provided to and received from a user. Baseband subsystem 3808 may also be connected to memory 3404, which is configured to store data and/or instructions to facilitate operation of wireless device 3801 and/or provide storage of information to a user. there is.

In the example wireless device 3801, the outputs of the PAs 3110a-3110d are matched (via matching circuits 3820a-3820d) and their respective duplexers 3812a-3812d and band-select switches. It is shown routed via 3814 to antenna 3816. Band-select switch 3814 may be configured to allow selection of an operating band. In some embodiments, each duplexer 3812 may allow transmit and receive operations to be performed simultaneously using a common antenna (e.g., 3816). In Figure 45, the received signals are shown as being routed to “Rx” paths (not shown), which may include, for example, a low noise amplifier (LNA).

Many wireless device configurations may utilize one or more features described herein. For example, a wireless device does not need to be a multi-band device. In another example, a wireless device may include additional antennas, such as a diversity antenna, and additional connectivity features such as Wi-Fi, Bluetooth, and GPS.

Unless the context clearly requires otherwise, throughout the description and claims, the words “comprise,” “including,” etc. are used in an inclusive, as opposed to exclusive or exhaustive sense; In other words, it should be interpreted to mean “including, but not limited to.” The word “coupled,” as commonly used herein, refers to two or more elements that may be connected directly or connected by one or more intermediate elements. Additionally, the words “herein,” “above,” “hereinafter,” and words of similar prominence, when used in this application, refer to this application as a whole and not to any specific portions of this application. Where the context permits, words in the above description using the singular or plural number may also include the plural or singular number respectively. The word "or", with respect to a list of two or more items, covers all of the following interpretations of the word: any of the items in the list, all of the items in the list, and any combination of items in the list.

The above detailed description of embodiments of the invention is not intended to be exhaustive or to limit the invention to the precise form disclosed above. Although specific embodiments of the invention, and examples of the invention, have been described above for illustrative purposes, various equivalent modifications are possible within the scope of the invention, as those skilled in the art will recognize. For example, although processes or blocks are presented in a given order, alternative embodiments may perform routines with steps in a different order, or use systems with blocks, and some processes or blocks are deleted. , can be moved, added, subdivided, combined and/or modified. Each of these processes or blocks may be implemented in a variety of different ways. Additionally, although processes or blocks are sometimes shown as being performed in series, these processes or blocks may instead be performed in parallel, or may be performed at different times.

The teachings of the invention provided herein may be applied to other systems, not necessarily the system described above. The elements and operations of the various elements described above may be combined to provide additional embodiments.

Although some embodiments of the inventions have been described, these embodiments have been presented by way of example only and are not intended to limit the scope of the disclosure. Indeed, the novel methods and systems described herein may be implemented in a variety of different forms; Additionally, various omissions, substitutions and changes in the form of the methods and systems described above may be made without departing from the spirit of the disclosure. The appended claims and their equivalents are intended to cover such forms or modifications as fall within the scope and spirit of the disclosure.

Claims (20)

As a signal combiner,
A balun transformer circuit having a first coil and a second coil, wherein the first coil is implemented between a first port and a second port, and the second coil is implemented between a third port and a fourth port. The first port and the third port are coupled by a first capacitor, the second port and the fourth port are coupled by a second capacitor, and the first port receives the first signal. configured to, the fourth port is configured to receive a second signal, the second port is configured to calculate a combination of the first signal and the second signal, the capacitance of the first capacitor and the second signal The capacitance of the capacitor is the same -; and
A termination circuit coupling the third port to ground and including a third capacitor.
Including,
The termination circuit has a reactance equal to the impedance at the second port, and the capacitance of the third capacitor is twice the capacitance of the first capacitor and the second capacitor. The method of claim 1, wherein the first port is configured to receive a carrier-amplified signal from a carrier amplifier of the Doherty power amplifier, and the fourth port is configured to receive a peaking-amplified signal from a peaking amplifier of the Doherty power amplifier. Consisting of a signal combiner. The signal coupler of claim 2, wherein the termination circuit has a capacitive reactance equal in magnitude to the impedance at the second port. delete delete The signal combiner of claim 1, wherein an S-parameter between a first one of the ports and a second one of the ports is equal to (1+j)/2. The signal combiner of claim 1, wherein the S-parameter between a first one of the ports and a second one of the ports is equal to (1-j)/2. 2. The signal combiner of claim 1, wherein the S-parameter matrix of S-parameters between the ports includes only the values 0, (1+j)/2, and (1-j)/2. A power amplifier module, comprising:
A packaging substrate configured to receive a plurality of components; and
Signal coupler implemented on the packaging substrate
Including,
The signal coupler includes a balun converter circuit having a first coil and a second coil, wherein the first coil is implemented between a first port and a second port, and the second coil is implemented between a third port and a fourth port. Implemented in, the first port and the third port are coupled by a first capacitor, the second port and the fourth port are coupled by a second capacitor, and the first port is connected to the first signal. configured to receive, wherein the fourth port is configured to receive a second signal, the second port is configured to produce a combination of the first signal and the second signal, and the signal combiner is configured to receive a second signal, and the signal combiner is configured to receive a second signal. and a termination circuit coupled to ground and including a third capacitor, wherein the termination circuit has a reactance equal in magnitude to the impedance at the second port, and the capacitance of the first capacitor and that of the second capacitor. The power amplifier module has the same capacitance, and the capacitance of the third capacitor is twice the capacitance of the first capacitor and the second capacitor. 10. The power amplifier module of claim 9, wherein the balun converter circuit is implemented as an integrated passive device. 11. The power amplifier module of claim 10, wherein the integrated passive device further implements an impedance matching circuit including one or more automatic transducers. The method of claim 9, wherein the power amplifier module further comprises a Doherty power amplifier implemented on the packaging substrate, wherein the Doherty power amplifier produces a carrier amplifier and a peaking-amplified signal. Power amplifier module with peaking amplifier. 13. The power amplifier module of claim 12, wherein the first port is configured to receive the carrier-amplified signal and the fourth port is configured to receive the peaking-amplified signal. The power amplifier module of claim 13, wherein the termination circuit has a capacitive reactance equal in magnitude to the impedance at the second port. delete delete 10. The power amplifier module of claim 9, wherein the S-parameter matrix of S-parameters between the ports includes only the values 0, (1+j)/2, and (1-j)/2. As a wireless device,
A transceiver configured to generate a radio-frequency signal;
A power amplifier module in communication with the transceiver, the power amplifier module comprising an input circuit configured to receive the radio-frequency signal and split the radio-frequency signal into a first portion and a second portion, the power amplifier module comprising: further comprising an amplifier circuit including a carrier amplifier coupled to the input circuit to receive the first portion and a peaking amplifier coupled to the input circuit to receive the second portion, wherein the power amplifier module further comprising an output circuit coupled to the amplifier circuit, the output circuit comprising a balun converter circuit having a first coil and a second coil, the first coil being implemented between the first port and the second port; , the second coil is implemented between the third port and the fourth port, the first port and the third port are coupled by a first capacitor, and the second port and the fourth port are coupled by a second capacitor. coupled by, wherein the first port is configured to receive a first amplified signal from one of the carrier amplifier or the peaking amplifier, and the fourth port is configured to receive a first amplified signal from the other of the carrier amplifier or the peaking amplifier. configured to receive an amplified signal, wherein the second port is configured to produce a combination of the first amplified signal and the second amplified signal as an amplified version of the radio-frequency signal, and the third port is configured to: , coupled to ground through a termination circuit that includes a third capacitor and has a reactance equal in magnitude to the impedance at the second port, the capacitance of the first capacitor and the capacitance of the second capacitor are the same, and the The capacitance of the 3 capacitor is twice the capacitance of the first capacitor and the second capacitor; and
Antenna in communication with said power amplifier module
Including,
The wireless device of claim 1, wherein the antenna is configured to facilitate transmission of an amplified version of the radio-frequency signal. 19. The method of claim 18, wherein the first port is configured to receive the first amplified signal from the carrier amplifier, the fourth port is configured to receive the second amplified signal from the peaking amplifier, and the termination A wireless device wherein the circuit has a capacitive reactance equal in magnitude to the impedance at the second port. delete

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