JPH0137711B2 - - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION The present invention relates to a radar device that accurately detects and tracks targets flying at low altitude.

Conventionally, this type of device is a monopulse radar (MN
OPULSE RADAR).
Figure 1a is a block diagram showing the principle of monopulse radar, in which 1 is a monopulse antenna, 2
is a hybrid circuit, 3 is an amplifier, 4 is an automatic gain control device, 5 is a phase detector, 6 is the central axis of monopulse antenna 1, 7 is a target, 8 is an image of target 7, 9 is a reflected wave from target 7 ( 10 is a reflected wave from the image 8, that is, a reflected wave from the target 7 due to the multipath effect (indirect wave), and 11 is the sea surface. FIG. 1b shows the monopulse antenna 1
FIG. 2 is an antenna pattern diagram. Monopulse radar simultaneously receives reflected waves from a target using two partially overlapping antenna patterns as shown in Figure 1b, and calculates the sum signal Σ and difference signal of the two received signals obtained from each antenna pattern. Δ is obtained by the hybrid circuit 2. As a result, the output signal of the hybrid circuit is completely equivalent to the signal received by the two antenna patterns as shown in FIG. In the figure, the horizontal axis shows the reception angle θ (the angle between the antenna center axis 6 and the received reflected wave), and the vertical axis shows the reception voltage V.
12 is a sum pattern, 13 is a difference pattern, 14
is the received voltage Σ _d due to the pattern 12 of the sum of the reflected waves 9, 15 is the received voltage Δ _d due to the pattern 13 of the difference between the reflected waves 9, 16 is the received voltage Σ _i according to the pattern 12 of the sum of the reflected waves 10, and 17 is the reflected wave This is the received voltage _Δi due to the pattern 13 of the difference between the waves 10. however,
Σ, Δ, Σ _d , Δ _d , Σ _i , Δ _i are complex signal voltages including amplitude and phase.

First, consider the case where image 8 does not exist. At this time, the monopulse antenna 1 receives only the reflected wave 9 from the target 7, so the received signal voltage Σ due to the sum pattern 12 and the received signal voltage △ due to the difference pattern are respectively Σ=Σ _d = Σ(θ _d ) (1) △=△ _d = △(θ _d ) (2). Here, θ _d is the receiving angle of the reflected wave 9.
The gain A of the amplifier 3 is controlled by the automatic gain control device 4 as shown in the following equation.

A=V _s /|Σ| (3) Here, V _s is a constant determined by the feedback amount of the automatic gain control device.

Therefore, the received signal voltage after passing through the amplifier 3 is Σ′=V _s Σ _d / |Σ _d |=V _s Σ(θ _d )/|Σ(θ _d )| (4) △′= V _s △ _d / |Σ _d | = V _s △(θ _d ) / |Σ(θ _d ) | (5). The phase detector 4 is a detector that maintains the amplitude and phase difference of two input signals.If the two complex input signals are I and Q, the output voltage E is E=|I| ² Re[Q/I] (6) becomes. Here, Re [ ] represents the real part. Therefore, the two signals expressed by equations (4) and (5) are detected by the phase detector 5.
, the output voltage E becomes E=V ² _s Re [Δd/Σd] = V ² _s Re [Δ(θ _d )/Σ(θ _d )] (7).

FIG. 3 is a diagram in which the relationship between the reception angle θ _d and the phase detector output voltage E is calculated based on equation (7) and the antenna pattern shown in FIG. As shown in FIG. 3, the output voltage _E and the reception angle θ _d have a proportional relationship in the region where the angle θ _d is not very large. The positional direction θ _d can be accurately known. Furthermore, by feeding back the output voltage E given by equation (7) to the drive section of the monopulse antenna 1 as an angular error signal, the target can be tracked with high accuracy.

On the other hand, when the image 8 exists, that is, when the reflected wave 10 due to the multipath effect and the reflected wave 9 from the target are simultaneously received by the monopulse antenna 1, the voltage Σ received in the sum pattern 12 and the difference pattern The voltage Δ received at 11 is a composite value of two voltages as shown below.

Σ=Σ _d + Σ _i = Σ (θ _d ) + Σ (θ _i ) (8) △=△ _d + △ _i = △ (θ _d ) + △ (θ _i ) (9) Here, θ _i is the reflected wave 10 is the receiving angle.

By substituting equations (8) and (9) into equation (7), the phase detector output voltage E when there is a multipath effect is similarly given by the following equation.

E=V ² _s Re [△ _s + △ _i / Σ _d + Σ _i ] = V ² _s Re [
△(θ _d )/Σ(θ _d )・1+△(θ _i )/(△(θ _d )/
1 + Σ (θ _i ) / Σ (θ _d )] (10) In the absence of multipath effects, the phase detection output voltage E is determined only by the receiving angle of the reflected wave from the target if the antenna pattern is determined. However, in this case, due to the influence of the reflected wave from the image, as shown in equation (10), in addition to the above parameters, it is also affected by the amplitude ratio and phase difference of the two reflected waves. This means that the reception angle of the reflected wave from the target cannot be determined from the phase detector output voltage alone, and essentially monopulse radar cannot accurately detect and track the target under the influence of multipath effects. It becomes impossible.

However, conventional radar equipment uses a monopulse radar to detect and track targets flying at low altitude. As a result, the target detection and tracking accuracy is very poor, and in extreme cases, the target may be lost.

In order to solve these drawbacks, this invention arranges a plurality of antennas in a straight line in the vertical direction,
The purpose of this system is to perform signal processing on received data received independently by each antenna according to a predetermined method to remove multipath effects and improve target detection and tracking accuracy.The drawings will be described in detail below.

FIG. 4 shows an embodiment of the present invention.
8 is a transmitting/receiving antenna, 19 is a receiving antenna,
20 is a reception antenna switching device, 21 is a receiver, 2
2 is a reference signal generator, 23 is a transmitter, 24 is a transmission/reception switch, 25 is a distance measuring device, 26 is a target speed measuring device, 27 is a signal processing device, and 28 is a pulse radio wave. FIG. 5 is a diagram showing a transmission signal waveform,
In the figure, 29 is a reference signal, and 30 is a transmission signal.

Here, for convenience of explanation, the total number of antennas 18 and 19 is assumed to be N, and each antenna is called antenna #1, #2, . . . , #N in the order of distance from the sea surface 11. The reference signal 29 is generated by the reference signal generator 22 and is a continuous sine wave signal whose frequency is equal to the transmission frequency. The transmitter 23 pulse-modulates the reference signal 29 with a period Tr as shown in FIG. 5, and outputs a series of transmission signals 30 (#1,
#2, #3,...) are generated. The generated transmission signal 30 is radiated to the outside space as a pulse radio wave 28 via the transmission/reception switch 24 and the transmission/reception antenna 18. Among the pulses 28, the pulse radio waves reflected by the target 7 and the image 8 are transmitted to the antennas 18, 1 designated by the reception antenna switching device 20.
9 and is received by a particular one of receiver 2
1 is input. The receiver 21 performs phase detection on the input signal using the reference signal 29 and inputs the detected complex signal to the signal processing device 27 . As the phase of the complex signal, only the phase corresponding to the propagation distance of the pulse radio wave is preserved by the phase detection, regardless of the phase of the pulse radio wave to be transmitted.

The reception antenna is switched by the reception antenna switching device 20 in synchronization with the generation of the transmission signal 30. Various methods can be considered for the synchronization timing, but here, the reflected wave generated from the pulse radio wave #1 is received by the antenna #1, the reflected wave generated from the pulse radio wave #2 is received by the antenna #2, Similarly, the reflected wave generated from #N pulse radio wave is #N
The following describes the case of receiving with an antenna. The essence is the same in other cases as well.

The reflected waves received by antennas #1 to #N are reflected wave 9 from target 7 and reflected wave 1 from image 8.
The complex signal Si, which is a composite value of 0 and is therefore input to the signal processing device 27, is expressed as Si=a _i +b _i (i=1, 2, . . . , N) (11). Here, i indicates the number of the antenna from which the complex signal Si was obtained, and a _i and b _i correspond to reflected waves 9 and 10, respectively. Further, the reflected wave received by the #1 antenna is processed by the distance measuring device 25 and the target speed measuring device 26, and the distance and relative speed between the target 7 and this radar device are calculated.
The signals are input to the signal processing device 27 as a distance signal R _p and a relative speed signal υ. Here, the distance measuring device measures the slant range to the target using a split gate method. Further, the target speed measuring device 26 measures the relative speed of the target in the distance direction using a pulse Doppler method. The distance data R _p and the relative velocity data υ measured here include errors due to the influence of the reflected waves 10, but these errors can be ignored if the altitude of the target 7 is low enough to cause a multipath effect.

FIG. 6 is a diagram showing the principle of the processing method performed by the signal processing device 27. In the figure, 31 is a virtual target for signal processing, 32 is the surface where the target 7 and image 8 are present, 33 is the antenna 18, 19 is arranged, and 34 is an axis indicating the centers of antennas #1 to #N.

The signal processing device 27 processes the complex signal given by equation (11)
Arithmetic processing as shown in the following equation is performed on Si.

V=| _N 〓 ⁱ⁼¹ S _i t _i | (12) Here, t _i (i=1, 2,..., N) is a phase correction coefficient, which is calculated by the signal processing device 27 as follows. Ru. First, the virtual target 31 is set at a position at an elevation angle θ _p (θ _p is the angle formed by the central axis 34 and the direction of the virtual target 31). Here, θ _p is set to a value at which target 7 is considered to exist. At this time, the phase correction coefficient t _i is calculated by the following equation.

ti = K・exp [−j2π/λUi] (13) Here, λ is the transmission wavelength, K is an arbitrary constant that is not zero, and ui (i = 1, 2, ..., N) is the round-trip radio wave It is expressed as the propagation path length, and can be expressed by the following formula using the distance Ri between the virtual target 31 and each receiving antenna 18, 19 at the time of reception. ui=R ₁ + Ri (14a) = {(h _N −h ₁ /2 + R _p tanθ _p ) ² + (R _p −υ(i−
1) Tr) ² } ^1/2 + {(h _N +h ₁ /2+R _p tanθ _p −hi) ² + (R _p −υ(i−1)Tr) ² } ^1/2 (14b) Here h _i (i=1, 2, . . . , N) is the height at which the receiving antenna is arranged, and h _M is given by the following equation.

h _M = (h _N + h ₁ )/2 (15) The signal processing device 27 performs the above calculation on the virtual target 31
This is done by sequentially changing the position of , that is, the elevation angle θ _p .
At this time, let the elevation angles for target 7 and image 8 be θ _t and θ _i , respectively, and the elevation angle θ _p satisfies |θ _p −θ _t <△θ (16) or |θ _p −θ _i <△θ (17). _Let the calculation result of _equation (12) be V=V _IN when V _IN /V _OUT ≫1 (18) holds true.

Here, Δθ is an equivalent antenna beam width obtained by performing the above-described signal processing on the received signals obtained by antennas #1 to #N.

The phase correction coefficient shown by equation (13) takes into consideration the movement of the target during the time it takes to obtain all the received signals necessary for the processing, and adjusts the reflected waves from the virtual target 31 to #1 to #N. This is to correct the phase difference in the received signal due to the difference in distance between the virtual target and each of the antennas #1 to #N, assuming that the received signal is received by the antenna #1 to #N, so that the received signal is in the same phase. The situation is shown in FIG.
In the figure, 35 is an equiphase plane from the virtual target.
Now, assuming that the position of the virtual target 31 matches the position of the target 7, the calculation expressed by equation (12) is #1
Since the ~#N antennas are arranged on the 35 plane and the reflected waves from the target 7 are simultaneously received and integrated, the calculation result is the reflected wave from the target using a real aperture antenna with the 35 plane as a mirror surface. It is exactly equivalent to receiving . Therefore, to sequentially change the elevation angle θ _p of the signals received by antennas #1 to #N and perform the arithmetic processing shown in equation (12), it is necessary to sequentially change the beam direction of the real aperture antenna and achieve the target. Alternatively, it is equivalent to receiving the reflected wave from the image, and if equations (16) and (17) hold, it is equivalent to receiving the reflected wave from the target or image with the main beam of the real aperture antenna. , Conversely, if the above two equations do not hold, this corresponds to the fact that the reflected waves from the target and the image are received by the side ropes, and the relationship shown in equation (18) holds true. Furthermore, the beam width Δθ can be considered in the same way as in the case of a real aperture antenna, and is given by the following equation from a known formula that gives the beam width of a real aperture antenna.

Δθ=λ/(h _N −h ₁ ) (19) Thus, according to the radar device according to the present invention,
The same angular resolution can be obtained when receiving radio waves from a target with a radar device that has a real aperture antenna with a large aperture length in the vertical direction, but the beam scanning is much more extensive than that of a radar device that has a real aperture antenna. This has the advantage that it can be performed simply and at high speed by changing the phase correction coefficient given by equation (13), compared to the need for a separate drive device. In addition, the beam width given by equation (19) is obtained when the target is sufficiently far away with a real aperture antenna, but in this radar system, the phase correction coefficient of equation (13) is calculated in consideration of the distance to the target. is determined, the beam width given by equation (19) can be obtained regardless of the distance to the target.

Furthermore, the phase correction coefficient used in this radar device takes into account only the horizontal movement of the target, and does not take into account the vertical movement. On the other hand, the movement of the image is similar to that of the target in the horizontal direction, but it constantly fluctuates in the vertical direction due to fluctuations in the sea surface 11. As a result, the error in the low phase correction for the image increases, and the image The calculation result will be smaller than the target.

This effect is produced as a result of receiving complex received signals to be processed in time series, and cannot be obtained in this type of radar apparatus configured to receive complex received signals simultaneously.

Therefore, by making the range of change in the elevation angle θ _p sufficiently small,
If the calculation of equation (12) is carried out, a calculation result as shown in FIG. 8 can be obtained.

FIG. 8 is a diagram showing the effect of the present invention, and 3
6 and 37 are calculation results corresponding to target 7 and image 8, respectively. As shown in Fig. 8, the target and image can be separated by applying the signal processing described above to the received data received by multiple antennas, and by detecting the peak value of the calculation result V, the target can be separated. The presence of the target can be detected accurately by removing the influence of multipath effects, and the target can also be accurately tracked by extracting the angle information of the peak position of the calculation result V.

Now, in the above explanation, it was assumed that the measured data R _p gives the horizontal distance between the target 7 and the antenna _element 18 as shown in FIG. A slant range between elements 18 is provided. Even if R _p is not a horizontal distance but a slant range, such an error in R _p will be Things that do not affect the effects of the present invention will be explained below.

First, the error occurring in the phase correction coefficient ti calculated using equations (13) and (14) will be explained.

In equation (14), h _M + R _p tanθ _p is the virtual target 3
1 from sea level 11, which is h _p
far.

h _p = h _M + R _p tanθ _p (20) Rewriting equation (14) using equation (20) yields the following equation.

ui=[(h _p −h ₁ ) ² + {R _p −υ(i−1) _Tr } ² ] ^1/2 +[h _p −hi) ² +{R _p −υ(i−1)Tr } ² ] ^1/2 i=1, ~, N (21) When the radar device according to the present invention observes a target flying at a low altitude where multipath effects are a problem, the following equation generally holds true.

R _p ≫ (h _p −hi) ² (22) Also, the distance υ(N−1) Tr that the target moves within the time (N−1) Tr required to obtain the received signal necessary for signal processing is , is sufficiently small compared to the size of R _p . At this time, using equation (22), equation (21) is expressed as the following equation.

u _i =2R _p +1/2R _p (h _p −h ₁ ) ² −2υ(i−1)Tr+1/2R _p (h _p −h ₁ ) ² (23) The first term on the right side of equation (23) R _p and the second term, 1/2R _p (h _p −h ₁ ) ² , are constants with respect to changes in i, and even if R _p is a slant range rather than a horizontal distance, it completely eliminates the error in phase correction. I won't give it. Also,
The third term on the right side, 2υ(i-1)Tr, corrects the phase change of the received signal due to movement of the target during pulse radio wave transmission, and is not related to R _p . Next, consider the fourth term on the right side. Here, the horizontal distance is R _H
The slant range R _p is expressed by the following formula.

R _p =R _H +△R (24) △R is the difference between the slant range R _p and the horizontal distance, and when the height of the target 7 from the sea surface 11 is expressed by ht, it is expressed by the following formula.

△R=√ _H ² + ² −R _H (25) In the radar device according to the present invention, R _H =10Km, ht=
Since the purpose is search and tracking at a low altitude of about 100 meters, equation (25) can be approximated by the following equation.

ΔR~ht ² /2R _H (26) From equation (26), it is clear that ΔR/R _H ≪ (27). Using equation (27), the fourth term on the right side of equation (23) can be expressed as the following equation.

Fourth term on the right side of equation (23) = 1/2R _H (h _p −hi) ² −△R/2R _H ² (h _p −hi) ² (28) The second term on the right side of equation (28) is the distance This provides a phase correction due to the fact that the data is a slant range rather than a horizontal distance. The magnitude of the phase correction error depends on the transmission wavelength λ. That is, in a field where the second term on the right side of equation (28) is sufficiently small with respect to λ, it does not matter that the distance data is in the slant range. In the radar device according to the present invention, the transmission wavelength is λ
= 0.3m is often used, and since it is aimed at searching and tracking low-altitude flying targets,
At most R _H = 10 Km, ht = 100 m, and h _p -hi = 200 m. At this time, it is clear that the following equation holds true.

△R/2R _H ² (h _p-hi ) ² ≪λ (29) In this way, the radar device according to the present invention can generate phase correction coefficients with sufficient accuracy even when using slant range instead of horizontal distance. Then you can
Effects as shown in FIG. 8 can be obtained.

Then the altitude ht of the target measured in this way
The error when the target elevation angle θt is derived from is explained below. Given the target altitude ht, the target elevation angle θt is expressed by the following equation based on the geometrical relationship shown in FIG.

θt=tan ^-1 ht−h _M /R _p (30) Substituting equation (24) into equation (30),
Using the equation and equation (27), equation (30) can be expressed as the following equation.

θ _t = tan ^-1 ((h _t −h _M )/R _H −h _t ² (h _t −h _M )/2R _H ³ )
(31) Equation (31) is expressed as Equation (32) using h _t <<R _H and h _M <<R _H.

θ _t h _t −h _M ／R _t −h _t ² (h _t −h _M )／2R _H ³ (32) The second term of equation (32) indicates that the distance measurement data R _p is not the horizontal distance but the slant range. It is clear that the second term is sufficiently smaller than the first term.

In this way, the radar device according to the present invention can generate the phase correction coefficient with sufficient accuracy and measure the target elevation angle even when using the slant range instead of the horizontal distance, and the effect shown in FIG. 8 can be obtained. I can do it.

As described above, according to the present invention, it is possible to eliminate the influence of multipath effects on targets flying at low altitude and to accurately know the target position.
It is highly effective when used in radar equipment that detects and tracks targets at low elevation angles.

[Brief explanation of drawings]

Figure 1a is a block diagram showing the principle of monopulse radar, Figure 1b is a diagram showing the monopulse antenna pattern, Figure 2 is a diagram of sum and difference monopulse antenna patterns, and Figure 3 is the reception angle and phase. FIG. 4 is a block diagram showing an embodiment of the present invention; FIG. 5 is a diagram showing the transmitted signal waveform; FIG. 6 is a diagram showing the principle of the signal processing method. FIG. 7 is a diagram showing the equivalence between this invention device and a real aperture antenna, and FIG. 8 is a diagram showing the effects of this invention device. In the figure, 1 is a monopulse antenna, 2 is a hybrid circuit, 3 is an amplifier, 4 is an automatic gain control device,
5 is a phase detector, 6 is the central axis of the antenna, 7 is a target, 8 is an image, 9 and 10 are reflected waves, 11 is a sea surface, 12 is a sum pattern, 13 is a difference pattern,
14, 15, 16, 17 are reception voltages, 18 is a transmission/reception antenna, 19 is a reception-only antenna, 20 is a reception antenna switching device, 21 is a receiver, 22 is a reference signal generator, 23 is a transmitter, and 24 is a transmission/reception switching device. 25 is a distance measuring device, 26 is a target speed measuring device,
27 is a signal processing device, 28 is a pulse radio wave, 29 is a reference signal, 30 is a transmission signal, 31 is a virtual target, 3
2 is a plane where the target and image are present, 33 is a plane where the antennas are arranged, 34 is the central axis of the antenna, 35 is an equal phase plane, and 36 and 37 are calculation results. Note that the same or corresponding parts in the figures are indicated by the same reference numerals.

Claims (1)

[Scope of Claims] 1. An antenna in which a plurality of antenna elements are arranged vertically in one row at predetermined intervals, one of the antenna elements is used as a transmitting antenna, and a target is sent to the transmitting antenna at regular intervals. a transmitter that emits radio waves in a direction; a switching device that uses the antenna element as a receiving antenna to selectively output signals received by each receiving antenna in time series; and a selection signal from the switching device. A receiver provided, a distance measuring device that measures a slant range to the target based on a signal output from the receiver, and a target speed measuring device that measures the relative speed to the target;
λ is the transmission wavelength, i is the splint that distinguishes the received signal from the receiving antenna element, N is the number of receiving antenna elements, and h _i (i=1, 2, ..., N) is the height at which the receiving antenna element is arranged. , R _p is the distance measurement data, v is the relative velocity data, T _r is the transmission period T _r of the above radio wave, θ _p
When is a signal processing variable and K is a non-zero arbitrary constant, t _i = KeXP [−j2π/λ] [{(h _N −h ₁ /2 + R _p tan
_p ) ² + (R _p −υ(1−1)T _r ) ² } ^1/2 + {(h ₁ +h _N /2+R _p tanθ _p +hi) ² + (R _p −υ
(i-1) T _r ) ² } ^1/2 )] Calculate the correction coefficient t _i expressed by i=1, 2,...,N for each receiving antenna, and further calculate the signal obtained by each receiving antenna. S _i
When (i=1,...,N), calculate the operation result V expressed as V=| _N 〓 ⁱ⁼¹ S _i t _i |, change the variable θ _p in the above signal processing, a signal processing device that outputs the peak value of the result V, and the time-series signal selection by the switching device synchronizes with the transmission timing of the radio wave by the transmitter, and also sets a signal processing variable θ _p that provides the peak value. A radar device characterized in that a target is detected as angle information of the target position.