JPH04275745A - Orthogonal modulation circuit - Google Patents

Abstract

PURPOSE:To supply a direct modulator compensating an image signal appearing in the output of the orthogonal modulator owing to the orthogonal phase error of the orthogonal modulator for changing the carrier frequency of the input of the orthogonal modulator through the use of a direct modulation method. CONSTITUTION:A first waveform shaping means 3 which limits band width of an inputted signal, a second waveform shaping means 4 which limits band width of the inputted signal and outputs a signal which is frequency-offset in correspondence with plural channels, an input switch means 2 changing over the input signal, the orthogonal modulation means 6 and an orthogonal phase error compensation means 5 which obtains the amplitude of an interference signal component from the output signal of the second waveform shaping means 4 and the output signal of the orthogonal modulation means and previously compensates the orthogonal phase error of the signal from the first waveform shaping means 3 by the orthogonal phase error are given.

Description

[Detailed description of the invention]

0001

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a quadrature modulation circuit, and more particularly to a quadrature modulation circuit for digital wireless communication.

0002

2. Description of the Related Art In mobile communications, current analog mobile telephones and portable devices have a frequency configuration of single conversion at the transmitter and double conversion at the receiver. FIG. 5 shows a block diagram of the configuration of a single conversion type transmitting circuit. The configuration of this transmitting circuit is as follows: an input terminal 201, an output terminal 202, a serial/parallel (S/P) converter 203, ROM filters 204, 2
05, D/A converters 206 and 207, low-pass filters 208 and 209, quadrature modulator 221, bandpass filters 215 and 219, amplifier 216, mixer 217, frequency sensor 218, and power amplifier 220. Among these, the quadrature modulator 221 is the mixer 210, 21
1, adder 213, 90 degree phase shifter 212, local oscillator 2
14. The bandpass filter 215 is a filter for removing unnecessary harmonic components generated from the output of the quadrature modulator 221.

The operation of the transmitting circuit with this configuration is as follows:
A signal input from 1 is input to ROM filters 204 and 205 via an S/P converter 203, and is band-limited. Next, the output signals from the ROM filters 204 and 205 are converted into analog signals by D/A converters 206 and 207, harmonic components are removed by low-pass filters 208 and 209, and the signals are input to a quadrature modulator 221. . Quadrature modulator 2
The signal modulated by 21 is passed through a bandpass filter 215 to remove unnecessary harmonic components, amplified by an amplifier 216, multiplied by the output of a frequency synthesizer 218 and a multiplication circuit 217, and frequency-converted to a radio frequency (RF) band signal. be done. The frequency-converted RF modulation signal is amplified by power amplifier 220 and output.

Next, FIG. 6 shows the frequency relationship of a single conversion type transmitter circuit. ROM filter 2
The baseband signal 61 waveform-shaped at 04 and 205 is converted into a modulated signal of an intermediate frequency (IF) 62 by a quadrature modulator 221. The frequency synthesizer 218 generates a high carrier band noise ratio (CNR) radio frequency signal (hereinafter referred to as an RF signal) that oscillates at an oscillation frequency of 63 channels and a frequency interval corresponding to the channel interval. The oscillation frequency and intermediate frequency (IF) of the frequency synthesizer 218 according to each channel
) performs a multiplication operation, and an RF modulation signal is generated.

On the other hand, as a technique for reducing the size of a transmitting circuit, there is a direct modulation method in which an RF modulated signal is directly obtained from a baseband signal. FIG. 7 shows a circuit configuration of a conventional direct modulation transmitter. The circuit of this transmitter includes an input terminal 401, an output terminal 402, an S/P converter 403, and a ROM filter 4.
04, 405, digital to analog converter 406, 40
7. Consists of a quadrature modulator 420, a frequency synthesizer 414, a bandpass filter 415, and a power amplifier 416. In this direct modulation, the S/P converter 403 directly modulates the baseband signals of the I and Q channels into RF signals.
5. The amplifier 216 and mixer circuit 217 can be omitted. It is possible to significantly downsize the transmitter circuit. Recently, 800 MHz band quadrature modulators have been fabricated into monolithic ICs using a gallium arsenide process.

FIG. 8 shows the configuration of a quadrature modulator. The same figure (A
), the quadrature modulator includes a 90 degree phase shifter 705 for shifting the carrier signal by 90 degrees, and a differential amplifier 706.
, 707, I channel double balanced mixer 708,
It is composed of a Q channel double balanced mixer 709, a 0 degree synthesizer 710, and a buffer amplifier 711. In the case of the single conversion type transmitting circuit shown in FIG. 5, the carrier frequency of the quadrature modulator 221 is fixed, so the frequency-dependent 90 degree phase shifter 212 is used to oscillate the local oscillator 214, which is a fixed oscillator. It is sufficient to set it for the frequency, and the quadrature phase error of the quadrature modulator 221 is fixed and small.

[0007]

However, in the case of direct modulation as shown in FIG. 7, the input carrier frequency of the quadrature modulator 420 is changed by the frequency synthesizer 414 according to the center frequency of each channel. Therefore, mainly by this 90 degree phase shifter 412, the in-phase component of the carrier signal,
Quadrature phase errors of orthogonal components will occur.

This quadrature phase error will be explained. First, I-channel data and Q-channel data input to the quadrature modulator are expressed as shown in the following equation.

I(t)=Kcos [φi+Δωnt]

(1) Q(t)=K(1+ΔM) sin[φ
i+Δωnt]
(2) Here, Δωn=2πΔfn, ΔM is the amplitude error of the orthogonal component. When I-channel data and Q-channel data are orthogonally modulated using an in-phase carrier signal cos [ωct] and an orthogonal carrier signal sin [ωct + Δθ], S(t)=K cos [φi + Δωt] cos [ω
ct] −K(1+ΔM) sin[φ
i +Δωt] sin [ωct+Δθ]


(3), and further rearranging equation (3), S(t)=[K(1+
cos Δθ+ΔMcosΔθ) /2] cos [φ
i+(ωc+Δω)t) −[K (si
n Δθ+ ΔMsin Δθ)/2] sin [φ
i+( ωc+Δω)t) +[K (
1-cos Δθ- ΔMcos Δθ) / 2]
cos [−φi + (ωc−Δω)t]
+[K (sin Δθ+ ΔMsin Δθ)/2
]sin [−φi+(ωc−Δω)t)


(4) If the number of output bits of the ROM filter is 10 bits, the amplitude error is expressed by one least significant bit (LSB), and the amplitude ratio of the image component to the desired signal due to the amplitude error is -60 dB or less. , ignoring the term including the amplitude error ΔM of the orthogonal component, S(t) = [K(1 + cos Δθ)/2] cos
[φi + (ωc +Δω)t) − [K
(sin Δθ)/2] sin[φi +( ωc
+Δω)t) +[K (1 −co
s Δθ) / 2] cos[−φi + (ωc
-Δω)t] +[K (sin Δθ)
/2] sin [−φi+( ωc −Δω)t)


(5) It becomes.

Furthermore, in the configuration of the orthogonal modulator shown in FIG.
Since only one differential amplifier 716 is required for the 18 inputs, the consumption of the circuit can be reduced. In this configuration, it is assumed that the in-phase channel and the orthogonal channel each have a phase error of Δθ/2 with respect to the local carrier signal. In this case, the quadrature phase error is expressed by the following equation.

[0011] S(t)=[K(cos Δθ /2] co
s [φi+(ωc+Δωt)t]
−K [sin(Δθ)/2)]sin [−φ
i+(ωc −Δωt)t]


(6) In any case, if the orthogonality of the quadrature modulator is incomplete, the desired wave component [φi+(ωc+Δω
t)t], the image component [−φi+(ωc −
Δω)t] occurs, and this component becomes a spurious wave to its own channel.

For example, in a quadrature modulator IC for mobile communication, image suppression is about 30 dB with respect to a center frequency of 140 MHz using a bipolar process, and it is necessary to reduce image components to about 60 dB.

[0013] The present invention has been made in view of the above points.
In order to downsize mobile devices, portable devices, etc., a direct modulation method is used that can omit IF band amplifiers, filters, and mixer circuits, and is advantageous in downsizing the transmitter circuit. It is an object of the present invention to provide a direct modulator that compensates for the image signal appearing at the output of a quadrature modulator due to the quadrature phase error of the quadrature modulator due to frequency changes.

[0014]

[Means for Solving the Problems] FIG. 1 is a diagram showing the basic configuration of a quadrature modulator according to the present invention. a first waveform shaping means (3) for band-limiting the channel data inputted from the input terminal (1); A second waveform shaping means (4) outputs channel data whose frequency is offset by inputting the channel data to the second waveform shaping means (4) when a channel is connected, and when transmitting a signal, the channel data is inputted to the first waveform shaping means (4). input switching means (2) for switching the input signal to be input to the means (3);
), orthogonal modulation means (6) that receives channel data from the first waveform shaping means (3) and the second waveform shaping means (4), performs phase modulation, and generates and outputs a modulated signal; By detecting the amplitude of the interference signal component from the output channel data of the waveform shaping means (4) and the output signal of the orthogonal modulation means, a quadrature phase error is detected, and the first waveform shaping means (3) uses the orthogonal phase error. and a quadrature phase error compensating means (5) for compensating the quadrature phase error in advance to the channel data outputted from the channel data. In addition, quadrature phase error compensation means (5
) is a first multiplication means (7) which inputs the output signal of the orthogonal modulation means (6) and the local oscillation signal of the orthogonal modulation means and outputs a frequency-converted signal; ) converts the output signal into a digital value and outputs a digital signal, the output of the second waveform shaping means (4), and the output of the digital conversion means (8) are input, and an orthogonal a second multiplication means (
9), and quadrature phase error output means (11) for outputting quadrature phase compensation data in response to the output of the second multiplication means (9).
and a calculation means (12) for performing quadrature phase error compensation calculation using quadrature phase compensation data on the channel data output from the first waveform shaping means (3).

[0015]

According to the present invention, when a channel is initially set, input channel data is input to the second waveform shaping means, and the input signal is separated into a desired wave signal component and an interference signal component by the orthogonal modulation means and output. Further, during channel data transmission, channel data is input to the first waveform shaping means. The quadrature phase error is detected by multiplying the output signal from the second waveform shaping means by the output signal from the quadrature modulation means and detecting the amplitude of the interference signal component. This quadrature phase error is used to perform a quadrature phase error compensation calculation with the output signal of the first waveform shaping means, and by compensating the output signal of the first waveform shaping means in advance, it is possible to eliminate imperfections in the quadrature modulator. It is possible to suppress interference signal components caused by orthogonality.

[0016]

DESCRIPTION OF THE PREFERRED EMBODIMENTS FIG. 2 shows the basic configuration of a quadrature modulator according to the present invention. The orthogonal modulator shown in the figure has an input terminal 101, an output terminal 102, a serial/parallel converter (hereinafter referred to as an S/P converter) 103, a switch circuit 60, and a first waveform shaping circuit 10.
, second waveform shaping circuit 20, quadrature phase error compensation circuit 50
, D/A converter 104, low pass filter 105, quadrature modulator 30, mixers 31, 32, adder 33, 90 degree phase shifter 34, frequency synthesizer 35, mixers 106, 1
09, low-pass filters 107 and 110, an A/D converter 108, a memory circuit 40, and a latch circuit 111.

The operation of this configuration will be explained. In FIG. 2, a first waveform shaping circuit 10 band-limits I and Q channel data input from an input terminal 101. Second
The waveform shaping circuit 20 band-limits the I and Q channel data and offsets the frequency for each channel. The switch circuit 60 inputs data to the second waveform shaping circuit 20 at the initial stage of channel connection, and further inputs channel data to the first waveform shaping circuit 10 at the time of channel data transmission. D/A converter 104 converts I and Q channel data into analog values. Low pass filter 105 removes harmonic components. The quadrature modulator 30 includes mixers 31 and 32, an adder 33, a 90 degree phase shifter 34, and a frequency synthesizer 35.
Consisted of. The quadrature modulator 30 receives the output of the low-pass filter 105, generates a modulated signal, and outputs the modulated signal. The mixer 106 receives the output signal of the quadrature modulator 30 and the output signal of the frequency synthesizer 35 as input. Low-pass filter 107 removes harmonic components of the output signal of mixer 106. The A/D converter 108 is a low pass filter 10
The output signal of 7 is converted into a digital value. Mixa 109
inputs the output signal of the second waveform shaping circuit 20 and the digital signal that is the output signal of the A/D converter 108, and calculates the orthogonal phase error component. A low-pass filter 110 removes harmonic components from the calculated quadrature phase error component, and inputs the resultant to the memory circuit 40 . The memory circuit 40 is the mixer 10
Quadrature phase error compensation data is output in accordance with the quadrature phase error component which is the output signal of No. 9. Next, the quadrature phase error compensation circuit 50 performs quadrature phase error compensation calculation using the output data of the memory circuit 40 and the I and Q channel data output signals of the first waveform shaping circuit 10 using the quadrature phase error compensation data.

FIG. 3 shows an explanatory diagram of quadrature phase error compensation of the quadrature modulator of the present invention. If the baseband signal (a) has no frequency offset, the image signal component caused by the quadrature phase error of the quadrature modulator falls within the band of its own channel, and therefore the signal component and image component cannot be separated. Therefore, the frequency of the baseband signal (a) is offset by digital processing. When a signal with a Δf frequency offset is orthogonally modulated, the synthesizer also has an oscillation frequency (b)
A desired wave component (d) is generated at a frequency +Δf away from the desired wave component (d), and an image component (c) is generated at a frequency −Δf apart from the desired wave component (d). The present invention extracts this image component (c), detects the quadrature phase error corresponding to each element of the quadrature modulator, and pre-compensates signal data for each frequency-offset I channel and Q channel data. It performs calculations and compensates for the orthogonal error of the orthogonal modulator using baseband digital processing.

As described above, in the present invention, first, the second waveform shaping circuit 20 frequency-offsets the baseband signal, and the orthogonal modulator 30 separates the desired signal component and the image signal component in the RF band. The phase error Δ in the quadrature modulator 30
When there is θ, the output modulation signal of the orthogonal modulator 30 is expressed by equation (5) when the orthogonal modulator shown in FIG. 8 is used.

[0020] The signal after quadrature modulation is converted into an in-phase carrier wave signal co
Multiplying by s[ωct] and removing high frequency components with a low-pass filter yields R(t)=K cos[φi +Δωt]−K(
sin Δθ) sin [φi +Δωt]


(7) becomes. To explain this using the configuration of FIG. 2, the output component of equation (7) is converted into digital data by the A/D converter 108, and this digital data and the output of the ROM filter of the second waveform shaping circuit 20 are used to generate In the mixer 109, K sin [φi +Δω
t], the quadrature phase error component T(t), which is the output of the mixer 109, becomes T(t)=K2/2 sin[2 φi +2
Δωt] −K2( sinΔ
θ) /2{1-cos[2φi+2Δωt]
(8) becomes. By band-limiting the output signal T(t) of the mixer 109 with a low-pass filter 110, it is possible to detect a DC component of orthogonal transfer error compensation that is proportional to the orthogonal phase error angle of the orthogonal modulator 30. This DC component U(t
) is U(t)=-K2(sinΔθ)/2

(9) Next, in the memory circuit 40, an actual correction value (calculation is performed on the band-limited data using this value) is obtained from the orthogonality compensation component sinΔθ of the orthogonal modulator 30: α= tanΔθ β=−1+1/ cosΔθ can get. The input data is band-limited, and the offset frequency corresponding to each channel of the first waveform shaping circuit 10 and the correction values α and β obtained in the memory circuit 40 are multiplied by the quadrature phase error compensation circuit 50. , and the output signal of the first waveform shaping circuit 10 is added to that value to compensate for the quadrature phase error.

FIG. 4 shows the configuration of an embodiment of a quadrature modulator according to the present invention. In the figure, the same components as those in FIG. 2 are given the same reference numerals. The configuration in the figure includes a data input signal terminal 101, an S/
An orthogonal phase filter consisting of a P converter 103, a data input changeover switch 60, a waveform shaping ROM filter 11 that performs band limitation, a ROM filter 22 that performs band limitation and frequency offset, an adder 21, a multiplier 51, and an adder 52. Error compensation circuit 50, D/A converter 104, low pass filter 105, mixers 31, 32, 0 degree synthesizer 33, 90
A quadrature modulator 30 composed of a degree phase shifter 34, a frequency sensor 35, a mixer 106, and a low-pass filter 10
7, an A/D converter 108, a mixer 109, a low-pass filter 110, a ROM 41, and a latch circuit 111.

In this embodiment, the communication channel is determined and the oscillation frequency of the frequency synthesizer 35 is set to the center frequency. In this case, the channel data signal is transmitted to the switch circuit 6
0 is input to the second waveform shaping circuit 20. The signal input to the second waveform shaping circuit 20 is band-limited and Δf
frequency offset. After the frequency-offset signal is converted into an analog signal by the D/A converter 104, the low-pass filter 105 removes harmonic components, and the signal is orthogonally modulated by the quadrature modulator 30. When the quadrature modulator 30 has a quadrature phase error, an image signal component appears at a frequency −Δf apart from the local frequency in the RF band, as shown in FIG.

Next, the mixer 106 frequency-converts the output signal of the quadrature modulator 30 and the output signal of the frequency synthesizer 35, removes high frequency components from the Δf frequency offset signal with a low-pass filter 107, and performs A/D conversion. It is converted into a digital value by a converter 108. This digitally converted signal is multiplied by the output data of the second waveform shaping circuit 20 by the mixer 109. The output signal of the mixer 109 is band-limited by a low-pass filter 110, and a DC component proportional to the quadrature phase error angle of the quadrature modulator 30 is obtained. The output signal of low-pass filter 110 is input to memory circuit 40 . Memory circuit 40 determines quadrature phase error compensation data.

Next, the switch circuit 60 is switched, and the input data is input to the first waveform shaping circuit 10. The input data is band-limited by the ROM filter 11 of the first waveform shaping circuit 10 and is input to the quadrature phase error compensation circuit 50 . The input data input to the quadrature phase error compensation circuit 50 is multiplied and added to the quadrature phase error data obtained by the memory circuit 40 in a mixer 51 and an adder 52. The output signal of the quadrature phase error compensation circuit 50 is a D/A
The signal is converted into an analog signal by a converter 104, band-limited by a low-pass filter 105, orthogonally modulated by a quadrature modulator 30, and output.

Note that the RF mixer 106 is a quadrature modulator 3.
It is made into a monolithic IC together with the 0-degree mixers 31 and 32, the 90-degree phase shifter 34, and the 0-degree synthesizer 33. Furthermore, the newly added baseband circuit, excluding the low-pass filter 107, is integrated into a one-chip LSI using a CMOS process, together with a ROM filter for band limitation and a D/A converter. Therefore, the quadrature phase error compensation circuit newly added in this embodiment does not cause a large increase in circuit scale compared to the conventional direct modulation circuit.

[0026]

As described above, according to the present invention, in the direct modulation method, the image signal caused by the quadrature phase error of the quadrature modulator, which is a problem, can be corrected by quadrature phase error that appears in the output of the quadrature modulator. By correcting with the phase error compensation circuit, unnecessary components that cause interference to the same channel can be reduced. Further, direct quadrature modulation in RF is possible using a monolithic IC quadrature modulator, and the IF band circuit in the conventional double conversion circuit can be omitted, so the transmitting circuit can be made smaller.

[Brief explanation of the drawing]

FIG. 1 is a diagram showing the principle configuration of a quadrature modulator according to the present invention.

FIG. 2 is a basic configuration diagram of a quadrature modulator according to the present invention.

FIG. 3 is an explanatory diagram of quadrature phase error compensation of the quadrature modulator of the present invention.

FIG. 4 is a configuration diagram of an embodiment of a quadrature modulator of the present invention.

FIG. 5 is a block diagram of the configuration of a single conversion type transmitting circuit.

FIG. 6 is a diagram showing the frequency relationship of a single conversion type transmitter circuit.

FIG. 7 is a circuit configuration diagram of a conventional direct modulation transmitter.

FIG. 8 is a configuration diagram of a quadrature modulator.

[Explanation of symbols]

1 Input terminal 2 Input switching means 3 First waveform shaping means 4 Second waveform shaping means 5 Quadrature phase error compensation means 6 Orthogonal modulation means 7 First multiplication means 8 Digital conversion means 9 Second multiplication means 10 First waveform shaping circuit 11 Quadrature phase error output means 12 Calculation means 13 Output terminal 20 Second waveform shaping circuit 30 Quadrature modulator 31, 32 mixer 33 0 degree synthesizer 34 90 degree phase shifter 35 Frequency synthesizer 40 Memory circuit 50 Quadrature phase error compensation circuit 51 Multiplier 52 Adder 60 Switch circuit 106 Multiplication circuit 107 Low pass filter 108 A/D converter 109 Multiplication circuit 110 Low pass filter

Claims (2)

[Claims] 1. A first waveform shaping means for band-limiting channel data input from an input terminal; and a first waveform shaping means for band-limiting channel data input from the input terminal, and a frequency offset corresponding to each of a plurality of channels. a second waveform shaping means for outputting the channel data, and inputting the channel data to the second waveform shaping means at the time of channel connection, and inputting the channel data to the first waveform shaping means at the time of signal transmission. input switching means for switching input signals; orthogonal modulation means for taking in the channel data from the first waveform shaping means and the second waveform shaping means, performing phase modulation, and generating and outputting a modulated signal; By detecting the amplitude of the interference signal component from the channel data output from the second waveform shaping means and the output signal from the orthogonal modulation means, a quadrature phase error is detected, and the first waveform shaping is performed using the quadrature phase error. A quadrature modulation circuit comprising quadrature phase error compensating means for compensating in advance a quadrature phase error to the channel data outputted from the means. 2. The orthogonal phase error compensation means includes a first multiplication means that inputs the output signal of the orthogonal modulation means and the local oscillation signal of the orthogonal modulation means and outputs a frequency-converted signal; Digital conversion means converts the output signal of the first multiplication means into a digital value and outputs a digital signal, the output of the second waveform shaping means and the output of the digital conversion means are input, and the quadrature phase error component is a second multiplication means for outputting quadrature phase compensation data corresponding to the output of the second multiplication means, and a quadrature phase error output means for outputting quadrature phase compensation data corresponding to the output of the second multiplication means, and the channel data output from the first waveform shaping means. 2. The orthogonal modulation circuit according to claim 1, further comprising arithmetic means for performing orthogonal phase error compensation calculation using the orthogonal phase compensation data.