JP5841098B2 - Zero current switching circuit and full bridge circuit - Google Patents

Description

  The present invention relates to a zero current switching circuit, and more particularly, to a zero current switching circuit that can reduce the switching loss of a switching element without using the resonance characteristics of an LC circuit.

  FIG. 24 shows an example of a conventional load drive circuit that controls pulse width modulation (PWM) of power supplied to a load. In such a load control circuit for PWM control, in order to control the power supplied to the electric load M, a switching element Q such as a MOSFET (metal oxide semiconductor field-effect transistor) is ON / OFF controlled at a high frequency. Is done. Normally, in the ON / OFF control of the switching element Q, a switching loss occurs during the transition period of the switching element Q. Switching loss is determined by the product of the voltage across the switching element and the current flowing through the switching element.

  FIG. 25A shows temporal changes in voltage and current when the switching element Q transitions from the OFF state to the ON state (when turned on). When the voltage indicated by the solid line I starts to decrease, the current indicated by the broken line II rapidly increases to the maximum value and then decreases. Thus, when the switching element Q transitions from the OFF state to the ON state, the current increases before the voltage becomes 0V.

  FIG. 25B shows temporal changes in voltage and current when the switching element Q transitions from the ON state to the OFF state (when turned off). After the voltage indicated by the solid line III rises to some extent, the current indicated by the broken line IV decreases. Thus, when the switching element Q transitions from the ON state to the OFF state, the voltage rises before the current becomes 0A.

  As described above, during the transition period of the switching element Q, a current flows when the voltage is not 0 V, so that a switching loss occurs. In the pulse width modulation circuit, since the switching element is repeatedly ON / OFF controlled at a high frequency, the switching loss increases.

  As a technique for reducing the switching loss, it is effective to shorten the switching time (rise or fall transition time). In that case, a contradiction occurs that noise performance deteriorates. For this reason, it is necessary for the system to adjust the switching time and the switching frequency in order to achieve both the target noise performance and temperature performance conditions. However, in general, it is difficult to achieve both noise performance and temperature performance goals simply by adjusting the switching time and switching frequency. For this reason, it is necessary to add a further circuit as a separate measure, which causes a rise in the price of the system.

  As another technique for reducing switching loss, the resonance characteristics of the inductor (L) and the capacitor (C) are used to delay the rise or fall of the current or voltage at the time of switching and It is known to shift each other. For example, when the switching element transitions from the OFF state to the ON state, the rise of the current starts to rise with a delay after the voltage drops to 0V. Similarly, when the switching element transitions from the ON state to the OFF state, the rising of the voltage starts rising after a delay after the current decreases to 0A.

  Further, as a switching circuit for reducing switching loss of a switching element, Patent Document 1 below describes a zero current switching (ZCS) device at the time of switching on. Patent Document 2 below describes zero voltage switching (ZVS) when the switch is turned on.

JP 2011-239527 A JP 2011-244661 A

  The voltage phase and the current phase are not shifted from each other at the time of switching, and the conventional technology uses the resonance characteristics of the inductor (L) and the capacitor (C). However, in the technology using such LC resonance characteristics, the switching frequency and driving duty of the switching element are limited, and complicated timing control of the switching element is necessary.

  Further, in the power conversion circuit described in Patent Document 1, it is necessary to further provide a snubber circuit around the switching element in order to protect the switching element, and the circuit configuration is complicated. Further, in the switching power supply device described in Patent Document 2, it is necessary to further provide another switching element in order to realize ZVS (zero voltage switching), and the circuit configuration is complicated.

  SUMMARY OF THE INVENTION Accordingly, the first aspect of the invention is to provide a zero current switching circuit that can reduce the switching loss of the switching element with a simple circuit configuration without using the LC resonance characteristics. Another object of the second invention is to provide a full bridge circuit using a zero current switching circuit that does not use LC resonance characteristics and can reduce the switching loss of the switching element with a simple circuit configuration. Yes.

  In order to achieve the above object, a first invention is a zero current switching circuit for controlling power supplied to a load, the load having one terminal connected to a first potential source, and the other of the loads. A switching element in which one terminal is connected via an inductor connected to the other terminal and the other terminal is connected between the second potential source and the load or the switching element and the inductor in parallel. And a first return circuit connected in parallel to the load, and a second return circuit connected in parallel to the inductor.

  As described above, according to the first aspect of the present invention, the load driving circuit that controls the power supplied to the load is configured by the zero current switching circuit. In this zero current switching circuit, when the switching element is turned on and off, an inductor having a characteristic that the current is delayed from the voltage and a capacitor having a characteristic that the current is earlier than the voltage operate with these characteristics.

  As a result, when the switching element is turned on, even if the voltage between both terminals of the switching element starts to drop, the amount of current flowing through the switching element does not increase immediately due to the action of the inductor, and the voltage becomes 0V. Start increasing. In this way, when the switching element is turned on, the rise of the current is delayed from the fall of the voltage due to the characteristics of the inductor, so that the voltage drops when the current is 0 A and then increases when the voltage is 0 V. . Since switching loss does not occur when the voltage or current is 0, the switching loss when the switching element is turned on can be reduced.

  Also, when the switching element is turned off, even if the current flowing through the switching element starts to decrease, the voltage across the switching element does not increase immediately due to the action of the capacitor, but starts increasing after the current reaches 0A. . In this way, when the switching element is turned off, the current falls faster than the voltage rise due to the characteristics of the capacitor, so that the current decreases when the voltage is 0 V, and then the voltage rises when the current is 0 A. . When the voltage or current is 0, no switching loss occurs, so that the switching loss when the switching element is turned off can be reduced.

Thus, according to the first invention, the switching loss of the switching element can be reduced without using the LC resonance characteristics and with a simple circuit configuration. Therefore, according to the first invention, the switching loss can be reduced without being restricted by the switching speed and switching frequency of the switching element.
In addition, the delay time of the current generated at the time of turn-on and the delay time of the voltage generated at the time of turn-off can be adjusted to independent amounts by adjusting the values of the inductor and the capacitor. Noise characteristics generated by the switching circuit can be easily set to a desired limit value or less.

  Preferably, in the first aspect of the present invention, a first rectifier element that is connected between the first potential source and the capacitor and selectively passes a current from the second potential source side to the first potential source side; , Connected from a first node between the first rectifier element and the capacitor and a second node between the switching element and the inductor, from the second node side to the first node side. A second rectifier and a third rectifier connected in series with each other, a third node between the second rectifier and the third rectifier, and the first or second potential source And a second capacitor connected between the first and second capacitors.

  As a result, when the switching element is turned on, a part of the current flowing through the inductor can be guided to charge the capacitor. For this reason, at the time of turn-on, the efficiency can be further improved by transferring the energy stored in the inductor to the capacitor and regenerating the energy transferred to the capacitor and regenerated to the load.

  According to a second aspect of the present invention, there is provided a full bridge circuit including four switching circuits, wherein the four switching circuits are configured by the zero current switching circuit according to claim 1 or 2.

  As described above, according to the second invention, the four switching circuits constituting the bridge can reduce the switching loss of the switching element with a simple circuit configuration without using the LC resonance characteristics. With this configuration, switching loss in the entire full bridge can be greatly reduced.

1 is a circuit diagram of a zero current switching circuit according to a first embodiment of the first invention. FIG. 2 is a timing chart of the zero current switching circuit shown in FIG. 1. (A) And (B) is operation | movement explanatory drawing of the zero current switching circuit shown in FIG. (A) And (B) is operation | movement explanatory drawing of the zero current switching circuit shown in FIG. 1 following FIG. 3 (B). (A) And (B) is operation | movement explanatory drawing of the zero current switching circuit shown in FIG. 1 following FIG. 4 (B). It is a circuit diagram of the zero current switching circuit by 2nd Embodiment of 1st invention. It is a circuit diagram of the zero current switching circuit by 3rd Embodiment of 1st invention. It is a circuit diagram of the zero current switching circuit by 4th Embodiment of 1st invention. It is a circuit diagram of the zero current switching circuit by 5th Embodiment of 1st invention. It is a circuit diagram of the zero current switching circuit by 6th Embodiment of 1st invention. It is a circuit diagram of the zero current switching circuit by 7th Embodiment of 1st invention. It is a circuit diagram of the zero current switching circuit by 8th Embodiment of 1st invention. It is a circuit diagram of the zero current switching circuit by 9th Embodiment of 1st invention. 14 is a timing chart of the zero current switching circuit shown in FIG. 13. It is operation | movement explanatory drawing of the zero current switching circuit shown in FIG. FIG. 15B is an operation explanatory diagram of the zero current switching circuit shown in FIG. 13 following FIG. 15A. FIG. 15B is an operation explanatory diagram of the zero current switching circuit shown in FIG. 13 following FIG. 15B. FIG. 15 is an operation explanatory diagram of the zero current switching circuit shown in FIG. 13 following FIG. 15C. FIG. 15D is an operation explanatory diagram of the zero current switching circuit shown in FIG. 13 following FIG. 15D. It is a circuit diagram of the zero current switching circuit by 10th Embodiment of 1st invention. It is a circuit diagram of the zero current switching circuit by 11th Embodiment of 1st invention. It is a circuit diagram of the zero current switching circuit by 12th Embodiment of 1st invention. It is a circuit diagram of the zero current switching circuit by 13th Embodiment of 1st invention. It is a circuit diagram of the zero current switching circuit by 14th Embodiment of 1st invention. It is a circuit diagram of the zero current switching circuit by 15th Embodiment of 1st invention. It is a circuit diagram of the zero current switching circuit by 16th Embodiment of 1st invention. FIG. 5 is a circuit diagram of a full bridge circuit according to an embodiment of the second invention. It is a circuit diagram of a conventional load drive circuit for PWM control. FIG. 24A is a graph showing temporal changes in voltage and current when the load driving circuit shown in FIG. 24 is turned on. FIG. 24B is a graph showing voltage and current changes when the load driving circuit shown in FIG. It is a graph which shows a time change.

Hereinafter, embodiments of the present invention will be described with reference to the drawings.
FIG. 1 shows a circuit diagram of a zero current switching circuit according to a first embodiment of the first invention. The zero current switching circuit shown in FIG. 1 is a load drive control circuit for controlling the power supplied to the load M. Between the high potential source V _IN (the terminal on the high potential side of the DC power supply) as the first potential source and the low potential source (GND) (the terminal on the low potential side of the DC power supply) as the second potential source, A load M, an inductor 12 and a switching element 10 are sequentially connected in series.

More specifically, one terminal ma of the load M is connected to the high potential source V _IN , and one terminal 10a of the switching element 10 is connected to the low potential source (GND). An inductor 12 is connected between the other terminal mb of the load M and the other terminal 10b of the switching element 10.

  Power supply to the load M is controlled by ON / OFF of the switching element 10. A PWM control signal is input from a PWM circuit (not shown) to the gate electrode of the MOSFET of the switching element 10.

  The load M is, for example, a motor, and the switching element 10 is, for example, a MOSFET. Further, the switching element 10 may be configured by a bipolar transistor or an insulated gate bipolar transistor (IGBT) instead of the MOSFET.

  As will be described later, when the switching element 10 is turned on, even if the voltage between both terminals of the switching element starts to decrease due to the action of the inductor 12, the amount of current flowing through the switching element does not increase immediately, Starts rising after the voltage reaches 0V.

The capacitor 14 is connected in parallel with the load M, that is, in series with the switching element 10 and the inductor 12. Further, a resistor 20 and a diode 22 connected in parallel with each other are connected between the capacitor 14 and the high potential source V _IN .

  As will be described later, when the switching element 10 is turned off, even if the current flowing through the switching element starts to decrease due to the function of the capacitor 14, the voltage across the switching element immediately increases due to the function of the capacitor. First, the current starts to rise after the current reaches 0A.

  A first return diode 16 as a first return circuit is connected to the load M in parallel. The first freewheeling diode 16 circulates the regenerative current due to the induced electromotive force of the load M when the switching element 10 is turned off.

  A second return diode 18 as a second return circuit is connected in parallel to the inductor 12. The second freewheeling diode 18 circulates a current caused by the induced electromotive force of the inductor 12 when the switching element 10 is turned off.

An operation example of the zero current switching circuit shown in FIG. 1 will be described with reference to the timing chart of FIG. 2 and FIGS. 3A to 5B.
Curve I in Figure 2 shows the time variation of the gate voltage V _G of the switching element 10, the curve II shows the time variation of the potential V _Q of the other terminal 10b of the switching element 10, the curve III is the load M represents the potential V _OUT of the other terminal mb, curve IV shows the time variation of the current I _Q flowing through the switching element 10, the curve V represents the time variation of the current I _L flowing through the inductor 12, curve VI is a capacitor 14 shows the time change of the current I _C flowing through the first free-wheeling diode 16, the curve VII shows the time change of the current I _D flowing through the load M, and the curve VII shows the time change of the current I _OUT flowing through the load M.

Since one terminal 10 a of the switching element 10 is grounded (GND), the potential V _Q of the other terminal 10 b of the switching element 10 forms a voltage between both terminals of the switching element 10, that is, the switching element 10. This corresponds to the voltage between the drain (D) and the source (S) of the MOSFET. Further, I _OUT flowing through the load M is constant due to the inductance component included in the load M and PWM control.

  Here, among the ON / OFF operations of the switching element 10 repeated for PWM control, a pair of operations at the time of turn-on and turn-off will be described. In the timing chart of FIG. 2, the switching element 10 is turned on at times t0 to t1, and is turned off at times t4 to t5.

  3A shows the operation of the zero current switching circuit at time t0 to t1 in the timing chart of FIG. 2, FIG. 3B shows the operation at time t1 to t2, and FIG. 4B shows the operation from time t3 to t4, FIG. 5A shows the operation from time t4 to t5, and FIG. 5B shows the operation from time t5 to time t5. Represents the operation of t6.

First, the operation of the zero current switching circuit at times t0 to t1 in the timing chart of FIG. 2 will be described. Prior to time t0, the gate voltage V _G of the switching element 10 starts to rise from “L” to “H”. When the gate voltage V _G reaches the turn-on threshold value of the MOSFET of the switching element 10 at time t0, the drain-source voltage V _{Q of the} MOSFET starts to decrease. However, due to the characteristics of the inductor 12, the current I _L flowing through the inductor 12 remains at 0A at times t0 to t1. Therefore, current I _Q flowing through the switching element 10 connected in series to the inductor 12 which is still 0A. Therefore, the switching loss in the switching element 10 at time t0 to t1 is zero.

FIG. 3A shows the current flowing through the zero current switching circuit at times t0 to t1. The current I _OUT due to the induced electromotive force of the load M when the switching element 10 is turned off before the time t 0 circulates through the first freewheeling diode 16. At time t0 to t1, no current flows through the switching element 10 or the inductor 12 as described above.

Next, the operation of the zero current switching circuit at times t1 to t2 in the timing chart of FIG. 2 will be described. By time t1, the drain-source voltage V _Q of the MOSFET of the switching element 10 is reduced to 0V. Then, over a period from time t1 to time t2, the current flowing through the inductor 12 I _L, and the current I _Q flowing through the switching element 10 gradually increases. After time t1, the current circulating through the first return diode 16 gradually decreases to 0A. Subsequently, a current I _C for charging the capacitor 14 also flows. Therefore, current I _Q and the current I _L is increased to the current value current I _C is added to charge the capacitor 14. However, the drain-source voltage V _Q of the switching element 10 at time t1 to t2 has already been 0V. Therefore, the switching loss in the switching element 10 at time t1 to t2 is also zero.

FIG. 3B shows a current flowing through the zero current switching circuit at time t1 to t2. And a charging current Ic of current I _OUT and capacitor 14 through the load M is joined, the current I _L flowing through the inductor 12. Further, the current I _L, the current I _Q flowing through the switching element 10 connected in series directly to the inductor 12.
As described above, the switching element 10 can be turned on from the OFF state to the ON state with substantially no switching loss from time t0 to time t2.

Next, the operation of the zero current switching circuit at times t2 to t3 in the timing chart of FIG. 2 will be described. From time t2 to t3, the zero current switching circuit is in the ON state. After time t2, the potential V _OUT of the other terminal mb of the load M gradually decreases, and at the same time, the charging current Ic of the capacitor 14 gradually decreases. Accordingly, the current flowing through the inductor 12 I _L, and also the current I _Q flowing through the switching element 10 decreases. When the potential V _OUT becomes 0 V, the charging current I _C also becomes 0 A, and the current I _L and the current _IQ are in a steady state.

FIG. 4A shows the current flowing through the zero current switching circuit at times t2 to t3. Current flowing through the inductor 12 I _L, and the current I _Q flowing through the switching element 10, when a steady state, the current I _OUT through the load M is directly current I _L, and the further the current I _Q.
The MOSFET constituting the switching element 10 has an ON resistance of several milliohms. Therefore, when the switching element 10 is in an ON state, a slight loss occurs due to the the on-resistance and current I _Q.

Subsequently, the switching element 10 is turned off from time t3 to time t5.
The operation of the zero current switching circuit at times t3 to t4 in the timing chart of FIG. 2 will be described. Prior to time t3, the gate voltage V _G of the switching element 10 starts to decrease from “H” to “L”. At time t3, when the gate voltage V _G reaches the MOSFET turn-off threshold of the switching element 10, the current I _Q flowing through the switching element 10 starts to descend. However, at the same time as the falling start of the current I _Q, since the discharge current I _C from the capacitor 14 starts to rise, the potential V _OUT of the other terminal mb load M remains 0V. At the same time, the drain-source voltage V _Q of the MOSFET of the switching element 10 does not become equal to or higher than V _{OUT due to} the action of the second freewheeling diode 18 and therefore remains substantially 0V. Therefore, the switching loss in the switching element 10 at time t3 to t4 is zero.

FIG. 4B shows a current flowing through the zero current switching circuit at times t3 to t4. When the current I _Q flowing through the switching element 10 is reduced, since the induction by inductor 12 current flows through the second reflux diode 18, current I _L that flows through inductor 12 is maintained constant. The current I _OUT through the load M is divided into a current I _Q flowing in the current I _C and the inductor 12 through the capacitor 14.

Next, the operation of the zero current switching circuit at times t4 to t5 in the timing chart of FIG. 2 will be described. By time t4, the 0A switching element 10 flows a current I _Q is reduced. Thereafter, from time t4 to time t5, the capacitor 14 starts charging while maintaining the output current _IOUT flowing through the load M constant. As a result, the source of the switching element 10 - together with the drain voltage V _Q gradually increases, the potential V _OUT of the other terminal of the load M gradually increases. However, current I _Q flowing through the switching element 10 at time t4~t5 is already 0A. Therefore, the switching loss in the switching element 10 from time t4 to t5 is also zero.
In the period from time t4 to time t5, the induced electromotive force of the inductor 12 is generated, so that the current I _L flowing through the inductor 12 gradually decreases.

FIG. 5A shows the current flowing through the zero current switching circuit at times t4 to t5. The output current I _OUT flowing through the load M becomes the charging current I _C of the capacitor 14. The current I _L by the induced electromotive force of the inductor 12 is flowing through the second reflux diode 18.
As described above, the switching element 10 can be turned off from the ON state to the OFF state without switching loss from time t3 to time t5.

Next, the operation of the zero current switching circuit at times t5 to t6 in the timing chart of FIG. 2 will be described. At time t5 to t6, the switching element 10 is already in the OFF state. When the capacitor 14 is fully charged, the charging current I _C flowing through the capacitor 14 gradually decreases. Extent that the charging current I _C is decreased, the regenerative current I _D flowing through the first freewheeling diode 16 is increased, as a result, the output current I _OUT of the load M is kept constant. The current I _L by the induced electromotive force of the inductor 12, the time until after t6 has elapsed, continues to flow while gradually decreases.

FIG. 5B shows the current flowing through the zero current switching circuit at times t5 to t6. Current I _L by the induced electromotive force of the inductor 12 is gradually decreased and eventually becomes 0A. Further, the ratio of the current I _C flowing through the capacitor 14 to the output current I _OUT of the load M decreases, while the current I _D flowing through the first return diode 16 increases. As a result, the current flowing through the zero current switching circuit is in the state shown in FIG.

  As described above, according to the zero current switching circuit of the present embodiment, the switching loss at the time of turning on and turning off the switching element can be reduced with a simple circuit configuration without using the LC resonance characteristics.

Next, a second embodiment of the first invention will be described.
FIG. 6 is a circuit diagram of a zero current switching circuit according to the second embodiment. This zero current switching circuit is obtained by replacing the first freewheeling diode 16 with the second switching element 16a in the circuit shown in FIG. The rest of the configuration is the same as that shown in FIG.

  Since the second switching element 16a also performs substantially the same function as the first freewheeling diode 16 in the OFF state, the operation of the zero current switching circuit of the second embodiment is substantially the same as that according to the first embodiment. The same.

Next, a third embodiment of the first invention will be described.
FIG. 7 is a circuit diagram of a zero current switching circuit according to the third embodiment. In this zero current switching circuit, in the circuit shown in FIG. 1, the capacitor 14 is connected in parallel with the switching element 10 and the inductor 12, not in parallel with the load M. Further, a resistor 20 and a diode 22 connected in parallel with each other are connected between the capacitor 14 and the low potential source GND. Other than this, the configuration is the same as that shown in FIG.

In the third embodiment, when the switching element 10 is turned on, a discharge current flows through the capacitor 14 via the inductor 12 and the switching element 10. Further, when the switching element 10 is turned off, a charging current flows from the high potential source _VIN to the low potential source GND side through the load M and the capacitor 14 in the capacitor 14.

Next, a fourth embodiment of the first invention will be described.
FIG. 8 is a circuit diagram of a zero current switching circuit according to the fourth embodiment. This zero current switching circuit is obtained by replacing the first freewheeling diode 16 with the second switching element 16a in the zero current switching circuit according to the third embodiment shown in FIG. The rest of the configuration is the same as that shown in FIG.

  Since the second switching element 16a also performs substantially the same function as the first freewheeling diode 16 in the OFF state, the operation of the zero current switching circuit of the fourth embodiment is substantially the same as that according to the third embodiment. The same.

Next, a fifth embodiment of the first invention will be described.
FIG. 9 is a circuit diagram of a zero current switching circuit according to the fifth embodiment. The zero current switching circuit of this embodiment is such that the switching element 10 or the like is a high-side drive circuit for a load M, compared to the one according to the first embodiment shown in FIG. That is, in this embodiment, contrary to the first embodiment, the first potential source is the low potential source GND, and the second potential source is the high potential source _VIN .

  The zero current switching circuit of this embodiment performs the same operation as that of the first embodiment except that the direction of current is opposite to that of the first embodiment. Therefore, also in the zero current switching circuit of this embodiment, it is possible to reduce the switching loss when the switching element is turned on and off with a simple circuit configuration without using the LC resonance characteristics.

Next, a sixth embodiment of the first invention will be described.
FIG. 10 is a circuit diagram of a zero current switching circuit according to the sixth embodiment. This zero current switching circuit is obtained by replacing the first freewheeling diode 16 with the second switching element 16a in the circuit shown in FIG. The rest of the configuration is the same as that shown in FIG.

  Since the second switching element 16a also performs substantially the same function as the first freewheeling diode 16 in the OFF state, the operation of the zero current switching circuit of the sixth embodiment is substantially the same as that according to the fifth embodiment. The same.

Next, a seventh embodiment of the first invention will be described.
FIG. 11 is a circuit diagram of a zero current switching circuit according to the seventh embodiment. In the zero current switching circuit, in the circuit shown in FIG. 9, the capacitor 14 is connected in parallel with the switching element 10 and the inductor 12 instead of in parallel with the load M. Further, a resistor 20 and a diode 22 connected in parallel with each other are connected between the capacitor 14 and the high potential source V _IN . Other than this, the configuration is the same as that shown in FIG.

In the seventh embodiment, when the switching element 10 is turned on, a discharge current flows through the capacitor 14 via the inductor 12 and the switching element 10. Further, when the switching element 10 is turned off, a charging current flows from the high potential source _VIN to the low potential source GND side through the capacitor 14 and the load M.

Next, an eighth embodiment of the first invention will be described.
FIG. 12 is a circuit diagram of a zero current switching circuit according to the eighth embodiment. This zero current switching circuit is obtained by replacing the first freewheeling diode 16 with the second switching element 16a in the zero current switching circuit according to the seventh embodiment shown in FIG. The rest of the configuration is the same as that shown in FIG.

  Since the second switching element 16a also performs substantially the same function as the first freewheeling diode 16 in the OFF state, the operation of the zero current switching circuit of the fourth embodiment is substantially the same as that according to the seventh embodiment. The same.

Next, a ninth embodiment of the first invention will be described. This embodiment is characterized in that the power loss at the time of switching is greatly improved by regenerating energy stored in the inductor and the capacitor to the load particularly when performing zero current switching.
FIG. 13 is a circuit diagram of a zero current switching circuit according to the ninth embodiment. The zero current switching circuit shown in FIG. 13 is a load drive control circuit for controlling the power supplied to the load M. Between the high potential source V _IN (the terminal on the high potential side of the DC power supply) as the first potential source and the low potential source (GND) (the terminal on the low potential side of the DC power supply) as the second potential source, A load M, an inductor 12 and a switching element 10 are sequentially connected in series.

More specifically, one terminal ma of the load M is connected to the high potential source V _IN , and one terminal 10a of the switching element 10 is connected to the low potential source (GND). An inductor 12 is connected between the other terminal mb of the load M and the other terminal 10b of the switching element 10.

  Power supply to the load M is controlled by ON / OFF of the switching element 10. A PWM control signal having an arbitrary duty is input from a PWM circuit (not shown) to the gate electrode of the MOSFET of the switching element 10.

  Note that when the switching element 10 is turned on, the current flowing through the switching element does not increase immediately even if the voltage between both terminals of the switching element starts to decrease due to the action of the inductor 12, and the voltage becomes 0V. Starting to rise. Thereby, the switching element 10 can be turned on from the OFF state to the ON state without switching loss.

The capacitor 14 is connected in parallel with the load M, that is, in series with the switching element 10 and the inductor 12. Further, a diode 22 as a first rectifying element is connected between the high potential source V _IN and the capacitor 14. The diode 22 selectively allows a current to flow from the capacitor 14 to the high potential source _VIN side.

  Between the first node n1 between the diode 22 and the capacitor 14 and the second node n2 between the switching element 10 and the inductor 12, diodes 24 and 26 as second and third rectifier elements are provided. They are connected in series with each other. The diodes 24 and 26 selectively pass current from the second node n2 side to the first node n1 side.

Further, a second capacitor 28 is connected between the third node n3 between the diode 24 and the diode 26 and the low potential source GND.
It is desirable that the capacitor 28 and the capacitor 14 have substantially the same capacitance value.

  When the switching element 10 is turned off, the voltage across the switching element does not increase immediately even if the current flowing through the switching element starts to decrease due to the function of the capacitor 28, and starts increasing after the current reaches 0A. To do. Thereby, the switching element 10 can be turned off from the ON state to the OFF state without switching loss.

  Furthermore, in this embodiment, when the switching element 10 is turned on, a part of the current flowing through the inductor 12 can be led to charge the capacitor 14. For this reason, at the time of turn-on, an increase in current flowing through the switching element 10 connected in series to the inductor 12 can be suppressed. In addition, by returning the energy stored in the capacitor 14 to the load M, it is possible to reduce power loss and configure a more efficient zero current switching circuit.

  A first free-wheeling diode 16 as a first free-wheeling circuit is connected to the load M in parallel. The first freewheeling diode 16 prevents the generation of a surge voltage by circulating a regenerative current due to the induced electromotive force of the load M when the switching element 10 is turned off.

  As a second reflux circuit, a resistor 30 and a capacitor 31 connected in series with each other are connected in parallel to the inductor 12. The resistor 30 and the capacitor 31 function as a damper circuit that prevents the high-frequency alternating current from flowing due to resonance of the voltage and current of the inductor 12, the capacitor 28, and the capacitor 14 when the switching element 10 is turned on and off.

Hereinafter, with reference to the timing chart of FIG. 14 and FIGS. 15A to 15E, an operation example in the present embodiment illustrated in FIG.
15A shows the operation of the zero current switching circuit at time t0 to t1 in the timing chart of FIG. 14, FIG. 15B shows the operation at time t1 to t2, FIG. 15C shows the operation at time t2 to t3, FIG. 15D represents the operation from time t3 to t4, and FIG. 15E represents the operation from time t4 to t5.

First, the operation of the zero current switching circuit at times t0 to t1 in the timing chart of FIG. 14 will be described. Prior to time t0, the gate voltage V _G of the switching element 10 starts to rise from “L” to “H”. When the gate voltage V _G reaches the turn-on threshold value of the MOSFET of the switching element 10 at time t0, the drain-source voltage V _{Q of the} MOSFET starts to decrease. However, due to the characteristics of the inductor 12, the current I _L flowing through the inductor 12 remains at 0A at times t0 to t1. Therefore, current I _Q flowing through the switching element 10 connected in series to the inductor 12 which is still 0A. Therefore, the switching loss in the switching element 10 at time t0 to t1 is zero.

Then, at time t0 before, since the current I _OUT by the induced electromotive force of the load M is refluxed a first reflux diode 16 in the forward direction, MOSFET drain of the switching element 10 - source voltage V _Q is It is approximately equal to the potential of the high potential source V _IN . Therefore, the diode 24 is forward-biased and charged so that the voltage V _C (28) of the capacitor 28 is substantially equal to the voltage of the high potential source V _IN . At the same time, the potential difference between both ends of the capacitor 14 and the diode 22 connected in series is equal to the forward voltage drop amount of the first freewheeling diode 16, and the potential of the node n3 on the anode side of the diode 26 is the drain potential of the switching element 10. Since it is approximately equal to V _Q , the voltage V _C (14) across the capacitor 14 is approximately 0V.

FIG. 15A shows the current flowing through the zero current switching circuit at times t0 to t1. The current I _OUT due to the induced electromotive force of the load M when the switching element 10 is turned off before the time t 0 circulates through the first freewheeling diode 16. Therefore, at time t0 to t1, no current flows through the switching element 10 or the inductor 12 as described above.

  From time t0 to t1, the switching element 10 can be turned on from the OFF state to the ON state with substantially no switching loss.

Next, the operation of the zero current switching circuit at times t1 to t2 in the timing chart of FIG. 14 will be described. By time t1, the drain-source voltage V _Q of the MOSFET of the switching element 10 is reduced to 0V. Then, over a period from time t1 to time t2, the current flowing through the inductor 12 I _L, and the current I _Q flowing through the switching element 10 gradually increases. After time t1, the current circulating through the first return diode 16 gradually decreases to 0A. Then, the current to discharge the capacitor 28 I _C (28), by which are connected in series through the diode 26, the current I _C (14 to charge the capacitor 14 with a current value equal the current I _C (28) ) Also flows. Therefore, the current I _Q and the current I _L are added to the current I _C (28) for discharging the capacitor 28 and the current I _C (14) for charging the capacitor 14 with the same value as the current I _C (28). Rises to the current value. However, the drain-source voltage V _Q of the switching element 10 at time t1 to t2 has already been 0V. Therefore, the switching loss in the switching element 10 at time t1 to t2 is also zero.

  Here, the relationship among the inductor 12, the capacitor 28, and the capacitor 14 will be described. Capacitor 28 is connected to one side of capacitor 14 at node n3 with diode 26 in the forward direction. The other side of the capacitor 14 is connected to the drain of the MOSFET of the switching element 10 via the inductor 12.

As described above, the initial charge states of the capacitor 28 and the capacitor 14 at the timing immediately before the switching element 10 is turned on are V _IN (= capacitor 28) and 0 (= capacitor 14), respectively.
Here, when the switching element 10 is turned on, the energy (1/2 × C × V ² ) charged in the capacitor 28 becomes a current I _L and moves to the inductor 12.
Furthermore, the current energy of the inductor 12 charges the capacitor 14 and is stored in the capacitor 14 as voltage energy.

As will be described later, in order to realize zero current switching when the switching element 10 is turned off, it is necessary that the capacitor 28 is discharged and the potential of the node n3 is low immediately before the switching element 10 is turned off.
Therefore, in this embodiment, when the switching element 10 is turned on, the capacitor 14 is charged after the electric charge of the capacitor 28 is replaced with the current of the inductor 12. If this is expressed by movement of energy, it is moved in the order of capacitor 2 → inductor 12 → capacitor 14.

On the other hand, in the first embodiment of the first invention shown in FIG. 1, the current charged in the capacitor 14 flows through the inductor 12 when the switching element 10 is turned on. However, since the current once stored in the inductor 12 circulates through the diode 18, it is released as thermal energy by the internal resistance of the inductor 12 and the power loss obtained by multiplying the forward voltage drop of the diode 18 and the circulated current. It will be.
Compared with this, according to the present embodiment, since there is no free wheel diode in parallel with the inductor 12, the current energy of the inductor is transferred to the capacitor 14 without loss.

FIG. 15B shows the current flowing through the zero current switching circuit at times t1 to t3. The current I _OUT flowing through the load M, the discharge current I _C (28) of the capacitor 28, and the charging current I _C (14) of the capacitor 14 are combined to form a current I _L flowing through the inductor 12. Further, the current I _L, the current I _Q flowing through the switching element 10 connected in series directly to the inductor 12.

Next, the operation of the zero current switching circuit at time t12 in the timing chart of FIG. 14 will be described. Since the current of the inductor 12 increases from the time t1 and the current that discharges the capacitor 28 charges the capacitor 14, the voltage V _C (14) of the capacitor 14 increases. At this time, the voltage of V _OUT connected to the capacitor 14 decreases while passing 0 V at time t12. Incidentally, the average value of the current I _OUT flowing through the load M is generally proportional to the multiplication of the voltage between the high potential source V _IN and the low potential source GND and the PWM drive duty of the switching element 10.

I _OUT average value = (V _IN −GND) × DUTY / R
R is the DC resistance of load M

However, since the potential of V _OUT drops to a potential lower than the potential of the low potential source GND after time t12, the instantaneous value of the current I _OUT of the load M reaches from time t12 to time t4 as shown in FIG. It becomes larger than the above I _OUT average value.

As described above, the voltage (energy) stored in the capacitor 28 before the switching element 10 is turned on is transferred to the capacitor 14 and applied to the load M in the process after the switching element 10 is turned on. V _OUT is changed in the direction in which the voltage increases. As a result, the current I _{OUT of the} load M increases so that the energy of the capacitor 28 acts as the power consumption of the load M.

Next, the operation of the zero current switching circuit at times t2 to t3 in the timing chart of FIG. 14 will be described. From time t2 to t3, the switching element 10 is in the ON state. The voltage V _C (28) of the capacitor 28 becomes 0 V at time t2. On the other hand, the current I _{L of the} inductor 12 works to maintain the current value when the capacitor 28 is discharged, and continues to flow after time t2. Then, current I _Q of the switching element 10 connected to the current I _L and series continue to decrease while charging the capacitor 14, a steady state at time t3. Therefore, the capacitor 14 is charged between the time t2 and the time t3, and the voltage V _C (14) of the capacitor 14 continues to increase to twice the voltage of the high potential source V _IN . As a result, the potential V _OUT of the other terminal mb of the load M continues to decrease, and at time t3, it decreases to the voltage of the high potential source V _IN × (−1). At this time, the current I _{OUT of the} load M becomes the maximum value, and the energy stored in the capacitor 28 is regenerated to the load M.

FIG. 15B shows the current flowing through the zero current switching circuit at times t1 to t3. The current I _OUT flowing through the load M, the discharge current I _C (28) of the capacitor 28, and the charging current I _C (14) of the capacitor 14 are combined to form a current I _L flowing through the inductor 12. Further, the current I _L, the current I _Q flowing through the switching element 10 connected in series directly to the inductor 12.

Next, the operation of the zero current switching circuit at times t3 to t4 in the timing chart of FIG. 14 will be described. Current I _Q of the current I _L and the switching element 10 of the inductor 12 at the time t3~t4 is steady state. The voltage V _C (28) of the capacitor 28 has already been discharged and is 0 V, and after the voltage V _C (14) of the capacitor 14 has been charged to a value twice that of the high potential source V _IN at time t3, During t4, the current I _C (14) of the capacitor 14 is discharged while flowing through the load M. At a time t4, a steady state is reached at a voltage substantially equal to the voltage of the high potential source V _IN .

FIG. 15C shows the current flowing through the zero current switching circuit at times t3 to t4. The current I _OUT flowing through the load M is a combination of the current I _Q flowing through the switching element 10 (= current I _L flowing through the inductor 12) and the current I _C (14) flowing through the capacitor 14.

In the zero current switching circuit at times t4 to t5 in the timing chart of FIG. 14, since the switching element 10 is in the ON state and the voltages of the capacitors 28 and 14 are in the saturated state, the capacitor current is 0A. The current I _{Q of} 10 and the current I _{L of the} inductor 12 are equal to the current I _{OUT of the} load M and are in a steady state.

FIG. 15D shows the current flowing through the zero current switching circuit at times t4 to t5. Current I _OUT through the load M is equal to the current I _L flowing through the current I _Q, and the inductor 12 flows through the switching element 10.

Next, the operation of the zero current switching circuit at times t5 to t7 in the timing chart of FIG. 14 will be described.
Prior to time t5, the gate voltage V _G of the switching element 10 starts to decrease from “H” to “L”. Then, at time t5, the gate voltage V _G reaches the MOSFET turn-off threshold of the switching element 10, the current I _Q flowing through the switching element 10 starts to descend. However, since the voltage V _C (28) of the capacitor 28 is 0 V, the drain-source voltage V _Q of the MOSFET of the switching element 10 remains substantially 0 V. Therefore, during time t5 switching element 10 is turned off immediately after the current I _Q of the switching element 10 is to reach 0A, drain - because source voltage V _Q is 0V, the switching loss in the switching element 10 0 It is.

Subsequently, when the capacitor 28 is charged by the current I _{OUT of the} load M via the inductor 12 and the voltage V _C (28) of the capacitor 28 becomes higher than the high potential source V _{IN at} time t6, the diode 24 → the diode 26 → diode 22 → while forward current I _L in the load M → inductor 12 flows, current I _L becomes 0A at gradually reduced to continue the time t7. At the same time, the voltage of the capacitor 14 V _C (14) but is substantially equal to the high potential source V _IN at time t5 before, as the current I _L of the inductor 12 decreases, the current I _OUT of the load M is a capacitor 14 Since the flow starts to charge, the voltage V _C (14) of the capacitor 14 gradually decreases and becomes 0 V at time t7.

The potential V _OUT of the other terminal mb of the load M rises as the voltage V _C (14) of the capacitor 14 decreases, and becomes substantially equal to the potential of the high potential source V _IN at time t7. It acts so that the reflux current of the load M flows through 16.

FIG. 15E shows the current flowing through the zero current switching circuit at times t5 to t7. When the current I _Q flowing through the switching element 10 is reduced, the current I _L of the inductor 12 to charge the capacitor 28 flows through the capacitor 28. When the voltage V _C (28) of the capacitor 28 reaches a voltage that forward biases the diode 26 and the diode 22, the current I _L passes through the diode 26, the diode 22, and the load M and returns to the inductor 12 again. Decrease. At the same time, a current I _C (14) for charging the capacitor 14 flows while the voltage V _{OUT at} the terminal mb of the load M rises.

Then, when the voltage V _OUT rises to become equal to the voltage of the high potential source V _IN, the inductor I _L current 0A next, the voltage V _C (28) of the capacitor 28 is substantially equal to the voltage of the high potential source V _IN , the voltage V _C (14) to 0V capacitor 14, the current I _Q of the switching element 10 at 0A, the current of the load M is circulated through the first freewheeling diode 16, the state shown in FIG. 15A .

  In this manner, the switching element 10 can be turned off from the ON state to the OFF state without switching loss from time t5 to time t7.

  As described above, according to the zero current switching circuit of the present embodiment, the inductor that does not use the LC resonance characteristic, has a simple circuit configuration, and suppresses the increase in current and voltage when the switching element is turned on and off. Since the energy stored in the capacitor can be regenerated to the load side, the switching loss of the switching element can be reduced, and at the same time, the energy stored in each element can be consumed by the load without loss. A high-efficiency zero current switching circuit device can be provided.

Further, as shown in FIGS. 2 and 14, the zero current switching circuit of the present invention arbitrarily sets the time for the voltage signal _{VOUT at} the output terminal to rise from 0V and the time for the voltage signal _VOUT to fall from the power supply voltage depending on the values of the inductor and the capacitor. Since it is possible, it has an excellent feature that it can easily cope with the increase in the transition time of the rectangular wave signal in the known technique as one of the techniques for reducing the radiation noise emitted from the wiring to the load M.

Next, a tenth embodiment of the first invention will be described.
FIG. 16 is a circuit diagram of a zero current switching circuit according to the tenth embodiment. This zero current switching circuit is obtained by replacing the first freewheeling diode 16 with the second switching element 16a in the circuit shown in FIG. The rest of the configuration is the same as that shown in FIG.

  Since the second switching element 16a also performs substantially the same function as the first freewheeling diode 16 in the OFF state, the operation of the zero current switching circuit of the tenth embodiment is substantially the same as that according to the ninth embodiment. The same.

Next, an eleventh embodiment of the first invention will be described.
FIG. 17 is a circuit diagram of a zero current switching circuit according to the eleventh embodiment. The zero current switching circuit of this embodiment is such that the switching element 10 and the like is a high-side drive circuit of a load M, compared to the one according to the first embodiment shown in FIG. That is, in this embodiment, contrary to the first embodiment, the first potential source is the low potential source GND, and the second potential source is the high potential source _VIN .

  The zero current switching circuit of this embodiment performs the same operation as that of the first embodiment except that the direction of current is opposite to that of the first embodiment. Therefore, also in the zero current switching circuit of this embodiment, it is possible to reduce the switching loss when the switching element is turned on and off with a simple circuit configuration without using the LC resonance characteristics.

Next, a twelfth embodiment of the first invention is described.
FIG. 18 is a circuit diagram of a zero current switching circuit according to the twelfth embodiment. This zero current switching circuit is obtained by replacing the first freewheeling diode 16 with the second switching element 16a in the circuit shown in FIG. Other configurations are the same as those shown in FIG.

  Since the second switching element 16a also performs substantially the same function as the first freewheeling diode 16 in the OFF state, the operation of the zero current switching circuit of the twelfth embodiment is substantially the same as that according to the eleventh embodiment. The same.

Next, a thirteenth embodiment of the first invention is described.
FIG. 19 is a circuit diagram of a zero current switching circuit according to the thirteenth embodiment. This zero current switching circuit is obtained by changing the connection destination of the capacitor 28 from the low potential source GND to the high potential source V _IN in the circuit shown in FIG. The rest of the configuration is the same as that shown in FIG.

Since the function of the capacitor is to integrate the current flowing through the capacitor and replace it with a change in voltage, the same regardless of whether the steady-state charging voltage of the capacitor 28 at a predetermined timing is 0 V or V _IN. In order to operate, the operation of the zero current switching circuit of the thirteenth embodiment is substantially the same as that of the ninth embodiment.

Next, a fourteenth embodiment of the first invention is described.
FIG. 20 is a circuit diagram of a zero current switching circuit according to the fourteenth embodiment. This zero current switching circuit is obtained by changing the connection destination of the capacitor 28 from the low potential source GND to the high potential source V _IN in the circuit shown in FIG. The rest of the configuration is the same as that shown in FIG.

Since the function of the capacitor is to integrate the current flowing through the capacitor and replace it with a change in voltage, the same regardless of whether the steady-state charging voltage of the capacitor 28 at a predetermined timing is 0 V or V _IN. In order to operate, the operation of the zero current switching circuit of the thirteenth embodiment is substantially the same as that of the tenth embodiment.

Next, a fifteenth embodiment of the first invention is described.
FIG. 21 is a circuit diagram of a zero current switching circuit according to the fifteenth embodiment. This zero current switching circuit is obtained by changing the connection destination of the capacitor 28 from the high potential source _VIN to the low potential source GND in the circuit shown in FIG. The rest of the configuration is the same as that shown in FIG.

Since the function of the capacitor is to integrate the current flowing through the capacitor and replace it with a change in voltage, the same regardless of whether the steady-state charging voltage of the capacitor 28 at a predetermined timing is 0 V or V _IN. In order to work, the operation of the zero current switching circuit of the thirteenth embodiment is substantially the same as that of the eleventh embodiment.

Next, a sixteenth embodiment of the first invention will be explained.
FIG. 22 is a circuit diagram of a zero current switching circuit according to the sixteenth embodiment. This zero current switching circuit is obtained by changing the connection destination of the capacitor 28 from the high potential source _VIN to the low potential source GND in the circuit shown in FIG. The rest of the configuration is the same as that shown in FIG.

Since the function of the capacitor is to integrate the current flowing through the capacitor and replace it with a change in voltage, the same regardless of whether the steady-state charging voltage of the capacitor 28 at a predetermined timing is 0 V or V _IN. In order to operate, the operation of the zero current switching circuit of the thirteenth embodiment is substantially the same as that of the twelfth embodiment.

  Next, an embodiment of the second invention will be described. FIG. 23 shows a circuit diagram of a full bridge circuit according to an embodiment of the second invention. In the full bridge circuit shown in FIG. 23, the first low-side switching circuit 34 and the second low-side switching circuit 38 are configured by the zero current switching circuit shown in FIG. 13, and the first high-side switching circuit 32 and the second high-side switching circuit are configured. The circuit 36 includes the zero current switching circuit shown in FIG.

PWM control signals V _G1 and V _G4 synchronized from a PWM circuit (not shown) are input to the gates of the FETs of the switching elements 10 of the first high-side switching circuit 32 and the second low-side switching circuit 38, respectively. In addition, PWM control signals V _G2 and V _G3 inverted in synchronization are input from the PWM circuit to the FET gates of the switching elements 10 of the second high-side switching circuit 36 and the first low-side switching circuit 34, respectively.

  As described above, each of the switching circuits 32, 34, 36, and 38 constituting the full bridge circuit is constituted by a zero current switching circuit including a switching circuit having no switching loss. For this reason, the switching loss in the whole full bridge is greatly reduced.

In each embodiment mentioned above, although the example which constituted the present invention on specific conditions was explained, the present invention can perform various change and combination, and is not limited to this. For example, in the above-described embodiments, the present invention has been described as a zero current switching circuit and a full bridge circuit as a load driving device for PWM control of power supplied to a load. It is not limited to the load driving device.
In addition, it should be easily assumed that the power loss due to the forward voltage drop of the diode can be reduced by replacing each rectifier element used in the zero switching circuit disclosed in the present invention with a MOSFET.

DESCRIPTION OF SYMBOLS 10 Switching element 12 Inductor 14 Capacitor 16 1st free-wheeling diode 16a 2nd switching element 18 2nd free-wheeling diode 20 Resistance 22, 24, 26 Diode 28 2nd capacitor 30 Resistance 31 Capacitor 32 1st high side switching circuit 34 1st low side switching Circuit 36 Second high side switching circuit 38 Second low side switching circuit

Claims (3)

A zero current switching circuit for controlling power supplied to a load,
A switching element connected between the other terminal of the load having one terminal connected to the first potential source and the second potential source;
An inductor connected between the other terminal of the switching element having one terminal connected to the second potential source and the other terminal of the load;
A capacitor connected in parallel with the load or the switching element and the inductor;
A first reflux circuit connected in parallel with the load;
A second reflux circuit connected in parallel with the inductor;
A zero current switching circuit comprising: A zero current switching circuit for controlling power supplied to a load,
A switching element connected between the other terminal of the load having one terminal connected to the first potential source and the second potential source;
An inductor connected between the other terminal of the switching element having one terminal connected to the second potential source and the other terminal of the load;
A capacitor connected in parallel with the load;
A reflux circuit connected in parallel with the load;
A first rectifier element connected between the first potential source and the capacitor and configured to selectively pass a current from the second potential source side to the first potential source side;
From the second node side to the first node side connected between a first node between the first rectifier element and the capacitor and a second node between the switching element and the inductor. A second and a third rectifying element connected in series with each other to selectively pass a current;
A second node connected between a third node between the second rectifying element and the third rectifying element and the first or second potential source;
Features and to Ruze Russia current switching circuit that example Bei a. A full bridge circuit comprising a first high side switching circuit, a first low side switching circuit, a second high side switching circuit, and a second low side switching circuit,
The first and second high-side switching circuits are respectively
A first switching element connected between the output terminal and the first potential source;
A first inductor connected between the other terminal of the first switching element having one terminal connected to the first potential source and the output terminal;
A first capacitor connected between the output terminal and a second potential source;
A first reflux circuit connected in parallel with the inductor;
With
The output terminal of the first high-side switching circuit is connected to one terminal of the load, and the output terminal of the second high-side switching circuit is connected to the other terminal of the load,
The first and second low-side switching circuits are respectively
A second switching element connected between the output terminal and the second potential source;
A second inductor connected between the other terminal of the second switching element having one terminal connected to the second potential source and the output terminal;
A second capacitor connected between the output terminal and a first potential source;
A second reflux circuit connected in parallel with the inductor;
With
The output terminal of the first low side switching circuit is connected to one terminal of the load, and the output terminal of the second low side switching circuit is connected to the other terminal of the load.
A full bridge circuit characterized by that.

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