JP5102185B2 - Signal conditioning device for interfacing an intravascular sensor with variable operating characteristics to a physiological monitor - Google Patents

Description

  The present invention relates generally to the field of medical diagnostic devices, and more particularly to a diagnostic device that identifies suspicious blocks in a coronary artery by a sensor attached to the end of a flexible elongated member such as a guide wire.

  Over the past decade, innovations in cardiovascular disease diagnosis have shifted from external imaging to internal catheterization-based diagnostic methods. Diagnosis of cardiovascular disease is performed by vascular mapping in which a radiopaque dye is injected into the blood vessel and a live x-ray image of the portion of the cardiovascular system of interest is taken. Magnetic resonance imaging (MRI) has also been used. More recently, however, vasculature blocks and other vasculatures have been developed by microsensors placed at the ends of flexible elongate members such as catheters or guide wires used for catheterization procedures. Diagnosis devices and methods for diagnosing tube structure diseases have been developed.

  One such microsensor device is a pressure sensor attached to the end of a guide wire. An example of such a pressure sensor is described in US Pat. No. 6,106,476 by Corr et al., The entire teaching of which is hereby incorporated by reference. Such intravascular pressure sensors are used in the vasculature to facilitate locating and determining the severity of blood flow constrictions or other disruptors in the blood vessels of the human body. Measure blood pressure at various points. Such devices are commonly used to determine the effectiveness of vascular treatment by placing a pressure sensor near the stenosis and measuring a pressure differential indicative of a partial block of the blood vessel.

  As can be imagined, the intravascular pressure sensors described above are used in operating room environments and include many types of sensors and devices for diagnosing and treating cardiovascular diseases. Obviously, the room for error is very limited. Therefore, it is substantially important to simplify every aspect of the operating room so as to reduce the incidence of errors.

  Despite the importance of keeping the device simple, it is between a compression sensor attached to an intravascular guidewire and a physiological monitor that displays a human readable output corresponding to the sensed pressure. It is necessary to provide an interface device. This interface device receives synchronization information from the monitor in the form of an excitation signal and provides a regulated standardized output in the form of an analog voltage signal. The interface device drives a pressure sensor attached to the guide wire with the sensor current, adjusts the sensed analog sensor input signal, and performs a mathematical transformation to provide a standardized output to the physiological monitor ( By microcontroller). In this way, the interface device provides a means of connecting many types of sensor devices to the physiological monitor so that the inputs to the physiological monitor are standardized and independent of the sensing device's signal requirements and operating characteristics.

  In a known conventional intravascular pressure sensor and physiological monitor interface device sold by JOMED (Rancho Cordova, CA) shown in FIG. 1a, an amplifier module 10 (eg, model 7000 patient cable). And a WAVEMAP ™ processor box 12 is inserted between the physiological monitor 14 and the WAVEWIRE ™ pressure sensing guide wire 16. The guide wire 16 is a disposable device connected to the amplifier module 10 via the connector 15. The amplifier module 10 receives power and excitation signals through two separate and different conductive lines in a cable 17 that is connected to another output conductor of the WAVERMAP ™ processor box 12. This WAVEMAP (TM) processor box receives power from a standard wall outlet 18 via a standard three (grounded) power card 20 plugged into it. Although not shown in the drawings, the physiological monitor is also powered via a standard AC wall outlet.

  The WAVEMAP ™ processor box 12 includes a separate and distinct signal interface connected to a physiological monitor 14. The WAVEMAP ™ processor box receives a differential voltage excitation signal (either AC or DC) from the physiological monitor 14 via the cable 22. The power of the excitation signal transmitted via the cable 22 is significantly lower than the AC power that can be transmitted from the wall outlet 18 to the WAVEMAP ™ processor box 12 via the power card 20. Cable 22 also transmits a signal (5 microvolts / mmHG) representing the sensed pressure from WAVERMAP ™ processor box 12 to physiological monitor 14. Yet another cable 24 transmits the aortic pressure (Pa) sensed by another device from the physiological monitor 14 to the WAVEMAP ™ processor box 12. Physically setting up an intravascular pressure reading device is complex due to the many devices and separate power supplies required for the known device of the prior art, while at the same time the numerous codes required by this known device. And cumbersome for connection. Also, once set up, many codes are scattered around the patient.

  The temperature compensation / signal conditioning scheme currently used as a signal conditioning interface (eg, the WAVEMAP ™ processor 12 described above) to compensate for temperature and pressure effects on an intravascular pressure sensor attached to a guidewire. Rely on digital processors. The compensation equation includes a set of six coefficients for temperature compensation, pressure sensitivity, and a polynomial that includes the effect of temperature on pressure sensitivity for each of the two resistive elements in the characterized sensor device. The compensation value is calculated for each pressure reading (a constant term is pre-calculated to reduce the processing load as much as possible). By calculating a polynomial result for each pressure reading, a significant processing load is imposed on the signal conditioning interface processor.

Summary of the Invention

  The present invention provides a signal conditioning device having a simplified connection scheme that interfaces with an intravascular diagnostic device, such as a pressure sensor attached to the distal end of a guide wire, with low power requirements, and an excitation signal to the intravascular diagnostic device. A physiologic monitor to supply.

  The present invention includes a signal conditioning device connected by a cable to an intravascular measurement device and a physiological monitor. The signal conditioning device includes a number of analog and digital circuits that cooperate to perform amplification, filtering and / or compensation on the signal sent between the pressure sensor and the physiological monitor.

  The signal conditioning device includes a sensor interface circuit that provides a sensor drive signal for exciting a sensor supported by the attachable intravascular measurement device and provides a measurement signal. The signal conditioning device also includes a physiological monitor interface. The physiological monitor interface transmits an input for receiving a sensor excitation signal from the physiological monitor and an output signal corresponding to the sensed measurement provided by the attached sensor resulting from the sensor drive signal. Output.

  The signal conditioning device also includes a power supply circuit disposed between the physiological monitoring interface of the signal conditioning device and the signal conditioning circuit. The power supply circuit receives a part of the power supplied by the sensor excitation signal and supplies the power obtained from the part of the power supplied by the sensor excitation signal to at least a part of the signal processing circuit in the signal conditioning device. Includes a signal converter.

  According to another feature of the new signal conditioner, the temperature compensated current source in the signal conditioner adjusts the current supplied to at least one of the pair of resistive sensor elements of the attached sensor to provide a resistance It compensates for the difference between temperature changes on the resistive sensor element pair, thereby facilitating the elimination of temperature effects on the resistive sensor element.

Detailed description

The features of the invention are set forth in the appended claims, but the invention and its objects and advantages can best be understood from the following detailed description and the accompanying drawings.
In general, an exemplary signal conditioning device using the present invention described below with respect to FIG. 1b is designed to interface a pressure sensor attached to a guide wire to a standard physiological (eg, blood pressure) monitor. Has been. The signal conditioner processes the signal received from the pressure sensor attached to the guidewire and provides a normalized signal to any of a number of different physiological monitors with potentially different signal requirements.

  From an overall system mechanics perspective, the exemplary signal conditioning device reduces the number of power sources and the number of different cables and physically different devices required to perform intravascular blood pressure measurements. These desirable attributes are achieved by having the conditioning device receive and / or use a differential sensor excitation signal transmitted in a new way by a known physiological monitor.

  Known signal conditioning devices use the excitation signal as a reference voltage that generates an output signal scaled according to the sensed pressure. However, in the exemplary signal conditioning device, a power circuit that rectifies and converts from AC to DC draws current from the received excitation signal. The drawn current provides power to processors, small scale integrated circuits and discrete circuit elements that perform signal generation / amplification / conditioning functions within the signal conditioning device. Such functions include driving output current to a polysilicon pressure sensor mounted on the guide wire. An example of such a polysilicon pressure sensor is described in US Pat. No. 6,106,476 by Corr et al., Which is hereby incorporated by reference in its entirety. As an example, the signal conditioning device provides an output signal to a physiological monitor having a sensitivity of about 5 microvolts per volt (input) / mmHg. The signal conditioner also drives an LCD display that shows the high and low pressures sensed during the 2 second interval.

  As in conventional signal conditioning interface circuits, a portion of the input excitation signal from the physiological monitor drives the physiological monitor with a differential voltage output signal transmitted by the signal conditioning device representing the sensed pressure (ie, Give a voltage reference to the differential voltage output signal). The differential voltage output signal is generated, for example, by a pair of digital to analog converters. The generated differential voltage output signal generally includes a base (ie, reference) differential voltage signal corresponding to the input differential signal from the physiological monitor. The reference differential voltage is multiplied by a scalar value to represent the sensed and regulated (eg, filtered) pressure value provided by the signal conditioning processor. Thus, the disclosed embodiments of the present invention perform signal generation, conditioning and amplification without relying on another signal source to provide DC power to the circuit of the signal conditioning device.

  Referring to FIG. 1b, a signal conditioner 50 using the present invention connects to a physiological monitor 52 via a five-line connector cable 54. The 5-line connector cable 54 includes a pair of excitation signal lines driven by the physiological monitor 52. The excitation signal line is driven as a differential voltage pair with 2.4 to 11 Vdc, 2.4 to 8 Vrms sine wave (1 kHz to 5 kHz), or 2.4 to 8 Vrms square wave (DC to 5 kHz) as an example. The sinusoidal input is further limited in its range due to the droop between the peak voltages at low frequencies. The rectified square wave has very little gap, so droop is not a problem.

  In one embodiment of the invention, the electronic elements of the signal conditioning device 50 are powered by a current drawn from the excitation signal supplied on the excitation signal line of the cable 54. Although not shown in the exemplary embodiment of the present invention, in another embodiment, the signal conditioning device is when power from an external source is insufficient or unavailable to the signal conditioning device 50. Including battery as auxiliary / backup power source. In a preferred embodiment, the signal conditioning device 50 is capable of operating at less than about 20 mArms due to its design, and such power requirements are related to the Association for Advancement of Medical Instruments for Sensor Excitation Power (“AAMI”). There is no battery as it is satisfied by a physiological monitor that satisfies the standard. Examples of physiological monitors 50 that satisfy the power requirements described above include pressure sensors that satisfy the American National Standards Institute (“ANSI”) / AAMI BP22-1994; Nihon Kohden America, Inc. (Foothill Ranch, CA) Model RM-6000, RMC-2000, RMC-3100, Lifescope-S, RMC-1100, models EP-1102 and EP-1600 manufactured by NEC (Tokyo, Japan), and Fukuda Denshi (Tokyo, Japan) ) Total hemodynamic instrument with model MCS-5500, MCS-7000, DS-3300 manufactured by the Company) may be included.

  The 5-line connector cable 54 includes a pair of differential output signal lines. This output signal line is driven by an output digital-to-analog converter (which will be described in more detail below) of the signal conditioner 50. As an example, the differential output signal operates at 5 microV per volt / mmHg. Therefore, an operating range of -150 microV / V to 1650 microV / V represents a sensed pressure range of -30 to 330 mmHg. An exemplary resolution (minimum step) for the differential output signal is 0.2 mmHg.

  The fifth line of the 5-line connector cable 54 transmits a ground signal. Thus, all signal / power requirements for the signal conditioner 50 are satisfied by the standard 5-line output of the physiological monitor 52. Therefore, the need for any interface device (such as processor box 12 of FIG. 1a) is eliminated and the complexity of the pressure sensing system mechanism is reduced.

  On the patient side, the signal conditioning device 50 is connected to a replaceable guide wire 56 via a connector 58 and a corresponding static cable 59. A connector 58 connects a set of 10 lines in a static cable 59 that transmits signals between the replaceable guide wire 56 and the signal conditioner 50. The five lines of the first set of connectors 58 are used to generate and receive signals associated with the pressure sensor. The five lines of the second set of connectors 58 are electrically erasable programs for the characterization of a guide wire sensor mounted on a static cable 59 that stores a set of values relating to the characteristics of the mounted sensor. It relates to an interface to possible read-only memory (“EEPROM”).

  With respect to the five lines of the second set of connectors 58, four of the five lines of the ten-line connector 58 (the fifth line is not used) were attached to a guide wire, which is a pressure sensor as an example. It facilitates reading of the characteristic description data from the EEPROM transmitted over the static cable for the sensor device 60. The EEPROM contains temperature compensation, gain and offset values used by the signal conditioning device 50 to process the sensed signal from the sensor device 60. The power and ground lines are connected to the EEPROM via connector 58 by a signal conditioner. The clock and data lines for reading the EEPROM data constitute the last two lines.

  The first set of five lines associated with connector 58 includes, by way of example, a voltage reference line connected to each of two pressure sensitive polysilicon resistive sensor elements on pressure sensor 60 attached to a guide wire. Is included. The remaining four lines contain two sets of excitation / sense signal pairs. In one embodiment of the invention, the first current flows over the first shorted excitation / sense line pair. A second current, separately adjustable with respect to the first current, flows over the second shorted excitation / sense line pair of connector 58. In the configuration of FIG. 1b, first and second currents pass through first and second resistive sensor elements of a pressure sensor 60 attached to the distal end of a replaceable guide wire 56. A pressure sensing circuit including a resistive sensor element is completed by connecting the remaining two terminals of the resistive sensor element to a voltage reference line.

  In operation, the electrical sensing circuit functions as follows. The polysilicon sensor element on the pressure sensor 60 is pressure sensitive. In certain embodiments having a pair of resistive elements, one element increases resistance and another element decreases resistance in response to a change in pressure. For example, in one embodiment of the present invention, each resistive element has a pressure sensitivity of 15 to 35 microohms per ohm / mmHg (at 25 degrees Celsius and 100 mmHg). By supplying a steady current through the resistive element, the pressure change results in a resistance change that changes the voltage between the resistive sensor elements.

  A common voltage reference obtained by measuring the voltage between both terminals of the first and second resistive elements is supplied by the signal conditioning device 50 to the first terminal of each resistive sensor element of the sensor 60. Set by connecting to a common reference voltage. The differential amplifier in the signal conditioner 50 senses a voltage difference corresponding to the voltage at the second terminal of each resistive sensor element through the excitation / sensing line and sets a voltage differential signal. An analog-to-digital converter (“ADC”) in the signal conditioning device 50 converts the amplified analog voltage differential signal to a digital value. This digital value is received by the processor and filtered in a known manner to obtain a filtered digital pressure value based on previous calibration of the sensor 60 (eg, finite impulse response filter or “FIR” filtering). ) The filtered digital pressure value is then used to drive the digital input to a pair of output digital-to-analog converters (“DACs”). The output DAC pair provides a differential output signal corresponding to the output signal transmitted over cable 54 to physiological monitor 52.

  As an example, the driving current for the polysilicon resistive element of each sensor 60 is 30 to 90 micro A alternating current (square wave) operating at a frequency of about 630 Hz. The time-varying nature of the square wave signal facilitates AC connection between amplifier stages within the signal conditioner. Thus, the AC connection reduces the DC signal drift effect.

  The polysilicon resistive element has a temperature sensitivity in the range of about 2.0 to 3.6 m ohms per ohm / ° C, for example. Since the temperature sensitivity of the resistive element is not guaranteed to be the same, at least one of the two excitation lines will carry an independently adjustable current, thereby allowing temperature compensation of the pressure sensor and possibly an accurate pressure sensor Facilitates the other characterization-based adjustment applied by the signal conditioner to provide a reading of A separate sensor drive current facilitates compensating for differences in changes to resistance in the sensor element over the range of sensor 60 operating temperatures. Temperature compensation is at least for the two excitation lines to the pressure sensor so that the change in voltage between the sensor elements is substantially the same over the entire operating temperature range (ie, within acceptable error limits). This is done by adjusting the excitation current driven on one line. Hereinafter, temperature compensation of the polysilicon resistive sensor element will be described with reference to FIG.

  Although the configuration of the signal transmitted by the line connecting the signal conditioning device to the pressure sensor 60 attached to the guide wire has been described, it will be appreciated that the above-described line configuration for the cable connector 58 is only one example. The configuration of the sensor to which the signal conditioner 50 can be attached and the line between the sensor and the signal conditioner 50 will vary according to design considerations and functional requirements associated with other embodiments of the present invention. For example, other physiological sensors such as speed, flow rate and temperature sensors may be used in place of pressure sensor 60 in accordance with the present invention. The configuration of signals when 10 lines are connected is different in various other embodiments of the present invention.

  Referring to FIG. 2, a sensor / drive circuit of signal conditioner 50 and a polysilicon sensor 60 attached to a guide wire are illustratively shown. Polysilicon sensor 60 includes a first resistive polysilicon element 61 and a second resistive polysilicon element 62. Polysilicon elements 61 and 62 share a common reference voltage supplied from connector 58 via line 63. The first excitation current is supplied to the first polysilicon element 61 via line 64. A second adjustable excitation current is supplied to the second polysilicon element 62 via line 65. The current that has passed through each of the two resistive elements 61 and 62 causes a voltage drop across the resistive element. Since line 63 is connected to sensor elements 61 and 62, the voltage difference between lines 64 and 65 connected to the terminals of the first and second polysilicon sensor elements 61 and 62 is the output of amplifiers 66 and 67. Forwarded respectively. Differential amplifier 70 then senses the difference between the output voltages of amplifiers 66 and 67 on lines 68 and 69, respectively.

  When a pressure change is applied to the polysilicon sensor 60, the resistances of the first and second polysilicon elements 61 and 62 react in a complementary manner. In other words, when the applied pressure changes, one of the resistances increases and the other resistance decreases.

The voltage drops between both terminals of each resistive sensor element according to equation (1):
V = I _(excite) x R _(sensor) (1)

Assuming that the excitation current is stable, the voltage change between the two terminals of each resistive sensor element as a result of the change in resistance of the sensor element is:
V = I _(excite) x R _(sensor) (2)
The change in differential voltage (sum of both voltage changes) between output terminals 68 and 69 (input to differential amplifier 70) corresponds to a given pressure.

  Ideally, the R value for each sensor is solely due to the change in pressure applied to the sensor. However, temperature changes to the sensor elements also change their resistance. Therefore, even if there is no pressure change, the resistance (and thus the voltage drop) between both terminals of the two resistive elements 61 and 62 changes in response to a temperature change.

  However, the difference in voltage change between the two terminals of the sensor element is more important than the voltage change itself of the resistive element. Thus, if the resistances of elements 61 and 62 have changed exactly the same over the temperature range of interest (or the difference was negligible), no temperature compensation is necessary. However, this coincidence of resistance changes is very impractical.

  The signal conditioning device 50 senses the differential voltage from the sensor element of the sensor 60. The voltage is the product of the resistance and the current through the resistance. In one embodiment of the present invention, by adjusting the current through at least one of the resistive elements to compensate for the difference in temperature sensitivity of the two resistive sensor elements, rather than a matching resistance change over the temperature change range. Rather, the voltage change between the resistive elements induced by the temperature change is compensated. Changes to the current through the resistive sensor element 62 are provided by the temperature compensating DAC 71.

With continued reference to FIG. 2, the following equations (3-6) characterize the temperature compensation achieved by equalizing the voltage change induced by the temperature change over the temperature range of interest:
V _a = V _b (for the temperature range of interest) (3)

Assuming that the above voltage change is due to a temperature induced change in resistance,
(I _a ΔR _a ) = (I _b ΔR _b ); therefore (4)
I _b = (I _a ΔR _a ) / ΔR _b ; and (5)
I _b / I _a = ΔR _a / ΔR _b (6)

  In order to compensate for temperature effects, the resistance change for resistors 61 and 62 over the operating temperature range of the pressure sensor is evaluated as substantially linear. Thus, by modifying the ratio of the current through each of elements 61 and 62 to be inversely proportional to the resistance change of elements 61 and 62 over the operating temperature range, the change in voltage across each of elements 61 and 62 is specified. Remain substantially equal (within acceptable limits) over the compensated operating temperature range. In examples where the effect of temperature is not linear (or linear approximation is not acceptable), a polynomial compensation formula (based on temperature) and / or division of the compensated range into sub-ranges may be used with the temperature sensor. it can.

  Referring briefly to FIG. 3, an exemplary physical device / user interface for the signal conditioning device 50 is shown. The exemplary user interface includes a 120 segment, eight character alphanumeric LCD display 80. This LCD display conveys various states of the device during operation. The user interface also includes three instantaneous, normally open switches 82, 84 and 86. The select button 82 allows the unit to be awakened when it is connected to the physiological monitor 52 and has entered sleep mode without being used for a period of time. Select button 82 facilitates selection of the type of signal represented on the output from signal conditioning device 50. An exemplary set of output signal modes includes zero (0 mm Hg), 200 mm Hg, and a calibration pulse sequence (stepping from 0 to 200 Hg in steps of 10 mm Hg every 1/2 second time). When the signal conditioner 50 is in active running mode, the up arrow (+) button 84 allows the pressure output to be adjusted in steps of 1 mm Hg (eg, up to 30 mm Hg). The down arrow (-) button 86 facilitates a complementary function that allows its output to be adjusted downward.

  Attention is now directed to FIG. 4, in which the main functional blocks of a signal conditioning device 50 using the present invention are schematically shown. The power supply circuit 100 receives differential excitation voltages on lines 102 and 104 from a connector 106 that interfaces (via cable 54) to the physiological monitor 52. The power supply circuit 100 converts the differential excitation voltage to 3 volts DC from a variety of different forms including, as an example, DC, sinusoidal and square wave AC signals (as described above). Although the typical input is an AC signal, the power supply circuit 100 can also convert the DC differential input received by lines 102 and 104 to a 3 volt DC power supply. The 3 volt DC supplies operating power to all circuits in the signal conditioner. The power lines to the individual functional circuit blocks have been removed in the drawing to reduce congestion.

  Referring briefly to FIG. 5 where a set of circuits including power supply circuit 100 is schematically shown, if the excitation signals on lines 102 and 104 are alternating current, the signals are rectified by full-wave bridge rectifier 110. An unfiltered full wave DC voltage is generated without a filter capacitor. Large filter capacitors in the rectifier generate excessive surge and waveform distortion due to capacitive loads, especially when powering up. If the excitation signals on lines 102 and 104 are direct current, full wave bridge rectifier 110 directs the largest negative terminal to ground and forwards the positive terminal to subsequent current regulator 112. The differential amplifier 114 monitors the full-wave direct current by the voltage drop of the resistor 116. Differential amplifier 114 controls a p-channel field effect transistor (“PFET”) 118 that limits the current to a peak below 25 mA.

  Filter capacitor 120 connected to the output of PFET 118 is charged to approximately the peak of the excitation voltage on differential input lines 102 and 104 by a controlled current. When the voltage on the filter capacitor 120 is higher than 3.5 volts, Vcc is adjusted to 3.3 volts by a low dropout (“LDO”) regulator 122. The current drain for this mode is less than 6 mA. When the voltage on filter capacitor 120 is lower than the 3 volt requirement for LDO regulator 122, charge pump 124 is powered by PFET switch 126 to boost the voltage to 3.3 volts. The current required by the circuit to maintain a 3.3 volt output (at the input to steering diodes 128 and 130) is high in this mode, but the Vcc output voltage and power remain constant It is still below 15 mA. The increase in current causes a high voltage drop across the bridge rectifier 110 and slightly reduces the voltage on the filter capacitor 120. This provides hysteresis to the PFET switch 126. A pair of steering diodes 128 and 130 transfer the highest voltage output (LDO or charge pump) to the circuit powered by the signal conditioning device 50. A capacitor 132 connected in parallel with the load removes ripple and crossover spikes from the 3.0V output power signal on line 136.

  A small power supply (not shown) provides bias and Vcc power to operate the current regulating differential amplifier 114. The drop across the full-wave rectifier pair that supplies power to the differential amplifier 114 is the pair connected between the input lines 102 and 104 and the current regulator because the current required for the bias circuit is low. Much smaller than. The voltage regulation scheme will provide a high output voltage and will allow proper operation of the power circuit below 2 volts. The resistor limits the load by a filter cap (not shown) that causes surge and excitation distortion in the bias current for amplifier 114. A portion of the filtered low power Vcc supplied as a bias input to the differential amplifier 114 is sent to the 1.234V reference integrated circuit. The 1.234 V reference voltage is divided by resistors into 50 mV and is used to set the current limit in the comparator including amplifier 114 and PFET 118. The 1.234V voltage also sets a trip point (low voltage switch) for the crossover from high to low excitation voltage in the differential amplifier 134. When the voltage set by the divider including a pair of resistors is equal to the 1.234V reference voltage, the output of differential amplifier 134 switches and turns PFET 126 on or off.

  The 3.0V output of the circuit shown in FIG. 5 is used to generate two accurate voltages. Returning to FIG. 4, the accurate voltage reference 138 receives a 3.0V reference voltage from the power supply circuit 100 via line 139 and sets two accurate voltage output signals. Initially, an accurate reference output signal of 2.5V is generated from the 3.0V output by the integrated circuit. The 2.5V accurate reference is used whenever high accuracy is required. The 1.5V reference output signal is then derived from the 2.5V reference via an accurate voltage divider. The 1.5V reference is used to center the operating voltage range of the amplifier in the signal conditioner 50 and as a return path (Vref) for the pressure sensor resistance.

  With continued reference to FIG. 4, a microcontroller 140 such as the microcontroller MSP430P337A from Texas Instruments (Dallas, TX) is powered by the 3.0 V power signal output on line 136 of the power supply circuit 100. The The microcontroller 140 operates away from a 32,768 Hz watch crystal that is internally multiplied above 1 MHz. Microcontroller 140 provides timing and data signals for driving the circuit shown in FIG. Microcontroller 140 also receives a filtered digital signal corresponding to the sensed pressure and processes the received pressure value (eg, performs FIR filtering). Hereinafter, the output and input signals of the microcontroller 140 will be described with reference to the functional blocks shown in FIG.

  The square wave signal supplied to sensor current source 160 via line 152 and to temperature compensated DAC 162 via line 154 is accurately adjusted. Such accuracy is desirable because it is proportional to the signal supplied with the current across the sensor that measures the pressure, in the excitation signal on line 152 to the sensor current source 160 or on line 154 to the temperature compensation DAC 162. Any inaccuracies in this will affect the accuracy of the signal conditioning device. Thus, in one embodiment of the present invention, a pair of accurate square wave generators 150 are driven by a timing signal on line 142 from microcontroller 140. The 2.5V output of the accurate voltage reference 138 provides an accurate power signal to the square wave generator 15, which supplies an accurate 627Hz square wave signal to the sensor current source 160 and the temperature compensation DAC 162. Make it possible to do.

  The 627 Hz square wave received from the microcontroller 140 by the square wave generator 150 is approximately 3V. The desired voltage level for the sensor current source 160 is a 0.600 volt peak (1200 Vp-p), 1.5 volt to keep the operational amplifier in the sensor current source 160 within the linear operating range. Centered by DC offset. The VMOS FET in the square wave generator 150 accurately adjusts the square wave signal having the above-described characteristics provided to the sensor current source 160 and the temperature compensation DAC 162. The VMOS FET saturates when the gate is driven high by 3V signal from the microcontroller 140. Due to the low “on” resistance of the FETs and the high value of the pull-up resistance connected to the drain of each FET, this essentially forces the drain to 0 volts. As described above, an accurate reference signal of 2.500V from an accurate voltage reference 138 provides power to the FET. With respect to the FET driving the square wave input signal to the sensor current source 160, the accurate resistor divider sets a “high” level square wave input when the FET is off. The output voltage on line 152 is 2.106V. When the FET is on, the output voltage on line 152 drops to 0.904V. Thus, the peak-to-peak voltage of the square wave on line 152 is 1.202V and the square wave signal on line 152 is centered at 1.5V.

  A similar square wave generator for line 154 for a temperature compensated (“TC”) DAC 162 develops a square wave having a magnitude between 1.200V peaks. All of the DACs in the exemplary circuit including TC DAC 162 (eg, TI5616 12-bit DAC from Texas Instruments) have an internal gain of two. Thus, the DC offset of the signal driven on line 154 is half of the desired output DC offset of 1.5V, ie 0.750V. Therefore, the high and low voltages (high / low) of the square wave are 1.350V and 0.150V, respectively.

  Sensor current source 160 receives a square wave input signal on line 152 and a temperature compensated square wave input on line 164 from temperature compensated DAC 162. Sensor current source 160 supplies first and second excitation currents to sensor interface 170 via lines 166 and 168. This sensor interface 170 sends the signals received on lines 166 and 168 to resistive sensor elements 61 and 62 on sensor 60 (see FIG. 2). The sensor interface 170 also provides an accurate voltage reference of 1.5V to the sensor 60 to complete the sensor circuit path through the resistive sensor elements 61 and 62. The sensor output signals on lines 172 and 174 provide a differential voltage signal corresponding to the change in pressure applied to pressure sensor 60. As described above, the change in differential voltage results from the change in resistance of the resistance sensor elements 61 and 62 due to the applied pressure change.

  Line 166 (corresponding to line 64 in FIG. 2) includes a fixed current source that supplies 60 microampere (peak) current to resistive sensor element 61. The current on line 166 is proportional to the input voltage (0.600 Vpeak as shown above). A peak 0.600 input voltage is developed across the 10K current setting resistor. This sets the current to (0.600 / 10K) = 60 μA peak.

  Line 168 (corresponding to line 65 in FIG. 2) includes a variable current source that supplies 30 to 90 microamperes of current to resistive sensor element 62. This variable current source is derived from a square wave signal on line 152 based on a programmable digital input value transmitted on a data line (not shown) along with a load signal on line 176 from microcontroller 140. The resulting fixed 1.2 Vp-p square wave is summed with the variable square wave generated from TC DAV162. This sum is developed across the 10K current setting resistor. The current varies by TC DAC 162 between 30 μA (peak), which is programmed by the microcontroller with a DAC 162 value of zero, and 90 μA (peak), where the DAC 162 is loaded with a full output digital input value (eg, 0 FFF). Be made.

  The amplifier (see amplifiers 66 and 67 in FIG. 2) buffers the voltage sensed by lines 166 and 168 (lines 68 and 69 in FIG. 2). A pull-up resistor is also connected to lines 166 and 168 to ensure proper detection when the sensor 60 is removed. The pull-up resistor is a very large resistance to reduce the accuracy error when a wire is present (eg, 10M ohms in parallel with a nearly 3.5K sensor is 0.04% error).

  The buffered sensed voltage on lines 166 and 168 is sent to the input terminal of differential amplifier circuit 180 by lines 172 and 174 as a differential voltage pair. As an example, the differential amplifier circuit 180 is a small signal amplifier having a gain of 25, and performs excellent common mode rejection. Feedback resistors and capacitors are included in a known manner to provide stability and reduce the response of the output of differential amplifier 180 on line 182 to a small amount of phase delay in the wire.

The differential sensor output voltage of sensor current source 160 on lines 172 and 174 is also sent to fault detection circuit 184. The resistance range of each sensor circuit is 2500 to 5000 ohms. With a current of 60 μA through sensor element 61 and a current of 30 to 90 μA applied to sensor element 62, the minimum and maximum voltages across the resistive element are as follows:
For resistive sensor element 61:
V (min) = 60 μA * Rmin = 150 mV
V (max) = 60 μA * Rmax = 300 mV
For resistive sensor element 62:
Vb (min) = 30 μA * Rmin = 75 mV
Vb (max) = 90 μA * Rmax = 450 mV

  A window detection circuit within fault detection circuit 184 monitors the minimum and maximum sensor square wave voltages. This voltage is compared to set a limit defined by the resistor divider network. For fault conditions, Rmin and Rmax limits are guaranteed to be abnormal (eg, 8.7 K ohms and 1.5 K ohms) and corresponded to maximum and minimum voltages (eg, 2.02 V and 1.55 V) Set to value. The voltage limit sets the window comparator range. When the input voltage exceeds the window range, the output of the comparator is switched. Four comparators are supplied to the microcontroller and indicate a failure. The output of the comparator has a long time delay to prevent momentary failures from causing troublesome failures. The fault detection circuit provides three fault status signals to the microcontroller 140 via line 186. The two lines identify a short circuit for each of the two resistive sensor elements 61 and 62 on sensor 60. The third line identifies the case where the guide wire 56 is not connected to the signal conditioner 50.

  The variable offset stage 190, the next stage of the signal conditioning device 50, receives the amplified differential output signal from the differential amplifier 180 via line 182. This variable offset stage 190 also receives an offset voltage signal via line 192 from an offset DAC 194 (programmed by the value sent by the data line along with the load select line 196 from the microcontroller 140). The variable offset stage 190 facilitates zeroing out the offset due to imbalance resulting from the temperature compensation performed by the TC DAC 162. After compensating for the temperature of the resistive sensor elements 61 and 62, the current passed through each element 61 and 62 is not equal at most moments. As a result, there is a differential voltage between the signals transmitted on lines 172 and 174 when there is no applied pressure. Since the offset voltage supplied via line 192 by offset DAC 194 causes the voltage difference to be zero, the input to analog-to-digital converter 220 is set to a voltage representing 0 mmHg (approximately 729 mV). An acceptable window for 0 mm Hg that the microcontroller 140 can accommodate is 0.5 to 1.0 volts. Microcontroller 140 internally corrects the voltage within this window. The offset value is affected by the gain of variable gain stage 200. The microcontroller 140 takes this gain stage into account when setting the offset DAC 194. In one embodiment of the signal conditioning device, the offset stage 190 also has a fixed gain of 5, thereby reducing the gain otherwise required by the differential amplifier 180 and the bandwidth of the differential amplifier 180. Decrease.

  The required offset depends on the worst case study of the sensor specification. In one embodiment of the signal conditioning device, the maximum offset from the sensor is 33 mV. After 25 differential amplifier gains, the offset increases to 825 mV. Taking into account changes in atmospheric pressure and pressure measurements, the offset range is increased to 1.0 V, for example, for margin.

  The output signal on line 202 from variable offset stage 190 is received by variable gain stage 200. The variable gain stage 200 provides a variable gain that is determined by the input sent from the gain DAC 204 via line 206. The gain DAC 204 receives the programmed gain value (calibration value provided by the EEPROM of the sensor 60) along with the selection signal transmitted on line 208 from the microcontroller 140.

  In one embodiment of signal conditioner 50, the total system gain (based on sensor specifications) is between 125 and 2500. Thus, taking into account the gains of the previous two amplifier stages (ie 25 and 5), the last stage must have a gain of 1-20. Microcontroller 140 acquires the gain for the connected sensor by reading the EEPROM of sensor 60 and sends the corresponding value to the data input of gain DAC 204, selectable by a data line (not shown). The output of variable gain stage 200 is provided on line 210.

  The synchronous demodulation circuit 212 extracts a voltage peak from a square wave signal resulting from a square wave input to a sensor operating on the sensed pressure signal and the signal conditioning circuit. In one embodiment of the present invention, an AC system was generated to block DC signal components rather than operating a DC connection system that is prone to drift and high cumulative offset. In the exemplary embodiment, a square wave is employed because its level is more easily measured on the oscilloscope and is not prone to generate phase errors associated with the sine wave when the voltages are summed. In the exemplary embodiment of the invention, the AC coupled waveform, which is the peak level of the square wave input sent via line 210, is demodulated by the synchronous demodulation circuit 212 to represent a DC level signal.

  Demodulation is performed by synchronously sampling the last 50% of each peak (positive and negative) of the square wave (see FIG. 6). Under the control of the timing signal from the microcontroller 140, the synchronous demodulation circuit 212 inverts the square wave and half peaks that are not inverted (shown as signal waveforms 270, 272, and 274 on the left side of FIG. 6). In addition, both halves of the full square wave cycle are sampled by sampling the half peak of the inverted signal (shown as waveforms 276, 278 and 280 on the right side of FIG. 6). The sampled pulses (active high) are shown as waveforms 282 and 284 as non-inverted and inverted waveforms, respectively.

  The positive and negative peak voltages are stored in sampling and holding capacitors within the synchronous demodulation circuit 212. The charge accumulated on the capacitor represents a DC voltage corresponding to the square wave peak value. The capacitor accumulates the charge acquired between samples (by ADC 220), although a small discharge of the capacitor during the non-sampling period causes ripple. Referring to FIG. 7, the inverted sampling scheme allows two samples to be obtained from each period of the input waveform 286 according to the sample waveform 288 (inverted samples are indicated by dashed lines), thereby It further increases accuracy and reduces ripple in the output signal on line 214 (shown as waveform 290 and dashed actual sample input 292) from synchronous demodulation circuit 212.

  Demodulator filter stage 216 is a low pass filter that removes 627 Hz ripple in the sampling and holding circuit and reduces 50 Hz and 60 Hz noise. The corner frequency is set by the FIR filter in the microcontroller 140 firmware to provide a system bandwidth of 25 Hz.

  The low pass filtered output on line 218 is received by analog to digital converter 220. The output of the ADC 220 is sampled at a rate of 256 Hz (interrupt driven by the microcontroller 140) by the clock, control and data lines 222 that connect the ADC 220 to the microcontroller 140. After receiving the filtered and digitized signal from the ADC 220 via line 222, the microcontroller 140 performs additional operations on the received data (eg, FIR filtering) and then inputs the value to the line. The output is output to the output stage 230 via 224. An output stage 230 that includes two digital-to-analog converters receives power from the differential excitation signal by lines 102 and 104 of cable connector 106 and receives digital control data (to the DAC) by line 224. The output stage generates a differential output voltage on lines 232 and 234 and provides it to the cable connector 106.

  Referring briefly to FIG. 8, output stage 230 modulates the excitation signal, which can be either DC or AC, received from physiological monitor 52 via cable 54 with a pressure waveform to produce a magnitude of the excitation signal. Develop a signal proportional to height and sensed pressure. Microcontroller 140 receives the digitized pressure waveform input from ADC 220 via line 222, adapts the FIR filter, adapts offset and gain adjustments to the output stage, and outputs digital information via line 224. Send to a pair of DACs 236 and 238 in stage 230.

  The digitized pressure waveform values sent to the DACs 236 and 238 by the microcontroller 140 are the excitation signals (buffered by the buffer stage 240) sent to the reference inputs of the DACs 236 and 238 of the output stage 230 via lines 242 and 244. Modulated and inverted buffered). The two DACs 236 and 238 generate a differential output that duplicates the excitation voltage but is opposite in polarity. The differential signal output from the two DACs 236 and 238 is subtracted from the differential excitation signal sent via lines 242 and 244. Since the signal is a differential signal, the DC offset introduced by DACs 236 and 238 or the excitation signal does not cause the problem of biasing the output amplifiers to their linear range. Thus, the output stage 230 is DC connected, which is a general requirement for examples where a DC excitation signal source is used.

  Before the differential pressure signal is output by cable 54, additional signal conditioning is applied via lines 242 and 244 to the modulated excitation signal. Initially, the differential signal passes through a buffer / inverting buffer stage 246. The buffered output of buffer stage 246 is then attenuated by resistor network 248. When the sensed pressure is 0 mmHg, the differential output across the output attenuator stage 248 is zero volts because the DACs 236 and 238 have zero excitation voltage. The resistance of the attenuator stage 248 is selected to allow the circuit to meet AAMI requirements for a low output impedance and differential voltage equal to 5 μV / V / mmHg. The amplifier circuits in buffer stages 240 and 246 have a bandwidth that is greater than the maximum excitation frequency. Feedback capacitors on the buffer amplifiers in stages 240 and 246 limit the maximum bandwidth (frequency response characteristic) while ensuring output stability.

  Microcontroller 140 interfaces with several peripheral components. A set of data / clock lines 250 interface to calibration information. The signal conditioning device 50 includes a calibration EPROM 252 that includes a set of values entered during manufacture that characterizes circuit operation. The set of values stored in EPROM 252 during calibration of signal conditioner 50 includes calibration data for the input stage (eg, gain DAC), calibration to adjust the gain of the output stage (eg, output DAC). Data, and data checking (eg, checksum) are included. A set of data / clock lines 250 is also connected to an external sensor interface 254 that facilitates extracting calibration / characterization data sets for resistive sensor elements 61 and 62 on pressure sensor 60 attached to the guidewire. Has been. Information stored in the EEPROM includes temperature compensation offset, gain, and offset value. The microcontroller 140 reads the EEPROM value once during the setup of the TC DAC 162, offset DAC 194 and gain DAC 204. Calibration / characteristic data is stored, for example, in an EEPROM connected to a guide wire that supports the sensor 60.

  The additional data / control lines support the signal conditioning device user interface elements described with reference to FIG. A set of lines 260 provides data / control to the LCD output circuit 262. A second set of lines 264 interfaces the microcontroller 140 to the touchpad 266.

  Having described the components and functional blocks of the signal conditioner 50, the following focuses on the calibration, setup and operation of the system including the signal conditioner 50.

[Manufacturer's signal conditioner calibration]
The signal conditioning device 50 includes both an input (patient side) and an output (physiological monitor side) that are calibrated during manufacture. The sensor input of the signal conditioning device 50 is calibrated during manufacturing test so that the accuracy of the display and calibration transfer function is high. In particular, after connecting a calibration standard (eg, a mock sensor providing a differential resistance) to the signal conditioning device 50 and entering the input calibration mode, the signal conditioning device 50 polls the analog differential voltage input. The calibration input voltage is read and checked. Thereafter, the signal conditioner 50 performs an automatic zero function on the differential voltage input corresponding to 0 mmHg. The sensor input of the signal conditioning device 50 is then calibrated against the signal input intended to represent 200 mmHg. If the difference between the 200 mmHg reading and the zero reading is greater than 3% of the reading (ie, +/− 6 mmHg at 200 mmHg), a critical error is registered. If the difference is within 3% (ie +/− 6 mmHg), the difference between the actual value at 200 mmHg and the theoretical value (based on zero point) is stored and connected to the signal conditioner Used to modify the gain DAC code for each wire made.

  Output calibration has two modes of operation: manufacturing test and field adjustment. During manufacture, the output of the signal conditioning device 50 to the physiological monitor is calibrated to provide a standard 5 μV per volt / mmHg output signal. In the production test mode, the technician adjusts the scale factor up / down to achieve the desired output. The adjustment coefficient set during the test is stored in the EEPROM of the signal conditioner 50. In field mode, the output is adjusted to meet the signal input requirements for a particular physiological monitor. After connecting the signal conditioner 50 to the physiological monitor 52, the user is prompted to press the up / down arrows until the output on the physiological monitor shows 0 mmHg. Once the 0 mmHg output is set, the user is prompted to press the up / down arrows until the output signal is properly scaled so that the physiological monitor shows 200 mmHg.

[Guide Wire EEPROM Base Signal Conditioner Calibration]
With respect to “input” calibration of the guide wire 56, the guide wire 56 provides sensor characterization data from the EEPROM when connected to the signal conditioner 50. The EEPROM read / write functions are performed via a standard two-wire serial interface (data or "SDA" and clock or "SCL") well known to those skilled in the art. Each connected guidewire attachment sensor includes a piezoresistive pressure sensing element having specific pressure / temperature response characteristics. During manufacturing, the pressure / temperature response characteristics are determined and the signal adjustment values that produce consistent outputs are stored in an EEPROM mounted on the guide wire housing. These values include temperature coefficient offset current, signal conditioning gain, position offset default value, and checksum. These values are applied to the signal conditioning DAC described above to modify the sensor current and differential voltage representing the sensed pressure.

After reading the characteristic description data from the EEPROM, the signal conditioning device 50 provides the calibration information provided to the adjustment circuit. The calibration information includes a temperature compensation value adapted to the temperature compensation DAC 69 that modifies the input current to the compensated resistive sensor element 62. As described above with reference to FIG. 2, the modification to the current through the resistive sensor “R _b ” 62 substantially does not effectively remove the temperature effect on the differential signal read from lines 68 and 69. Reduce this over the specified operating temperature range. The calibration information also includes gain and offset values adapted to the gain and offset DAC that modify the analog differential voltage derived from lines 68 and 69. Temperature compensation and gain factors are fixed during testing by the manufacturer. The offset factor is a default value that can be modified when loaded during startup.

  After adapting the calibration information to the adjustment circuit, on the basis of the calibration data supplied by the guide wire sensor EEPROM, the signal conditioning device 50 sends a square wave excitation pulse to the AC-connected sensor 60 for differential through the sensor interface. Read the voltage signal. The excitation pulse is a square wave driven at approximately 630 Hz by the timer output of the microcontroller. The demodulated pulse is driven at the same frequency as the excitation pulse, but has a different duty cycle. The sensor voltage is measured to ensure a signal within the expected range. Detected errors include the presence of “no wire” and “shorted wire”.

[User calibration of signal conditioner]
The signal conditioning device 50 feeds the contents of the EEPROM attached to the guide wire to the DAC, confirms that the sensor 60 is properly connected, and then feeds it to the physiological monitor 52 via cable 54. Test the output. The output from the signal conditioner 50 to the physiological monitor 52 is 10 mmHg every 1/2 second from 0 mmHg to 200 mmHg to allow the operator to verify the offset and gain via the output of the physiological monitor 52. Pulsed to increase.

  With respect to the connected guide wire 56 and connector cable 54, the signal conditioner 50 performs an automatic zeroing operation. Auto-zeroing sets the currently sensed pressure as zero, ie the reference pressure. The output of the signal conditioning device via the cable 54 to the physiological monitor 52 is a voltage corresponding to the 0 mmHg level. Thereafter, in response to the disconnection and reconnection of the guide wire 56, the signal conditioner initializes the output again to zero.

  The auto-zeroing routine that is executed in response to the first sensing of the guide wire connected to the signal conditioner 50 includes two main stages. During the first stage of the automatic zeroing routine, the microcontroller determines whether the amplifier stage has “railed”. Analog-digital mapping in the signal conditioner is from -210 mmHg (zero count) to 510 mmHg ($ FFF count). The full scale range is only a part of this area (eg, -30 to 330 mmHg). One possible cause for “railing” is the use of the sensor device at an altitude that is significantly different from the altitude of the initial sensor calibration position. If the amplifier rails, the microcontroller will attempt to return it to its linear gain region. Therefore, during the first stage, it was required to sample the sensor over a 1/2 second period, average the samples, and place a “zero” reading within 50 mmHg of the preferred zero point (eg, $ 4 AA). The amplifier is de-railed by calculating the step in the offset DAC ("count"). The count value is adjusted repeatedly until the target area is reached.

After the amplifier is derailed, during the second phase, a final adjustment is made to set the zero input reading. The microcontroller also checks for a changing input signal indicating that the guidewire sensor is prematurely placed in the body. During the second phase, sampling is performed at a cycle of 4 seconds. During zeroing, the gain DAC 204 code is read and the microcontroller calculates the adjustment transfer function according to equation (7):
Step = ΔADC count * (volt / ADC count) * (step / volt) * 1 / configuration gain

(7)
Here, configuration gain = base of variable gain + (gain DAC code * gain / DAC code).

For example, if a 12-bit DAC is used and the gain span is 5 to 100, then gain / DAC code = 95/4096, and the $ 1A5 DAC code is
Configuration gain = 5 + (421 * 95/4096) = 15 (approximately)
Give rise to

1 / Gain = $ 8000 / (Gain configured with Hex) * Note: This is the number of Q15 (ie, binary fractional value with sign bit and 15 bits of resolution).

Step / Volt = 1 / (2 * Ref Volt * (offset circuit gain) * 1/4096
Volts / ADC count = Ref Volt / 4096
Here, bolt / (ADC count) = 2.5 / 4096
Conversion to Q15 is possible with zero point representing a value of $ 0014-measured zero = ΔADC count, and number of steps (+/-) = ΔADC count / ADC count adjustment per step (ie DAC Is currently 8, a value of minus 10 represents a negative (errored) value for the DAC) and the calculated number of steps is compared to the current offset.

  After automatic zeroing, the output of the signal conditioning device 50 provided to the physiological monitor 52 via cable 54 is adjusted using up / down (+/−) arrow buttons 84 and 86. This adjustment, called “normalization”, modifies the internal offset (normalization) variable in the microcontroller. The internal offset variable is added to the calculated output pressure sample before sending the output sample value to the output DAC. Up / down adjustments are typically made after the pressure sensing guidewire 56 has been inserted into the body (eg, near the aortic pressure sensor).

[Running mode]
In the run mode, the signal conditioner 50 receives a differential analog voltage signal indicative of the pressure change applied to the sensor 60, calculates the corresponding pressure signal, and outputs it to the physiological monitor 52 via the cable 54. The LCD 80 indicates “running”. The sampling and update rate for the signal conditioning device 50 is high enough to accommodate the blood pressure change rate and low enough to ensure that there is sufficient power to operate the microcontroller. An exemplary sampling rate is approximately every 4 milliseconds. However, those skilled in the art will recognize that a wide range of sampling rates are satisfactory when considering the factors described above.

  The microcontroller applies FIR filtering to the sampled data to improve signal quality. In the exemplary embodiment, the MSP430P337A microcontroller from Texas Instruments includes a built-in signed multiply-accumulate peripheral that enables fast multiplication. The FIR coefficients are stored in a dedicated memory (“ROM”) on the device. The number of coefficients is determined by the response requirement factor and the time to perform the filter function on the microcontroller operating at a relatively slow clock cycle to reduce power consumption.

  Microcontroller 140 scales the output before sending values to output DACs 236,238. The mapping of input values from the ADC converter 220 is applied to the output DACs 236, 238. The input zero from the ADC converter 220 is allowed to float between +/− 100 mmHg near the ideal zero point, while the output zero is absolute. During zeroing, the signal conditioner sets the operating range of the amplifier (via the offset DAC) and averages over the received zero input signal. The averaged value is used as a reference for subsequent measurements (this reference is subtracted from the reading during the running mode). This value is multiplied by the output attenuation factor before being sent to the DACs 236,238.

  The LCD 80 of the signal conditioner 50 can display the maximum and minimum pressures over the latest completed sample interval (eg, 4 seconds). The microcontroller converts the digital output value (count) into mmHg (binary number), and then converts the binary mmHg value into binary coded decimal numbers (100, 10 and 1) in a known manner. This number is converted to a segment in the LCD display and latched to turn on the appropriate LCD 80 segment.

  Referring to FIG. 9, the flowchart summarizes a set of functional operations performed by the signal conditioner 50. Initially, in step 300, signal conditioning device 50 is connected to physiological monitor 52 via cable 54. In response to the power supplied by the physiological monitor 52 with the excitation signal transmitted through the cable 54, in step 302, the signal conditioning device 50 performs a power-on self-test and outputs a value of 0 mmHg to the physiological monitor. If the signal conditioner 50 detects an operational error during step 304, control proceeds to step 306 where an error message is displayed on the LCD output of the signal conditioner 50. Control returns to self-test step 302.

  However, if the self-test is successful, control proceeds to a state where the signal conditioner 50 checks the connection of the guide wire cable connector 58 (indicating that the guide wire is connected), which is the interface button 82, 84. And it becomes the mode which responds to selection of 86. If a calibration sequence is selected using the interface button during step 310, control proceeds to step 312 where the user is prompted to adjust the output signal. Output calibration modes include 0 mmHg, 200 mmHg, and a pulse calibration mode that alternates between 0 mmHg and 200 mmHg for a period of time for output to the physiological monitor. The user makes a button selection input to exit this calibration mode and proceeds to step 314.

  In this step 314, the signal conditioner determines whether a sensor static cable (including feature description EEPROM) is connected to the signal conditioner 50 (step 314 also if a calibration button selection was not sensed). Is also reached from step 310). If the signal conditioning device does not sense the connected sensor static cable 59, the LCD display indicates “no cable” and control returns to step 310.

  If the signal conditioner senses a connected sensor static cable, control proceeds to step 316, where the signal conditioner describes the operational characteristics of the sensor supported by the connected guide cable. The contents of the feature description EEPROM are read. During step 318, if the EEPROM value is invalid (eg, a checksum error), an error message is displayed during step 320 and control returns to step 316. However, if the value is valid, it is used during step 322 to configure the signal conditioning device 50 circuitry. Thereafter, during step 324, if the signal conditioner determines that the user has entered the calibration mode button selection sequence, control proceeds to step 326 where the endpoint and intermediate steps are within range of output to the signal conditioner. A step of calibrating is performed. The user exits this mode by entering a predetermined button selection sequence, and control proceeds to step 328 (also reached when no calibration button sequence is sensed during step 324).

  The static cable 59 can be connected while the guide wire 56 is removed. Accordingly, if the guide wire 56 is not connected during step 328, control returns to step 324 and "no wire" is displayed on the LCD output of the signal conditioner. Otherwise, if the guide wire 56 is connected, control proceeds to step 330 where the signal conditioner checks the sensor wire and pulses the output between zero and 200 mm Hg for 10 seconds; A run mode is entered in which automatic zeroing is performed on the sensor input, the input value is sensed and an output value is generated on the physiological monitor.

  The signal conditioner also responds accordingly in response to input from the select “+” and “−” buttons. The select button causes the signal conditioner 50 to output the currently calculated and sensed pressure (maximum and minimum pressures calculated in the last 2 seconds) on the LCD screen. In other cases, the LCD simply outputs “running” and provides a differential output signal to the physiological monitor 52 via cable 54.

  If any of the up / down buttons are pressed during step 332, control proceeds to step 334 where the output is adjusted up or down according to the pressed button. If the up / down button has not been pressed during step 332, control proceeds to step 336. If a sensor wire break is detected during step 336, control proceeds to step 338 where the signal conditioner 50 provides an output representing zero output via cable 54 and the LCD display displays "no wire". To do. Control continues to step 340.

  If no wire reconnection is detected at step 340, control proceeds to step 342. If the static cable is not interrupted during step 342, control returns to step 338. Otherwise, if a cable break is detected, control returns to step 310. If wire reconnection is detected at step 340, control proceeds to step 344. If a wire break is sensed during step 344, control proceeds to step 310. Otherwise, if the static cable is still connected, control returns to step 332.

  While an exemplary set of steps (stages or states) related to the operation of the signal conditioning device 50 using the present invention has been described, in another embodiment, these steps include additional functionality. It will be appreciated that modifications may be made to eliminate functionality or to reconstruct the sequence of operations performed by the device. Such modifications are considered in view of the flexibility to program such devices in a manner well known to those skilled in the art.

  Another feature of an exemplary system using the present invention is that it is used to treat a patient in a medical facility. Some such treatments include determining the severity of vascular occlusion and determining the treatment strategy. Referring to FIGS. 10 and 11, the front and rear surfaces of a slide rule type device 400 and the slide insert are shown. A slide rule type device is used with a pressure reading provided by a guidewire attached pressure sensing system that is signal-tuned to represent a fractional flow reserve value. Thereafter, the fractional flow reserve values calculated on the front of the slide rule type device 400 shown in FIG. 10 are treated according to the chart shown on the rear surface of this device 400 shown in FIG. Used to express guidance.

  Referring initially to FIG. 10, fractional flow reserve (“FFR”) represents the degree of occlusion in a blood vessel that has occurred due to a stenosis as represented by a pressure measurement from within the blood vessel. FFR is calculated as the ratio of the two pressure sensor readings in the artery. The first value represents the sensed pressure measured from the side of the blood vessel in which blood is flowing towards the stenosis. This is the higher of the two pressure readings. The second pressure value divides the first value and is measured on the opposite side of the stenosis. The slide rule device performs division by matching the first and second values in the designated slide rule scale portion. The FFR value is then read from the corresponding result scale. The slide rule device shown in FIGS. 10 and 11 consists of three parts: a front plate 402, a rear plate 404 and a slide insert 406 (visible through a pull tab 405 cut out in the front plate 402). Become. The front and rear plates 402 and 404 are fastened together, for example, by rivets 408, 410, 412 and 414 located at the four corners of the plates 402 and 404. Other means for securing the front and back plates together are well known. Thereafter, the slide insert 406 is disposed between the front and back plates 402 and 404. Although not shown in the drawings, an inner groove guides the slide insert 406 within the device 400.

  Referring to FIG. 10, in one embodiment of the invention, this ratio represents the mean aortic pressure (Pa) divided by the mean end pressure (Pd) measured in the vessel. The upper slide rule 420 includes a molecular scale 422 representing the Pa measurement, which is printed on the front plate and ranges from 1 to 40 mmHg. The upper slide rule 420 also includes a denominator scale 424 (representing a Pd value) and a calculated FFR scale 426 (Pa / Pd) printed on a slide insert 406 that can be viewed through a window 428 in the front plate 402. Contains. The printed values on scales 422, 424 and 426 facilitate division by subtracting the values on denominator scale 424 from the values on numerator scale 422 and reading the results on logarithmic FFR scale 426. Logarithmic.

  The extended slide rule 430 is provided to handle high pressures in the range of 40 to 330 mmHg. To accommodate a wide range of aortic pressure readings, the lower scale is divided into two parts. The scale configuration of the extended slide rule 430 corresponds to the configuration described with reference to the scale of the upper slide rule 420. The Pa pressure value is printed on the front plate 402. Pd and the calculated FFR value are printed on the slide insertion unit 406. It will be appreciated that this configuration is modified so that the sheet on which the scales are arranged is different from that shown as an example in FIG. If space permits, instructions 440 for using the slide rule device 400 are printed on the front plate 402. It should be noted that the FFR may instead be calculated digitally and / or displayed instead.

  Referring to FIG. 11, a set of charts are shown that assist in decision making after determining a patient's FFR for a particular blood vessel. The first section 460 provides a stent grading system based on FFR values determined after performing balloon vascular treatment and / or stent deployment. Values between 1.0 and 0.9 represent optimal results. An FFR value between 0.9 and 0.75 characterizes satisfactory results, but it also indicates a risk of restenosis. A value below 0.75 is characterized as an unsatisfactory result (eg, a serious lesion is still present).

  The second section 470 provides guidance on reading the results of the FFR calculation. A first graph 472 labeled “Normal FFR Results” represents an example of an intermediate angiographic lesion. Pressure tracing indicates that the lesion was rated using a pressure guide wire and that the FFR was found to be higher than 0.75 (ie, FFR = 0.98). Therefore, the lesion was not critical for hemodynamics. The second graph labeled “Abnormal FFR Result” shows an example of an angiographic intermediate lesion assessed by a pressure wire, and the displayed FFR pressure tracing result is from 0.75 It turned out to be low. This lesion is therefore hemodynamically significant and restricts blood flow. The third section 480 provides a set of study results regarding FFR values, treatment, and patient events and survival. It will be appreciated that the following is an example of the type of information that can be provided on the rear plate 404. In another embodiment of the invention, other information is provided.

  Hereafter, the usage method of the apparatus of this invention is demonstrated. With respect to sensor 60 and physiological monitor 52 connected to signal conditioner 50, sensor 60 is advanced to a target location within the patient's vasculature (not shown). For example, physiological parameters such as pressure, temperature, velocity, or flow rate of blood flowing through the patient's blood vessel are measured by sensor 60 and sent to signal conditioning device 50. The device 50 adjusts the signal and sends it to a physiological monitor 52, which displays the signal as a physiological parameter measure at the target location. Measurements can be taken at various locations within the patient's blood vessel to facilitate the determination of a medical treatment modality suitable for use at that target location. For example, if the treatment location includes a constricted region of the patient's blood vessel, as described above with reference to FIGS. 10 and 11, the sensor 60 may apply pressure across the stenosis to determine a fractional flow reserve. Can be measured. Additional techniques will be apparent to those skilled in the art.

  Exemplary embodiments of the invention and certain variations thereof are shown in the drawings and the accompanying description. Those of skill in the art will readily appreciate from the above disclosure that many variations to the disclosed embodiments are possible in other embodiments of the invention. Such modifications include, by way of example, modifications to the form and / or content of the disclosed circuits and functional blocks. The invention is not limited to the disclosed embodiments. Rather, the invention covers the disclosed embodiments and others within the scope of the invention, as permitted by the invention defined by this disclosure and the appended claims. It is.

2 is a schematic diagram illustrating a conventional connection scheme between a diagnostic pressure sensing guidewire and a physiological monitor and an exemplary connection scheme according to the present invention between a diagnostic pressure sensing guidewire and a physiological monitor, respectively. 1 is a schematic circuit diagram showing a part of a pressure sensor that performs temperature compensation of a pressure sensor element according to an embodiment of the present invention. 1 is a schematic perspective view of an exemplary packaging signal conditioner for commercial use. FIG. 1 is a schematic diagram of main functional elements of an exemplary signal conditioner. FIG. 1 is a schematic circuit diagram of a power supply circuit incorporated in a signal conditioning device using the present invention. The wave form diagram which shows the demodulation system which extracts a peak voltage from the output waveform of the analog signal adjustment element of a signal conditioner exemplarily. The wave form diagram which shows another characteristic of the demodulation system which enables the signal sampling in the rate of 2 times the input waveform repetition rate exemplarily. FIG. 3 is a schematic circuit diagram of an output stage that supplies a differential output from a signal conditioner to a physiological monitor. 6 is a flow chart summarizing the operation of an exemplary signal conditioning apparatus using the present invention. 1 is a front view of a slide rule device used to calculate a blood flow restriction measure known as fractional flow reserve. FIG. The rear view of a slide rule apparatus.

Claims (8)

Signal adjustment is configured to be insertable between an intravascular measurement device having a pair of resistive sensing elements and a physiological monitor, and facilitates setting of compatibility between the physiological monitor and the intravascular measurement device In the device
A sensor current source that provides two separate excitation signals that independently drive a pair of resistive sensing elements of an intravascular measurement device;
Data input for receiving a fixed temperature compensation calibration value for the intravascular measurement device;
A variable current source for supplying a regulated current to at least one of the pair of resistive sensing elements to compensate for temperature changes in the pair of resistive sensing elements;
Adjusting current is obtained from a fixed temperature compensated calibration values, signal conditioner.   The signal conditioning apparatus of claim 1, wherein the variable current source is powered by an excitation signal supplied by a physiological monitor.   The signal conditioning apparatus according to claim 1, wherein the variable current source includes a digital-to-analog converter.   The signal adjustment device according to claim 1, wherein the intravascular measurement device includes a pressure sensor having a size suitable for sensing pressure in the blood vessel. The signal conditioning apparatus of claim 2 , wherein the excitation signal provided by the physiological monitor is less than about 20 mArms. The signal conditioning apparatus of claim 2, wherein the excitation signal provided by the physiological monitor is an alternating current signal. 3. A signal conditioning device according to claim 2, wherein the excitation signal supplied by the physiological monitor is a square wave signal. The signal conditioning device of claim 1, wherein the pair of resistive sensing elements is a component of a pressure sensor sized appropriately to sense pressure in a blood vessel.

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