JP4989992B2 - Matching circuit, transmitter, receiver, transceiver, and radar device - Google Patents

Description

  The present invention relates to a matching circuit having an impedance adjustment function used in high-frequency circuits such as millimeter waves and microwaves, and a transmitter, a receiver, a transceiver, and a radar apparatus including the matching circuit.

  In an advanced information society in recent years, in order to transmit a large amount of data at a high speed, application systems such as information communication devices using high frequencies from a microwave region of 1 to 30 GHz to a millimeter wave region of 30 to 300 GHz are used. It has been proposed. For example, a system using millimeter waves such as a radar device for measuring a distance between vehicles has been proposed.

In a system using such a high frequency, a high frequency circuit is used. In the technical field of high-frequency circuits, due to the high frequency of signals used, there is a problem that impedance mismatch occurs at the connection part of high-frequency components, and reflection of high-frequency signals is increased. For example, since the reactance of a bonding wire increases due to an inductance component as the frequency increases, MMIC (Microwave Monolithic Integrated)
When a high-frequency component such as a circuit) and a transmission line designed for 50Ω are connected by a bonding wire, there is a problem that impedance mismatch occurs and reflection of the high-frequency signal increases. In order to reduce reflection at the connection portion, a matching circuit for matching impedance is provided on the transmission line side, or the length of the bonding wire is adjusted.

  Even when such a matching circuit is provided, the size of the pattern constituting the matching circuit varies when manufacturing a high-frequency circuit, or the mounting position of the MMIC shifts and the inductance component of the bonding wire varies. As a result, the manufactured high-frequency circuit deviates from the design value, and impedance mismatch occurs. In addition, when manufacturing a high frequency circuit, if the length of the bonding wire deviates from the design value, the magnitude of the inductance component of the bonding wire deviates from the design value, causing impedance mismatch.

  In order to eliminate impedance mismatch caused by variations in manufacturing a high-frequency circuit, there is a matching circuit capable of adjusting impedance. In the prior art, an impedance adjustment step is provided to adjust the impedance of the matching circuit.

  In the first conventional technique, a plurality of impedance elements having different impedances are formed in advance, and a specific impedance element is connected to a matching circuit in the step of adjusting the impedance to achieve impedance matching (for example, patents). Reference 1).

  The second conventional technique uses a matching circuit having a spherical conductor formed so that a groove is formed in the direction in which the microstrip line extends and is movably engaged along the groove. In this technique, in the step of adjusting the impedance, the impedance is adjusted by adjusting the position of the spherical conductor along the groove, and the spherical conductor is fixed to the position where the impedance is matched by a non-conductive adhesive (for example, Patent Document 2).

  In the third conventional technique, a plurality of pads for impedance correction are formed in advance on a substrate, and a specific pad is connected to a matching circuit using solder in a process of adjusting impedance, thereby matching impedance. (For example, refer to Patent Document 3).

JP 2005-117111 A JP-A-9-252207 Japanese Patent Laid-Open No. 2005-5788

  In the first conventional technique, when adjusting the impedance, a step of connecting a specific impedance element to the matching circuit is required, so that the time required for adjusting the impedance becomes long. After connecting the impedance element, it is necessary to check whether impedance matching is achieved. If the impedance is not matched, it is necessary to repeat the process of connecting another impedance element to the matching circuit. The time required for the adjustment becomes longer, and the productivity is reduced, thereby increasing the manufacturing cost. In addition, since the impedance is adjusted by connecting the impedance element provided in advance to the matching circuit, the impedance cannot be adjusted by continuously changing the impedance, and the impedance cannot be adjusted with high accuracy. is there.

  In the second conventional technique, the position of the spherical conductor is moved to check whether impedance matching is achieved. If the impedance is not matched, another impedance element is connected to the matching circuit. Need to be repeated. Furthermore, when the adjustment of the position of the spherical conductor is completed, a process of fixing the position of the spherical conductor with a non-conductive adhesive is required, so that the time required for adjusting the impedance becomes long, and the productivity is reduced, Manufacturing cost increases.

  In the third conventional technique, when impedance adjustment is performed, a step of connecting a specific pad to the matching circuit with solder is required, so that the time required for impedance adjustment becomes long. After connecting the pads, it is necessary to check whether impedance matching is achieved. If the impedance is not matched, it is necessary to repeat the process of connecting another pad to the matching circuit. The time required increases and the productivity decreases, resulting in an increase in manufacturing cost. In addition, since impedance adjustment is performed by connecting a pad provided in advance to the matching circuit, there is a problem that impedance cannot be adjusted by continuously changing impedance and impedance adjustment cannot be performed with high accuracy. .

  Accordingly, an object of the present invention is to provide a matching circuit capable of easily and accurately adjusting the impedance, and a transmitter, a receiver, a transceiver, and a radar apparatus including the matching circuit.

The present invention includes a first transmission line,
A second transmission line;
Two first connection ends, one first connection end of the two first connection ends is connected to one end portion of the first transmission line, and the other end is connected to one end portion of the second transmission line A first phase shift circuit connected to the first connection end and capable of adjusting a phase of an electromagnetic wave passing therethrough by applying a voltage;
A third transmission line extending from the second transmission line and extending;
A second phase shift circuit having at least one second connection end, the second connection end being connected to one end of the third transmission line, and capable of adjusting a phase of an electromagnetic wave reflected by applying a voltage; Including
The first phase shift circuit includes:
A magic T having first to fourth input / output terminals, in which electromagnetic waves input from the first input / output terminals are distributed and output from the second and fourth input / output terminals;
A first variable capacitance element extending from the second input / output terminal and having an electric capacity that changes in accordance with an applied voltage is provided at an end opposite to the end connected to the second input / output terminal. 1 extension part,
A second variable capacitance element extending from the fourth input / output terminal and having an electric capacity that changes in accordance with an applied voltage is provided at an end opposite to the end to which the fourth input / output terminal is connected. And 2 extending parts,
The first transmission line is connected to the first input / output terminal;
The second transmission line is connected to the third input / output terminal;
The second phase shift circuit includes a third variable capacitance element whose electric capacity changes according to an applied voltage,
A first electrode connected to the first variable capacitance element for applying a voltage to the first variable capacitance element;
A second electrode connected to the second variable capacitance element for applying a voltage to the second variable capacitance element;
The matching circuit further includes a third electrode connected to the third variable capacitance element and configured to apply a voltage to the third variable capacitance element.

In the present invention, the first to third transmission lines and the first and second phase shift circuits are provided, and an electrically insulating substrate;
Formed on one surface in the thickness direction of the substrate, it has conductivity, and component placement unit electronic components are arranged,
On the one surface, a projecting portion that is formed to extend from the component placement portion toward a region where the first transmission line is provided, and has conductivity,
A back electrode formed on the other surface in the thickness direction of the substrate;
It further includes a connection part for electrically connecting the protruding part and the back electrode.

Furthermore, in the present invention, when an electronic component is electrically connected to the first transmission line and electromagnetic waves are input from the electronic component,
Electromagnetic wave inputted from said electronic component to the first transmission line, a reflection coefficient which is reflected by the connecting member for connecting the electronic component the first transmission line and gamma _1,
The magnitude of the transmission coefficient through which the electromagnetic wave input to the first transmission line passes to the branching portion where the third transmission line of the second transmission line branches is defined as | T _M1 |
The phase that changes when the electromagnetic wave input to the first transmission line travels to the branch portion is θ,
When the electromagnetic wave input to the second transmission line, the reflection coefficient is reflected by the branch section and gamma _2,
In a range where Γ ₂ is variable in the amount of phase shift of the first and second phase shift circuits,

The line lengths of the first and third transmission lines and the line length between the first phase shift circuit and the branching part of the second transmission line are selected so as to satisfy Expression (1). .
In the present invention, the first transmission line is connected to one end of the connecting body that connects the first transmission line and the electronic component, and the one end connected to the first connection end. And an impedance changing portion having an impedance different from that of the connection portion.
Furthermore, the present invention, when electromagnetic waves are input from the electronic component through the connection body,
The reflection coefficient of the electromagnetic wave input from the electronic component side to the connection block constituted by the connection body, the connector portion from the free end portion of the first transmission line to the impedance change portion, and the impedance change portion. Is Γ ₃ ,
Transmission through which electromagnetic waves input from the connection block to the remaining part of the first transmission line excluding the connector part and the impedance changing part pass to the branch part of the second transmission line where the third transmission line branches. When the magnitude of the coefficient is | T _M2 |
In the range where Γ ₃ is variable in the amount of phase shift of the first and second phase shift circuits,
| Γ ₃ | <| T _M2 | ² (2)
The line lengths of the first and third transmission lines, the impedance of the impedance changing unit, and the line length between the first phase shift circuit and the branching unit of the second transmission line are selected so as to satisfy Expression (2) It is characterized by that.
Furthermore, the present invention provides an electromagnetic wave input from an electronic component to a connection circuit constituted by the connection body and the matching circuit in a range in which a phase shift amount of the first and second phase shift circuits is variable. Between the first phase shift circuit and the branch part of the second transmission line, and the line lengths of the first and third transmission lines, the impedance of the impedance changing part, and the branching part. The track length is selected.

  Further, according to the present invention, the line length between the fourth input / output terminal and the second variable capacitance element is such that the line length between the second input / output terminal and the first variable capacitance element is (2n− 1) It is characterized by being selected differently by the length of λ / 4 (n represents a natural number).

Furthermore, the present invention provides a high frequency oscillator for generating a high frequency signal,
A transmission line connected to the high-frequency oscillator and transmitting a high-frequency signal from the high-frequency oscillator;
The matching circuit inserted into the transmission line;
A transmitter including an antenna connected to the transmission line and radiating a high frequency signal.

The present invention further includes an antenna for capturing a high-frequency signal;
A transmission line connected to the antenna and transmitting a high-frequency signal captured by the antenna;
The matching circuit inserted into the transmission line;
And a high-frequency detector connected to the transmission line for detecting a high-frequency signal transmitted to the transmission line.

Furthermore, the present invention provides a high frequency oscillator for generating a high frequency signal,
A fourth transmission line connected to the high-frequency oscillator for transmitting a high-frequency signal;
The first terminal is connected to the fourth transmission line, and a high-frequency signal applied to the first terminal is selectively applied to the second terminal or the third terminal. An output branching device;
A fifth transmission line connected to the second terminal and transmitting a high-frequency signal applied from the second terminal;
A fourth, fifth, and sixth terminal that outputs a high-frequency signal to the fourth terminal via the fifth transmission line, and outputs a high-frequency signal to the fifth terminal; A duplexer that outputs to the sixth terminal;
A sixth transmission line connected to the fifth terminal for transmitting a high-frequency signal output from the fifth terminal and transmitting a high-frequency signal to the fifth terminal;
An antenna connected to the sixth transmission line for radiating and capturing high-frequency signals;
A seventh transmission line connected to the third terminal and transmitting a high-frequency signal output from the third terminal;
An eighth transmission line connected to the sixth terminal and transmitting a high-frequency signal output from the sixth terminal;
A mixer that is connected to the seventh and eighth transmission lines, mixes high-frequency signals given from the seventh and eighth transmission lines, and outputs an intermediate frequency signal;
A transceiver including the matching circuit inserted into at least one of the fourth to eighth transmission lines.

Furthermore, the present invention provides a high frequency oscillator for generating a high frequency signal,
A fourth transmission line connected to the high-frequency oscillator for transmitting a high-frequency signal;
The first terminal is connected to the fourth transmission line, and a high-frequency signal applied to the first terminal is selectively applied to the second terminal or the third terminal. An output branching device;
A fifth transmission line connected to the second terminal and transmitting a high-frequency signal applied from the second terminal;
A transmitting antenna connected to the fifth transmission line and emitting a high-frequency signal;
A receiving antenna for capturing high-frequency signals;
A sixth transmission line connected to the receiving antenna and transmitting the captured high-frequency signal;
A seventh transmission line connected to the third terminal and transmitting a high-frequency signal output from the third terminal;
A mixer that is connected to the sixth and seventh transmission lines, mixes high-frequency signals given from the sixth and seventh transmission lines, and outputs an intermediate frequency signal;
A transceiver including the matching circuit inserted into at least one of the fourth to seventh transmission lines.

Furthermore, the present invention includes the transceiver,
A radar apparatus comprising: a distance detector that detects a distance from the transceiver to a detection target based on an intermediate frequency signal from the transceiver.

  According to the present invention, the second phase shift circuit is connected to the third transmission line. The transmission line constituted by the third transmission line and the second phase shift circuit functions as, for example, a stub. The impedance of the stub depends on the length in relation to the frequency of the electromagnetic wave passing through. Specifically, it depends on the electrical length of the stub. Since the amount of phase shift in the second phase shift circuit changes according to the applied voltage, the electrical length of the second phase shift circuit changes in accordance with the applied voltage. That is, when the phase of the electromagnetic wave passing through is considered as a unit, the length of the second phase shift circuit changes according to the applied voltage. Therefore, the electrical length of the transmission line constituted by the third transmission line and the second phase shift circuit also changes according to the applied voltage. As a result, even if the length of the manufactured third transmission line deviates from the design value, the voltage applied to the second phase shift circuit can be adjusted by the third transmission line and the second phase shift circuit. The impedance of the constructed transmission line can be adjusted as designed.

  A first phase shift circuit is provided between the first transmission line and the second transmission line. Similarly to the above, the transmission line constituted by the first transmission line, the first phase shift circuit, and the portion from the branching portion where the third transmission line of the second transmission line branches to the portion from the first phase shift circuit. The electrical length varies depending on the voltage applied to the first phase shift circuit. Therefore, even if the length of the manufactured first transmission line and the portion near the first phase shift circuit from the branch portion of the second transmission line are out of the design value, the voltage applied to the first phase shift circuit is Just by adjusting, the electrical length of the transmission line composed of the first transmission line, the first phase shift circuit, and the branching portion of the second transmission line closer to the first phase shift circuit is adjusted as designed. can do.

  In this way, by adjusting the voltages applied to the first and second phase shift circuits, the impedance of the entire matching circuit can be adjusted.

  Further, when the electronic component is connected to the first transmission line with a bonding wire, the impedance of the circuit including the bonding wire and the matching circuit depends on the mounting state of the electronic component, the length of the bonding wire, the arrangement of the bonding wire, etc. Depends on the bonding state of the bonding wire. Even after mounting electronic components and connecting them with bonding wires, the impedance of the matching circuit can be adjusted by adjusting the voltages applied to the first phase shift circuit and the second phase shift circuit, respectively. The impedance when the bonding wire and the matching circuit are viewed from the electronic component can be adjusted. For example, the power output from the electronic component is adjusted by adjusting the voltage applied to the first phase shift circuit and the second phase shift circuit so as to match the impedance between the circuit including the bonding wire and the matching circuit and the electronic component. Can be prevented from being reflected by the matching circuit. Here, since the electrical length from the bonding wire to the stub can be adjusted independently by the first phase shift circuit and the electrical length of the stub by the second phase shift circuit, no matter how the impedance of the bonding wire changes. The impedance can be matched by adjusting the voltages applied to the first phase shift circuit and the second phase shift circuit, respectively.

  Since the voltage changes continuously, the impedance of the matching circuit can be changed continuously by continuously changing the voltage applied to the first and second phase shift circuits. As a result, the impedance of the matching circuit can be adjusted with higher accuracy than the first and third conventional techniques that can change the impedance of the matching circuit discontinuously.

Further, unlike the first to third conventional techniques, a process such as a bonding process using a non-conductive adhesive and a soldering process are not required. This can be done easily, and the time required for adjusting the impedance can be shortened to reduce the cost.
Furthermore, in the magic T constituting the first phase shift circuit, the electromagnetic wave input from the first input / output terminal is distributed and output from the second and fourth input / output terminals, and is not output from the third input / output terminal. The phase of the electromagnetic wave output from the second input / output terminal is different from the phase of the electromagnetic wave output from the fourth input / output terminal by πrad. The first phase shift circuit is provided with a first extension portion extending from the second input / output terminal of the magic T and a second extension portion extending from the fourth input / output terminal. The first extension part and the second extension part respectively include first and second variable capacitance elements whose electric capacitance changes according to the applied voltage. The matching circuit further includes first and second electrodes for applying a voltage to the first and second variable capacitance elements, respectively. By adjusting the voltage applied to the first and second electrodes, the impedance of the first and second extending portions can be adjusted. By adjusting the impedances of the first and second extending portions, the phase of the electromagnetic wave reflected by the first and second extending portions can be changed. As a result, a phase shift circuit capable of adjusting the phase of the electromagnetic wave passing through the first phase shift circuit can be realized.
The second phase shift circuit includes a third variable capacitance element, and the matching circuit includes a third electrode for applying a voltage to the third variable capacitance element. By adjusting the voltage applied to the third electrode, the impedance of the third variable capacitance element can be adjusted, whereby the phase of the electromagnetic wave reflected by the second phase shift circuit can be adjusted.

  Further, according to the present invention, a conductive projecting portion is formed extending from a component placement portion formed on one surface in the thickness direction of the substrate toward a region where the first transmission line is provided. Is done. The protruding portion and the back electrode formed on the other surface in the thickness direction of the substrate are electrically connected via a connecting portion. If the protrusion is not provided, the transmission mode is coupled to the parallel plate mode formed by the component placement portion and the back electrode, resulting in leakage and an increase in transmission loss. By providing the protrusion, transmission in the parallel plate mode can be suppressed and transmission loss can be suppressed, so that a high-frequency signal can be transmitted efficiently.

Further, according to the present invention, the first and _second reflection coefficients Γ ₂ reflected by the branching portion satisfy the expression (1) in a range where the phase shift amounts of the first and second phase shift circuits are variable. The line length of the three transmission lines and the line length between the first phase shift circuit and the branching part of the second transmission line are selected. When the first to third transmission lines having such a line length are formed, a matching circuit satisfying the expression (1) is realized by adjusting voltages applied to the first phase shift circuit and the second phase shift circuit. Can do. When the matching circuit satisfies the formula (1), when the input / output terminal of the electronic component is connected to the first transmission line via the connection body, the high-frequency signal input from the input / output terminal is reflected by the connection body, The phase difference between the reflected wave reflected at the branching portion where the third transmission line branches from the second transmission line is equal and the reflected wave reflected at the connection body and the reflected wave reflected at the branching portion is πrad. It is overwhelmed with. That is, the reflected wave reflected at the connection portion between the connection body and the first transmission line and the reflected wave reflected at the branching portion overlap and cancel each other. As a result, reflection of high-frequency signals input from the electronic components can be reduced.

  Furthermore, according to this invention, an impedance change part is provided in a 1st transmission line. The impedance changing unit has an impedance different from that of the connection portion of the first transmission line to which one end of the connection body that connects the first transmission line and the electronic component is connected. That is, the impedance of the first transmission line changes in a path from the connection site to one end connected to the first phase shift circuit. In the impedance changing portion where the impedance changes, a part of the electromagnetic wave input from the electronic component through the connection body is reflected and a part thereof is transmitted. Part of the electromagnetic wave transmitted through the impedance changing portion is reflected by a branching portion of the second transmission line where the third transmission line branches, and returns to the impedance changing portion. The electromagnetic wave reflected by the impedance changing part and the electromagnetic wave reflected after passing through the impedance changing part are superimposed. The phase of the electromagnetic wave that is reflected through the impedance changing unit and then returned to the impedance changing unit is as follows: the first transmission line, the first phase shift circuit, and the portion of the second transmission line that is closer to the first phase shift circuit. Depends on two parameters: the electrical length of the transmission line constituted by the above and the impedance of the transmission line functioning as a stub by the third transmission line and the second phase shift circuit. These two parameters can be set to intended values by adjusting the voltages applied to the first and second phase shift circuits. As a result, the phase of the electromagnetic wave that is reflected after passing through the impedance changing portion and reflected back to the impedance changing portion can be adjusted, and the superimposed electromagnetic wave can be weakened. As a result, it is input from the electronic component to the matching circuit via the connection body. The reflection coefficient of the electromagnetic wave can be reduced. The role of the impedance changing unit will be described. Part of the electromagnetic wave input from the electronic component is reflected by the connection body and part of the electromagnetic wave is transmitted. A part of the electromagnetic waves transmitted through the connection body is reflected by the impedance changing portion and returns to the connection body. The electromagnetic wave reflected by the connecting body and the electromagnetic wave reflected by the impedance changing portion after passing through the connecting body are superimposed, and the phase of the electromagnetic wave reflected by the impedance changing portion after passing through the connecting body and returning to the connecting body is determined. If appropriately selected, the magnitude of the reflected wave can be reduced as compared with the case where there is no impedance change portion. Then, when matching, the magnitude of reflection from the transmission line constituted by the third transmission line functioning as a stub and the second phase shift circuit may be small, so that loss is small and broadband matching is possible.

  Furthermore, according to the present invention, in the range in which the phase shift amount of the first and second phase shift circuits is variable, the line lengths of the first and third transmission lines, the impedance of the impedance changing unit, so as to satisfy the formula (2), In addition, the line length between the first phase shift circuit and the branch part of the second transmission line is selected.

The characteristics of the matching circuit that satisfies Equation (2) will be described. The reflection coefficient viewed from the electronic component of the connection block constituted by the connection body, the connector portion from the free end portion of the first transmission line to the impedance change portion, and the impedance change portion is Γ ₃ , the reflection coefficient as viewed from the first phase-shift circuit and gamma _4, the transmission coefficient from the electronic component side of the connection block to the first phase-shift circuit side is T _43, the electronic component side from the first phase shift circuit side of the connecting block Let T _{34 be} the transmission coefficient to the transmission block, and let | T _M2 | be the magnitude of the transmission coefficient through which the electromagnetic wave input from the connection block passes to the branch portion where the third transmission line of the second transmission line branches. and .theta.1, the third transmission line and reflection coefficient as viewed from the first phase-shift circuit side of the stub of a second phase-shift circuit and gamma _2, the input terminal of the electronic component side of the connection block and port 1, second transmission The first phase shift circuit of the line and The contralateral output end as port 2, the reflection coefficient S ₁₁ when looking at a circuit constituted by the connecting member and the matching circuit from the electronic component obtains the conditions that can be zero. First, as a preparation for simplifying the equation, a relational expression established between Γ ₃ , Γ ₄ , Τ ₃₄ , and Τ ₄₃ is derived. In order to simplify the derivation of the equation, there is no loss term in the connection block, and the load condition of only the reactance component is obtained.

FIG. 17 is a diagram for explaining the reflection coefficient and the transmission coefficient of the connection block. In the case of a load with only the reactance component X, the following equations (3), (4), (5), and (11) are established for the reflection coefficient and the transmission coefficient. Since the view from the port 3 side is symmetrical to the view from the port 4 side, the following equation (3) is established.
Γ ₄ = Γ ₃ (3)

Further, according to the condition of the reversible circuit, the following equation (4) is established.
_{３４ 34} = Τ ₄₃ (4)

FIG. 18 is a Smith diagram. Since Γ ₃ can be regarded as a reflection coefficient when a load having an impedance R ₀ + jX is connected to a transmission line having a characteristic impedance R ₀ (see FIG. 17), a constant resistance circle having a normalized impedance 1 (see FIG. 18) on the Smith diagram. On the broken line). 2Γ ₃ and 2Γ ₃ −1 exist on the solid line and the alternate long and short dash line in FIG. 18, respectively, and are represented by the following expression (5).
| 2Γ ₃ -1 | = 1 (5)
According to the conditions of the lossless circuit, the following expressions (6) and (7) are established.
| Γ ₃ | ² + | Τ ₄₃ | ² = 1 (6)
Γ ₃ ^* Τ ₃₄ + Τ ₄₃ ^* Γ ₄ = 0 (7)

In Expression (7), Γ ₃ ^* and Τ ₄₃ ^* represent conjugate complex numbers of Γ ₃ and Τ ₄₃ , respectively. Substituting equation (3) into equation (7), the following equation (8) is established.
Γ ₃ ^* _{４３ 43} + _{４３ 43} ^* Γ ₃ = 0 (8)

When Expression (8) is expressed by dividing it into an absolute value and a phase, it is transformed into the following Expression (9).
| Γ ₃ || Τ ₄₃ | exp {j (∠Τ ₄₃ −∠Γ ₃ )}
+ | Γ ₃ || Τ ₄₃ | exp {j (∠Γ ₃ −∠Τ ₄₃ )} = 0 (9)

In Equation (9), _{４３ 43} and ∠Γ ₃ represent the phase angles of Τ ₂₁ and Γ ₃ , respectively, and the symbol “j” represents the imaginary unit. From Expression (9), the following Expression (10) is established.
_{４３ 43} = ∠Γ ₃ −π / 2 (10)

_On the Smith diagram, Γ ₃ exists on the broken line in FIG. 18, and from equations (6) and (10), vector Γ ₃ , vector _{２１ 21,} and vector 1 form a right triangle (see FIG. 18). The following equation (11) is established.
Γ ₃ + Τ ₄₃ = 1 (11)
Reflection coefficient S ₁₁ when viewed connecting circuit from the electronic component is expressed by the following equation (12).

  Since the following expression (13) is established for | x | <1, the expression (12) is transformed using the expressions (3), (4), (11), and (13). 14).

When this formula (14) is arranged with respect to Γ ₂ , the following formula (1) in which Γ ₁ , T _M1 , and θ are replaced with Γ ₃ , T _M2 , and θ1 in the formula (1) as a condition that the reflection coefficient S ₁₁ can be set to 0 ( 15) is derived.

  Taking the absolute values of both sides and using equation (5), the following equation (16) is established.

Unless | Γ ₂ | <1, electromagnetic waves cannot be transmitted from port 1 to port 2, so substituting this condition into equation (16) yields equation (2). The equation (2) is a necessary condition for the reflection coefficient S ₁₁ 0. In the present invention, since the matching circuit is produced so as to satisfy the formula (2) in the range in which the phase shift amount of the first and second phase shift circuits is variable, the connection constituted by the connection body and the matching circuit from the electronic components the reflection coefficient S ₁₁ when viewed circuit can be adjusted to zero. Accordingly, the electronic component can be used by being connected to the designed load, and the characteristics as designed of the electronic component can be obtained.

Further, according to the present invention, in the range in which the phase shift amount of the first and second phase shift circuits is variable, an electromagnetic wave input from an electronic component to the connection circuit constituted by the connection body and the matching circuit is connected to the connection circuit. as the reflection coefficient S _11, which is reflected by the circuit is 0, the line length of the first and third transmission lines, impedance of the impedance change portion, and the branch with the first phase shift circuit of the second transmission line section The track length between is selected. This makes it possible to adjust the reflection coefficient S ₁₁ to 0, the electromagnetic wave that is input from the electronic component can be reduced attenuation to be output to the port 2.

  Further, according to the present invention, the line length between the fourth input / output terminal and the second variable capacitance element is equal to the line length between the second input / output terminal and the first variable capacitance element (2n -1) It is selected differently by the length of λ / 4. Therefore, the phase of the electromagnetic wave input and reflected by the first extension portion is different from the phase of the electromagnetic wave input by the second extension portion and reflected by π rad. The phase of the electromagnetic wave reflected from the first extending portion and output from the third input / output terminal is equal to the phase of the electromagnetic wave reflected from the second extending portion and output from the third input / output terminal. And output from the third input / output terminal. When the first phase shift circuit designed in this way is used, even if the first and second variable capacitance elements that cause the phase change of the reflected wave are used, the electromagnetic wave input from the first input / output terminal is input to the third input circuit. A phase shift circuit that outputs from the output terminal can be realized.

Furthermore, according to the present invention, since a matching circuit is inserted in the transmission line, it is caused by the transmission line due to, for example, variations in the shape of bonding wires and bumps for connecting a high-frequency oscillator and variations in the width of the transmission line. Therefore, it is possible to realize a transmitter having a stable oscillation characteristic and a high transmission output since the insertion loss is suppressed to a small level.

  Furthermore, according to the present invention, the high frequency signal captured by the antenna is transmitted to the transmission line and detected by the high frequency detector. In the present invention, since a matching circuit is inserted in the transmission line, it is caused by the transmission line due to, for example, variations in the shape of bonding wires or bumps for connecting a high-frequency detector or variations in the width of the transmission line. Therefore, it is possible to realize a receiver having a high detection output because the phase shift can be individually adjusted to achieve matching and has a stable detection characteristic and an insertion loss can be suppressed to a small level.

  Further, according to the present invention, the high-frequency signal generated by the high-frequency oscillator is transmitted to the fourth transmission line and given to the first terminal of the branching device, and is given to the fifth transmission line from the second terminal of the branching device. It is given to the fourth terminal of the duplexer, given from the fifth terminal of the duplexer to the sixth transmission line, and radiated from the antenna. The high-frequency signal received by the antenna is given to the sixth transmission line, given to the fifth terminal of the duplexer, given from the sixth terminal of the duplexer to the eighth transmission line, and given to the mixer. . In addition, a high-frequency signal generated by the high-frequency oscillator is supplied as a local signal from the third terminal of the branching unit to the mixer via the seventh transmission line. The mixer mixes the high-frequency signal generated by the high-frequency oscillator and the high-frequency signal received by the antenna and outputs an intermediate frequency signal, thereby obtaining information contained in the received high-frequency signal.

  By inserting a matching circuit into at least one of the fourth to eighth transmission lines, the phase of a high-frequency signal that changes undesirably due to the transmission line due to, for example, variations in wiring width is adjusted. Thus, for example, a transmitter / receiver having a stable oscillation characteristic and a high transmission output can be realized because the insertion loss is suppressed to a small value. For example, it has a stable detection characteristic and a small insertion loss. Therefore, it is possible to realize a transmitter / receiver having a high detection output, to improve the reliability of a received high-frequency signal, and to improve the reliability of an intermediate frequency signal generated by, for example, a mixer .

  Further, according to the present invention, the high-frequency signal generated by the high-frequency oscillator is transmitted to the fourth transmission line and given to the first terminal of the branching device, and is given to the fifth transmission line from the second terminal of the branching device. Radiated from the antenna. The high-frequency signal received by the receiving antenna is given to the sixth transmission line and given to the mixer. In addition, a high-frequency signal generated by the high-frequency oscillator is supplied as a local signal from the third terminal of the branching unit to the mixer via the seventh transmission line. The mixer mixes the high-frequency signal generated by the high-frequency oscillator and the high-frequency signal received by the receiving antenna and outputs an intermediate frequency signal, thereby obtaining information contained in the received high-frequency signal.

  By inserting a matching circuit into at least one of the fourth to seventh transmission lines, the phase of a high-frequency signal that changes undesirably due to the transmission line due to, for example, variations in wiring width is adjusted. Thus, for example, a transmitter / receiver having a stable oscillation characteristic and a high transmission output can be realized because the insertion loss is suppressed to a small value. For example, it has a stable detection characteristic and a small insertion loss. Therefore, it is possible to realize a transmitter / receiver having a high detection output, to improve the reliability of a received high-frequency signal, and to improve the reliability of an intermediate frequency signal generated by, for example, a mixer .

Furthermore, according to the present invention, since the distance detector detects the distance from the transceiver to the detection object based on the intermediate frequency signal from the transceiver, the distance to the detection object can be accurately detected. Radar device.

  FIG. 1 is a perspective view showing a matching circuit 1 according to an embodiment of the present invention. The matching circuit 1 is electrically connected to the electronic component 2 via the bonding wire 3. The matching circuit 1 has an impedance adjusted so that an electromagnetic wave (hereinafter referred to as a high frequency signal) input from the electronic component 2 through the bonding wire 3 corresponding to the connection body is not reflected by the connection portion 4 with the bonding wire 3. The

  The matching circuit 1 is formed on the dielectric substrate 5 in the present embodiment, and includes first to third transmission lines 6, 7, 8, a first phase shift circuit 11, and a second phase shift circuit 12. Composed.

  The first phase shift circuit 11 has two first connection ends 13a and 13b. The first transmission line 6 is connected to one first connection end 13a, and is formed to extend from the first connection end 13a toward the component placement table 14 on which the electronic component 2 is disposed. The second transmission line 7 is connected to the other first connection end 13 b and extends from the first connection end 13 b in a direction away from the component placement table 14.

  The second phase shift circuit 12 has at least one second connection end, and has two second connection ends 15a and 15b in the present embodiment. The third transmission line 8 extends from the second transmission line 7 and is connected to one second connection end 15 a of the second phase shift circuit 12.

  The first phase shift circuit 11 can adjust the phase of the electromagnetic wave passing therethrough by applying a voltage. The second phase shift circuit 12 can adjust the phase of the reflected electromagnetic wave by applying a voltage.

The dielectric substrate 5 has electrical insulation, is formed of a dielectric, and is formed of glass, single crystal, ceramics, resin, or a composite thereof. As glass, quartz glass, crystallized glass, or the like is used. As the single crystal, Si, GaAs, quartz, sapphire, MgO, LaAlO ₃ or the like is used. As the ceramic, alumina, aluminum nitride, glass ceramic, forsterite, cordierite, or the like is used. As the resin, an epoxy resin, a fluorine-containing resin, a liquid crystal polymer, or the like is used. The dielectric substrate 5 is realized by a single layer substrate or a multilayer substrate. In the case of a multilayer substrate, the thickness in the thickness direction Z of each layer is selected to be 100 μm to 150 μm.

  On the surface of one side Z <b> 1 in the thickness direction Z of the dielectric substrate 5, a component placement portion 16 that has conductivity and forms the component placement base 14 is formed. The component placement portion 16 is in the form of a thin film plate in the present embodiment, the surface in the thickness direction Z is formed in a square shape, and is placed as close as possible to the first transmission line 6. Thus, by arranging the component placement portion 16 close to the first transmission line 6, the length of the bonding wire 3 can be made as short as possible. The length of the bonding wire 3 is designed to be about 300 μm to 700 μm, and the diameter is about 25 μm. The electronic component 2 is arranged in the component arrangement unit 16. The electronic component 2 is, for example, a high-frequency oscillator, a mixer, and an amplifier realized by MMIC. On the surface of the other Z2 of the dielectric substrate 5 in the thickness direction Z, a conductive back electrode 17 is formed over the entire surface. In addition to the component placement unit 16, the component placement table 14 further includes a plurality of connection portions 18 that electrically connect the component placement unit 16 and the back surface electrode 17, and the back surface electrode 17. The connecting portion 18 is formed by filling a via extending through the dielectric substrate 5 in the thickness direction Z with a conductive material. The plurality of connecting portions 18 are arranged so as to surround the electronic component 2 when viewed from one side Z1 in the thickness direction Z of the dielectric substrate 5 when the electronic component 2 is arranged.

  The first to third transmission lines 6, 7, and 8 are realized by a microstrip line, a coplanar line, a strip line, a dielectric waveguide, an image guide, a nonradiative dielectric line, and the like, respectively. In the present embodiment, the first to third transmission lines 6, 7, and 8 are realized by microstrip lines, each including a conductive line, a dielectric substrate 5, and a back electrode 17. . The conductive line is selected to have a line width of about 100 μm to 500 μm so that transmission loss of a high-frequency signal to be transmitted is reduced.

  The conductive lines constituting the component placement portion 16, the back electrode 17, the connection portion 18, and the first to third transmission lines 6, 7, and 8 are mainly Cu (copper), Ag (silver), and W (tungsten). , Mo (molybdenum), Al (aluminum), Ni (nickel), and Au (gold).

  In the present embodiment, the transmission line constituted by the third transmission line 8 and the second phase shift circuit 12 functions as a stub. The impedance of the stub varies depending on the line length. In other words, the impedance of the stub changes depending on the electrical length. Since the second phase shift circuit 12 can adjust the phase of the reflected electromagnetic wave by applying a voltage as described above, the third transmission line 8 and the second phase shift circuit can be adjusted by changing the phase of the reflected wave. 12 and the entire electrical length of the transmission line constituted by the circuit 12 changes. That is, changing the phase of the electromagnetic wave reflected by the second phase shift circuit 12 is equivalent to changing the line length of the transmission line that functions as a stub in a pseudo manner. In this way, by adjusting the voltage applied to the second phase shift circuit 12, the overall impedance of the transmission line constituted by the third transmission line 8 functioning as a stub and the second phase shift circuit 12 is adjusted. can do.

  Similarly, the third transmission line 8 of the first transmission line 6, the first phase shift circuit 11, and the second transmission line 7 is adjusted by adjusting the voltage applied to the first phase shift circuit 11. The impedance is adjusted by adjusting the overall pseudo length of the transmission line constituted by the branching portion 19 and the portion (hereinafter referred to as one of the second transmission lines 7) 7a close to the first phase shift circuit 11 can do.

  The line length of the first and third transmission lines 6 and 8 and the line length of one of the second transmission lines 7 are within the range in which the phase shift amounts of the first and second phase shift circuits 11 and 12 are variable. It is chosen to satisfy (1).

In Equation (1), Γ ₁ is a high-frequency signal input from the electronic component 2 to the first transmission line 6 is reflected by the bonding wire 3 corresponding to a connection body that connects the electronic component 2 and the first transmission line 6. | T _M1 | represents the transmission coefficient through which the electromagnetic wave input to the first transmission line 6 passes to the branching portion 19, and θ represents the high-frequency signal input to the first transmission line 6. , Γ ₂ represents a reflection coefficient at which the electromagnetic wave input to the second transmission line 7 is reflected by the branch portion 19.

  FIG. 2 is a plan view showing the matching circuit 1. The first phase shift circuit 11 includes a rat race 26 having first to fourth input / output terminals 21 to 24, a first extending portion 27 formed extending from the second input / output terminal 22, and a fourth And a second extending portion 28 formed extending from the input / output terminal 24.

  The rat race 26 is formed so that the length of one round is 3 (2n-1) λ / 2, where λ is the wavelength of the electromagnetic wave input from the first input / output terminal 21, that is, the high-frequency signal output from the electronic component 2. The ring-shaped transmission line 31 and the fifth input / output terminal 25 are further included (n represents a natural number). The ring-shaped transmission line 31 is realized by a microstrip line, is formed in a substantially circular shape including a circular shape and an elliptical shape, and is formed in a circular shape in the present embodiment. The first to fifth input / output terminals 21 to 25 are provided in this order in one direction in the circumferential direction in the ring-shaped transmission line 31. Line length between the first and second input / output terminals 21, 22, line length between the second and third input / output terminals 22, 23, line length between the third and fourth input / output terminals 23, 24 The line length between the fourth and fifth input / output terminals 24 and 25 is selected as (2n-1) λ / 4, and the line length between the fourth and first input / output terminals 24 and 21 is 3 (2n-1) λ / 4. There are two line lengths between the terminals. If one line length is x, the other line length is 3λ / 2−x. However, the line length between the terminals is shorter. The line length is explained. In the present embodiment, the symbol “n” is selected as the numerical value “1”.

  In the rat race 26 having such a configuration, when a high frequency signal having a wavelength λ is input from the first input / output terminal 21, the input high frequency signal is distributed to the second input / output terminal 22 and the fourth input / output terminal 24. And is not output from the third and fifth input / output terminals 23 and 25.

  The end of the first transmission line 6 opposite to the end connected to the bonding wire 3 is connected to a first input / output terminal 21 corresponding to one first connection end 13a. The second transmission line 7 is connected to a third input / output terminal 23 corresponding to the other first connection end 13b.

  The first extending portion 27 includes a transmission line 27 a extending from the second input / output terminal 22 and a first variable capacitance element provided at an end opposite to the end where the transmission line 27 a is connected to the rat race 26. 27b. The second extending portion 28 includes a transmission line 28 a extending from the fourth input / output terminal 24, and a second variable capacitance element provided at an end opposite to the end where the transmission line 28 a is connected to the rat race 26. 28b. In the present embodiment, the transmission lines 27a and 28a constituting the first and second extending portions 27 and 28 are realized by microstrip lines, and the first and second variable capacitance elements 27b and 28b apply. It consists of an element whose capacitance changes according to the voltage.

In the present embodiment, the second phase shift circuit 12 includes a third variable capacitance element 32 whose electric capacity changes according to an applied voltage. The first to third variable capacitance elements 27b, 28b, and 32 are semiconductor elements such as varactor diodes, ferroelectric elements, piezoelectric elements, and MEMS (
This is realized by a voltage controlled variable capacitor including a microelectromechanical system) element, and in the present embodiment, realized by a varactor diode.

  The matching circuit 1 includes a first electrode 33 for applying a voltage to the first variable capacitor 27b, a second electrode 34 for applying an electrode to the second variable capacitor 28b, and a third variable capacitor 32. A third electrode 35 for applying a voltage, and a reference electrode 36 corresponding to a fourth electrode for applying a reference potential for determining a voltage applied to the first to third variable capacitance elements 27b, 28b, 32. In addition. The potential difference between the reference electrode 36 and the first to third electrodes 33, 34, and 35 becomes a voltage applied to the first to third variable capacitance elements 27b, 28b, and 32.

  The first to third electrodes 33, 34, and 35 extend from the first variable capacitance element 27 b, the second variable capacitance element 28 b, and the second connection end 15 b of the second phase shift circuit 12, respectively. An electrode pad is formed. Between the electrode pad of the first electrode 33 and the first variable capacitance element 27b, between the electrode pad of the second electrode 34 and the second variable capacitance element 28b, and between the electrode pad of the third electrode 35 and the second phase shift. A fan-shaped radial stub is provided between the circuit 12 (third variable capacitance element 32) and the second connection end 15b. This radial stub is provided to prevent leakage of a high-frequency signal and to obtain a large amount of phase change by making the reactance components of the first to third variable capacitance elements 27b, 28b, 32 close to 0, respectively. Instead of providing radial stubs for the first to third electrodes 33, 34, and 35, low-pass filters may be provided.

  The reference electrode 36 extends from the fifth input / output terminal 25, and an electrode pad is formed at the end. A low-pass filter 38 that blocks high-frequency signals is formed between the electrode pad and the fifth input / output terminal 25. The low-pass filter 38 is formed so as to block at least a high-frequency signal output from the electronic component 2. The low-pass filter 38 is configured to include two square portions formed in a quadrilateral shape with the line width widened in the transmission line. Instead of providing a low-pass filter on the reference electrode 36, a radial stub may be provided. The reference electrode 36 is formed to penetrate the dielectric substrate 5 in the thickness direction Z, and includes a through electrode 37 that electrically connects the electrode pad and the back electrode 17. Therefore, the reference voltage of the voltage applied to the first to third variable capacitance elements 27b, 28b, 32 is set to the same value as the voltage of the back electrode 17.

  By providing the low-pass filter 38, it is possible to reliably prevent a high frequency signal from being input from the fifth input / output terminal 25 to the reference electrode 36. Thereby, transmission loss of the high frequency signal can be suppressed. Further, since the reference electrode 36 is connected to the fifth input / output terminal 25 from which the high frequency signal input from the first input / output terminal 21 is not output, the provision of the reference electrode 36 does not affect the transmission of the high frequency signal. . Accordingly, the reference electrode 36 for determining the reference voltage can be provided without affecting the impedance of the first phase shift circuit 11.

  The first extending portion 27 is formed to function as a stub that reflects a high-frequency signal that is input from the second input / output terminal 22 through the first input / output terminal 21. The second extending portion 28 is formed so as to function as a stub that reflects a high-frequency signal input from the fourth input / output terminal 24 through the first input / output terminal 21. The impedance of the first extension part 27 depends on the line length of the transmission line 27a of the first extension part 27 and the electric capacity of the first variable capacitance element 27b. Since the electric capacity of the first variable capacitance element 27b is variable according to the applied voltage, it can be adjusted by adjusting the voltage applied to the first electrode 33. Similarly, the impedance of the second extending portion 28 can be adjusted by adjusting the voltage applied to the second electrode 34.

  The transmission line 28a between the fourth input / output terminal 24 and the second variable capacitance element 28b has a transmission line length of the transmission line 27a between the second input / output terminal 22 and the first variable capacitance element 27b. (2n-1) λ / 4 is selected to be different. In the present embodiment, the transmission line 28a between the fourth input / output terminal 24 and the second variable capacitance element 28b has a transmission line length between the second input / output terminal 22 and the first variable capacitance element 27b. It is selected longer than the line length of 27a by the length of λ / 4. As described above, without being reflected by the first extending portion 27 or the second extending portion 28, the ring-shaped transmission line 31 that is input from the first input / output terminal 21 is divided into one and the other in the circumferential direction. The phases of the high-frequency signals that are transmitted and merged at the third input / output terminal 23 are different from each other by π rad, but the transmission line 27a of the first extension part 27 and the transmission line 28a of the second extension part 28 are Since it is formed in such a line length, it is inputted from the first input / output terminal 21 and branches, and is reflected by the first extension part 27 and the second extension part 28 and merges at the third input / output terminal 23. The phases of the high frequency signals are equal. As a result, the high-frequency signal input from the first input / output terminal 21 is output from the third input / output terminal 23.

Since the impedance of the first and second extending portions 27 and 28 changes according to the voltage applied to the first and second electrodes 33 and 34, the voltage applied to the first and second electrodes 33 and 34 is adjusted. By doing so, the phase of the high-frequency signal passing through the first phase shift circuit 11 can be adjusted. Thereby, a phase shift circuit capable of adjusting the phase of the high-frequency signal passing through the first phase shift circuit 11 can be realized.

  Further, since the electric capacity of the third variable capacitance element 32 changes according to the voltage applied to the third electrode 35, the high frequency reflected by the second phase shift circuit 12 by adjusting the voltage applied to the third electrode 35. A phase shift circuit capable of adjusting the phase of the signal can be realized.

  Of the second transmission line 7, at least one of a part (hereinafter referred to as the other of the second transmission lines 7) 7 b that is farther from the first phase shift circuit 11 than the branching part 19 and the first transmission line 6. Has a DC blocking capacitor. In the present embodiment, a DC blocking capacitor 39 is provided on both the first transmission line 6 and the other 7 b of the second transmission line 7. The DC blocking capacitor 39 functions as a filter that allows a high frequency signal output from the electronic component 2 to pass therethrough and blocks the DC signal. The DC blocking capacitor 39 is realized by an interdigital capacitor, a ceramic capacitor, a single plate capacitor, a thin film capacitor, or the like, in which a thin insulating film formed of silicon nitride, silicon oxide, or the like is sandwiched between transmission lines. In the present embodiment, the DC blocking capacitor 39 is realized by an interdigital capacitor.

  In the present embodiment, since the characteristic impedance of the line to which the electronic component 2 is connected is designed to be different from the characteristic impedance of the first input / output terminal 21 of the rat race 26, the first transmission line 6 is And an impedance transformer. For example, in this embodiment, the characteristic impedance of the line to which the electronic component 2 is connected is designed to be 50Ω, and the characteristic impedance of the first input / output terminal 21 of the rat race 26 is designed to be 35Ω. The line width of the first transmission line 6 is selected to be 50Ω in the vicinity of the first input / output terminal 21, and is selected to be 35Ω at the portion where the bonding wire 3 of the first transmission line 6 is connected. . The first transmission line 6 is provided with a λ / 4 impedance transformer at the center for matching and connecting a 35Ω transmission line and a 50Ω transmission line.

FIG. 3 is a graph showing S ₁₁ (reflection coefficient viewed from the electronic component 2) of the S parameters. In FIG. 3, the center represents the numerical value “0”, the horizontal axis represents the real axis, and the vertical axis represents the imaginary axis. S ₁₁ is a ratio of reflection of the high frequency signal at the connecting portion between the electronic component 2 and the bonding wire 3 when a high frequency signal is input from the electronic component 2 to the matching circuit 1 through the bonding wire 3, that is, a reflection coefficient. To express. If taken impedance matching, since S ₁₁ is small, the measurement approach the center of the graph. In Figure 3 illustrates the measured values of S ₁₁ in the frequency range including the frequency range of the high frequency signal inputted to the matching circuit 1.

FIG. 3A is a graph showing S ₁₁ at 76.5 GHz when the matching circuit is not used. FIG. 3B is a graph showing the first phase shift circuit 11 and the S ₁₁ using the matching circuit of the present embodiment. it is a graph showing the _{S 11} in 76.5GHz when changing the voltage applied to the second phase-shift circuit 12. 3 (2) is closer to the center of the measurement data than FIG. 3 (1), so that impedance matching is achieved and reflection is reduced compared to the case where no matching circuit is provided. To express. Specifically, it was shown that the absolute value of the reflection coefficient viewed from the electronic component 2 was improved from −7 dB to −12 dB by using the matching circuit of the present embodiment. Furthermore, it can be seen that S ₁₁ is changed by changing the voltages of the first phase shift circuit 11 and the second phase shift circuit 12. Specifically, by changing the voltage applied to the first phase shift circuit 11 and the second phase shift circuit 12, the absolute value of the reflection coefficient viewed from the electronic component 2 is −9.5 dB to −13.5 dB, It was shown that the phase changed from -148 ° to 167 °. As a result, when the high frequency circuit is manufactured, the size of the pattern constituting the matching circuit varies, or the mounting position of the MMIC shifts and the inductance component of the bonding wire varies, thereby designing the manufactured high frequency circuit. Even when the value deviates from the value, it can be seen that adjustment is possible by changing the voltage applied to the first phase shift circuit 11 and the second phase shift circuit 12.

  According to the matching circuit 1 of the present embodiment described above, the first transmission line 6, the first phase shift circuit 11, and the second transmission line are adjusted by adjusting the voltage applied to the first phase shift circuit 11. 7 can be adjusted by adjusting the pseudo length of the entire transmission line constituted by the one 7a. In addition, by adjusting the voltage applied to the second phase shift circuit 12, the overall impedance of the transmission line constituted by the third transmission line 8 functioning as a stub and the second phase shift circuit 12 can be adjusted. it can. That is, the impedance of the entire matching circuit 1 can be adjusted by adjusting the voltages applied to the first phase shift circuit 11 and the second phase shift circuit 12 respectively.

  Further, when the electronic component 2 is connected to the first transmission line 6 with the bonding wire 3, the impedance of the circuit including the bonding wire 3 and the matching circuit 1 depends on the mounting state of the electronic component 2 and the length of the bonding wire 3. And the connection state of the bonding wire 3 such as the arrangement of the bonding wire 3. Even after the electronic component 2 is mounted and connected by the bonding wire 3 as described above, the impedance of the matching circuit 1 is adjusted by adjusting the voltage applied to the first phase shift circuit 11 and the second phase shift circuit 12 respectively. Therefore, the impedance when the bonding wire 3 and the matching circuit 1 are viewed from the electronic component 2 can be adjusted. For example, by adjusting the voltage applied to the first phase shift circuit 11 and the second phase shift circuit 12 so as to match the impedance between the circuit including the bonding wire 3 and the matching circuit 1 and the electronic component 2, the electronic component It is possible to prevent the power output from 2 from being reflected by the matching circuit 1.

  Further, since the voltage changes continuously, the impedance of the matching circuit 1 can be changed continuously by continuously changing the voltage applied to the first and second phase shift circuits 11 and 12. As a result, the impedance of the matching circuit 1 can be adjusted with higher accuracy than the first and third conventional techniques that can discontinuously change the impedance of the matching circuit 1. Here, since the electrical length from the bonding wire 3 to the stub can be independently adjusted by the first phase shift circuit 11 and the electrical length of the stub can be independently adjusted by the second phase shift circuit 12, how the impedance of the bonding wire 3 is adjusted. Even if it changes, the impedance can be matched by adjusting the voltages applied to the first phase shift circuit 11 and the second phase shift circuit 12, respectively.

  In addition, the matching circuit 1 of the present embodiment has the line lengths of the first and third transmission lines 6 and 8 and the line length of one of the second transmission lines 7 a of the first and second phase shift circuits 11 and 12. In a range where the amount of phase shift is variable, it is selected so as to satisfy formula (1). When the matching circuit 1 satisfies the formula (1), when the input / output terminal of the electronic component 2 is connected to the first transmission line 6 via the bonding wire 3, the high-frequency signal input from the input / output terminal is reflected by the bonding wire 3. The reflection coefficient reflected by the branch portion where the second transmission line 7 branches from the third transmission line 8 are equal, and the reflected wave reflected by the bonding wire 3 and the reflection reflected by the branch portion Waves are superimposed with a phase difference of π rad. That is, the reflected wave reflected by the bonding wire 3 and the reflected wave reflected by the branching portion are overlapped and canceled. Thereby, reflection of the high frequency signal inputted from the electronic component 2 can be reduced.

  Since matching circuit 1 of the present embodiment includes DC blocking capacitor 39, low-frequency and DC signals passing through matching circuit 1 can be removed, and first and second phase shift circuits 11 and 12 are connected to each other. The applied voltage can be prevented from being applied to the electronic component 2 connected to the matching circuit 1. As a result, it is possible to avoid destruction of the electronic component 2 and unstable characteristics caused by applying a voltage to the electronic component 2.

  Furthermore, according to the matching circuit 1 of the present embodiment, the line length of the transmission line 28a between the fourth input / output terminal 24 and the second variable capacitance element 28b is the same as that of the second input / output terminal 22 and the first variable capacitance element. It is selected to be longer by the length of λ / 4 than the line length of the transmission line 27a with respect to 27b. Therefore, the amount of phase change between the high-frequency signal input to the first extending portion 27 and reflected and the high-frequency signal input to the second extending portion 28 and reflected are different by π rad. The phase of the electromagnetic wave reflected from the first extending portion 27 and output from the third input / output terminal 23 and the electromagnetic wave reflected from the second extending portion 28 and output from the third input / output terminal 23 are They are equal, overlapped, and output from the third input / output terminal 23. When the first phase shift circuit 11 designed in this way is used, even if the first and second variable capacitance elements 27b and 28b that cause the phase change of the reflected wave are used, the first input / output terminal 21 is used. A phase shift circuit in which electromagnetic waves are output from the third input / output terminal 23 can be realized.

  In the present embodiment, the bonding wire 3 has been described as a connection body that connects the matching circuit 1 and the electronic component 2, but the connection body is not limited to the bonding wire 3. For example, a bump may be formed as a connection body on the surface of the electronic component 2 facing the dielectric substrate 5, and the bump and the first transmission line 6 of the matching circuit 1 may be connected by surface mounting such as flip chip. . Even in this case, as described above, the characteristic impedance of the matching circuit 1 can be easily adjusted after the electronic component 2 is surface-mounted, and the reflection of the high-frequency signal generated between the electronic component 2 and the matching circuit 1 is reduced. be able to.

  FIG. 4 is a plan view showing a matching circuit 40 according to another embodiment of the present invention. The matching circuit 40 of the present embodiment is different from the configuration of the matching circuit 1 of the above-described embodiment only in the connection portion of the first transmission line 6 and the second transmission line 7 to the rat race 26, and thus the corresponding configuration. The same reference numerals are assigned to the above, and overlapping description is omitted, and only the connection portion of the first transmission line 6 and the second transmission line 7 to the rat race 26 will be described.

  The first and second transmission lines 6 and 7 of the present embodiment have the same configuration as the first and second transmission lines 6 and 7 of the matching circuit 1 of the above-described embodiment. The end of the first transmission line 6 opposite to the end connected to the bonding wire 3 is connected to the third input / output terminal 23. One side 7 a of the second transmission line 7 is connected to the first input / output terminal 21.

  The matching circuit 40 of the present embodiment includes a first extending portion 27 and a second extending portion 28 as in the matching circuit 1 of the above-described embodiment. The high-frequency signal output from the electronic component 2 is input to the third input / output terminal 23 and branches. When the high-frequency signal branched at the third input / output terminal 23 is transmitted to the one and the other in the circumferential direction through the ring-shaped transmission line 31 and merged at the first input / output terminal 21, the phases are different from each other by πrad. As described above, when the light is reflected by the first extending portion 27 and the second extending portion 28 and merges at the first input / output terminal 21, the phases are equal to each other. As a result, a high frequency signal input from the third input / output terminal 23 is output from the first input / output terminal 21.

  As described above, since the impedance of the matching circuit 40 can be easily adjusted by adjusting the voltage applied to the first phase shift circuit 11 and the second phase shift circuit 12, the power output from the electronic component 2 can be adjusted. Can be prevented from being reflected by the bonding wire 3 and the matching circuit 40.

FIG. 5 is a plan view showing a matching circuit 41 according to still another embodiment of the present invention. The matching circuit 41 of the present embodiment includes a component arrangement table 42 on which the electronic component 2 is arranged, and performs impedance matching with the electronic component 2 together with the component arrangement table 42. The matching circuit 41 of the present embodiment is different only in the configuration of the matching circuit 1 of the above-described embodiment and the component placement table 42, and therefore, the corresponding configuration is denoted by the same reference numeral and redundant description is omitted. Only the component placement table 42 will be described.

  The component placement table 42 has a dielectric substrate 5, a component placement portion 43 on which the electronic component 2 is placed, a conductive projection 44, a back electrode 17, a projection 44, and a back surface. And a first connecting portion 45 that electrically connects the electrode 17.

  The component placement portion 43 is formed in a thin rectangular shape, and is perpendicular to the extending direction Y of the first transmission line 6 and the thickness direction Z of the dielectric substrate 5 on one surface in the thickness direction Z of the dielectric substrate 5. Extending in the direction X. The component placement unit 43 is placed so that the center in the longitudinal direction X overlaps the extension line of the first transmission line 6 and is placed as close to the first transmission line 6 as possible. The width of the component placement portion 43 in the longitudinal direction X is such that when viewed from one side Z1 in the thickness direction Z of the dielectric substrate 5, the electronic component 2 does not overlap one side and the other in the longitudinal direction X when the electronic component 2 is placed. It is selected wider than the electronic component 2 so that the regions 47a and 47b are formed. In the electronic component 2, when viewed from one side Z <b> 1 in the thickness direction Z of the dielectric substrate 5, the input / output terminal 46 overlaps the extension line of the first transmission line 6 and is close to the first transmission line 6 in the component placement portion 43. It is arranged near the end. Thereby, the length of the bonding wire 3 can be made as short as possible.

  The component placement table 42 includes one or a plurality of second connection portions 48 that electrically connect the component placement portion 43 and the back surface electrode 17. The second connection portion 48 is formed by filling a via having a conductive property between the back electrode 17 and the component placement portion 43 so as to penetrate the dielectric substrate 5. In the present embodiment, the second connection portion 48 is arranged at a predetermined interval in the entire component arrangement portion 43 as viewed from one side in the thickness direction Z of the dielectric substrate 5. Thus, the component placement unit 43 is set to the same potential as the back electrode 17.

  The protruding portion 44 is formed on one surface in the thickness direction Z of the dielectric substrate 5 so as to extend from the component placement portion 43 toward a region where the first transmission line 6 is provided. In the present embodiment, the projecting portions 44 are arranged at intervals in the longitudinal direction X of the component arranging portion 43 that is perpendicular to the extending direction Y of the first transmission line 6 and the thickness direction Z of the dielectric substrate 5, respectively. It consists of a plurality of projecting portions 49 extending in the short direction Y of the component placement portion 43. In the present embodiment, the projecting portion 44 is configured to include four projecting portions 49. The four protruding portions 49 are arranged symmetrically with respect to the extension line of the first transmission line 6 when viewed from one side Z <b> 1 in the thickness direction Z of the dielectric substrate 5. Of the four projecting portions 49, a pair of projecting portions 49 arranged on the inner side are referred to as first and second projecting portions 49a and 49b, respectively, and a pair of projecting portions 49 arranged on the outer side are respectively described. It describes as the 3rd and 4th protrusion part 49c, 49d.

  The first connection portion 45 electrically connects each protruding portion 49 and the back surface electrode 17. The first connection portion 45 is formed by filling a through hole penetrating the dielectric substrate 5 between each protruding portion 49 and the back electrode 17 with a conductive material. Thereby, each protruding portion 49 is set to the same potential as the back surface electrode 17.

  The line width W1 of the first transmission line 6 and the distance W3 between the first and second projecting portions 49a, 49b and the first transmission line 6 are the signal lines, the first and second projecting lines. When viewed as a coplanar line with the portions 49a and 49b as the ground, the characteristic impedance viewed from the first transmission line 6 side is set to 50Ω including the bonding wire 3. The three bonding wires 3 are set to be parallel. Accordingly, the line width W1 of the first transmission line 6 and the distance W3 between the first and second projecting portions 49a and 49b and the first transmission line 6 depend on the output pad distance of the electronic component 2.

The width W2 in the extending direction Y of the first and second projecting portions 49a and 49b is such that the impedance including the bonding wire 3 and the matching circuit 41 matches the electronic component 2, and the input / output terminal 46 of the electronic component 2 Set to reduce reflection. The width W2 in the extending direction Y of the first and second projecting portions 49a and 49b is such that at least the end portion of the first transmission line 6 near the component placement portion 43 is sandwiched between the first and second projecting portions 49a and 49b. To be elected. Preferably, the width W2 in the extending direction Y of the first and second projecting portions 49a and 49b is selected to be slightly larger than λ / 4 of the wavelength λ of the high-frequency signal output from the electronic component 2. For example, when the wavelength of the high-frequency signal is 1500 μm, the width W2 in the extending direction Y of the first and second projecting portions 49a and 49b is selected to be 410 μm.

  Further, the interval W4 between the first protruding portion 49a and the third protruding portion 49c, and the interval W4 between the second protruding portion 49b and the fourth protruding portion 49d are impedances including the bonding wire 3 and the matching circuit 41. It is set so as to match with the electronic component 2 and reduce reflection at the input / output terminal 46 of the electronic component 2, and is selected to be, for example, about λ / 8 or more and less than λ / 4. In the present embodiment, in the direction X in which the protruding portions 49 are arranged, the pair of third and fourth protruding portions 49c and 49d are provided outside the first and second protruding portions 49a and 49b. The protruding portions 49 may be provided, and the interval between the protruding portions 49 in that case is also set in the same manner as W4.

  The width W5 in the extending direction Y of the third and fourth projecting portions 49c, 49d is selected to be about λ / 4, for example. For example, when the wavelength of the high frequency signal is 1500 μm, the width W5 in the extending direction Y of the third and fourth projecting portions 49c and 49d is selected to be 380 μm.

  According to the matching circuit 41 of the present embodiment described above, if the protrusion 44 is not provided, the transmission mode is coupled to the parallel plate mode formed by the component placement portion 43 and the back electrode 17. Leakage and transmission loss increase. By providing the protrusion 44, transmission in the parallel plate mode can be suppressed and transmission loss can be suppressed, so that a high-frequency signal can be transmitted efficiently. Here, the component placement portion 43 is placed on one surface in the thickness direction Z of the dielectric substrate 5, but may be inside the dielectric substrate 5.

  FIG. 6 is a plan view showing a matching circuit 51 according to still another embodiment of the present invention. The matching circuit 51 of this embodiment is different from the configuration of the matching circuits 1, 40, and 41 of the above-described embodiments only in the first phase shift circuit 52. Therefore, the description which overlaps is abbreviate | omitted and only a different part is demonstrated.

  The first phase shift circuit 52 includes a branch line 57 having first to fourth input / output terminals 53 to 56, a first extending portion 27 formed extending from the second input / output terminal 54, and a third And a second extending portion 28 formed extending from the input / output terminal 55.

  The branch line 57 is a square formed with a length of one circle of (2n−1) λ, where λ is the wavelength of the electromagnetic wave input from the first input / output terminal 53, that is, the high-frequency signal output from the electronic component 2. A transmission line 58 is included. The rectangular transmission line 58 is realized by a microstrip line, and is formed in a square shape or a substantially rounded square shape. In the present embodiment, the square transmission line 58 is formed in a square shape. The first to fourth input / output terminals 53 to 56 are provided in this order in one direction in the circumferential direction in the rectangular transmission line 58. The line length between the output terminals adjacent to each other is selected as (2n−1) λ / 4. In the present embodiment, the symbol “n” is selected as the numerical value “1”. The characteristic impedance of the transmission line between the first input / output terminal 53 and the second input / output terminal 54 is Z1, and the characteristic impedance of the transmission line between the second input / output terminal 54 and the third input / output terminal 55. Is Z2, the characteristic impedance of the transmission line between the third input / output terminal 55 and the fourth input / output terminal 56 is Z3, and transmission between the fourth input / output terminal 56 and the first input / output terminal 53 is performed. Assuming that the characteristic impedance of the line is Z4, the rectangular transmission line 58 is formed so that Z1, Z2, Z3, and Z4 each satisfy the relationship of the following equation (17).

  In the branch line 57 having such a configuration, when a high frequency signal having a wavelength λ is input from the first input / output terminal 53, the input high frequency signal is distributed to the second input / output terminal 54 and the third input / output terminal 55. And is not output from the fourth input / output terminal 56. Further, the phase of the high-frequency signal output from the second input / output terminal 54 is different from the phase of the high-frequency signal output from the third input / output terminal 55 by π / 2 rad.

  The end of the first transmission line 6 opposite to the end connected to the bonding wire 3 is connected to a first input / output terminal 53 corresponding to one first connection end 13a. The second transmission line 7 is connected to a fourth input / output terminal 56 corresponding to the other first connection end 13b.

  The first extending portion 27 includes a transmission line 27 a extending from the second input / output terminal 54, and a first variable capacitance element provided at an end opposite to the end where the transmission line 27 a is connected to the branch line 57. 27b. The second extending portion 28 includes a transmission line 28 a extending from the third input / output terminal 55, and a second variable capacitance element provided at the end opposite to the end where the transmission line 28 a is connected to the branch line 57. 28b.

  The first to third electrodes 33, 34, and 35 have the same configuration as that of the first to third electrodes 33, 34, and 35 described above, and thus redundant description is omitted. The reference electrode 36 for determining the reference voltage of the voltage applied to the first to third variable capacitance elements 27b, 28b, 32 is only different in connection point from the reference electrode 36 of the above-described embodiment. Only will be described. The reference electrode 36 extends in a branched manner from between the DC blocking capacitor 39 and the first input / output terminal 53 of the first transmission line 6.

  The first extending portion 27 is formed so as to function as a stub that reflects a high-frequency signal input from the second input / output terminal 54 through the first input / output terminal 53. The second extending portion 28 is formed so as to function as a stub that reflects a high-frequency signal input from the third input / output terminal 55 through the first input / output terminal 53. The first and second extending portions 27 and 28 are different from the first and second extending portions 27 and 28 of the above-described embodiment only in the line lengths of the transmission lines 27a and 28a. To do.

  The line length of the transmission line 27a between the second input / output terminal 54 and the first variable capacitance element 27b is substantially the same as the line length of the transmission line 28a between the third input / output terminal 55 and the second variable capacitance element 28b. They are chosen to be equal or different by a length of nλ / 2. Since the transmission line 27a of the first extension part 27 and the transmission line 28a of the second extension part 28 are formed to have the above-described line length, they are input from the first input / output terminal 53, and the first extension part 27 The phases of the high frequency signals reflected by the existing portion 27 and the second extending portion 28 and joined at the fourth input / output terminal 56 are equal. As a result, the high frequency signal input from the first input / output terminal 53 is output from the fourth input / output terminal 56. When the first phase shift circuit 52 designed in this way is used, even if the first and second variable capacitance elements 27b and 28b that cause the phase change of the reflected wave are used, the electromagnetic wave input from the first input / output terminal is used. However, a phase shift circuit that outputs electromagnetic waves from the fourth input / output terminal 56 can be realized.

Since the impedance of the first and second extending portions 27 and 28 changes according to the voltage applied to the first and second electrodes 33 and 34, the voltage applied to the first and second electrodes 33 and 34 is adjusted. By doing so, the amount of change in phase of the high-frequency signal passing through the first phase shift circuit 52 can be adjusted.

  Further, since the electric capacity of the third variable capacitance element 32 changes according to the voltage applied to the third electrode 35, the high frequency reflected by the second phase shift circuit 12 by adjusting the voltage applied to the third electrode 35. The amount of change in the phase of the signal can be adjusted. Therefore, by adjusting the voltage applied to the third electrode 35, the impedance of the transmission line including the third transmission line 8 functioning as a stub and the second phase shift circuit 12 can be adjusted.

  According to the matching circuit 51 of the present embodiment described above, by using the branch line 57, the voltage applied to the first and second electrodes 33 and 34 is adjusted to pass through the first phase shift circuit 52. Thus, it is possible to realize a phase shift circuit capable of adjusting the phase of the high-frequency signal. In addition, by adjusting the voltage applied to the third electrode 35, a phase shift circuit capable of adjusting the phase of the high-frequency signal reflected by the second phase shift circuit 12 can be realized. As a result, the impedance of the matching circuit 51 can be easily adjusted by adjusting the voltage applied to the first phase shift circuit 52 and the second phase shift circuit 12 as described above, and is output from the electronic component 2. Can be prevented from being reflected by the matching circuit 51.

  FIG. 7 is a plan view showing a matching circuit 61 according to still another embodiment of the present invention. Since the matching circuit 61 of the present embodiment is different from the configuration of the matching circuits 1, 40, 41, 51 of the above-described embodiments and the first phase shift circuit 62, and the second reference electrode 81 is added, Corresponding components will be denoted by the same reference numerals and redundant description will be omitted, and only different parts will be described.

  The first phase shift circuit 62 includes a directional coupler 67 having first to fourth input / output terminals 63 to 66, a first extending portion 27 formed extending from the second input / output terminal 64, And a second extending portion 28 formed extending from the third input / output terminal 65.

  When a high frequency signal having a wavelength λ is input from the first input / output terminal 63, the directional coupler 67 is distributed to the second input / output terminal 64 and the third input / output terminal 65 and output. The fourth input / output terminal 66 is configured not to output the signal. Further, the phase of the high frequency signal output from the second input / output terminal 64 is different from the phase of the high frequency signal output from the third input / output terminal 65 by π / 2 rad. The degree of coupling of the directional coupler 67 of the present embodiment is selected to be 2 dB or more and 4 dB or less, and preferably 3 dB.

  The directional coupler 67 is realized by a long type directional coupler in the present embodiment. The first to fourth input / output terminals 63 to 66 are provided at the vertices of the quadrangle, and the first input / output terminal 63 and the third input / output terminal 65 are provided diagonally. In the present embodiment, a direction in which a straight line connecting the first input / output terminal 63 and the second input / output terminal 64 extends is described as a lateral direction X, and the first input / output terminal 63 and the fourth input / output terminal 66 are connected. A direction in which the straight line extends is referred to as a vertical direction Y.

  The first input / output terminal 63 and the third input / output terminal 65 are connected by a transmission line. The transmission line includes a transmission line 68 extending in the lateral direction X from the first input / output terminal 63 to the middle between the first input / output terminal 63 and the second input / output terminal 64, and the first input / output terminal 63 and the second input / output. A first vertical transmission line 69 extending in the vertical direction Y from the middle of the terminal 64 to the middle of the fourth input / output terminal 66 and the third input / output terminal 65; a fourth input / output terminal 66; And a transmission line 70 extending in the lateral direction X from the middle to the third input / output terminal 65.

  A transmission line 71 extends in the lateral direction X from the second input / output terminal 64 to a position close enough not to be connected to the transmission line 68 extending from the first input / output terminal 63, and the first vertical line extends from the tip of the transmission line 71. A second vertical transmission line 72 extends in the vertical direction Y along the transmission line 69. The tip of the second vertical transmission line 72 extends in the vertical direction Y to the extent that it is not connected to the transmission line 70 extending from the third input / output terminal 65.

  A transmission line 73 extends in the lateral direction X from the fourth input / output terminal 66 to a position close enough not to connect to the transmission line 70 extending from the third input / output terminal 65. A third vertical transmission line 74 extends in the vertical direction Y along the transmission line 69. The tip of the third vertical transmission line 74 extends in the vertical direction Y to the extent that it is not connected to the transmission line 68 extending from the first input / output terminal 63.

  From the transmission line 68 extending in the horizontal direction X from the first input / output terminal 63, a fourth vertical transmission line 75 is branched and extends in the vertical direction Y. The fourth vertical transmission line 75 is provided close to the third vertical transmission line 74 so as to sandwich the third vertical transmission line 74 with the first vertical transmission line 69.

  From the transmission line 70 extending in the horizontal direction X from the third input / output terminal 65, a fifth vertical transmission line 76 is branched and extends in the vertical direction Y. The fifth vertical transmission line 76 is provided close to the second vertical transmission line 72 so as to sandwich the second vertical transmission line 72 with the first vertical transmission line 69.

  The line lengths of the first to third vertical transmission lines 69, 72, and 74 are selected to be approximately λ / 4, and the line lengths of the fourth and fifth vertical transmission lines 75 and 76 are selected to be approximately λ / 8. Each transmission line is realized by a microstrip line.

  The directional coupler 67 is a wire that electrically connects the middle in the longitudinal direction Y of the first vertical transmission line 69, the free end of the fourth vertical transmission line 75, and the free end of the fifth vertical transmission line 76. The bridge 77 is further included. The directional coupler 67 further includes a wire-like bridge 78 that electrically connects the free end of the third vertical transmission line 74 and the end opposite to the free end of the second vertical transmission line 72. . Further, the directional coupler 67 further includes a wire-like bridge 79 that electrically connects an end portion of the third vertical transmission line 74 opposite to the free end and the free end of the second vertical transmission line 72. .

  The end of the first transmission line 6 opposite to the end connected to the bonding wire 3 is connected to a first input / output terminal 63 corresponding to one first connection end 13a. The second transmission line 7 is connected to a fourth input / output terminal 66 corresponding to the other first connection end 13b.

  The first extending portion 27 is provided with a transmission line 27 a extending from the second input / output terminal 64 and a first variable portion provided at an end opposite to the end where the transmission line 27 a is connected to the directional coupler 67. And a capacitive element 27b. The second extending portion 28 is provided with a transmission line 28 a extending from the third input / output terminal 65 and a second variable portion provided at an end opposite to the end where the transmission line 28 a is connected to the directional coupler 67. And a capacitive element 28b.

  The first to third electrodes 33, 34, and 35 have the same configuration as that of the first to third electrodes 33, 34, and 35 described above, and thus redundant description is omitted. The reference electrode 36 for determining the reference voltage of the voltage applied to the first to third variable capacitance elements 27b, 28b, 32 is the same as the configuration of the above-described embodiment, and the first transmission line Branching between the DC blocking capacitor 39 and the first input / output terminal 63. The matching circuit 61 includes a second reference electrode 81 in addition to the matching circuits 1, 40, 41, and 51 of the above-described embodiments. The second reference electrode 81 has the same configuration as that of the reference electrode 36, and extends from the branch portion 19 and the DC blocking capacitor 39 in the other 7 b of the second transmission line 7. Similar to the reference electrode 36, the second reference electrode 81 determines a reference voltage for a voltage applied to the first to third variable capacitance elements 27b, 28b, 32.

  The first extending portion 27 is formed so as to function as a stub that reflects a high-frequency signal input from the second input / output terminal 64 through the first input / output terminal 63. The second extending portion 28 is formed so as to function as a stub that reflects a high-frequency signal input from the third input / output terminal 65 through the first input / output terminal 63. The line length of the transmission line 27a between the second input / output terminal 64 and the first variable capacitance element 27b is substantially equal to the line length of the transmission line 28a between the third input / output terminal 65 and the second variable capacitance element 28b. They are chosen to be equal lengths or different so as to differ by a length of nλ / 2. Since the transmission line 27a of the first extension part 27 and the transmission line 28a of the second extension part 28 are formed to have the above-mentioned line length, they are input from the first input / output terminal 63, and the first extension part The phases of the high frequency signals reflected by the existing portion 27 and the second extending portion 28 and joined at the fourth input / output terminal 66 become equal. As a result, a high frequency signal input from the first input / output terminal 63 is output from the fourth input / output terminal 66. When the first phase shift circuit 62 designed in this way is used, even if the first and second variable capacitance elements 27b and 28b that cause a phase change of the reflected wave are used, the electromagnetic wave input from the first input / output terminal is used. However, a phase shift circuit that outputs electromagnetic waves from the fourth input / output terminal 66 can be realized.

  Since the impedance of the first and second extending portions 27 and 28 changes according to the voltage applied to the first and second electrodes 33 and 34, the voltage applied to the first and second electrodes 33 and 34 is adjusted. By doing so, the amount of change in the phase of the high frequency signal passing through the first phase shift circuit 62 can be adjusted.

  In addition, since the electric capacity of the third variable capacitance element 32 changes according to the voltage applied to the third electrode 35, the high frequency passing through the second phase shift circuit 12 is adjusted by adjusting the voltage applied to the third electrode 35. The amount of change in the phase of the signal can be adjusted. Therefore, by adjusting the voltage applied to the third electrode 35, the impedance of the transmission line including the third transmission line 8 functioning as a stub and the second phase shift circuit 12 can be adjusted.

  According to the matching circuit 61 of the present embodiment described above, the first phase shift circuit 62 is adjusted by adjusting the voltage applied to the first and second electrodes 33 and 34 by using the directional coupler 67. It is possible to realize a phase shift circuit capable of adjusting the phase of the high-frequency signal passing through. In addition, by adjusting the voltage applied to the third electrode 35, a phase shift circuit capable of adjusting the phase of the high-frequency signal reflected by the second phase shift circuit 12 can be realized. As a result, the impedance of the matching circuit 61 can be easily adjusted by adjusting the voltages applied to the first phase shift circuit 62 and the second phase shift circuit 12 as described above, and is output from the electronic component 2. Can be prevented from being reflected by the matching circuit 61.

  FIG. 8 is a plan view showing a matching circuit 84 according to still another embodiment of the present invention. Since the matching circuit 84 of the present embodiment is different from the configuration of the matching circuits 1, 40, 41, 51, 61 of the above-described embodiments and the first phase shift circuit 85, the same reference numerals are used for the corresponding configurations. The description which overlaps is abbreviate | omitted and only a different part is demonstrated.

  The first phase shift circuit 85 includes a circulator 89 having first to third input / output terminals 86, 87, 88 and an extending portion 91 formed extending from the second input / output terminal 87. Is done.

When a high frequency signal having a wavelength λ is input from the first input / output terminal 86, the circulator 89 outputs the input high frequency signal from the second input / output terminal 87 and does not output from the third input / output terminal 88. Composed. The high frequency signal input from the second input / output terminal 87 is output from the third input / output terminal 88 and is not output from the first input / output terminal 86. The circulator 89 is realized, for example, by applying a static magnetic field having an appropriate strength to ferrite sandwiching the transmission line.

  The end of the first transmission line 6 opposite to the end connected to the bonding wire 3 is connected to a first input / output terminal 86 corresponding to one first connection end 13a. The second transmission line 7 is connected to a third input / output terminal 88 corresponding to the other first connection end 13b.

  The extending portion 91 includes a transmission line 91a extending from the second input / output terminal 87, and a first variable capacitance element 91b provided at an end opposite to the end where the transmission line 91a is connected to the circulator 89. Consists of including. In the present embodiment, the transmission line 91a constituting the extending portion 91 is realized by a microstrip line, and the first variable capacitance element 91b is composed of an element whose electric capacity changes according to an applied voltage, for example, a varactor Realized by a diode.

  The second phase shift circuit 12 includes a second variable capacitance element 92 whose electric capacity changes according to an applied voltage, and is realized by, for example, a varactor diode.

  The first electrode 33 has the same configuration as the first electrode 33 described above, is connected to the first variable capacitance element 91b, and applies a voltage to the first variable capacitance element 91b. The second electrode 34 has the same configuration as the second electrode 34 described above, is connected to the second variable capacitance element 92, and applies a voltage to the second variable capacitance element 92.

  From the transmission line 91a of the extension part 91, it is the structure similar to the above-mentioned reference electrode 36, Comprising: The reference electrode equivalent to the 3rd electrode for determining the reference voltage of the voltage applied to the 1st variable capacitance element 91b 36 branches and extends. Further, the other part 7b of the second transmission line 7 has the same configuration as the second reference electrode 81 described above, and corresponds to a fourth electrode for determining the reference voltage of the voltage applied to the second variable capacitance element 92. The second reference electrode 81 is extended.

  Matching circuit 84 further includes a DC blocking capacitor 93 that blocks a low-frequency signal having a lower frequency than a high-frequency signal passing through matching circuit 84 and a DC signal. The DC blocking capacitor 93 is provided between the circulator 89 and a branch portion of the extending portion 91 where the reference electrode 36 corresponding to the third electrode branches. The DC blocking capacitor 93 is provided between the circulator 89 and the branch portion of the second transmission line 7 where the third transmission line 8 branches. Further, the DC blocking capacitor 93 is provided in a portion of the second transmission line 7 that is further away from the first phase shift circuit 85 than the branching portion where the second reference electrode 81 corresponding to the fourth electrode branches.

  The extending portion 91 is formed so as to function as a stub that reflects a high-frequency signal input from the second input / output terminal 87 through the first input / output terminal 86. As a result, the high frequency signal input from the second input / output terminal 87 through the first input / output terminal 86 is reflected by the extending portion 91 and input to the second input / output terminal 87 again, and the third input / output terminal 87 88.

  Since the impedance of the extending portion 91 changes according to the voltage applied to the first electrode 33, the phase of the high frequency signal passing through the first phase shift circuit 85 is adjusted by adjusting the voltage applied to the first electrode 33. The amount of change can be adjusted.

  In addition, since the electric capacity of the second variable capacitance element 92 changes according to the voltage applied to the second electrode 34, the high frequency reflected by the second phase shift circuit 12 by adjusting the voltage applied to the second electrode 34. The amount of change in the phase of the signal can be adjusted. Therefore, by adjusting the voltage applied to the second electrode 34, the impedance of the transmission line including the third transmission line 8 functioning as a stub and the second phase shift circuit 12 can be adjusted.

  According to the matching circuit 84 of the present embodiment described above, the voltage applied to the first and second electrodes 33 and 34 is adjusted by using the circulator 89, thereby passing through the first phase shift circuit 85. A phase shift circuit capable of adjusting the phase of the high-frequency signal can be realized. In addition, by adjusting the voltage applied to the third electrode 35, a phase shift circuit capable of adjusting the phase of the high-frequency signal reflected by the second phase shift circuit 12 can be realized. As a result, the impedance of the matching circuit 84 can be easily adjusted by adjusting the voltage applied to the first phase shift circuit 85 and the second phase shift circuit 12 as described above, and is output from the electronic component 2. Can be prevented from being reflected by the matching circuit 61.

  FIG. 9 is a perspective view showing a matching circuit 101 according to still another embodiment of the present invention. The matching circuit 101 of the present embodiment is different from the configuration of the matching circuits 1, 40, 41, 51, 61, and 84 of the above-described embodiments in the first phase shift circuit 102. A referential mark is attached | subjected and the overlapping description is abbreviate | omitted and only a different part is demonstrated.

  The first phase shift circuit 102 includes a magic T 107 having first to fourth input / output terminals 103 to 106, a first extending portion 27 formed extending from the second input / output terminal 104, and a fourth input And a second extending portion 28 formed extending from the output terminal 106.

  When the magic T 107 receives a high-frequency signal having a wavelength λ from the first input / output terminal 103, the input high-frequency signal is distributed to the second input / output terminal 104 and the fourth input / output terminal 106 and output. No output from the 3 input / output terminal 105. The magic T107 is realized by a waveguide formed on the dielectric substrate 5. In the present embodiment, the waveguide is filled with a dielectric made of the same material as the dielectric substrate 5. Furthermore, the phases of the high-frequency signals input from the first input / output terminal 103 and output from the second and fourth input / output terminals 104 and 106 are different from each other by π rad.

  The first, second, and fourth input / output terminals 103, 104, and 106 are formed on one surface of the dielectric substrate 5 in the thickness direction Z, respectively. The first input / output terminal 103 is formed between the second input / output terminal 104 and the fourth input / output terminal 106. In the present embodiment, the direction in which the first, second, and fourth input / output terminals 103, 104, and 106 are formed is referred to as a lateral direction X. In the present embodiment, a direction perpendicular to the lateral direction X and the thickness direction Z of the dielectric substrate 5 is referred to as a longitudinal direction Y.

  The magic T 107 includes a waveguide 108 extending in the lateral direction X in the other Z 2 in the thickness direction Z of the first, second, and fourth input / output terminals 103, 104, 106, and the first and second from the waveguide 108. , Waveguides 109, 110, and 111 extending toward one side Z <b> 1 in the thickness direction Z toward the fourth input / output terminals 103, 104, and 106. The magic T107 is formed in plane symmetry with respect to a virtual plane that passes through the center in the horizontal direction X and is perpendicular to the horizontal direction X. The third input / output terminal 105 is formed on one side in the vertical direction Y in the middle of the horizontal direction X of the waveguide 108 extending in the horizontal direction X.

  The end of the first transmission line 6 opposite to the end connected to the bonding wire 3 is connected to a first input / output terminal 103 corresponding to one first connection end 13a. The second transmission line 7 is connected to a third input / output terminal 105 corresponding to the other first connection end 13b. In the present embodiment, the second transmission line 7 is realized by a waveguide extending from the third input / output terminal 105 to one side in the longitudinal direction Y. The other bottom surface portion 113 of the second transmission line 7 in the thickness direction Z is constituted by a part of the back surface electrode 17. Therefore, each waveguide is set to the same voltage as the back electrode 17.

  The first extending portion 27 includes a transmission line 27a extending from the second input / output terminal 104, and a first variable capacitance element 27b provided at an end opposite to the end where the transmission line 27a is connected to the magic T107. It is comprised including. The second extending portion 28 includes a transmission line 28a extending from the fourth input / output terminal 106, and a second variable capacitance element 28b provided at the end opposite to the end where the transmission line 28a is connected to the magic T107. It is comprised including. In the present embodiment, the transmission lines 27a and 28a constituting the first and second extending portions 27 and 28 are realized by microstrip lines, and the first and second variable capacitance elements 27b and 28b apply. It consists of an element whose electric capacity changes according to the voltage, and is realized by, for example, a varactor diode.

  In the present embodiment, the third transmission line 8 includes a waveguide 112 that branches from the second transmission line 7 to one Z1 in the thickness direction Z of the dielectric substrate 5 and a free end portion of the waveguide 112. And a transmission line 114 extending from the line. The transmission line 114 constituting the third transmission line 8 is realized by a microstrip line, for example.

  In the present embodiment, the second phase shift circuit 12 includes a third variable capacitance element 32 whose electric capacity changes according to an applied voltage, and is realized by, for example, a varactor diode.

  The matching circuit 1 includes a first electrode 33 for applying a voltage to the first variable capacitor 27b, a second electrode 34 for applying an electrode to the second variable capacitor 28b, and a third variable capacitor 32. And a third electrode 35 for applying a voltage. In the present embodiment, since the voltage of each waveguide is set to the same voltage as that of the back electrode 17, the reference voltage serving as a reference for the voltage applied to the first to third variable capacitance elements 27b, 28b, 32 is The voltage is the same as that of the back electrode 17, and it is not necessary to provide the reference electrode 36 as in the above-described embodiments.

  The first extending portion 27 is formed so as to function as a stub that reflects a high-frequency signal input from the second input / output terminal 104 through the first input / output terminal 103. The second extending portion 28 is formed so as to function as a stub that reflects a high-frequency signal input from the fourth input / output terminal 106 through the first input / output terminal 103. The impedance of the first extension part 27 depends on the line length of the transmission line 27a of the first extension part 27 and the electric capacity of the first variable capacitance element 27b. Since the electric capacity of the first variable capacitance element 27b is variable according to the applied voltage, it can be adjusted by adjusting the voltage applied to the first electrode 33. Similarly, the impedance of the second extending portion 28 can be adjusted by adjusting the voltage applied to the second electrode 34.

  The line length of the transmission line 28a between the fourth input / output terminal 106 and the second variable capacitance element 28b is larger than the line length of the transmission line 27a between the second input / output terminal 104 and the first variable capacitance element 27b. , Λ / 4 is selected longer. Since the transmission line 27a of the first extension part 27 and the transmission line 28a of the second extension part 28 are formed to have the above-described line length, they are inputted from the first input / output terminal 103 and branched, The phases of the high-frequency signals that are reflected by the first extension portion 27 and the second extension portion 28 and merge at the third input / output terminal 105 are equal. As a result, the high frequency signal input from the first input / output terminal 103 is output from the third input / output terminal 105. When the first phase shift circuit 102 designed in this way is used, even if the first and second variable capacitance elements 27b and 28b that cause a phase change of the reflected wave are used, the electromagnetic wave input from the first input / output terminal is used. However, a phase shift circuit that outputs electromagnetic waves from the third input / output terminal 105 can be realized.

Since the impedance of the first and second extending portions 27 and 28 changes according to the voltage applied to the first and second electrodes 33 and 34, the voltage applied to the first and second electrodes 33 and 34 is adjusted. By doing so, the amount of change in the phase of the high-frequency signal passing through the first phase shift circuit 102 can be adjusted.

  Further, since the electric capacity of the third variable capacitance element 32 changes according to the voltage applied to the third electrode 35, the high frequency reflected by the second phase shift circuit 12 by adjusting the voltage applied to the third electrode 35. The amount of change in the phase of the signal can be adjusted. Therefore, by adjusting the voltage applied to the third electrode 35, the impedance of the transmission line including the third transmission line 8 functioning as a stub and the second phase shift circuit 12 can be adjusted.

  The side wall of each waveguide may be formed of an electrically conductive metal, and a plurality of columnar conductor columns formed of a conductive material are spaced from each other in the region where the side wall is formed. It may be formed with a gap.

  According to the matching circuit 101 of the present embodiment described above, by using the magic T107, the voltage applied to the first and second electrodes 33 and 34 is adjusted to pass through the first phase shift circuit 102. A phase shift circuit capable of adjusting the phase of the high-frequency signal can be realized. In addition, by adjusting the voltage applied to the third electrode 35, a phase shift circuit capable of adjusting the phase of the high-frequency signal reflected by the second phase shift circuit 12 can be realized. As a result, the impedance of the matching circuit 101 can be easily adjusted by adjusting the voltages applied to the first phase shift circuit 102 and the second phase shift circuit 12 as described above, and is output from the electronic component 2. Can be prevented from being reflected by the matching circuit 101.

  FIG. 10 is a perspective view showing the first phase shift circuit 122 of the matching circuit 121 according to still another embodiment of the present invention. In matching circuit 121 of the present embodiment, at least one of the first phase shift circuit and the second phase shift circuit includes a changing portion whose dielectric constant changes according to the applied electric field, and is a dielectric that propagates electromagnetic waves. It includes a body line and a pair of flat conductor portions that sandwich the dielectric line. In the present embodiment, the first phase shift circuit 122 is configured by a nonradiative dielectric line (NRD guide). The phase shift circuit may be a non-radiative dielectric line, a dielectric waveguide, an image guide, a microstrip line, a coplanar line, a strip line, or the like.

  One 123a of the pair of flat conductor portions 123a and 123b is realized by a conductor formed on one surface of the dielectric substrate 5 in the thickness direction Z, and the other 123b of the pair of flat conductor portions 123a and 123b is the back electrode 17. It is realized by. One of the pair of flat conductor portions 123a and 123b is mainly made of gold (Au), copper (Cu), aluminum (Al), platinum (Pt), titanium (Ti), silver (Ag), palladium (Pd), zinc. It is formed of a conductive metal such as (Zn), chromium (Cr), and Ni (nickel). The dielectric line 124 sandwiched between the pair of flat conductor parts 123 a and 123 b is formed to extend in a direction perpendicular to the thickness direction Z of the dielectric substrate 5. The dielectric line 124 is sandwiched by the dielectric substrate 5 at a side surface perpendicular to the extending direction of the dielectric line 124 and the thickness direction Z of the dielectric substrate 5.

In the present embodiment, the dielectric line 124 is formed by a changing portion, and is formed by a dielectric whose dielectric constant changes according to the applied electric field. The dielectric line 124 is formed of a material having a dielectric constant higher than that of a material in contact with the side surface perpendicular to the extending direction of the dielectric line 124 and the thickness direction Z of the dielectric substrate 5 in order to confine high-frequency signals. . In the present embodiment, the dielectric line 124 is realized by a substance having a dielectric constant higher than that of the dielectric substrate 5, for example, Ba (1-x) SrxTiO ₃ (strontium barium titanate, abbreviated as BST), Mg (1- x) formed of CaxTiO ₃ , Zn (1-x) SnxTiO ₃ , BaO—PbO—Nd ₂ O ₃ —TiO ₃ , SrBi ₂ Ta ₂ O ₉ or Bi _1.5 Zn _1.0 Nb _1.5 O ₇ Is done. The dielectric constant of the dielectric line 124 decreases as the applied electric field increases, that is, as the applied electric field strength increases.

  The voltage applied to the dielectric line 124 can be adjusted by, for example, the voltage applied to one 123a of the pair of flat conductor portions 123a and 123b. The high-frequency signal is confined in the dielectric line 124 and propagates in the direction in which the dielectric line 124 extends. The wavelength of the high-frequency signal propagating through the dielectric line 124 depends on the dielectric constant of the dielectric line 124 and depends on the voltage applied to one of the pair of flat conductor parts 123a and 123b. Therefore, the phase of the high-frequency signal passing through the first phase shift circuit 122 can be adjusted by the voltage applied to one 123a of the pair of flat conductor portions 123a and 123b, and the phase of the high-frequency signal passing therethrough can be adjusted. A phase shift circuit 122 can be realized.

  According to the matching circuit 121 of the present embodiment described above, the voltage applied to one of the pair of flat conductor portions 123a and 123b is adjusted by using the first phase shift circuit 122 realized by the NRD guide. Thus, a phase shift circuit capable of adjusting the phase of the high-frequency signal passing through the first phase shift circuit 52 can be realized. Further, by adjusting the voltage applied to the third electrode 35, a phase shift circuit capable of adjusting the phase of the high-frequency signal passing through the second phase shift circuit 12 can be realized. As a result, the impedance of the matching circuit 121 can be easily adjusted by adjusting the voltages applied to the first phase shift circuit 122 and the second phase shift circuit 12 as described above, and is output from the electronic component 2. Can be prevented from being reflected by the matching circuit 121. In addition, when the second phase shift circuit 12 is configured by an NRD that includes a changing portion as in the case of the first phase shift circuit 122, it functions as a stub together with the third transmission line 8, and depends on the voltage applied to the second phase shift circuit 12. Thus, it is possible to realize a phase shift circuit capable of adjusting the phase of the electromagnetic wave reflected.

  FIG. 11 is a perspective view showing a matching circuit 201 according to still another embodiment of the present invention. The matching circuit 201 of the present embodiment is different from the configuration of the matching circuits 1, 40, 41, 51, 61, 84, 101, 121 of the above-described embodiments in that the first transmission line 6 is different. For the above, the same reference numerals are attached, and redundant description is omitted, and only different portions will be described.

  The first transmission line 6 further includes an impedance changing unit 202. The impedance changing unit 202 is provided between the connection part 4 to which one end of the bonding wire 3 corresponding to the connection body is connected and the one end connected to the first connection end 13a. Furthermore, the impedance changing unit 202 is different in impedance from the connection part 4.

  The impedance changing portion 202 is a portion having a characteristic impedance different from that of the connector portion 203 from the free end portion 6 a to the impedance changing portion 202 in the first transmission line 6. In order to make the characteristic impedances of the impedance changing portion 202 and the connector portion 203 different, for example, the line widths of the connector portion 203 and the impedance changing portion 202 are made different, a stub is formed as the impedance changing portion 202, a transmission line A gap that divides the first transmission line 6 into two parts is formed in the extending direction of the dielectric substrate 5, or the relative dielectric constant of the dielectric substrate 5 between the impedance changing portion 202 and the back electrode 17 is changed between the connector portion 203 and the back electrode. Or a member having a relative dielectric constant different from that of air is disposed on the surface of the impedance changing portion 202 opposite to the dielectric substrate 5. Also good. Impedance changing section 202 in the present embodiment is realized by forming a stub.

  By providing the impedance changing unit 202, the impedance of the first transmission line 6 changes in a route from the connection site 4 to one end connected to the first phase shift circuit 11. A part of the high-frequency signal input from the electronic component 2 via the bonding wire 3 is reflected and partly transmitted by the impedance changing unit 202 where the impedance changes. Part of the high-frequency signal transmitted through the impedance changing unit 202 is reflected by the branching unit 19 where the third transmission line of the second transmission line branches and returns to the impedance changing unit 202. The high frequency signal reflected by the impedance changing unit 202 and the high frequency signal reflected after passing through the impedance changing unit 202 are superimposed. The phase of the high-frequency signal that is transmitted through the impedance changing unit 202 and then reflected back to the impedance changing unit 202 is changed from the first transmission line 6, the first phase shift circuit 11, and the second transmission line branching unit 19 to the first shift. It depends on two parameters: the electrical length of the transmission line constituted by the portion near the phase circuit 11, and the impedance of the transmission line functioning as a stub by the third transmission line 8 and the second phase shift circuit 12. These two parameters can be set to intended values by adjusting the voltages applied to the first and second phase shift circuits 11 and 12. As a result, the phase of the high-frequency signal that is reflected after being transmitted through the impedance changing unit 202 and returned to the impedance changing unit 202 can be adjusted, and the superimposed high-frequency signal can be weakened. As a result, the electronic component 2 is connected via the bonding wire 3. Thus, the reflection coefficient of the high-frequency signal input to the matching circuit 201 can be reduced.

The line lengths of the first and third transmission lines 6, 8, the impedance of the impedance changing unit 202, and the line length of one of the second transmission lines 7 a in this embodiment are the first and second phase shift circuits 11, In a range where the amount of phase shift of 12 is variable, it is selected so as to satisfy the following formula (2).
| Γ ₃ | <| T _M2 | ² (2)

In Equation (2), Γ ₃ represents the reflection coefficient of electromagnetic waves input from the electronic component 2 side to the connection block 204 constituted by the bonding wire 3, the connector portion 203, and the impedance changing portion 202, and | T _M2 | indicates that the electromagnetic wave input from the connection block 204 to the remaining portion 205 of the first transmission line 6 excluding the connector portion 203 and the impedance changing portion 202 is branched from the third transmission line 8 of the second transmission line 7. This represents the size of the transmission coefficient that passes to the branching section 19 that performs the transmission.

Furthermore, the line lengths of the first and third transmission lines 6 and 8, the impedance of the impedance changing unit 202, and the line length of one of the second transmission lines 7 in the present embodiment are the first and second phase shift circuits 11. , 12 in a range in which the phase shift amount is variable, the reflection coefficient S ₁₁ by which the high-frequency signal input from the electronic component 2 is reflected by the connection circuit 206 to the connection circuit 206 constituted by the bonding wire 3 and the matching circuit 201 is It is chosen to be zero.

When the matching circuit 201 is manufactured so as to satisfy the expression (2), bonding is performed by adjusting the voltages applied to the first and second phase shift circuits 11 and 12 as described above in the section “Effects of the Invention”. the reflection coefficient S ₁₁ of the high-frequency signal inputted from the electronic component 2 to the configured connection circuit 206 by the wire 3 and the matching circuit 201 is reflected by the connection circuit 206 can be adjusted to be 0. By adjusting in this way the reflection coefficient S ₁₁ to 0, that load as designed the electronic component 2 (for example, 50 [Omega) can be used in connection with, obtain a characteristic as designed electronic components 2 it can. Further, by adjusting the reflection coefficient S ₁₁ to 0, the high-frequency signal inputted from the electronic component 2 can be reduced attenuation to be outputted through the matching circuit 201.

FIG. 12 is a graph showing S ₁₁ (reflection coefficient viewed from the electronic component 2) of the S parameters. The matching circuit 201 of the present embodiment in which the impedance changing unit 202 is provided in the matching circuit 1 shown in FIG. represents the reflection coefficient S ₁₁ as seen from the electronic component 2 composed connecting circuit 206 by bonding wires 3 and. FIG. 12 shows S ₁₁ at 76.5 GHz when the voltage applied to the first phase shift circuit 11 and the second phase shift circuit 12 is changed. Compared with FIG. 3 (2), the measurement data of FIG. 12 is closer to the center, so that by providing the impedance changing unit 202, the impedance matching between the electronic component 2 and the connection circuit 206 can be further improved, and reflection can be achieved. It was shown to be reduced further. Further, it can be seen that S ₁₁ changes by changing the voltage applied to the first phase shift circuit 11 and the second phase shift circuit 12. As a result, when the connection circuit 206 is manufactured, the dimension of the pattern constituting the matching circuit (the width and length of the first to third transmission lines) varies, the mounting position of the electronic component 2 is shifted, and the length of the bonding wire is increased. Even if the manufactured connection circuit 206 deviates from the design value due to variations, adjustment is possible by changing the voltage applied to the first phase shift circuit 11 and the second phase shift circuit 12. I understand that.

  FIG. 13 is a schematic diagram illustrating a configuration of the transmitter 160 according to the embodiment of this invention. The transmitter 160 includes the matching circuit 1 of the embodiment shown in FIG. 1 described above, a high-frequency oscillator 161, a transmission line 162, and a transmitting antenna 163. The high frequency oscillator 161 includes a Gunn oscillator using a Gunn diode, an Impat oscillator using an Impat diode, or an MMIC (Microwave Monolithic Integrated Circuit) oscillator using a transistor such as an FET (Field Effect Transistor). Generate a signal. The transmission line 162 is configured by a microstrip line, a strip line, or a coplanar line. A first end 162 a of the transmission line 162 in the high-frequency signal transmission direction is connected to the high-frequency oscillator 161, and a second end 162 b of the transmission line 162 in the high-frequency signal transmission direction is connected to the transmitting antenna 163. The transmitting antenna 163 is realized by a patch antenna or a horn antenna. The transmission direction of the high frequency signal is the propagation direction of the electromagnetic wave. Although the first end 162a is configured by a microstrip line, a strip line, or a coplanar line, the first end 162a may be configured by a bonding wire 3 that electrically connects the high-frequency oscillator 161 and the matching circuit 1.

  The matching circuit 1 is inserted into the transmission line 162 so that the high-frequency signal passes. More specifically, the transmission line 162 electrically connects the transmission line that electrically connects the first transmission line 6 and the high-frequency oscillator 161, the other 7 b of the second transmission line 7, and the transmitting antenna 163. And a transmission line. Although the transmission line 162 is configured by a microstrip line, a strip line, or a coplanar line, the transmission line that electrically connects the first transmission line 6 and the high-frequency oscillator 161 is configured by the bonding wire 3 described above. Also good.

  A high-frequency signal generated by the high-frequency oscillator 161 passes through the transmission line 162 and the matching circuit 1, is given to the transmitting antenna 163, and is radiated as a radio wave from the transmitting antenna 163.

  As described above, since the matching circuit 1 is inserted into the transmission line 162, for example, due to variations in the shape of bonding wires and bumps for connecting the high-frequency oscillator 161, variations in the wiring width of the transmission line 162, etc. Thus, it is possible to achieve matching by individually adjusting the generated phase shift, and to realize a transmitter 160 having a stable oscillation characteristic and a high transmission output because the insertion loss is suppressed to a small level. .

  Although the transmitter 160 uses the matching circuit 1, any one of the matching circuits of the above-described embodiments may be used. Even if comprised in this way, the same effect can be achieved. In the transmitter 160, the transmission line 162 may be realized by a coplanar line, a coplanar line with a ground, a slot line, a waveguide, or a dielectric waveguide, in addition to the microstrip line and the strip line.

  FIG. 14 is a schematic diagram illustrating a configuration of the receiver 170 according to the embodiment of this invention. The same configurations as those of the transmitter 160 of the above-described embodiment illustrated in FIG. 13 are denoted by the same reference numerals, and description thereof may be omitted.

  The receiver 170 includes the matching circuit 1 according to the above-described embodiment, a high frequency detector 171, a transmission line 162, and a receiving antenna 173. The high frequency detector 171 is realized by, for example, a Schottky barrier diode detector, a video detector, a mixer MMIC, or the like.

  A first end 162 a of the transmission line 162 in the high-frequency signal transmission direction is connected to the high-frequency detector 171, and a second end 162 b of the transmission line 162 in the high-frequency signal transmission direction is connected to the receiving antenna 173. . The receiving antenna 173 is realized by a planar antenna such as a patch antenna, a horn antenna, a rod antenna, or the like.

  The matching circuit 1 is inserted into the transmission line 162 so that the high-frequency signal passes through the matching circuit 1.

  When a radio wave coming from the outside is captured by the receiving antenna 173, the receiving antenna 173 gives a high-frequency signal based on the radio wave to the transmission line 162, passes through the matching circuit 1, and gives the received high-frequency signal to the high-frequency detector 171. It is done. The high frequency detector 171 detects a high frequency signal and detects information included in the high frequency signal.

  In the receiver 170, the high frequency signal captured by the receiving antenna 173 is transmitted to the transmission line 162 and detected by the high frequency detector 171.

  As described above, since the matching circuit 1 is inserted into the transmission line 162, the transmission line 162 is connected to the transmission line 162 due to, for example, variations in the shape of bonding wires and bumps for connecting the high-frequency detector 171 and variations in the wiring width of the transmission line 162. Realizing receiver 170 having a high detection output because the phase shift caused by the adjustment can be individually adjusted to achieve matching, and has a stable detection characteristic and an insertion loss can be suppressed to a small level. Can do.

  In the receiver 170, the matching circuit 1 is used. However, instead of the matching circuit 1, any one of the matching circuits 1 of the above-described embodiments may be used. Even if comprised in this way, the same effect can be achieved. In the receiver 170, the transmission line 162 may be realized by a coplanar line, a coplanar line with a ground, a slot line, a waveguide, or a dielectric waveguide in addition to the microstrip line and the strip line.

  FIG. 15 is a schematic diagram illustrating a configuration of a radar apparatus 190 including the transceiver 180 according to the embodiment of this invention. In the radar apparatus 190, the same reference numerals are given to the same configurations as those of the transmitter 160 and the receiver 170 of the above-described embodiment shown in FIGS. 13 and 14, and the description thereof may be omitted. The radar apparatus 190 includes a transceiver 180 and a distance detector 191.

  The transceiver 180 includes the matching circuit 1, the high-frequency oscillator 161, the fourth to eighth transmission lines 181, 182, 183, 184, and 185, the branching unit 186, the branching unit 187, and the like. A transmission / reception antenna 188 and a mixer 189 are included. The transmission / reception antenna 188 is realized by a patch antenna or a horn antenna. The fourth to eighth transmission lines 181, 182, 183, 184, and 185 have the same configuration as the transmission line 162 described above.

  The first end 181a of the fourth transmission line 181 in the high-frequency signal transmission direction is connected to the high-frequency oscillator 161, and the second end 181b of the fourth transmission line 181 in the high-frequency signal transmission direction is connected to the branching device 186. Is done. The matching circuit 1 is inserted into the fourth transmission line 181 so that the high-frequency signal passes through the matching circuit 1.

  The branching unit (switching unit) 186 has first, second, and third terminals 186a, 186b, 186c, and a high-frequency signal supplied to the first terminal 186a is selectively applied to the second terminal 186b and the third terminal 186c. Output to. The branching device 186 is realized by, for example, a high frequency switching element. The branching unit 186 receives a control signal from a control unit (not shown), and selectively connects the first terminal 186a and the second terminal 186b or the first terminal 186a and the third terminal 186c based on the control signal. The radar device 190 is realized by a pulse radar. The controller connects the first terminal 186a and the second terminal 186b, outputs a pulsed high-frequency signal from the second terminal 186b, and then connects the first terminal 186a and the third terminal 186c, The signal is output from the third terminal 186c. A first end 182a of the fifth transmission line 182 in the transmission direction of the high frequency signal is connected to the second terminal 186b. A first end 184a of the seventh transmission line 184 in the transmission direction of the high frequency signal is connected to the third terminal 186c. The radar device 190 may be realized by an FM-CW radar using a voltage controlled oscillator as an oscillator.

  The duplexer 187 includes fourth, fifth, and sixth terminals 187a, 187b, and 187c. The duplexer 187 outputs a high-frequency signal applied to the fourth terminal 187a to the fifth terminal 187b and a high-frequency signal applied to the fifth terminal 187b. The signal is output to the sixth terminal 187c. A second end 182b of the fifth transmission line 182 in the high-frequency signal transmission direction is connected to the fourth terminal 187a. A first end 183a of the sixth transmission line 183 in the transmission direction of the high frequency signal is connected to the fifth terminal 187b. A second end 183 b of the sixth transmission line 183 in the transmission direction of the high frequency signal is connected to the transmission / reception antenna 188.

  The sixth terminal 187c is connected to the first end 185a of the eighth transmission line 185 in the high-frequency signal transmission direction. The second end 184 b of the seventh transmission line 184 in the high-frequency signal transmission direction and the second end 185 b of the eighth transmission line 185 in the high-frequency signal transmission direction are connected to the mixer 189. The duplexer 187 is realized by a hybrid circuit. The hybrid circuit is realized by a directional coupler, a branch line, a magic T, a rat race, or the like.

  The high-frequency signal generated by the high-frequency oscillator 161 passes through the fourth transmission line 181 and the matching circuit 1, and transmits / receives an antenna for transmission / reception via the branching unit 186, the fifth transmission line 182, the duplexer 187 and the sixth transmission line 183. 188 and radiated as radio waves from the transmitting / receiving antenna 188. The high-frequency signal generated by the high-frequency oscillator 161 passes through the fourth transmission line 181 and the matching circuit 1 and is given as a local signal to the mixer 189 via the branching unit 186 and the seventh transmission line 184.

  When a radio wave arriving from the outside is received by the transmission / reception antenna 188, the transmission / reception antenna 188 gives a high-frequency signal based on the radio wave to the sixth transmission line 183 and to the mixer 189 via the duplexer 187 and the eighth transmission line 185. It is done.

  The mixer 189 mixes the high frequency signals provided from the seventh and eighth transmission lines 184 and 185 and outputs an intermediate frequency signal. The intermediate frequency signal output from the mixer 189 is given to the distance detector 191.

  The distance detector 191 includes the high-frequency detector 171 described above, and is based on the intermediate frequency signal obtained by receiving the radio wave (echo) radiated from the transmitter / receiver 180 and reflected by the measurement object, Calculate the distance to the measurement object. The distance detector 191 is realized by a microcomputer, for example.

  In the transmitter / receiver 180, the matching circuit 1 is inserted into the fourth transmission line 181 so that a high-frequency signal passes through the matching circuit 1, so that the transmission / reception unit 180 is not caused by the transmission line 162 due to, for example, variations in wiring width. By adjusting the phase of a high-frequency signal that changes as desired, for example, it is possible to realize a transceiver 180 having a stable oscillation characteristic and a high transmission output because the insertion loss is suppressed to a small value. The transmitter / receiver 180 having a detection characteristic and a high detection output because the insertion loss is suppressed to be small can be realized, and the reliability of the intermediate frequency signal generated by the mixer 189 can be improved, for example. .

  In the radar device 190, the distance detector detects the distance from the transceiver 180 to the detection target, for example, the distance from the transmission / reception antenna 188 to the detection target, based on the intermediate frequency signal from the transmission / reception unit 180. The distance to the detection object can be accurately detected.

  The branching unit 186 may be realized by a hybrid circuit such as a directional coupler or a power divider. In this case, the high-frequency signal supplied to the first terminal 187a is branched to the second terminal 186b and the third terminal 186c. Is output. In this case, the power of the radio wave output from the transmission / reception antenna 188 is lower than that of the above-described configuration, but the control of the apparatus is simplified because it is not necessary to control the branching device 186.

  In the present embodiment, the matching circuit 1 is inserted into the fourth transmission line 181, but in still another embodiment of the present invention, the matching circuit 1 is at least one of the fourth to eighth transmission lines 181 to 185. In addition, the high-frequency signal may be inserted so as to pass through the matching circuit 1. Even if it is such a structure, the same effect can be achieved.

  Further, although the matching circuit 1 is used in the transceiver 180, any one of the matching circuits in the above-described embodiments may be used. Even if comprised in this way, the same effect can be achieved.

  In still another embodiment of the present invention, the duplexer 187 may be realized by a circulator or a high frequency switching element, and the same effect can be achieved even with such a configuration.

  FIG. 16 is a schematic diagram illustrating a configuration of a radar apparatus 196 including a transceiver 195 according to another embodiment of the present invention. In the radar apparatus 196, the same reference numerals are given to the same configurations as those of the transmitter 160 and the receiver 170 of the above-described embodiment shown in FIGS. 13 and 14, and the description thereof may be omitted. The radar device 196 includes a transceiver 195 and a distance detector 191.

  The transmitter / receiver 195 includes the matching circuit 1, the high-frequency oscillator 161, the fourth to seventh transmission lines 181, 182, 183, and 184, the branching unit 186, the transmitting antenna 163, and the receiving unit. An antenna 173 and a mixer 189 are included. The transmitting antenna 163 and the receiving antenna 173 are realized by a patch antenna or a horn antenna. The fourth to seventh transmission lines 181, 182, 183, and 184 have the same configuration as the transmission line 162 described above.

  The first end 181a of the fourth transmission line 181 in the high-frequency signal transmission direction is connected to the high-frequency oscillator 161, and the second end 181b of the fourth transmission line 181 in the high-frequency signal transmission direction is connected to the branching device 186. Is done. The matching circuit 1 is inserted into the fourth transmission line 181 so that the high-frequency signal passes through the matching circuit 1.

  The branching unit (switching unit) 186 has first, second, and third terminals 186a, 186b, 186c, and a high-frequency signal supplied to the first terminal 186a is selectively applied to the second terminal 186b and the third terminal 186c. Output to. The branching device 186 is realized by, for example, a high frequency switching element. The branching unit 186 receives a control signal from a control unit (not shown), and selectively connects the first terminal 186a and the second terminal 186b or the first terminal 186a and the third terminal 186c based on the control signal. The radar device 196 is realized by a pulse radar. The controller connects the first terminal 186a and the second terminal 186b, outputs a pulsed high-frequency signal from the second terminal 186b, and then connects the first terminal 186a and the third terminal 186c, The signal is output from the third terminal 186c. A first end 182a of the fifth transmission line 182 in the transmission direction of the high frequency signal is connected to the second terminal 186b. A first end 184a of the seventh transmission line 184 in the transmission direction of the high frequency signal is connected to the third terminal 186c. The radar device 196 may be realized by an FM-CW radar using a voltage controlled oscillator as an oscillator.

  The second end 182 b of the fifth transmission line 182 in the high-frequency signal transmission direction is connected to the transmission antenna 163.

  The receiving antenna 173 and the mixer 189 are connected by a sixth transmission line 183. The second end 184 b of the seventh transmission line 184 in the transmission direction of the high frequency signal is connected to the mixer 189.

  The high-frequency signal generated by the high-frequency oscillator 161 passes through the fourth transmission line 181 and the matching circuit 1, is given to the transmitting antenna 163 through the branching unit 186 and the fifth transmission line 182, and is transmitted from the transmitting antenna 163 to the radio wave. Is emitted as. The high-frequency signal generated by the high-frequency oscillator 161 passes through the fourth transmission line 181 and the matching circuit 1 and is given as a local signal to the mixer 189 via the branching unit 186 and the seventh transmission line 184.

  When receiving a radio wave coming from the outside by the receiving antenna 173, the receiving antenna 173 gives a high frequency signal based on the radio wave to the mixer 189 via the sixth transmission line 183.

The mixer 189 mixes the high frequency signals supplied from the sixth and seventh transmission lines 183 and 184 and outputs an intermediate frequency signal. The intermediate frequency signal output from the mixer 189 is given to the distance detector 191.

  The distance detector 191 includes the above-described high-frequency detector 171 and is based on the intermediate frequency signal obtained by receiving the radio wave (echo) radiated from the transmitter / receiver 195 and reflected by the measurement object, Calculate the distance to the measurement object. The distance detector 191 is realized by a microcomputer, for example.

  In the transmitter / receiver 195, the matching circuit 1 is inserted into the fourth transmission line 181 so that a high-frequency signal passes through the matching circuit 1, so that the transmitter / receiver 195 is not caused by the transmission line 162 due to, for example, variations in wiring width. By adjusting the phase of a high-frequency signal that changes as desired, for example, it is possible to realize a transceiver 195 having a stable oscillation characteristic and a high transmission output because the insertion loss is suppressed to a small value. A transceiver 195 having a detection characteristic and having a high detection output because the insertion loss is suppressed to a small level can be realized, and the reliability of an intermediate frequency signal generated by the mixer 189 can be improved, for example. .

  In the radar device 196, based on the intermediate frequency signal from the transmitter / receiver 195, the distance detector detects the distance from the transmitter / receiver 195 to the detection target, for example, the distance between the transmission and reception antennas 163 and 173 and the detection target. Therefore, the distance to the detection target can be accurately detected.

  The branching unit 186 may be realized by a hybrid circuit such as a directional coupler or a power divider. In this case, the high-frequency signal supplied to the fifth transmission line 182 branches to the second terminal 186b and the third terminal 186c. Is output. In this case, the power of the radio wave output from the transmitting antenna 163 is lower than that in the configuration described above, but the control of the apparatus is simplified because it is not necessary to control the branching device 186.

  In the present embodiment, the matching circuit 1 is inserted into the fourth transmission line 181, but in still another embodiment of the present invention, the matching circuit 1 is at least one of the fourth to seventh transmission lines 181 to 184. In addition, the high-frequency signal may be inserted so as to pass through the matching circuit 1. Even if it is such a structure, the same effect can be achieved.

  Further, although the matching circuit 1 is used in the transceiver 195, any one of the matching circuits of the above-described embodiments may be used. Even if comprised in this way, the same effect can be achieved.

1 is a perspective view showing a matching circuit 1 according to an embodiment of the present invention. 2 is a plan view showing a matching circuit 1. FIG. Is a graph showing the _{S 11} of S parameters. It is a top view which shows the matching circuit 40 of other embodiment of this invention. It is a top view which shows the matching circuit 41 of further another embodiment of this invention. It is a top view which shows the matching circuit 51 of further another embodiment of this invention. It is a top view which shows the matching circuit 61 of other embodiment of this invention. It is a top view which shows the matching circuit 84 of further another embodiment of this invention. It is a perspective view which shows the matching circuit 101 of other embodiment of this invention. It is a perspective view which shows the 1st phase shift circuit 122 of the matching circuit 121 of further another embodiment of this invention. It is a perspective view which shows the matching circuit 201 of further another embodiment of this Embodiment. Is a graph showing the _{S 11} of S parameters. It is a schematic diagram which shows the structure of the transmitter 160 of one Embodiment of this invention. It is a schematic diagram which shows the structure of the receiver 170 of one Embodiment of this invention. It is a schematic diagram which shows the structure of the radar apparatus 190 provided with the transmitter / receiver 180 of one Embodiment of this invention. It is a schematic diagram which shows the structure of the radar apparatus 196 provided with the transmitter / receiver 195 of other embodiment of this invention. It is a figure for demonstrating the reflection coefficient of a connection block, and a transmission coefficient. FIG.

Explanation of symbols

1, 40, 41, 51, 61, 84, 101, 121, 201 Matching circuit 2 Electronic component 3 Bonding wire 5 Dielectric substrate 6 First transmission line 7 Second transmission line 8 Third transmission line 11, 52, 62, 85, 102, 122 First phase shift circuit 12 Second phase shift circuit 13a One first connection end 13b The other first connection end 14, 42 Component placement table 15a One second connection end 15b The other second connection end 16, 43 Component placement portion 17 Back electrode 18 Connection portion 19 Branch portion 21, 53, 63, 86, 103 First input / output terminal 22, 54, 64, 87, 104 Second input / output terminal 23, 55, 65, 88 , 105 Third input / output terminal 24, 56, 66, 106 Fourth input / output terminal 25 Fifth input / output terminal 26 Rat race 27 First extending portion 27a Transmission line 27b First variable capacitance element 2 2nd extension part 28a Transmission line 28b 2nd variable capacity element 31 Ring-shaped transmission line 32 3rd variable capacity element 33 1st electrode 34 2nd electrode 35 3rd electrode 36 Reference electrode 39,93 DC blocking capacitor 44 Protrusion part 45 1st connection part 46 I / O terminal 48 2nd connection part 49 Projection part 57 Branch line 58 Square transmission line 67 Directional coupler 81 2nd reference electrode 89 Circulator 91 Extension part 91a Transmission line 91b 1st variable capacitance element 92 2nd variable capacitance element 107 Magic T
123a One of the pair of flat conductor portions 123b The other of the pair of flat conductor portions 160 Transmitter 161 High-frequency oscillator 162 Transmission line 170 Receiver 171 High-frequency detector 173 Reception antenna 180,195 Transceiver 181 Fourth transmission line 182 Fifth transmission Line 183 6th transmission line 184 7th transmission line 185 8th transmission line 186 Branching device 186a 1st terminal 186b 2nd terminal 186c 3rd terminal 187 Demultiplexer 187a 4th terminal 187b 5th terminal 187c 6th terminal 188 For transmission Antenna 189 Mixer 190, 196 Radar device 191 Distance detector 202 Impedance changing unit 203 Connector unit 204 Connection block 205 Remaining portion 206 Connection circuit

Claims (12)

A first transmission line;
A second transmission line;
Two first connection ends, one first connection end of the two first connection ends is connected to one end portion of the first transmission line, and the other end is connected to one end portion of the second transmission line A first phase shift circuit connected to the first connection end and capable of adjusting a phase of an electromagnetic wave passing therethrough by applying a voltage;
A third transmission line extending from the second transmission line and extending;
A second phase shift circuit having at least one second connection end, the second connection end being connected to one end of the third transmission line, and capable of adjusting a phase of an electromagnetic wave reflected by applying a voltage; Including
The first phase shift circuit includes:
A magic T having first to fourth input / output terminals, in which electromagnetic waves input from the first input / output terminals are distributed and output from the second and fourth input / output terminals;
A first variable capacitance element extending from the second input / output terminal and having an electric capacity that changes in accordance with an applied voltage is provided at an end opposite to the end connected to the second input / output terminal. 1 extension part,
A second variable capacitance element extending from the fourth input / output terminal and having an electric capacity that changes in accordance with an applied voltage is provided at an end opposite to the end to which the fourth input / output terminal is connected. And 2 extending parts,
The first transmission line is connected to the first input / output terminal;
The second transmission line is connected to the third input / output terminal;
The second phase shift circuit includes a third variable capacitance element whose electric capacity changes according to an applied voltage,
A first electrode connected to the first variable capacitance element for applying a voltage to the first variable capacitance element;
A second electrode connected to the second variable capacitance element for applying a voltage to the second variable capacitance element;
A matching circuit , further comprising a third electrode connected to the third variable capacitance element and for applying a voltage to the third variable capacitance element . A substrate having the first to third transmission lines and the first and second phase shift circuits, and having electrical insulation;
Formed on one surface in the thickness direction of the substrate, it has conductivity, and component placement unit electronic components are arranged,
On the one surface, a projecting portion that is formed to extend from the component placement portion toward a region where the first transmission line is provided, and has conductivity,
A back electrode formed on the other surface in the thickness direction of the substrate;
The matching circuit according to claim 1, further comprising a connecting portion that electrically connects the protruding portion and the back electrode. When an electronic component is electrically connected to the first transmission line and electromagnetic waves are input from the electronic component,
An electromagnetic wave input from the electronic component to the first transmission line is reflected by a connection coefficient that connects the electronic component and the first transmission line. ₁ age,
The magnitude of the transmission coefficient through which the electromagnetic wave input to the first transmission line passes to the branching portion where the third transmission line of the second transmission line branches | T _M1 │
The phase that changes when the electromagnetic wave input to the first transmission line travels to the branch portion is θ,
The reflection coefficient at which the electromagnetic wave input to the second transmission line is reflected by the branch portion is expressed as Γ ₂ Then,
Γ ₂ However, in a range where the amount of phase shift of the first and second phase shift circuits is variable,

The line lengths of the first and third transmission lines and the line length between the first phase shift circuit and the branching part of the second transmission line are selected so as to satisfy Expression (1). The matching circuit according to claim 1 or 2. The first transmission line is connected between a connection part to which one end of the connection body connecting the first transmission line and the electronic component is connected, and the one end part connected to the first connection end. The matching circuit according to claim 3 , further comprising an impedance changing unit having an impedance different from that of the connection part. When electromagnetic waves are input from the electronic component through the connection body,
The reflection coefficient of the electromagnetic wave input from the electronic component side to the connection block constituted by the connection body, the connector portion from the free end portion of the first transmission line to the impedance change portion, and the impedance change portion. Is Γ ₃ ,
Transmission through which electromagnetic waves input from the connection block to the remaining part of the first transmission line excluding the connector part and the impedance changing part pass to the branch part of the second transmission line where the third transmission line branches. When the magnitude of the coefficient is | T _M2 |
In the range where Γ ₃ is variable in the amount of phase shift of the first and second phase shift circuits,
| Γ ₃ | <| T _M2 | ² (2)
The line lengths of the first and third transmission lines, the impedance of the impedance changing unit, and the line length between the first phase shift circuit and the branching unit of the second transmission line are selected so as to satisfy Expression (2) The matching circuit according to claim 4 , wherein Reflection in which electromagnetic waves input from electronic components to the connection circuit constituted by the connection body and the matching circuit are reflected by the connection circuit within a range in which the phase shift amount of the first and second phase shift circuits is variable. The line length of the first and third transmission lines, the impedance of the impedance changing unit, and the line length between the first phase shift circuit and the branching unit of the second transmission line are selected so that the coefficient becomes 0 6. The matching circuit according to claim 5 , wherein: The line length between the fourth input / output terminal and the second variable capacitance element is equal to the line length between the second input / output terminal and the first variable capacitance element, and (2n-1) λ / 4. The matching circuit according to any one of claims 1 to 6 , wherein n is a natural number. A high-frequency oscillator that generates a high-frequency signal;
A transmission line connected to the high-frequency oscillator and transmitting a high-frequency signal from the high-frequency oscillator;
The matching circuit according to any one of claims 1 to 7 , which is inserted into the transmission line;
A transmitter comprising: an antenna connected to the transmission line and radiating a high frequency signal. An antenna that captures high-frequency signals;
A transmission line connected to the antenna and transmitting a high-frequency signal captured by the antenna;
The matching circuit according to any one of claims 1 to 7 , which is inserted into the transmission line;
And a high-frequency detector connected to the transmission line and detecting a high-frequency signal transmitted to the transmission line. A high-frequency oscillator that generates a high-frequency signal;
A fourth transmission line connected to the high-frequency oscillator for transmitting a high-frequency signal;
The first terminal is connected to the fourth transmission line, and a high-frequency signal applied to the first terminal is selectively applied to the second terminal or the third terminal. An output branching device;
A fifth transmission line connected to the second terminal and transmitting a high-frequency signal applied from the second terminal;
A fourth, fifth, and sixth terminal that outputs a high-frequency signal to the fourth terminal via the fifth transmission line, and outputs a high-frequency signal to the fifth terminal; A duplexer that outputs to the sixth terminal;
A sixth transmission line connected to the fifth terminal for transmitting a high-frequency signal output from the fifth terminal and transmitting a high-frequency signal to the fifth terminal;
An antenna connected to the sixth transmission line for radiating and capturing high-frequency signals;
A seventh transmission line connected to the third terminal and transmitting a high-frequency signal output from the third terminal;
An eighth transmission line connected to the sixth terminal and transmitting a high-frequency signal output from the sixth terminal;
A mixer that is connected to the seventh and eighth transmission lines, mixes high-frequency signals given from the seventh and eighth transmission lines, and outputs an intermediate frequency signal;
Transceiver characterized in that it comprises a matching circuit according to any one of claims 1 to 7 which is inserted into at least any one of the fourth to eighth transmission line. A high-frequency oscillator that generates a high-frequency signal;
A fourth transmission line connected to the high-frequency oscillator for transmitting a high-frequency signal;
The first terminal is connected to the fourth transmission line, and a high-frequency signal applied to the first terminal is selectively applied to the second terminal or the third terminal. An output branching device;
A fifth transmission line connected to the second terminal and transmitting a high-frequency signal applied from the second terminal;
A transmitting antenna connected to the fifth transmission line and emitting a high-frequency signal;
A receiving antenna for capturing high-frequency signals;
A sixth transmission line connected to the receiving antenna and transmitting the captured high-frequency signal;
A seventh transmission line connected to the third terminal and transmitting a high-frequency signal output from the third terminal;
A mixer that is connected to the sixth and seventh transmission lines, mixes high-frequency signals given from the sixth and seventh transmission lines, and outputs an intermediate frequency signal;
Transceiver characterized in that it comprises a matching circuit according to any one of claims 1 to 7 which is inserted into at least any one of the fourth to seventh transmission line. The transceiver according to claim 10 or 11 ,
A radar apparatus comprising: a distance detector that detects a distance from the transceiver to a detection target based on an intermediate frequency signal from the transceiver.

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