JP4966252B2 - Switching power supply - Google Patents

Description

  The present invention relates to a switching power supply device that drives a main switching element on / off based on a pulse width modulation (PWM) signal, converts an input voltage into a desired DC voltage, and outputs the voltage, and in particular, an output voltage by digital control. The present invention relates to a switching power supply device that controls the above.

  In recent years, there have been proposed a plurality of methods for realizing a highly intelligent switching power supply device by performing output voltage control using a digital processor. For example, as a switching power supply device that controls the output voltage by conventional digital control, the one disclosed in Patent Document 1 has been proposed.

In the switching power supply device of Patent Document 1, voltage conversion is performed by driving a switching element using a digital PWM signal created by a digital control circuit. Further, in this switching power supply device, an analog / digital conversion (hereinafter referred to as A / D conversion) circuit in the digital processor is used to control the output voltage of the switching power supply device to match the target value. The output voltage digital value is obtained by sampling the voltage at regular intervals and performing A / D conversion. The digital processor calculates the pulse width of the digital PWM signal by performing arithmetic processing from the output voltage digital value and the target value set in the digital processor, and digitally outputs the switching power supply device so that the output voltage becomes the target value. Feedback control by control is performed.
JP 2004-282916 A

  However, the switching power supply device having the configuration as in Patent Document 1 can perform high-speed processing, such as a digital signal processor (DSP), in order to solve the problems of output voltage setting accuracy and responsiveness to disturbance. It was necessary to use an expensive digital processor.

  First, the problem of output voltage setting accuracy in conventional digital control will be described below.

When output voltage control is performed by digital PWM, the resolution of the on-duty of the switching element is the resolution of the output voltage Vout. For example, when the output voltage Vout of a single-ended forward converter that is a switching power supply device that operates according to the equation (1) is set in increments of a% of the output voltage, the on-duty duty is expressed as Δduty shown in the equation (2). The resolution Rduty required there is expressed by equation (3).

  Here, Vout: output voltage, Vin: input voltage, N1: primary winding of transformer T1, N2: secondary winding of transformer T1, duty: on-duty of switching element, Rduty: resolution of duty.

  Taking specific numerical values as an example, the input voltage Vin = 48V, the output voltage Vout = 3.3V, the primary winding N1 = 6 turns of the transformer T1, and the secondary winding N2 = 1 turn of the transformer T1. In this case, if the step of the output voltage Vout is set to a value comparable to that of the analog-controlled switching power supply device, it is considered that a = 0.1% is necessary. At this time, from the equation (3), it is necessary for digital PWM. The resolution is Rduty≈2424.

  Usually, the digital PWM signal is generated based on the clock of the digital processor. Moreover, since a general-purpose switching power supply device in recent years is often set to a switching frequency of 500 kHz or more from the viewpoint of miniaturization of magnetic parts and the like, for example, when the switching frequency of the switching power supply device is 500 kHz, When calculating the clock of a digital processor capable of making the above-mentioned resolution digital PWM, it requires a clock frequency of 1.212 GHz or higher, which is a value obtained by multiplying the switching frequency of 500 kHz by the digital PWM resolution of 2424, and can operate at high speed. A high-performance digital processor is required.

  Next, a problem related to high-speed response of feedback control by digital control will be described.

  The switching power supply device is required to perform control to keep the output voltage Vout at a constant target value even when a disturbance in which the input voltage Vin suddenly increases occurs. For example, a single-ended forward converter that is a switching power supply device that operates according to Equation (1) performs control to reduce the on-duty duty of the switching element in accordance with Vin when the input voltage Vin increases rapidly. Thus, control is performed to keep the output voltage Vout at a constant target voltage, but the response speed becomes a problem. Specifically, in the switching power supply device, taking into account various processes in the digital processor, 50 clocks are calculated from the digital value and the target value to A / D convert the output voltage Vout to generate a digital value. It is considered that processing man-hours of about 200 clocks and a total of about 250 clocks are required to generate the digital PWM signal. Here, in order to obtain the same responsiveness as that of the analog-controlled switching power supply device, in order to complete these processes within one cycle of the switching operation, the clock frequency of the digital processor is set to 250 processing clocks and the switching frequency is 500 kHz. A clock frequency of 125 MHz or more, which is a value obtained by multiplying by, is required. In addition, since the digital processor performs various arithmetic processes in addition to the feedback control, it actually requires a clock frequency several times 125 MHz, and in this respect also, a high-speed and high-performance digital processor is required.

  Further, in the switching power supply device having the configuration as in Patent Document 1, in addition to the output voltage setting accuracy and the problem of responsiveness to disturbance, the output voltage caused by the switching operation of the switching element is used for sampling of the A / D converter circuit. There is a problem that an output voltage ripple having a frequency lower than the switching frequency is generated due to the influence of the switching ripple or the like which is a fluctuation or the influence of the quantization error of the A / D conversion circuit.

  The present invention has been made in view of the above-described background art, and includes a digital control circuit capable of realizing sufficient output voltage setting accuracy and high-speed response to disturbance using a low-speed and inexpensive general-purpose digital processor. Another object of the present invention is to provide a switching power supply device that can suppress output voltage ripple and obtain a stable and highly accurate output voltage.

  According to the present invention, a DC input voltage is switched by a drive pulse whose pulse width is modulated at a predetermined switching frequency, and an inverter circuit having a main switching element for generating an intermittent voltage, and an output voltage obtained by rectifying and smoothing the intermittent voltage. A rectifying and smoothing circuit to be obtained; a sawtooth wave generating circuit that sets the switching frequency and generates a sawtooth wave voltage having the frequency; and an analog / digital conversion circuit that converts a detected value of the output voltage into an output voltage digital value; Control pulse generation means for generating a control pulse voltage for controlling the output voltage by performing predetermined arithmetic processing based on the output voltage digital value, and generating a smoothed voltage by smoothing the control pulse voltage Comparing the smoothing voltage and the sawtooth wave voltage to the main smoothing circuit. A comparator circuit for outputting the drive pulse of the etching element, and the control pulse generating means adds a predetermined correction value to the output voltage digital value and compares the addition result with a predetermined target value. And when the addition result is higher than the target value, a low level voltage pulse is generated for a predetermined time, and when the addition result is lower than the target value, a high level voltage pulse is generated for a predetermined time. A pulse voltage generation unit that opens the output terminal to a high impedance when no voltage pulse is generated, and the predetermined correction value set in the calculation unit is determined by the addition result of the immediately previous calculation process being the target When it is higher than the value, it is stored as a negative value. When the addition result of the immediately preceding calculation process is lower than the target value, it is stored as a positive value. The calculation unit stores the corrected value. Is a switching power supply device which performs the next comparison operation using.

  The predetermined correction value set in the arithmetic unit is set to a magnitude corresponding to a value of half or more of a voltage fluctuation range generated by a ripple of the output voltage input to the analog / digital conversion circuit. Is. Further, the predetermined correction value set in the arithmetic unit may be set to a magnitude corresponding to a value equal to or greater than a quantization error width of the analog / digital conversion circuit.

  The rectifying / smoothing circuit may include a synchronous rectifying circuit including a switching element that is turned on / off in synchronization with the main switching element.

The sawtooth wave generation circuit may generate a rising portion of the sawtooth wave voltage with a slope proportional to the input voltage.

  According to the switching power supply device of the present invention, in a highly intelligent digitally controlled switching power supply device, it is possible to suppress the occurrence of low frequency ripple of the output voltage due to switching ripple or the like in a simple manner. Stable and highly accurate output voltage can be obtained by suppressing the above. Furthermore, it is possible to realize a switching power supply device that has a high-speed response to a disturbance and can set an output voltage with high accuracy even when configured using a low-speed clock and an inexpensive digital processor. it can.

  Hereinafter, a switching power supply device 10 according to an embodiment of the present invention will be described with reference to FIGS. As shown in FIG. 1, the switching power supply 10 is connected to a DC input power supply Ein, and performs rectification and smoothing of an inverter circuit 12 that performs a switching operation and an intermittent voltage generated by the switching operation to obtain an output voltage Vout. A smoothing circuit 14 is provided, and an output voltage Vout terminal is connected to the load 22.

  An output voltage detection circuit 15 is connected to the output voltage Vout terminal, an output of the output voltage detection circuit 15 is connected to an A / D conversion circuit 16, and an output of the A / D conversion circuit 16 is supplied to a control pulse generator 18. It is connected. The output voltage detection circuit 15 is, for example, a filter circuit composed of a resistor and a capacitor. The output voltage detection circuit 15 operates to reduce a ripple voltage superimposed on the output voltage Vout, or the output voltage Vout falls below the allowable input voltage of the A / D conversion circuit 16. In this way, the voltage is divided. The control pulse generation means 18 includes a calculation unit 18a and a pulse voltage generation unit 18b, and generates a predetermined control pulse voltage Va based on an output voltage digital value Vdig (n) that is an output signal of the A / D conversion circuit 16. To do. The output terminal 18c of the control pulse generator 18 is input to a pulse smoothing circuit 20 that smoothes the control pulse voltage Va and outputs a smoothed voltage Vs.

  Further, a sawtooth wave generation circuit 22 that generates a sawtooth wave voltage Vi that increases the voltage with a certain slope and determines the switching frequency of the main switching element TR1 is provided. Then, a smoothing voltage Vs and a sawtooth voltage Vi are input, a comparison circuit 24 is provided for comparing them and generating a main switching element drive pulse Vg. The output of the comparison circuit 24 is the main switching of the inverter circuit 12. It is connected to the control terminal of the element TR1.

  Next, details of the configuration will be described for each functional block. The inverter circuit 12 is connected to the primary winding T1a of the transformer T1 and the main switching element TR1 in series with the input power source Ein, and is intermittently connected to the secondary winding T1b of the transformer T1 by the on / off operation of the main switching element TR1. A voltage Vb is generated. The polarity of each winding of the transformer T1 is configured in a well-known single-ended forward system.

  A rectifying / smoothing circuit 14 is connected to the secondary winding T1b of the transformer T1. The rectifying / smoothing circuit 14 includes a forward rectifying element TR2 that conducts a pulse current when the main switching element TR1 is on, and a flywheel side rectifying element TR3 that is turned on / off complementarily with the forward rectifying element TR2. A synchronous rectification drive circuit 14a that drives the rectifying elements TR2 and TR3 in synchronization with the on / off operation of the main switching element TR1 and a smoothing circuit in which a choke coil Lo and a capacitor Co are connected in series are configured. The rectifying / smoothing circuit 14 rectifies and smoothes the voltage induced in the secondary winding T1b to generate an output voltage Vout at both ends of the smoothing capacitor Co, and power is supplied to the load 22 connected to both ends of the capacitor Co. The The rectifying elements TR2 and TR3 are preferably, for example, MOS-FETs or the like, and the rectification loss can be suppressed to a small value by using an element having a conduction resistance as small as possible.

  The A / D conversion circuit 16 receives an output voltage Vout, which is an analog value, via the output voltage detection circuit 15, converts the output voltage Vout to an output voltage digital value Vdig (n), and outputs it at a predetermined sampling frequency. An internal switch (not shown) provided at the input end of the A / D conversion circuit 16 is turned on for a short time, and an internal capacitor (not shown) is charged. Then, the voltage is quantized with a predetermined number of bits and corresponds to Vout (n). Output as output voltage digital value Vdig (n). This sampling frequency is relatively much lower than the switching frequency set by the oscillator 22a of the sawtooth wave generation circuit 22 described later.

  The control pulse generation means 18 is configured using, for example, a digital processor, and includes a calculation unit 18a and a pulse voltage generation unit 18b as shown in FIG. The calculation unit 18a corrects a correction value VK (n), which is a digital value, based on a correction amount ΔK, which is a predetermined digital value, and a comparison calculation result D (n) output by a digital comparison unit DCP described later. A value determining unit 32 and an adding unit 34 for adding the determined correction value VK (n) and the output voltage digital value Vdig (n) obtained from the A / D conversion circuit 16 are provided. Furthermore, digital comparison means DCP for comparing and calculating the addition result Vsu (n) and the target value C which is a predetermined digital value is provided, and the comparison calculation result D is directed toward the pulse voltage generation unit 18b and the correction value determination means 32. (N) is output. Here, the correction amount ΔK is set to, for example, a value that is half or more of the amplitude of the ripple component of the output voltage Vout that is input to the A / D conversion circuit 16 via the output voltage detection circuit 15. If the ripple component is sufficiently small, the ripple component may be set to a value equal to or greater than the quantization error of the A / D conversion circuit 16.

  The correction value determination means 32 performs the following operation as shown in the flowchart of FIG. For example, when the comparison result D (n) indicating that the addition result Vsu (n) is lower than the target value C is obtained in the n-th sampling, the next output voltage digital value Vdig (n + 1) in the addition means 34 is obtained. The correction value VK (n + 1) to be added to is determined as + ΔK. Conversely, when a comparison operation result D (n) indicating that the addition result Vsu (n) is higher than the target value C is obtained, a correction value to be added to the output voltage digital value Vdig (n + 1) in the next adding means 34. VK (n + 1) is determined to be −ΔK.

  The pulse voltage generator 18b includes switch elements S1 and S2 connected in series to the DC power supply Vcc2 and switch drive means 36 for driving the switch elements S1 and S2 based on the comparison calculation result D (n). The midpoint of the elements S1 and S2 is connected to the output terminal 18c.

  When the addition result Vsu (n) is lower than the target value C, the switch driving unit 36 turns on the switch element S1 for a predetermined time and then turns it off. At that time, the other switch element S2 is maintained in the OFF state. On the contrary, when the addition result Vsu (n) is higher than the target value C, the switch element S2 is turned on for a certain time and then turned off. At that time, the switch element S1 is maintained in the OFF state. That is, when the addition result Vsu (n) is higher than the target value C, the pulse voltage generation unit 18b generates a low level voltage (zero potential) for a predetermined time at the output terminal 18c, and when the addition result Vsu (n) is lower than the target value C. During the period in which a high level voltage (DC power supply Vcc2 potential) for a certain time is generated at the output terminal 18c and neither high nor low voltage is generated, the output terminal 18c is opened to a high impedance. Note that the potential of the output terminal 18c when the switch elements S1 and S2 are both opened is the charging potential of the capacitor C2 of the pulse smoothing circuit 20 described below.

  As shown in FIG. 1, the pulse smoothing circuit 20 forms a so-called integrating circuit by a resistor R2 and a capacitor C2. The integration circuit smoothes the control pulse voltage Va and outputs a smoothed voltage Vs that is a predetermined DC voltage. Accordingly, the pulse smoothing circuit 20 increases the smoothing voltage Vs when the control pulse voltage Va outputs a high level voltage pulse, and conversely outputs the low level voltage pulse when the control pulse voltage Va outputs a low level voltage pulse. During the period in which none of the voltage pulses is generated, the operation is performed so as to hold the previous smoothed voltage Vs.

  The sawtooth wave generation circuit 22 includes a constant voltage DC power source Vcc1, a charging resistor R1 having one end connected to the DC power source Vcc1, a timer capacitor C1 connected between the other end of the charging resistor R1 and the ground, and a timer capacitor. A reset element S3 connected to both ends of C1 and an oscillator 22a for controlling the reset element S3 are provided, and a sawtooth voltage Vi generated at both ends of the timer capacitor C1 is output.

  The oscillator 22a generates an impulse-like trigger pulse. The trigger pulse is a repetitive pulse having a constant period, and determines the switching frequency of the main switching element TR1 and the turn-on timing of the main switching element TR1. In addition, the reset element S3 has a function of short-circuiting both ends of the timer capacitor C1 when a trigger pulse is input, and is instantaneously opened, and continues to be opened until the next trigger pulse is input. Yes.

  The sawtooth wave generation circuit 22 configured as described above operates as follows. A trigger pulse is input from the oscillator 22a, the reset element S3 short-circuits both ends of the timer capacitor C1, and the electric charge is discharged so that Vi≈0. Further, the reset element S3 is instantaneously opened, and the charging current supplied via the charging resistor R1 flows into the timer capacitor C1, and Vi increases. At this time, since the time constant of the series circuit of the charging resistor R1 and the timer capacitor C1 is set to a value sufficiently larger than the cycle of the trigger pulse, Vi increases linearly with a substantially constant slope. Thereafter, when the next trigger pulse is input, the reset element S3 is short-circuited and the above operation is repeated. By such an operation, the sawtooth voltage Vi is generated at both ends of the timer capacitor C1.

  The comparison circuit 24 includes a comparator CP1 that receives the smoothing voltage Vs on the non-inverting input side, receives the sawtooth voltage Vi on the inverting input side, and outputs a drive pulse Vg for the main switching element TR1. The driving pulse Vg indicates a high level when the smoothing voltage Vs is higher than the sawtooth voltage Vi, and turns on the main switching element TR1, and in the opposite case, indicates a low level and turns off the main switching element TR1. is there.

  Next, the pulse width control (PWM control) operation of the switching power supply 10 will be described with reference to FIG. Here, Tsw: switching cycle of the switching power supply device 10, Ton1: ON time of the switching element TR1 when the input voltage Vin is low, duty1: ON duty of the switching element TR1 when the input voltage Vin is low, Ton2: input voltage ON time of the switching element TR1 when Vin is low, duty2: ON duty of the switching element TR1 when the input voltage Vin is low. When the switching power supply 10 operates ideally, the switching power supply 10 adjusts the time ratio of the drive pulse Vg of the main switching element TR1 so that the addition result Vsu (n) becomes equal to the target value C in the control pulse generating means 18. Thereby, the output voltage Vout is controlled.

  Since this switching power supply device is a synchronous rectification single-ended forward converter, the output voltage Vout is determined as shown in the above-described equation (1). As can be seen from Equation (1), for example, even if the input voltage Vin changes, the output voltage Vout becomes constant if the on-duty duty is changed in inverse proportion to the input voltage Vin. Specifically, as shown in the time chart of FIG. 4, in the period 1 where the input voltage Vin is low, the smoothing voltage Vs is increased to increase the on-duty as duty 1, and conversely, as in period 2 During period 2 when input power supply voltage Vin is high, control is performed so that the product of on-duty duty and input voltage Vin is always constant by lowering smoothing voltage Vs and reducing on-duty as duty2. The

  Here, the resolution of the output voltage control of the switching power supply device 10 will be described. In the switching power supply device 10, since the on-duty of the switching element TR1 is controlled by the smoothing voltage Vs, the resolution of the smoothing voltage Vs smoothed by the resistor R2 and the capacitor C2 is the resolution of the on-duty duty. decide.

For example, when the calculation unit 18a determines that the output voltage Vout should be increased, the output terminal 18c is supplied with a voltage pulse of Vcc2, which is a high level voltage pulse. Assuming that the smoothing voltage Vs immediately before the supply of the pulse voltage of Vcc2 to the output terminal 18b is Vs ₀ and the pulse voltage time width of Vcc2 is ts1, the smoothing voltage Vs immediately after the elapse of ts1 is expressed by Equation (4). Assuming that ts1 is small, Vs does not change greatly, so it is approximately calculated by equation (5).

When the calculation unit 18a determines that the output voltage Vout should be reduced, the output terminal 18c is short-circuited to a zero potential pulse that is a low level voltage pulse. When the smoothed voltage Vs immediately before the output terminal 18b outputs the zero potential pulse is Vs ₀ , and the pulse time width is ts2, the averaged voltage Vs immediately after the ts2 elapses is expressed by Expression (6), and ts2 is If it is assumed that Vs is small, Vs does not change greatly, and therefore is approximately calculated by Expression (7).

  As shown in equations (5) and (7), the amount of change in the smoothing voltage Vs is determined by the values of R2, C2, ts1, and ts2. The minimum values of ts1 and ts2 are determined depending on the processing speed of the digital processor to be used. By appropriately setting the values of the resistor R2 and the capacitor C2, the smoothing voltage Vs can be changed in a minute step. Even if the clock frequency of the digital processor used for the control pulse generating means 18 is low, a relatively high resolution can be obtained as compared with the background art. That is, the resolution of the smoothing voltage Vs, that is, the resolution of the on-duty duty is not affected by the clock frequency of the digital processor unlike the conventional digital control switching power supply. As described above, the voltage control of the switching power supply device 10 of the present embodiment makes it possible to obtain a high resolution by using a low-cost digital processor with a low-speed clock.

  Here, in the switching power supply that does not use this embodiment, a low-frequency ripple having a period Trip longer than the period Tsw is newly generated due to the influence of the switching ripple and noise of the period Tsw of the switching frequency superimposed on the output voltage Vout. appear.

  A general switching power supply device is required to control so that the DC component Vave of the output voltage Vout becomes a constant value. However, as can be seen from the operation of the A / D conversion circuit 16 described above, when a switching ripple component is superimposed on the output voltage Vout, the A / D conversion circuit 16 outputs the output voltage Vout (n) including the ripple component. Therefore, an error occurs in the control of the DC component Vave, and this error causes a low-frequency output voltage ripple.

  First, an operation for minimizing the influence of this error in the switching power supply device 10 of the present embodiment will be described.Next, in order to explain the effect of the switching power supply device 10 of this embodiment, A case will be described in which a control pulse generation means 40 having no correction value determination means 32 and addition means 34 is used in place of the control pulse generation means 18.

  Below, the operation | movement in the switching power supply device 10 of this embodiment is demonstrated based on the time chart of FIG. Note that the output voltage digital value Vdig (n), the correction amount ΔK, the correction value VK (n), the addition result Vsu (n), and the target value C are digital values, but for convenience of explanation, the same as the DC component Vave and the like. An analog value is illustrated. The correction amount ΔK is set to a value slightly larger than half the value corresponding to the amplitude of the switching ripple component of the output voltage Vout input to the A / D conversion circuit 16 via the output voltage detection circuit 15. Has been.

  At the first sampling timing, the A / D conversion circuit 16 samples the output voltage digital value Vdig (1) including the ripple component of the output voltage Vout indicated by the broken line. Here, for example, the DC component Vave is a value higher than the target value C in the calculation unit 18a of the control pulse generating means 18, but is sampled at a value lower than the target value C due to fluctuation due to ripple. Further, the correction value VK (1) is determined to be + ΔK based on the previous comparison calculation result D (0) (not shown). Therefore, the DC component Vave at this time is a value higher than the target value C, and the output voltage digital value Vdig (1) is a value lower than the target value C, but the correction value VK (1) is + ΔK, The addition result Vsu (1) becomes higher than the target value C by absorbing the fluctuation due to the ripple component. Accordingly, the switch element S2 of the pulse voltage generation unit 18b is turned on by the operation of the digital comparison unit DCP and the switch driving unit 36, and the averaged voltage Vs decreases by ΔVs. As a result, the DC component Vave is reduced according to the control target. Decrease by ΔV1. Even when the output voltage digital value Vdig (1) is higher than the target value C at the sampling timing, the addition result Vsu (1) is higher than the target value C, so the above control is performed. .

  At the second sampling timing, the output voltage digital value Vdig (2) including the ripple component is sampled at a value higher than the target value C. On the other hand, since the correction value VK (2) is determined to be −ΔK based on the previous comparison calculation result D (1), the addition result Vsu (2) is lower than the target value C. By the operation of the comparison unit DCP and the switch driving unit 36, the switch element S1 of the pulse voltage generation unit 18b is turned on, the average voltage Vs increases by ΔVs, and as a result, the DC component Vave increases by ΔV1 according to the control target. . Also in this case, the addition result Vsu (2) is lower than the target value C even when the output voltage digital value Vdig (2) is lower than the target value C depending on the sampling timing. Done.

  At the third sampling timing, the output voltage digital value Vdig (3) including the ripple component is sampled at a value lower than the target value C. On the other hand, since the correction value VK (3) is determined to be + ΔK based on the previous comparison calculation result D (2), the addition result Vsu (3) is higher than the target value C. Due to the operation of the means DCP and the switch driving means 36, the switch element S2 of the pulse voltage generator 18b is turned on, the averaged voltage Vs is reduced by ΔVs, and as a result, the direct current component Vave is reduced by ΔV1 according to the control target. This operation is the same as the operation in the first sampling.

  Since the switching power supply 10 performs control so that the average value Vave of the output voltage becomes the target value C of the output voltage, the ripple component of the output voltage included in Vdig (n) sets the target value C of the output voltage. The amplitude is centered.

  That is, the operation of the control pulse generator 18 of this embodiment is based on the previous comparison calculation result D with a correction value VK (n) greater than half the amplitude of the ripple component of the output voltage included in Vdig (n). Thus, the output is increased or decreased by forcibly sandwiching the target value C, and the influence of the switching ripple voltage on the sampling of the A / D conversion circuit 16 is eliminated. Then, by repeating the above operation, as shown in FIG. 5, the DC component Vave of the output voltage is suppressed to a small voltage ripple having a cycle Tsa that is twice the sampling cycle Tsa in the steady state.

  Next, in order to explain the effect of the switching power supply device 10 of this embodiment, as a comparison object, instead of the control pulse generating means 18, a control pulse generating means 40 having no correction value determining means 32 and adding means 34 is used. The case where it is used will be described. First, the configuration of the control pulse generator 40 will be described with reference to FIG. Here, the same components as those of the control pulse generator 18 are denoted by the same reference numerals, and description thereof is omitted.

  As shown in FIG. 6, the control pulse generating means 40 includes a calculation unit 40a and a pulse voltage generation unit 18b. The calculation unit 40a compares the output voltage digital value Vdig (n) obtained from the A / D conversion circuit 16 with a predetermined target value C set by the digital value by the digital comparison unit DCP, and generates a pulse voltage generation unit. The comparison calculation result D (n) is output toward 18b. The configuration and functions other than the calculation unit 40a are the same as those of the control pulse generating means 18.

  The operation of the switching power supply device 10 using the control pulse generating means 40 will be described based on the time chart of FIG.

  At the first sampling timing, the DC component Vave is higher than the target value C, and the output voltage digital value Vdig (1) including the ripple component is also sampled at a value higher than the target value C. Accordingly, the switch element S2 of the pulse voltage generation unit 18b is turned on by the operation of the digital comparison unit DCP and the switch driving unit 36, and the averaged voltage Vs decreases by ΔVs. As a result, the DC component Vave is reduced according to the control target. Decrease by ΔV1.

  At the second sampling timing, the DC component Vave is lower than the target value C, but the output voltage digital value Vdig (2) including the ripple component is sampled at a value higher than the target value C. The switch drive means 36 turns on the switch element S2 of the pulse voltage generator 18b, and the averaged voltage Vs decreases by ΔVs. As a result, the control of the direct current component Vave decreases the direct current component Vave by ΔV1, contrary to the target direction.

  At the third sampling timing, the DC component Vave is a value lower than the target value C, and the output voltage digital value Vdig (1) including the ripple component is also sampled at a value lower than the target value C. Accordingly, the switch element S1 of the pulse voltage generation unit 18b is turned on by the digital comparison unit DCP and the switch driving unit 36, and the averaged voltage Vs is increased by ΔVs. As a result, the DC component Vave is only ΔV1 according to the control target. To rise.

  As a result of repeating the above operation, as shown in FIG. 7, in this example, a low frequency ripple Trip having a period four times the sampling period Tsa is generated in the direct current component Vave of the output voltage. In FIG. 7, the switching elements S1 and S2 are turned on alternately two times continuously. However, the mutual relationship between the amplitude and phase of the switching ripple component, the switching period Tsw, the sampling period Tsa, and the like changes. In this case, the number of times that one switch element is continuously turned on may exceed, for example, 10 times. The low-frequency ripple has a period Trip of the low-frequency ripple and an amplitude that increases as the number of times one of the switch elements S1 and S2 is continuously turned on. The smoothing circuit of the choke coil Lo and the capacitor Co set for reducing the ripple can hardly be attenuated, which is a big problem.

  On the other hand, when the control pulse generator 18 according to the embodiment of the present invention is used, as described above, a low frequency ripple having a period twice as long as the sampling period Tsa of the output voltage Vave is generated. However, as described above, the correction amount ΔK is more than half the amplitude of the output ripple component of Vdig (n) generated from the output voltage Vout input to the A / D conversion circuit 16 via the output voltage detection circuit 15. Therefore, in a steady state, the switching elements S1 and S2 are alternately turned on, and the low-frequency ripple cycle Trip is not longer than this. The amplitude can be kept stable with a relatively small value. Further, by adjusting the values of the choke coil Lo and the capacitor Co and setting the sampling frequency to an appropriate frequency, it is possible to reduce the amplitude of the low frequency ripple to a level that can be almost ignored.

  In the case of control using the control pulse generating means 18 described above, for example, when the output voltage digital value Vdig (n) is outside the range of the target value C ± ΔK, the control pulse voltage Va is at a high level. Either the voltage or the low level voltage is continuous, and the operation is performed so that the output voltage digital value Vdig (n) converges within the range of the target value C ± ΔK. Therefore, for example, there is no adverse effect on the control of the output voltage Vout when the switching power supply device 10 is activated and the responsiveness to the fluctuation of the output voltage Vout caused by the fluctuation of the load current.

  Further, in the switching power supply device 10 using the control pulse generating means 40 of the comparative example, an analog signal is generated by the A / D converter circuit, in addition to the problem that the low frequency ripple is generated due to the switching ripple component described above. There is a problem that low-frequency ripple occurs due to a quantization error occurring at every timing of a sampling period for converting the signal into a digital signal. However, also for this problem, if the control using the control pulse generator 18 is performed, the low frequency ripple is suppressed by the same action.

  Next, another embodiment of the sawtooth wave generation circuit 22 in the switching power supply device 10 according to one embodiment of the present invention will be described. As shown in FIG. 8, in the sawtooth wave generation circuit 42 of this embodiment, one end of the charging resistor R1 opposite to the timer capacitor C1 side is connected to the input voltage Vin, and the sawtooth wave voltage output terminal 42b to the timer capacitor C1 is also provided. Is different from the sawtooth wave generation circuit 22 in that a voltage generation element Rb is inserted between the two.

  Since the constant is set so that a constant current J1 substantially proportional to the input voltage Vin flows in the charging resistor R1, the sawtooth voltage Vi increases linearly with a slope substantially proportional to the input voltage Vin. The voltage generating element Rb is an element that generates a voltage in proportion to the current J1, and uses, for example, a resistor. Since the current J1 has a value proportional to the input voltage, a voltage proportional to the input voltage Vin is generated in the voltage generating element Rb.

  The sawtooth wave generating circuit 42 operates so that the rising speed of the sawtooth wave voltage Vi generated at both ends of the timer capacitor C1 is slow when the input voltage Vin is low. Since the on-duty of the switching element TR1 is generated by comparing the sawtooth voltage Vi and the smoothed voltage Vs by the comparison circuit 24, when the smoothed voltage Vs is a constant value, the switching is performed when the input voltage Vin is low. The on-duty of the element TR1 becomes relatively wide. On the contrary, when the input voltage Vin is high, the operation is performed so that the rising speed of the sawtooth voltage Vi generated at both ends of the timer capacitor C1 increases, and when the smoothing voltage Vs is a constant value, the switching element TR1 is turned on / off. Duty becomes relatively narrow. With this operation, even when the smoothed voltage Vs is constant, the on-duty with respect to the input voltage Vin is controlled. Therefore, even if the input voltage fluctuates rapidly due to a disturbance, the output voltage is not affected.

  The voltage generating element Rb is an element for superimposing a DC voltage corresponding to the input voltage Vin on the sawtooth voltage Vi, and the ratio of the delay time of the comparison circuit 24 to the on-duty of the switching element TR1 is input. The operation of correcting the difference depending on the voltage can be performed, and the operation accuracy of the feedforward control can be improved.

  The above feedforward control is particularly effective in a switching power supply device that is always PWM controlled based on the formula (1) regardless of the load current. For example, when the rectifying / smoothing circuit 14 is configured by a diode element, when the current is equal to or lower than a predetermined load current, a current discontinuous mode in which the current of the choke coil Lo is not continuous is established, and the relationship of Expression (1) does not hold, and the control mode is Change nonlinearly. At this time, in order to keep the output voltage of the switching power supply 10 at the target value, the control pulse generating means 18 needs to perform a process of reducing the smoothed voltage Vs. On the other hand, in the case of a synchronous rectification circuit using a MOS-FET, the operation is always performed in the continuous current mode in which the current of the choke coil Lo is continuous regardless of the load current, and the relationship of the expression (1) is always established. Therefore, by applying the feedforward control to the switching power supply device using the synchronous rectifier circuit, the control pulse generating means 18 is applied over the entire range in which the load current changes in order to keep the output voltage of the switching power supply device 10 at the target value. Can control the smoothing voltage Vs to a substantially constant value, and can further improve the stability of the output voltage Vout.

  A technique for achieving high-speed response of output voltage control by using the sawtooth wave generation circuit 42 that responds to a change in the input voltage Vin exists as a kind of feedforward control applied to an analog-controlled switching power supply device. . For example, it is the same as that used for the analog-controlled switching power supply device described in Japanese Patent Application No. 2006-313089 by the present inventor. However, there is no case where the present invention is applied to a digitally controlled switching power supply device, and in conventional general digital control, PWM control itself is often performed by digital arithmetic processing. When applying feed-forward control by digital arithmetic processing, a large number of arithmetic processing steps are required, and a high-speed digital processor is required to follow a rapid change in input voltage.

  On the other hand, in the switching power supply of this embodiment, feedforward control can be easily applied to a switching power supply controlled by a low-speed digital processor.

  The present invention is not limited to the above embodiment, and can be applied to a push-pull method, a bridge method, a flyback method, various chopper methods, and the like as long as the inverter circuit 12 performs pulse width control. The main switching element TR1 is not limited to the MOS-FET, and may be a semiconductor element that can be driven by a pulse signal, such as a bipolar transistor or IGBT.

  The control pulse generator 18 and the A / D converter circuit 16 preferably operate synchronously. However, the A / D converter circuit 16 outputs the output voltage digital value Vdig (n) in a state where the control pulse generator 18 can operate. ) Can be output. If the low-frequency ripple of the switching power supply 10 satisfies a desired value, the correction amount ΔK set in the calculation unit 18a of the control pulse generating means 18 is set via the output voltage detection circuit 15 to the A / D conversion circuit 16. The value may be smaller than half the amplitude of the output ripple component of Vdig (n) generated from the output voltage Vout input to the output voltage Vout, and the operating state of the switching power supply device (the target of the input voltage Vin and output voltage Vout) Value, load current state, etc.). Furthermore, the correction amount ΔK may be changed depending on whether the correction value VK is set to a positive value or a negative value.

  Further, the pulse smoothing circuit 20 is not limited to the configuration of the primary filter constituted by one resistor R2 and one capacitor C2, but may be the configuration of a second-order or higher-order multi-order filter. It may be a passive filter or an active filter. The frequency characteristic of the smoothing may be set to an optimum one according to the environment (input condition, load condition, etc.) in which the switching power supply device is used.

It is a functional block diagram which shows one Embodiment of the switching power supply device of this invention. It is a circuit diagram which shows the internal structure of the control pulse generation means of this embodiment. It is a flowchart which shows the process of the control pulse generation means of this embodiment. It is a time chart explaining the ideal operation | movement of this embodiment. It is a time chart explaining the actual operation | movement of this embodiment. It is a circuit diagram which shows the internal structure of the control pulse generation means which does not have the correction value determination means etc. of the control pulse generation means of this embodiment. It is a time chart explaining operation | movement of a switching power supply device at the time of using the control pulse generation circuit which does not have the correction value determination means etc. of the control pulse generation means of this embodiment. It is a circuit diagram which shows the structure of other embodiment of the sawtooth wave generation circuit of this embodiment.

Explanation of symbols

DESCRIPTION OF SYMBOLS 10 Switching power supply device 12 Inverter circuit 14 Rectification smoothing circuit 16 A / D conversion circuit 18 Control pulse generation means 18a Operation part 18b Pulse voltage generation part 20, 44 Pulse smoothing circuit 44a Smoothing voltage fluctuation width compression circuit 22, 42 Sawtooth wave Generation circuit 32 Correction value determining means 34 Adding means TR1 Main switching element TR2 Forward side rectifying element TR3 Flywheel side rectifying element

Claims (5)

An inverter circuit having a main switching element that switches a DC input voltage with a drive pulse that is pulse-width modulated at a predetermined switching frequency and generates an intermittent voltage;
A rectifying / smoothing circuit for rectifying and smoothing the intermittent voltage to obtain an output voltage;
A sawtooth wave generating circuit that sets the switching frequency and generates a sawtooth voltage having the frequency;
An analog / digital conversion circuit for converting the detected value of the output voltage into an output voltage digital value;
Control pulse generating means for generating a control pulse voltage for controlling the output voltage by performing a predetermined calculation process based on the output voltage digital value;
A pulse smoothing circuit that smoothes the control pulse voltage to generate a smoothed voltage;
A comparison circuit that compares the smoothed voltage with the sawtooth voltage and outputs the drive pulse of the main switching element;
The control pulse generating means includes
A calculation unit that adds a predetermined correction value to the output voltage digital value and compares the addition result with a predetermined target value;
Based on the output of the arithmetic unit, a low level voltage pulse is generated for a predetermined time when the addition result is higher than the target value, and a high level voltage pulse is generated for a predetermined time when the result is lower than the target value. A pulse voltage generator that opens the output terminal to a high impedance when
The predetermined correction value set in the calculation unit is stored as a negative value when the addition result of the immediately preceding calculation process is higher than the target value, and the addition result of the immediately preceding calculation process is the target value. The switching power supply device is characterized in that a lower value is stored as a positive value, and the arithmetic unit performs a next comparison operation using the stored correction value.   The predetermined correction value set in the arithmetic unit is set to a magnitude corresponding to a value of half or more of a voltage fluctuation range generated by a ripple of the output voltage input to the analog / digital conversion circuit. The switching power supply device according to claim 1.   2. The switching according to claim 1, wherein the predetermined correction value set in the arithmetic unit is set to a magnitude corresponding to a value equal to or larger than a quantization error width of the analog / digital conversion circuit. Power supply.   4. The switching power supply apparatus according to claim 1, wherein the rectifying / smoothing circuit includes a synchronous rectifying circuit including a switching element that is turned on / off in synchronization with the main switching element. 4. The switching power supply device according to claim 1, wherein the sawtooth wave generating circuit generates a rising portion of the sawtooth wave voltage with a slope proportional to the input voltage.

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