JP4510566B2 - Digital converter and control method thereof - Google Patents

Description

  The present invention relates to a digital converter using a step-up chopper, and more particularly to an apparatus that can realize precise PWM control even when a general coil is used, and a control method therefor.

  Various proposals have been made as fully digitalized single-phase converters. For example, in Non-Patent Document 1, only the input current is is detected and the current control gain Kp is varied to detect the phase of the power supply voltage. There is also described a method of making the input current sinusoidal and controlling the amplitude. In principle, the ratio of the ON time of the switching element is defined as a flow rate d, and d = 1−Kp × ABS (is), and is = SQR (2) × Vs × SIN (ωt) / (Kp × Ed ).

  In the above equation, Kp is the current control gain, is is the instantaneous value of the input current, and Ed is the output DC voltage. ABS is an absolute value symbol, SQR is a root symbol, and SIN (ωt) is a sine wave having an angular frequency ω.

An input current is-current control gain Kp curve is created and used as a parameter corresponding to fluctuations in the input current.
Journal of the Institute of Electrical Engineers of Japan 2003 I-273-I-276

  However, the current control gain characteristics differ from coil to coil, and many common coils have non-linear characteristics in the current superposition characteristics of inductance values (see FIG. 6), so it is realistic to realize the above system. In addition, it is difficult to control the current and voltage stably because the ripple of the input current increases. In addition, the above system has a problem that it is difficult to specify the current control gain Kp in a region where the input current is small.

  The present invention has been made in view of the above problems, and is a digital converter and control capable of precise control over a wide range of input current while having a simple configuration in which an inexpensive general coil can be used. It aims to provide a method.

  In order to achieve the above object, the present invention provides a step-up chopper provided with a coil and a switching element, a rectifier circuit for supplying an input current to the coil, and a computer circuit for PWM-controlling the switching element in a predetermined control cycle. In a digital converter having a PWM, PWM control is performed with a different algorithm based on the determination result while determining whether the input current to the coil is a continuous mode in which the current is not interrupted during the control cycle or a discontinuous mode in which the current is interrupted in the middle of the control cycle It is characterized by that.

  In the present invention, the determination of the continuous mode or the discontinuous mode may be performed in every control cycle, or may be performed once in a plurality of control cycles. In any case, it is preferable to determine whether the current mode is the continuous mode or the discontinuous mode. In this control cycle, the coil charging start current Iv (n−2), the AC input voltage Vac (n−1) to the boost chopper, And the measured value of the DC output voltage Vdc (n−1) of the step-up chopper, the control time Ton (n−1) of the PMW wave in the current control cycle, and the inductance value of the coil.

  The control time of the PMW wave means the control on time Ton (n−1) in the embodiment, but is not particularly limited. If the OFF time of the switching element is PWM controlled, the control time is controlled. This is the off time. Further, the inductance value of the coil is preferably corrected in accordance with the measured value of the coil current.

  In the present invention, preferably, the control time Ton (n) of the PMW wave should be determined using an arithmetic expression corresponding to the determination result of the continuous mode or the discontinuous mode. More preferably, the control time Ton (n) of the PMW wave in the next control cycle is determined using the predicted value Iav (n) of the coil average current in the next control cycle as an arithmetic element. Here, the predicted value Iav (n) of the coil average current is determined by the integration of the predicted value Vac (n) of the AC input voltage and the integration parameter β, and the integration parameter β is the target value of the DC output voltage Vdc. It is preferably determined by PI (Proportional-Integral) control based on a deviation Verr from Vo.

  The present invention is also a control method for realizing the operation of each of the above inventions. In other words, the present invention is a control method for a digital converter including the technical elements according to claims 1 to 6.

  According to the present invention described above, it is possible to perform precise PWM control over a wide range of input current with a simple configuration. Further, since all can be realized by software control, a power factor correction circuit as in the case of hardware control can be omitted, and a general coil can be used, so that the entire apparatus can be downsized at low cost.

  Furthermore, according to the present invention, since control by instantaneous response is possible, it is possible to cope linearly with load fluctuations. Further, since control by feedforward is the main, divergence with respect to the target value can be suppressed, and control delay is small.

  Hereinafter, the present invention will be described in more detail based on examples. FIG. 1 is a circuit configuration diagram showing a digital converter under software control. The digital converter of the embodiment is incorporated as a part of a motor control system. That is, in the converter 1 of this embodiment, a single-phase AC voltage (for example, 200 V) is rectified by the full-wave rectifier circuit 2 to become a pulsating current, and then boosted by PWM (Pulse Width Modulation) control by the one-chip microcomputer 3 The chopper 4 converts the voltage into a predetermined DC voltage Vdc (for example, 350 V). The three-phase motor M is driven by an inverter circuit 5 controlled by the one-chip microcomputer 3.

  The step-up chopper 4 includes a coil L, a capacitor C, a switching element Q, and a shunt resistor r for current detection. The coil L, the capacitor C, and a shunt resistor r are connected in series, and the capacitor C is short-circuited to the capacitor C. Switching elements Q to be connected are connected in parallel. The input current of the boost chopper 4 flows toward the coil L and returns through the shunt resistor r. In this embodiment, an insulated gate bipolar transistor IGBT (Insulated Gate Bipolar Transistor) is used as the switching element Q.

  As shown in the figure, the input current I, the input AC voltage Vac, and the output DC voltage Vdc are input to the one-chip microcomputer 3 through the signal input units IN1 to IN3, respectively, and the built-in A / D converters AD1 to AD4 are used. , Each has been digitally converted. The A / D converter AD1 and the A / D converter AD4 acquire the same input current I at different timings. Here, the signal input unit IN1 is composed of an OP amplifier circuit that receives the voltage across the shunt resistor r, and functions as a current detection sensor together with the shunt resistor r. Further, the signal input unit IN2 and the signal input unit IN3 are configured by a resistance voltage dividing circuit and an OP amplifier amplifier circuit.

  The data acquired through the signal input units IN1 to IN3 and the A / D converters AD1 to AD4 are processed by the one-chip microcomputer 3 to calculate the PWM signal ON time (hereinafter referred to as control on time). The generated PWM signal is supplied from the output port PT to the gate of the switching element Q through the signal output unit OUT1. The one-chip microcomputer 3 is connected to the inverter circuit 5 through the signal input unit IN4 and the signal output unit OUT2, and controls the three-phase motor M by inverter.

  Hereinafter, the operation principle of this embodiment in which the switching element Q is ON / OFF controlled by the PWM wave will be described. FIG. 2 is a time chart illustrating the relationship between the PWM wave output from the one-chip microcomputer 3 and the current flowing through the coil L. As shown in the figure, a triangular wave due to the coil charging current and the coil discharging current flows through the coil L, but the energy stored in the coil L is sufficient and the continuous current flows (see FIG. 2A). Since there is insufficient energy, there is a discontinuous mode (see FIG. 2B) in which the current is interrupted in the middle.

  In this embodiment, since different PWM control is performed depending on which operation mode is in use, first, a determination formula for determining the operation mode will be described. In the time chart of FIG. 2, it is assumed that the current time is the control cycle (n−1). Then, it is considered that the control on time Ton (n) in the next control cycle (n) is determined based on the measured value in the control cycle (n−1). Note that the frequency of the AC input voltage is 50 Hz or 60 Hz, but in order to control it sufficiently quickly, in this embodiment, the control cycle T is set to 50 μS.

The circuit equation during coil charging (switching element ON) is
Vac (n−1) = L × {Ip (n−1) −Iv (n−2)} / Ton (n−1) (Expression 1) Here, Iv (n-2) is a coil charging start current, Ip (n-1) is a coil charging peak current, and Ton (n-1) is a control ON time, and each value in the control cycle (n-1). It is. Vac (n-1) is an input voltage in the control cycle (n-1). Since the control cycle (50 μS) is sufficiently short with respect to the power supply frequency, Vac (n-1) is regarded as a constant value. Can do.

On the other hand, the circuit equation at the time of coil discharge (switching element OFF) is
Vdc (n−1) −Vac (n−1) = L × {Ip (n−1) −Iv (n−1)} / Toff (n−1) (Expression 2) Here, Vdc (n-1) is the voltage across the capacitor C, Iv (n-1) is the coil current at the end of the current control cycle (coil charging start current of the next control cycle), Toff (n-1). Are OFF times, which are values in the control cycle (n−1), respectively.

If Ip (n-1) is eliminated from (Equation 1) and (Equation 2) and Iv (n-1) is solved,
Iv (n−1) = Iv (n−2) + T / L × [{Vac (n−1) −Vdc (n−1)} + Ton (n−1) × Vdc (n−1)]... (Equation 3) The control period T is T = Ton (n−1) + Toff (n−1).

  In the above (Formula 3), if Iv (n−1)> 0, the continuous mode is selected, and if Iv (n−1) = 0, the discontinuous mode is selected. However, (Formula 3) uses the coil charging start current Iv (n-2) in the current control cycle (n-1), and the coil charging start current Iv (n-1) in the next control cycle (n). Therefore, if the charging start current Iv (n−2) is not accurate, the determination of the continuous mode or the discontinuous mode will be wrong. That is, if Iv (n−2) is determined by a prediction calculation based on the control on time (Ton (n−2)) in the immediately preceding control cycle, an error is accumulated by the calculation of (Equation 3). As a result, the control instruction value itself may diverge from the target. Therefore, in this embodiment, the input current ad1 at the start of coil charging is measured for each control cycle.

  However, since a time delay Ts inevitably occurs from the start of the control cycle to the time of acquisition of the input current (see FIG. 4C), the coil charge is started by correcting the measured value ad1 in consideration of this time delay Ts. The current is Iv (n−2). Assuming that the current measurement value at the timing Ts of the control cycle (n−1) is ad1, the slope (Δi / Δt) of the coil charging current is Δi / Δt = e / from the relationship of L × Δi / Δt = e. L. Here, e is voltage, i is current, and t is time.

Therefore, the current increase amount at the time delay Ts can be calculated as Vac (n−1) / L × Ts, and using this value, Iv (n−2) = ad1−Vac (n−1) / L × Ts (Expression 4) And if this (Equation 4) is substituted into (Equation 3),
Iv (n-1) = ad1- [Vac (n-1) * Ts + T * {Vdc (n-1) -Vac (n-1)}-Ton (n-1) * Vdc (n-1)] / L (Expression 5), and the input current value Iv (n−1) at the final timing of the current control cycle (n−1) (= start timing of the next control cycle) is expressed as the current control cycle (n -1) can be accurately determined based on the measured value ad1 of the input current at the start timing.

  Then, depending on whether or not the coil charging start current Iv (n-1) of the next control cycle obtained in this way is positive, it is possible to accurately determine whether it is a continuous mode or a discontinuous mode, and an optimal control corresponding to that mode. Is possible. That is, the measured values of the AC input voltage Vac (n-1), the DC output voltage Vdc (n-1), and the input current ad1 in the current control cycle (n-1) and the previous control cycle are determined. On the basis of the control on time Ton (n−1), it can be determined whether the control for the continuous mode or the control for the discontinuous mode should be performed. In addition, the calculation procedure of the above (Formula 1)-(Formula 5) is supplementary description to FIGS.

  By the way, the DC output voltage Vdc (n−1) does not necessarily need to be updated every control cycle. Therefore, in this embodiment, Vdc (i) whose value is updated every 1 mS is used ( (See step ST30 in FIG. 5). Therefore, the discriminant of the present embodiment is, exactly, Iv (n-1) = ad1- [Vac (n-1) * Ts + T * {Vdc (i) -Vac (n-1)}-Ton (n −1) × Vdc (i)] / L (Expression 5 ′).

  Furthermore, the average value Vdc for the past 0.5 seconds calculated by the process of step ST38 in FIG. 5 may be used as the DC output voltage. In this case, Iv (n−1) = ad1− [Vac (n−1) × Ts + T × {Vdc−Vac (n−1)} − Ton (n−1) × Vdc] / L. The discriminant of equation 5 ″) is adopted.

<Discontinuous mode>
Next, a method for calculating the control on time Ton (n) based on the coil average current Iav during each control cycle will be described. First, the control on time Ton (n) in the discontinuous mode is calculated (see FIG. 2B).

  The circuit equation at the time of charging the coil is Vac (n) = L × {Ip (n) −Iv (n−1)} / Ton (n), but Iv (n−1) = 0 because of the discontinuous mode. In the end, Vac (n) = L × Ip (n) / Ton (n) (Formula 6). Here, Vac (n) is the AC input voltage, Ip (n) is the current value at the coil charging peak, Iv (n−1) is the current value at the start of coil charging, and Ton (n) is the control on time. .

On the other hand, the circuit equation at the time of coil discharge is Vdc (n) −Vac (n) = L / Tcut (n) × {Ip (n) −Iv (n)}. Tcut (n) is a time until the current value Ip (n) in the coil charge peak state is discharged and becomes zero (see FIG. 2B). Since the circuit equation in the discontinuous mode is considered here, Iv (n) = 0, and Vdc (n) −Vac (n) = L / Tcut (n) × Ip (n) (Expression 7) ) The average value Iav (n) of the input current in this control cycle is
Iav (n) = {Ip (n) × Ton (n) + Ip (n) × Tcut (n)} / (2 × T) (Expression 8).

And solving these (Equation 6) to (Equation 8) for Ton (n),
Ton (n) × Ton (n) = {2 × T × L × Iav (n) × (Vdc (n) −Vac (n))} / {Vac (n) × Vdc (n)} Equation 9) is calculated. In addition, the calculation process of (Formula 9) was shown in FIGS.

<Continuous mode>
Subsequently, the control ON time Ton (n) in the continuous mode is calculated (see FIG. 2A). The circuit equation during coil charging is Vac (n) = L × {Ip (n) −Iv (n−1)} / Ton (n) (Equation 10) On the other hand, the circuit equation at the time of coil discharge is Vdc (n) −Vac (n) = L / Toff (n) × {Ip (n) −Iv (n)} (Equation 11). Here, Toff (n) = T−Ton (n), which is the time from the start of coil discharge to the start of coil charge in the next control cycle.

The average current Iav (n) in this control cycle is Iav (n) =
[{Ip (n) + Iv (n−1)} × Ton (n) + {Ip (n) + Iv (n)} × Toff (n)] / {2 × T} (Expression 12) . Here, solving (Equation 10) to (Equation 12) for Toff (n) while erasing Ip (n) and Iv (n),
Toff (n) × Toff (n) = [2 × T × L × {Iv (n−1) −Iav (n)} / Vdc (n)] + T × T × Vac (n) / Vdc (n) · Since (Equation 13) is obtained, Ton (n) is eventually calculated as Ton = T−Toff (n) (Equation 14). Note that (Equation 14) and other calculation processes are supplemented in FIGS.

  In this embodiment, the control on-time Ton (n) is calculated using (Equation 9) or (Equation 14) according to the discontinuous mode or the continuous mode. In n), prediction parameters of the AC input voltage Vac (n), the DC output voltage Vdc (n), and the average input current Iav (n) are required.

The AC input voltage Vac (n) is predicted based on the current AC input voltage measurement value Vac (n-1) and the previous AC input voltage measurement value Vac (n-2). The difference between the control cycle (n-2) and the control cycle (n-1) measurement value is added to the current measurement value Vac (n-1) as follows.
Vac (n) = 2 × Vac (n−1) −Vac (n−2) (Equation 15)
On the other hand, for the DC output voltage Vdc (n), an average value Vdc of past measurement values for the DC voltage is adopted. Although the calculation method of the average value Vdc is determined as appropriate, in this embodiment, the measured values for the past 0.5 seconds are averaged by the averaging process executed every 0.5 seconds to obtain the DC output voltage Vdc. (See step ST38 in FIG. 5). This DC output voltage Vdc is stored in an appropriate work area of the memory, and the value Vdc of this work area is updated every 0.5 seconds.

  Therefore, in this case, in the discontinuous mode, Ton (n) × Ton (n) = {2 × T × L × Iav (n) × (Vdc−Vac (n))} / {Vac (n) × On the other hand, in continuous mode, Toff (n) × Toff (n) = [2 × T × L × {Iv (n−1) −Iav (n)} / Vdc. ] + T × T × Vac (n) / Vdc (Equation 13 ′), Ton = T−Toff (n) (Equation 14 ′).

  However, instead of using the average value Vdc for 0.5 seconds, the value of Vdc (i) obtained from the output value ad3 of the AD converter AD3 every 1 mS may be used. In this case, in the discontinuous mode, Ton (n) × Ton (n) = {2 × T × L × Iav (n) × (Vdc (i) −Vac (n))} / {Vac (n) × Vdc (i)} (Equation 9 ″) On the other hand, in continuous mode, Toff (n) × Toff (n) = [2 × T × L × {Iv (n−1) −Iav ( n)} / Vdc (i)] + T × T × Vac (n) / Vdc (i) (Equation 13 ″), Ton = T−Toff (n) (Equation 14 ″) Become. In FIG. 3, for convenience, the case where Vdc (i) is used is indicated by a solid line, and the case where the DC output voltage Vdc is used is indicated by a broken line.

  The predicted value of the average input current Iav (n) is Iav (n) = β × Vac (n) from the relationship with the predicted value of the AC input voltage Vac (n). Here, the gain β is adjusted by PI control so that the difference Verr becomes zero while comparing the reference value (target value) Vo of the DC output voltage with the above-mentioned averaged DC output voltage Vdc.

  That is, Verr = Vo−Vdc (Expression 16), and Vo = Vac (pk) + α (Expression 17). Here, the reference value Vo of the DC output voltage is set to a value obtained by adding the amount of boost α by the coil L to the peak value Vac (pk) of the AC input voltage (pulsating flow). By setting in this way, conversion with high efficiency according to the input voltage value is possible. In addition, since the amount of pressure boosted by the coil L can be reduced, it is possible to select an inexpensive and lightweight coil with an appropriate size that does not increase in size. The optimum inductance value of the coil L is generally determined based on a design formula of L = Vac × Vac × (Vdc−Vac) / {γ × Pac × Vdc / T}. Since the output voltage value Vdc is set in correspondence with the voltage value Vac, the inductance value of the coil always maintains an almost optimum value. In the above design formula, γ is the ripple content of the input current, T is the control period, Pac is the maximum input power, and Vac is the instantaneous value of the input voltage.

  FIG. 3 is a control block diagram for explaining the above-described control operation. 4 to 5 are flowcharts showing the processing contents of the one-chip microcomputer 3 that realizes the control operation of FIG. The control process shown in FIG. 3 includes a first timer interrupt INT1 (FIG. 4A) activated every 50 μS, a second timer interrupt INT2 (FIG. 5) activated every 1 mS, and a first timer interrupt INT1. The AD conversion end interrupt INT3 (FIG. 4B) which is activated when the AD conversion operation started in step S1 is completed is mainly configured. Note that the first timer and the second timer are asynchronous, and the first timer interrupt INT1 is set to a higher priority than the second timer interrupt INT2.

<First timer interrupt INT1>
Hereinafter, the operation content of the one-chip microcomputer 3 will be described with reference to FIGS. 4 and 5. As shown in FIG. 4A, when the first timer interrupt INT1 occurs, the register signal saving process (POP process) is finished, and then the PWM signal is raised to the H level and the previous control cycle (n The countdown operation of the set value Ton (n-1) of the PWM timer determined in -1) is started (ST1). This down-count operation thereafter proceeds independently from the processing of FIGS. 4 and 5, and PWM control is realized by returning the PWM signal to the L level after a predetermined time Ton (n−1) has elapsed. (See FIG. 4 (c)).

  Next, the one-chip microcomputer 3 operates the 4ch AD converters AD1 to AD4 in the continuous scan mode to start AD conversion processing, and ends the interrupt processing (ST2). When the AD conversion process executed in the continuous scan mode is completed, the CPU of the one-chip microcomputer 3 is set to receive an AD conversion end interrupt. As shown in FIG. 4C, there is a time delay Ts from the start of the control cycle by the interrupt of the first timer to the start of the first AD conversion, and from the start of the first AD conversion. There is a delay time Td until the start of the fourth AD conversion. The time delay Ts is, for example, about 2.2 μS, and the delay time Td is, for example, about 20 μS.

<AD conversion end interrupt INT3>
Thereafter, when the AD conversion operation is completed for all the AD converters, the control operation is executed by the interrupt process INT3 shown in FIG. First, the output values ad1, ad4 (input current to the coil L) of the AD converters AD1, AD4 are acquired (ST10). Next, the average value Iav (n−1) of the input current in the control cycle (n−1) is calculated by the average calculation (ad1 + ad4) / 2 (ST11). As shown in FIG. 4C, the input current value varies depending on the sampling point, so the accuracy as the average value (average current) of the input current is not high, but this average value Iav (n−1) Is used only for the correction of the inductance value described below, so that no particular problem occurs.

  When the average value Iav (n−1) of the input current is obtained, the inductance value of the coil L mounted on the circuit is specified based on the current value Iav (n−1). As shown in FIG. 6, the coil L often has an inductance value that changes in accordance with a DC superimposed current (average current) flowing therethrough. Therefore, in this embodiment, the characteristics of the coil L mounted in the circuit are stored in the memory in advance, and the inductance value corresponding to the average value Iav (n−1) of the input current is used in each arithmetic expression. I am doing so.

  Therefore, highly accurate control can be performed without using a large and expensive coil. In order to further increase the control accuracy, the current value Iav (n−1) obtained by the average calculation (ad1 + ad4) / 2 is used as the control on time Ton (n−1) in the control cycle (n−1). It may be corrected accordingly (see FIG. 3). That is, the two sampling points ad1 and ad4 may be divided into the coil charging current ad1 and the coil discharging current ad4, or the control on time Ton (n−1) may be long, and both may be the coil charging current. Therefore (see FIG. 4C), if the average current is corrected based on this point, more precise control can be realized.

  Subsequently, the output value ad2 (AC input voltage Vac (n-1)) of the AD converter AD2 is acquired (ST13). Then, Vac (n) is calculated based on the voltage prediction formula Vac (n) = 2 × Vac (n−1) −Vac (n−2) (ST14). Next, the output value ad3 (DC output voltage Vdc (n-1)) of the AD converter AD3 is acquired (ST15).

  Subsequently, the input current command value Iav (n) is calculated by calculating Iav (n) = β × Vac (n) (ST16). Note that. The necessary value of the integration parameter β is updated every 1 mS in the second interrupt process INT2 shown in FIG. 5 and stored in an appropriate work area.

  Next, the acquired value ad1 from the AD converter AD1, the inductance value L of the coil corrected in the process of step ST12, the AC input voltage value Vac (n−1) acquired in the process of step ST13, and this control Based on the control ON time Ton (n−1) in the cycle and the DC output voltage value Vdc (i), the coil charging start current Iv (n−1) is calculated based on the discriminant of (Expression 5 ′). (ST17). Then, according to the value (whether positive or not) of the coil charging start current Iv (n−1), it is determined whether the control should be controlled as the continuous mode or the discontinuous mode (ST17). As described above, the discriminant of (Expression 5 ″) may be used instead of (Expression 5 ′).

  If Iv (n−1) ≦ 0 and the discontinuous mode is set, Iv (n−1) = 0 is set (ST19), (Equation 9 ′) or (Equation 9 ″). ) To calculate the control on time Ton (n) (ST20). On the other hand, if Iv (n−1)> and the continuous mode is set, the control on-time Ton (n) is calculated based on the arithmetic expression of (Expression 14 ′) (Expression 14 ″) (ST21). ).

  Then, on the condition that the calculated control on time Ton (n) does not exceed the control upper limit value and the control lower limit value, the PWM timer uses Ton (n) as the control on time of the next control cycle (n). ) Is set (ST22, 23). When the calculated control ON time Ton (n) exceeds the upper limit value or the lower limit value, the control upper limit value or the control lower limit value is set in the PWM timer, respectively.

<Second timer interrupt INT2>
Next, the second timer interrupt INT2 started every 1 mS independently of the above-described first timer interrupt processing INT1 and AD conversion end interrupt INT3 will be described based on the flowchart of FIG.

  In the second timer interrupt INT2, first, Vdc (i) that is the output of the AD converter AD3 is acquired (ST30). The AD converter AD3 performs an AD conversion operation every 50 μS, but the second timer interrupt INT2 acquires the value of the output DC voltage Vdc (i) output based on the first timer interrupt INT1. .

  Next, the calculation of SUM ← SUM + Vdc (i) is executed, and the acquired value of the output DC voltage Vdc (i) is added to the average calculation buffer SUM in the work area (ST31). Further, the AC input voltage Vac (i), which is the output of the AD converter AD2, is acquired (ST32), and the AC input voltage Vac (i) is compared with the peak value Vac (pk) stored in the memory. (ST33).

  If Vac (i)> Vac (pk), the maximum wave height value Vac (pk) stored in the memory is updated by calculating Vac (pk) ← Vac (i) (ST34). After obtaining the peak value Vac (pk) of the AC input voltage in this way, the counter CT is decremented (−1) (ST35), and it is determined whether or not the counter value CT is zero (ST36).

  Here, when the counter value CT becomes CT = 0, the average value of the output DC voltage Vdc (i) is obtained by multiplying the value of the average calculation buffer SUM by 1/500 (ST37). Then, the DC output voltage Vdc is specified by this average value (ST38).

  When the DC output voltage Vdc is obtained in this way, the values of the average calculation buffer SUM and the counter CT are initialized (ST39), and the reference value (target value) Vo of the output DC voltage is calculated by calculating Vo ← Vac (pk) + α. Is calculated (ST40). α is a boosted amount in the coil L when compared with the peak value Vac (pk) of the input AC voltage. Then, the difference between the output reference voltage Vo and the output average voltage Vdc obtained from the measured value is calculated (ST41). Specifically, Verr (i) ← Vo−Vdc is calculated.

  Based on the above result, the command value β by the PI control is calculated, and the second timer interrupt processing INT2 is finished (ST42). Here, the calculation of the command value β is based on an arithmetic expression of β = Verr (i) × Kp + {Verr (i) × Ki + Verr (i−1) ′ × Ki}, where Verr (i−1) ′ × Ki is , The previous integration control value (i−1), which is a value calculated as Verr (i−1) ′ × Ki = Verr (i−1) × Ki + Verr (i−2) ′ × Ki.

  The contents of the control processing described above are summarized in FIG. 3 as a control block diagram, and the processing without INT1 and INT2 means the processing INT3 of the AD conversion end interrupt (FIG. 4B). Yes.

  As described above, the digital converter 1 according to the embodiment has no hardware control circuit and can be realized by software control. Therefore, the power factor correction circuit and the like can be omitted, and the size can be reduced at low cost. In addition, since a common coil is used and its inductance value is corrected in real time and reflected in the arithmetic expression, high-precision control is possible at a low cost. Furthermore, in this embodiment, since control is performed by instantaneous response, it is possible to cope linearly with load fluctuations. Further, as shown in (Equation 9) and (Equation 14), since control based on feedforward is mainly performed, divergence with respect to the target value can be suppressed and control delay is small.

It is a circuit block diagram which shows the digital converter which concerns on an Example. It is a time chart explaining the relationship between a control period and the charging / discharging operation | movement of a coil. It is a control block diagram explaining the control operation of the digital converter which concerns on an Example. It is a flowchart which shows the operation | movement content of a one-chip microcomputer. It is a flowchart which shows the operation | movement content of a one-chip microcomputer. It is a characteristic view which shows the relationship between the inductance value of a coil, and an average current. It is drawing explaining the derivation | leading-out process of (Formula 5). It is drawing explaining the derivation | leading-out process of (Formula 5). It is drawing explaining the derivation | leading-out process of (Formula 9). It is drawing explaining the derivation | leading-out process of (Formula 9). It is drawing explaining the derivation | leading-out process of (Formula 14). It is drawing explaining the derivation | leading-out process of (Formula 14).

Explanation of symbols

L Coil Q Switching element (IGBT)
4 Boost chopper 2 Rectifier circuit (Single-phase full-wave rectifier circuit)
3 Computer circuit (one-chip microcomputer)
1 Digital converter

Claims (7)

In a digital converter having a step-up chopper provided with a coil and a switching element, a rectifier circuit that supplies an input current to the coil, and a computer circuit that performs PWM control of the switching element in a predetermined control cycle,
Digital that performs PWM control with different algorithms based on the determination result while determining whether the input current to the coil is a continuous mode that is not interrupted during the control cycle or a discontinuous mode that is interrupted during the control cycle converter. Whether the current mode is the continuous mode or the discontinuous mode is determined by measuring the coil charging start current, the AC input voltage to the boost chopper, and the DC output voltage of the boost chopper in the current control cycle, and the PMW wave in the current control cycle. The digital converter according to claim 1, wherein the digital converter is determined based on a control time of the coil and an inductance value of the coil. The digital converter according to claim 2, wherein the inductance value is corrected corresponding to a measured value of the coil current. The digital converter according to any one of claims 1 to 3, wherein the control time Ton (n) of the PMW wave is determined using an arithmetic expression corresponding to a determination result of the continuous mode or the discontinuous mode. The control time Ton (n) of the PMW wave in the next control cycle is determined by using the predicted value Iav (n) of the coil average current in the next control cycle as an arithmetic element. The digital converter described. The predicted value Iav (n) of the coil average current is determined by the integration of the predicted value Vac (n) of the AC input voltage and the integration parameter β.
The digital converter according to claim 5, wherein the integration parameter β is determined by PI control based on a deviation Verr of the DC output voltage Vdc from the target value Vo. A method for controlling a digital converter that realizes the operation according to claim 1.