JP4173460B2 - Digital broadcast receiver - Google Patents

Description

  The present invention relates to a digital broadcast receiving apparatus, and more specifically, equalization that compensates for distortion caused by multipath in a received signal of an orthogonal frequency division multiplexing (OFDM) transmission method in demodulation of digital terrestrial broadcasting. Regarding the circuit.

  In recent years, OFDM transmission systems have attracted attention in digital audio broadcasting for mobile terminals and terrestrial digital television broadcasting.

  This OFDM transmission scheme is a scheme in which a large number of subcarriers (hereinafter also referred to as subcarriers) orthogonal to each other are modulated with digital data to be transmitted, and these modulated waves are multiplexed and transmitted. The OFDM transmission system has a feature that when the number of subcarriers used is as large as several hundred to several thousand, the symbol period of each modulated wave becomes extremely long, so that it is not easily affected by multipath interference.

  FIG. 14 is a waveform diagram for explaining an OFDM modulated signal.

  Referring to FIG. 14, in the OFDM transmission system, transmission data is distributed and modulated in hundreds to thousands of subcarriers, so that the modulation symbol rate of each subcarrier is extremely low, and one symbol period is extremely long. Become.

  Furthermore, by setting a guard period (hereinafter also referred to as a guard interval) before the effective symbol period, it is possible to effectively remove the influence of distortion due to multipath.

  As shown in FIG. 14, the guard interval G2 is formed by cyclically copying the second half portion G2 'of the effective symbol period (S2 + G2'). If the delay time of multipath interference is within the guard period, it is possible to prevent intersymbol interference due to delayed adjacent symbols by demodulating only the signal in the effective symbol period during demodulation.

  In the case of Japanese terrestrial digital TV broadcasting, there are three types of effective symbol period lengths. When the effective symbol period length is expressed by the number of subcarriers, mode 1 is 2048, mode 2 is 4096, and mode 3 is 8192.

  Also, the frequency band of Fast Fourier Transform (FFT) applied in one effective symbol period is set to, for example, 8.192 MHz in any mode. Since this frequency band is obtained by multiplying the basic subcarrier frequency by the number of subcarriers, the basic subcarrier frequency in each mode is 4 kHz, 2 kHz, and 1 kHz in Mode 1, Mode 2, and Mode 3, respectively. Therefore, the effective symbol period lengths of mode 1, mode 2, and mode 3 are 250 μs, 500 μs, and 1 ms, respectively.

  There are four types of guard period lengths. When the guard period length is hereinafter referred to as guard interval length, there are four types of guard interval lengths of 1/4, 1/8, 1/16, and 1/32 of the effective symbol period length.

  In the OFDM transmission method using the QAM modulation method for the modulation method of each subcarrier, when distortion due to multipath occurs, the amplitude and phase of each subcarrier differ from the amplitude and phase on the transmission side. For this reason, in the receiving apparatus, it is necessary to equalize (compensate) the signal affected by the distortion due to multipath so that they are equal. In the OFDM transmission system, the receiving side performs FFT (Fast Fourier Transform) processing and performs demodulation, so that pilot signals are scattered in the transmission signal, and the amplitude and phase of the pilot signal are monitored on the receiving side. The transmission path characteristics are estimated, and the received signal is equalized based on the estimated transmission path characteristics.

  FIG. 15 is a diagram showing an example of a pilot signal insertion pattern proposed in terrestrial digital television broadcasting.

  Referring to FIG. 15, pilot carrier signals are inserted in the frequency direction of one OFDM symbol at a ratio of one to twelve carriers. Further, in the time direction, for each OFDM symbol, the pilot carrier signal insertion position is shifted by three carriers. A pilot signal having such an arrangement is also referred to as a scattered pilot signal, and will be represented using the symbol SP below.

  FIG. 16 is a block diagram showing a configuration of a conventional digital broadcast receiving apparatus described in Patent Document 1, for example. This figure shows a part relating to equalization of an OFDM reception signal extracted from the overall configuration of the digital broadcast receiving apparatus.

  Referring to FIG. 16, an OFDM signal received via antenna 800 is converted into an intermediate frequency signal (IF signal) by tuner 802 and output to multipliers 804 and 806. The multipliers 804 and 806 are further supplied with the carrier waves generated by the carrier wave generation circuit 808 and having a phase difference of 90 degrees. This carrier wave is generated from the outputs of the multipliers 804 and 806 corresponding to the phase error detected by the FFT window circuit 812 using the guard interval correlation.

  Multipliers 804 and 806 multiply the intermediate frequency signal of the OFDM signal from tuner 802 and the carrier wave from carrier wave generation circuit 808, respectively. The multiplication result is given to the FFT circuit 810 as a baseband OFDM signal. The FFT circuit 810 performs FFT processing on the input baseband OFDM signal and demodulates the OFDM signal.

  The pilot signal extraction circuit 900 extracts a pilot signal from the output of the FFT circuit 810 and outputs it to the interpolation filter 902. The interpolation filter 902 performs interpolation processing on the extracted pilot signal, and estimates the amplitude and phase component of each carrier as the transmission path characteristics of the carrier. The estimated transmission line characteristics are given to the division circuit 904.

  The division circuit 904 divides the demodulated signal input from the FFT circuit 810 by the amplitude and phase supplied from the interpolation filter 902 as transmission path characteristics, and removes distortion components with respect to the transmission path characteristics. For example, when the amplitude of the carrier wave input from the FFT circuit 810 is ½ of the original amplitude, ½ is supplied as amplitude information from the interpolation filter 902. Therefore, if the division circuit 904 divides the amplitude of the signal input from the FFT circuit 810 by the amplitude information of the interpolation filter, the original signal having the amplitude of 1 (= (1/2) / (1/2)) is obtained. Obtainable. Similarly, the signal of the original phase can be obtained by performing a complex operation on the phase.

  The demapping circuit 818 demaps the signal point of the signal output from the division circuit 904. The TPS detection circuit 816 detects a transmission control signal (TPS: Transfer Parameter Signal) included in the signal output from the FFT circuit 810, detects information on the modulation scheme of the OFDM signal from the transmission control signal, and the detection result Is output to the demapping circuit 818. The demapping circuit 818 performs demapping processing corresponding to the modulation scheme information from the TPS detection circuit 816 and outputs the processing result.

  Here, as described above, four types of 1/4, 1/8, 1/16, and 1/32 of the guard interval length are defined as a ratio to the effective symbol period length. For this reason, the interpolation filter 902 is configured such that the bandwidth of the interpolation filter 902 is ¼ that has the longest guard interval so that equalization processing can be performed regardless of the length of the guard interval signal received. It was fixed to.

  However, when the interpolation filter 902 receives an OFDM signal whose guard interval length is shorter than ¼, the interpolation filter 902 processes a band of signal components that are not originally required, and there are many noise components accompanying the signal. Therefore, there is a problem that accurate transmission path estimation processing cannot be performed.

  Therefore, in the receiving apparatus described in Patent Document 1, as shown in FIG. 16, the FFT window circuit 812 detects the guard interval length, outputs the detection signal to the control circuit 820, and the control circuit 820 detects the guard interval length. It is configured to control the bandwidth of the interpolation filter 902 in response to the signal.

By adopting such a configuration, the interpolation filter 902 processes only the minimum necessary pilot signal by adjusting the bandwidth to the guard interval length of the received signal, and the influence of noise components in unnecessary bands is affected. Can be reduced.
Japanese Patent Laid-Open No. 11-239115

  Here, consider a case where a conventional digital broadcast receiving apparatus receives and receives an OFDM signal. In this case, the amount of fluctuation in the time direction varies depending on the moving speed of the apparatus main body. That is, the fluctuation in the time direction is severe when moving at high speed, and is gradual when moving at low speed.

  For this reason, in the transmission path estimation in the equalization circuit, if the interpolation filter is suitable for high speed movement, an optimal estimation result cannot be obtained for low speed movement. On the other hand, an interpolation filter suitable for low-speed movement causes a problem that an optimal estimation result cannot be obtained during high-speed movement.

  For this, it is necessary to construct an interpolation filter that can handle both high-speed movement and low-speed movement. However, in the conventional digital broadcast receiving apparatus as shown in FIG. 16, the interpolation filter is not configured to take into account mobile reception.

  SUMMARY OF THE INVENTION Therefore, an object of the present invention is to provide a digital broadcast receiving apparatus that stably realizes good reception performance by adapting an interpolation filter in an equalization circuit to a change in the state of a transmission path.

  According to an aspect of the present invention, there is provided a digital broadcast receiving apparatus capable of receiving a reception signal of an orthogonal frequency division division transmission scheme specified by a combination of an effective symbol period length and a guard interval length, after orthogonal detection Mode / guard interval determining means for receiving the in-phase and quadrature axis signals and determining the effective symbol period length and guard interval length of the received signal, and the in-phase and quadrature axis signals from the time domain to the frequency domain. And a Fourier transform means for converting the signal into a frequency domain, and an equalization means for equalizing the frequency domain signal based on the estimated transmission path characteristics. The equalizing means uses a extracting means for extracting a known pilot signal from the signal in the frequency domain, and an interpolation filter means for interpolating the transmission path characteristics from the pilot signal, so as to estimate the transmission path characteristics. And a dividing means for dividing the frequency domain signal by the estimated transmission line characteristic. The channel characteristic estimation unit performs interpolation based on the Doppler frequency estimation unit that estimates the Doppler frequency of the transmission channel from the extracted pilot signal, the estimation result of the Doppler frequency, and the determination result of the effective symbol period length and the guard interval length. Interpolation filter selection means for selecting filter coefficients corresponding to a predetermined number of tap outputs of the filter means.

  Preferably, the transmission path estimation means stores a plurality of sets of filter coefficients specified by a combination of a plurality of effective symbol period lengths and a plurality of guard interval lengths and a magnitude of the Doppler frequency, and provides instructions to the interpolation filter selection means. Correspondingly, it further includes storage means for selecting and outputting one set from a plurality of sets of filter coefficients.

  According to another aspect of the present invention, there is provided a digital broadcast receiving apparatus capable of receiving a reception signal of an orthogonal frequency division division transmission scheme specified by a combination of an effective symbol period length and a guard interval length, wherein the orthogonal detection is performed. A mode / guard interval determination means for receiving a later in-phase signal and a quadrature axis signal and determining an effective symbol period length and a guard interval length of the received signal; and an in-phase signal and a quadrature axis signal from the time domain to the frequency domain Fourier transform means for converting the signal into a signal, and equalization means for equalizing the frequency domain signal based on the estimated channel characteristics. The equalizing means uses a extracting means for extracting a known pilot signal from the signal in the frequency domain, and an interpolation filter means for interpolating the transmission path characteristics from the pilot signal, so as to estimate the transmission path characteristics. And a dividing means for dividing the frequency domain signal by the estimated transmission line characteristic. The channel characteristic estimation unit performs interpolation based on the delay time estimation unit that estimates the delay time of the transmission channel from the extracted pilot signal, the estimation result of the delay time, and the determination result of the effective symbol period length and the guard interval length. Interpolation filter selection means for selecting filter coefficients corresponding to a predetermined number of tap outputs of the filter means.

  Preferably, the transmission path estimation means holds, as stored data, a plurality of sets of filter coefficients specified by a combination of a plurality of effective symbol period lengths, a plurality of guard interval lengths, and a delay time, and an interpolation filter selection means Storage means for selecting and outputting one set from a plurality of sets of filter coefficients in response to the instruction.

  According to another aspect of the present invention, there is provided a digital broadcast receiving apparatus capable of receiving a reception signal of an orthogonal frequency division division transmission scheme specified by a combination of an effective symbol period length and a guard interval length, wherein the orthogonal detection is performed. A mode / guard interval determination means for receiving a later in-phase signal and a quadrature axis signal and determining an effective symbol period length and a guard interval length of the received signal; and an in-phase signal and a quadrature axis signal from the time domain to the frequency domain Fourier transform means for converting the signal into a signal, and equalization means for equalizing the frequency domain signal based on the estimated channel characteristics. The equalizing means uses a extracting means for extracting a known pilot signal from the signal in the frequency domain, and an interpolation filter means for interpolating the transmission path characteristics from the pilot signal, so as to estimate the transmission path characteristics. And a dividing means for dividing the frequency domain signal by the estimated transmission line characteristic. Transmission path characteristic estimation means includes Doppler frequency estimation means for estimating the Doppler frequency of the transmission path from the extracted pilot signal, delay time estimation means for estimating the delay time of the transmission path from the extracted pilot signal, Doppler frequency and delay time Interpolation filter selection means for selecting filter coefficients corresponding to a predetermined number of tap outputs of the interpolation filter means based on the estimation results of the above and the determination results of the effective symbol period length and the guard interval length.

  Preferably, the transmission path estimation means holds, as stored data, a plurality of filter coefficients specified by a combination of a plurality of effective symbol period lengths and a plurality of guard interval lengths, a Doppler frequency magnitude, and a delay time magnitude. And a storage means for selecting and outputting one set from a plurality of sets of filter coefficients in accordance with an instruction from the interpolation filter selection means.

  Preferably, the Doppler frequency estimation means includes a plurality of first pilot signals included in the first symbol and a plurality of pilot signals included in the second symbol and having the same subcarrier frequency as each of the plurality of pilot signals. The magnitude of the Doppler frequency depends on the means for calculating the correlation values with the second pilot signal, the adding means for adding the calculated plurality of correlation values, and whether or not the addition result of the correlation values exceeds a predetermined threshold value. Determining means for determining.

  Preferably, the delay time estimating means includes a plurality of first pilot signals included in a predetermined symbol and a predetermined symbol, each of which separates the data signal from each of the plurality of first pilot signals. Means for calculating correlation values with a plurality of adjacent second pilot signals, addition means for adding the calculated plurality of correlation values, and whether or not the addition result of the correlation values exceeds a predetermined threshold value And determining means for determining the magnitude of the delay time.

  Preferably, the delay time estimating unit averages the output signal of the second Fourier transform unit over a plurality of symbols, the second Fourier transform unit converting the extracted pilot signal from the time domain to the frequency domain signal. And averaging means for determining the magnitude of the delay time depending on whether or not the delay time of the output signal having the maximum level among the output signals of the averaging means exceeds a predetermined threshold value.

  Preferably, the mode / guard interval determining means includes delay means for receiving a in-phase signal and a quadrature axis signal after quadrature detection and delaying a period corresponding to one of a plurality of effective symbol period lengths, Correlation detection means for detecting the correlation between the quadrature axis signal and the delayed in-phase signal and quadrature axis signal, and the output of the correlation detection means, and moving average processing for a period corresponding to one of a plurality of guard interval lengths A moving average means for outputting, an absolute value adding means for outputting the sum of absolute values of the outputs of the moving average means, a peak value detecting means for detecting the maximum peak value of the output value of the absolute value adding means, and a peak value detecting means The settings for specifying the delay amount of the delay means and the period for performing the moving average processing of the moving average means according to the output are the effective symbol period length of the received signal and the guard interval. It determines whether to match the length, and a determination means for updating the set according to the result of the determination.

  According to an aspect of the present invention, by adopting a configuration in which a filter coefficient of an interpolation filter in transmission path estimation can be selected according to a Doppler frequency, high-precision equalization characteristics can be maintained even during mobile reception, and stable Reception quality can be obtained.

  According to another aspect of the present invention, the filter coefficient of the interpolation filter in the transmission path estimation can be selected according to the delay time, so that highly accurate equalization characteristics can be maintained even during mobile reception, and stable. Reception quality can be obtained.

  According to another aspect of the present invention, the filter coefficient of the interpolation filter in the transmission path estimation can be selected according to the Doppler frequency and the delay time, thereby further improving the transmission path estimation accuracy and further receiving quality. Can be stabilized.

  Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings. In the drawings, the same reference numerals indicate the same or corresponding parts.

[Embodiment 1]
FIG. 1 is a schematic block diagram showing the overall configuration of a digital broadcast receiving apparatus according to Embodiment 1 of the present invention.

  Referring to FIG. 1, in digital broadcast receiving apparatus 1000, an RF signal received from an antenna (not shown) is selected by tuner 100 and provided to OFDM demodulator 102.

  The demodulated signal from the OFDM demodulator 102 is supplied to a transport stream decoder (hereinafter also referred to as a TS decoder) 104 and is supplied to the MPEG decoder 110. That is, the TS decoder 104 extracts data streams such as video and audio from the transport stream data.

  The MPEG decoding unit 110 receives the data stream supplied from the TS decoder 104 and converts it into a video signal and an audio signal by using a random access memory (hereinafter also referred to as RAM) 112 as a buffer for temporarily storing data. To do.

  The digital broadcast receiving apparatus 1000 further stores the internal storage device 148 for receiving and storing the signal from the TS decoder 104 via the data bus BS1 and the internal storage device 148 via the data bus BS1. An arithmetic processing unit 144 for performing predetermined processing on the data and outputting it, a ROM 140 for recording a program in the arithmetic processing of the arithmetic processing unit 144, and a memory area for operation of the arithmetic processing unit 144 A RAM 142 to be provided and a high-speed digital interface 146 for performing data input / output between the data bus BS1 and the outside are provided.

  When the arithmetic processing unit 144 performs a predetermined process on the data stored in the built-in storage device 148 according to an instruction from the outside, the processed data is transferred from the on-screen display processing unit 130 to the combiner. 160.2.

  The synthesizer 160.2 synthesizes the output from the MPEG decoding unit 110 and the output from the on-screen display processing unit 130, and then gives them to the video output terminal 164. The output from the video output terminal 164 is given to the display unit 1004.

  Further, the digital broadcast receiving apparatus 1000 receives data obtained as a result of processing by the arithmetic processing unit 144 based on data stored in the built-in storage device 148, and outputs sound effects and the like for the video output from the display unit 1004. Generates an audio signal by receiving the additional sound generator 120 to be generated and given to the synthesizer 160.1 and the data processed by the arithmetic processing unit 144 based on the data stored in the built-in storage device 148 The PCM decoder 122 is provided to the combiner 160.1.

  The synthesizer 160.1 receives the output from the MPEG decoding unit 110 and the outputs from the additional sound generator 120 and the PCM decoder 122, and gives a synthesis result to the audio output terminal 162. The audio signal supplied to the audio output terminal 162 is output from the audio output unit 1002 as an audio signal.

  The digital broadcast receiving apparatus 1000 may include a modem 150 for exchanging data with the outside and an IC card interface 152 for receiving information from the IC card as necessary.

  Via the high-speed digital interface 146, for example, an external storage device 180 such as an HDD device for a home server, a remote control (or a keyboard or the like) as the external input device 182 and the data bus BS1 are connected.

  Further, the digital broadcast receiving apparatus 1000 may be integrated with a display unit 1004 that receives video output and displays it on a display, or an audio output unit 1002 such as a speaker that receives audio output signals and outputs audio. .

  FIG. 2 is a block diagram showing a configuration of OFDM demodulation section 102 in FIG.

  Referring to FIG. 2, OFDM demodulating section 102 has an A / D converter 201 that performs analog / digital conversion on the output of tuner 100, and an in-phase signal (I signal) and a quadrature axis signal (Q signal). And an I / Q separation unit 202 that separates the first carrier synchronization unit 204 and a first carrier synchronization unit 204 that corrects an error equal to or less than half of the carrier interval between the transmission carrier frequency and the reception carrier frequency.

  The OFDM demodulator 102 further includes a guard interval correlation circuit 206 that receives a I signal and a Q signal, detects a guard interval by delaying and taking a correlation for a predetermined period, and outputs a clock and a control signal to each block.

  The guard interval correlation circuit 206 receives the output of the first carrier synchronization unit 204, delays it, and looks at the correlation between the signals before and after the delay to determine the mode / guard interval determination circuit 300, and the mode / guard interval A symbol synchronization unit 302 that receives the output of the interval determination circuit 300 and generates a symbol synchronization pulse, and a clock synchronization unit 304 that receives the output of the symbol synchronization unit 302 and outputs a synchronization clock.

  The OFDM demodulator 102 further performs an FFT circuit 210 that performs fast Fourier transform with the number of points corresponding to the control signal output from the guard interval correlation circuit 206, and an error in units of carrier intervals between the transmission carrier frequency and the reception carrier frequency. A second carrier synchronization unit 212 for correction and an equalization circuit 214a for correcting distortion of a signal received on the transmission path are included.

  The OFDM demodulator 102 further includes a frequency deinterleave circuit 216 that cancels frequency-direction interleaving performed on the transmission side, a time deinterleave circuit 218 that cancels time-direction interleaving performed on the transmission side, and a transmission side. 2 includes a demapping circuit 220 that combines the data arranged according to the modulation method, and a bit deinterleaving circuit 222 that cancels the bit-wise interleaving performed on the transmission side.

OFDM demodulator 102 further includes a Viterbi decoder 224 for decoding the encoded convolutionally on the transmission side data, a byte deinterleaving circuit 226 to release the interleaving bytes that are applied in the transmission side, the transport stream A TS reproduction circuit 228 that reconstructs data so as to conform to the format and an RS decoding circuit 230 that decodes Reed-Solomon encoded data on the transmission side are included.

  The RS decoding circuit 230 outputs the Reed-Solomon decoded result to the TS decoder 104 shown in FIG.

  FIG. 3 is a block diagram showing a configuration of the guard interval correlation circuit 206 in FIG.

  Referring to FIG. 3, guard interval correlation circuit 206 receives a I / Q signal and a mode / guard interval determination circuit 300 for determining a mode and a guard interval, and an output of mode / guard interval determination circuit 300. A symbol synchronization unit 302 that detects a symbol period and outputs a symbol pulse, and a clock synchronization unit 304 that receives the outputs of the mode / guard interval determination circuit 300 and the symbol synchronization unit 302 and performs clock synchronization processing.

  The mode / guard interval determination circuit 300 receives the I signal and the Q signal, detects the correlation of the guard interval, and detects the correlation between the guard interval correlation detection unit 400 and the current setting based on the output of the guard interval correlation detection unit 400. It includes a mode / guard interval determination / setting unit 500 that determines whether the mode and guard interval coincide with each other and sets the circuit mode and guard interval.

  The guard interval correlation detection unit 400 continuously outputs a delay memory 402 that performs signal delay only during an effective symbol period, a correlator 404 that correlates signals before and after delay by complex multiplication, and an average value of guard period widths. It includes a moving average circuit 406 and an absolute value addition circuit 408 that converts the output of the moving average circuit 406 into a positive value and adds it.

  The mode / guard interval determination / setting unit 500 receives the output from the absolute value addition circuit 408 and detects a triangular wave peak value detecting circuit 502, and a plurality of modes / guard intervals corresponding to a combination of a plurality of mode / guard intervals. A peak value comparison circuit 504 for detecting the maximum peak value by comparing the peak values of the two, and a mode / guard interval setting circuit for setting the mode / guard interval by selecting a combination of the mode / guard interval indicating the maximum peak value 506.

  Here, the OFDM signal mode and guard interval determination method according to the present embodiment will be described.

  FIG. 4 is an operation waveform diagram for explaining the operation of the guard interval correlation circuit 206 of FIG.

  Referring to FIG. 4, in signal ADO given from A / D converter 201, guard intervals G1, G2,... Are added to the heads of the effective symbol periods S1, S2,. . As shown in FIG. 14, the guard intervals G1, G2,... Are copied from the last G1 ′, G2 ′,.

  Therefore, when the delay memory 402 delays the effective symbol period, as shown in the delay memory output MO, the output timing of the guard intervals G1, G2,... , G2 ′,. Since Gn and Gn ′ (n is a natural number) are in a copying relationship, the signal correlation during this period is high. On the other hand, in other periods, since the OFDM signal is a noise signal, the correlation is low.

  Therefore, as shown in FIG. 4, the signals Iav and Qav which are the I and Q signals of the moving average output and the signal Oabs which is the output signal of the absolute value adding circuit are represented by guard intervals G1, G2,. It gradually changes from the start timing, and takes a peak value at the end of the effective symbol period.

  The moving average outputs Iav and Qav are supplied to the first carrier synchronization unit 204. The absolute value addition output Oabs is supplied to the symbol synchronization unit 302 and the triangular wave peak value detection circuit 502, respectively.

  The absolute value addition output Oabs is detected by the triangular wave peak value detection circuit 502 for each corresponding mode / guard interval combination, and the subsequent peak value comparison circuit 504 compares the maximum peak value.

  The maximum peak value is equal to the delay amount of the effective symbol period length set in the delay memory 402 and the effective symbol period length of the received signal, that is, the circuit setting mode and the received signal mode match, and the moving average This output appears when the guard interval set in the circuit 406 matches the guard interval of the received signal. If either one of the mode guard intervals differs between the circuit setting and the received signal, the absolute value addition output Oabs has almost no peak. Accordingly, if the peak value comparison circuit 504 detects the maximum peak value, the mode guard interval of the received signal can be determined. The mode / guard interval setting circuit 506 selects a mode / guard interval having the maximum peak value from a plurality of combinations of the mode / guard intervals, and determines the mode / guard interval of the receiving circuit.

  Next, a method for equalizing the transmission path distortion generated in the OFDM signal will be described.

  FIG. 5 is a block diagram showing a configuration of equalization circuit 214a in FIG.

  Referring to FIG. 5, equalization circuit 214a is a pilot signal extraction circuit 600 that extracts pilot signal SP from the output of FFT circuit 210 (not shown), and the speed of temporal change in transmission path characteristics from pilot signal SP. The Doppler frequency estimation circuit 602 that estimates the Doppler frequency, the filter coefficient ROM 606a that stores the filter coefficient corresponding to the magnitude of the Doppler frequency as stored data, and the estimated Doppler frequency and mode / guard interval determination circuit 300 An interpolation filter selection circuit 604 that selects an optimum filter coefficient from the filter coefficient ROM 606a based on the mode guard interval, and a transmission path estimation circuit 608 that estimates the transmission path by inputting the pilot signal SP to the selected interpolation filter. Equipped with a.

  The equalization circuit 214a further includes a division circuit 610 that estimates the data signal by dividing the output of the FFT circuit 210 by the transmission path characteristic. The estimated data signal is supplied to a frequency deinterleave circuit 216 (not shown).

  The equalization circuit 214a according to the present embodiment is characterized by estimating the Doppler frequency of the received signal and selecting the optimum filter coefficient of the interpolation filter based on the estimation result. In this respect, it differs from a conventional equalization circuit (for example, refer to the equalization circuit 814 in FIG. 16) in which the filter coefficient of the interpolation filter is fixed regardless of the change in the transmission path. In describing the equalization processing according to the present embodiment, first, how the Doppler frequency affects the received signal when the OFDM signal is mobilely received will be described.

  As described above, the Doppler frequency is proportional to the speed of the temporal change in the transmission line characteristics. Now, assuming that the Doppler frequency is fd, the carrier frequency of the received signal from the station in the traveling direction when the OFDM signal is mobilely received increases by the Doppler frequency fd, and the carrier frequency of the received signal from the direction opposite to the traveling direction. Decreases by the Doppler frequency fd. That is, the carrier frequency of the received signal is translated in parallel by the Doppler frequency fd. Hereinafter, the phenomenon that the carrier frequency of the received signal shifts is also referred to as Doppler shift. For example, when there are 1024 transmission subcarriers of 1 kHz, 2 kHz,..., 1024 kHz, and mobile reception is performed at a speed at which the Doppler frequency is 10 Hz, the subcarrier frequencies at the time of reception are 1.01 kHz, 2.01 kHz. , ..., 1024.01 kHz.

  In the FFT circuit 210 of the digital broadcast receiving apparatus that receives an OFDM signal, if the frequency of the reception subcarrier is the same as the frequency of the transmission subcarrier due to the characteristics of the FFT processing, orthogonality is maintained, and interference with the subcarrier is prevented. It can be converted into a signal on no frequency axis.

  FIG. 6 is a schematic diagram for explaining the relationship between the input / output signal and the Doppler frequency in the FFT circuit 210. In the following, a subcarrier having a frequency of 1 kHz, which is a fundamental wave, is considered.

  As shown in FIG. 6A, when the Doppler frequency is 0, that is, when the frequencies of the transmission subcarrier and the reception subcarrier match, the frequency of the reception signal is 1 kHz. Since the effective symbol period is 1 msec, the received signal is exactly one cycle in the effective symbol period 1 msec. For this reason, the input signal of the FFT circuit 210 has a waveform at the boundary between the fundamental wave of 1 kHz in one effective symbol period 1 msec, which is the FFT capture period, and the same fundamental wave of 1 kHz copied before and after that. It becomes a continuous shape.

  Therefore, an output signal obtained by performing FFT processing on this input signal is a signal on the frequency axis that exists only at a frequency of 1 kHz.

On the other hand, when the Doppler frequency is 1 0H z, that is, when the frequency varies slightly between the received subcarrier and the transmission subcarrier frequency 1.01kHz next received signal, in that one cycle effective symbol period 1msec In contrast, it is slightly shorter. For this reason, as shown in FIG. 6B, in the input signal, the 1.01 kHz fundamental wave in the effective symbol period and the 1.01 kHz fundamental wave copied before and after it are discontinuous at the boundary portion. It becomes.

  When this discontinuous input signal is subjected to FFT processing, harmonic components such as 2 kHz, 3 kHz, 4 kHz,... Are generated in the output signal converted into the signal on the frequency axis in addition to the fundamental wave of 1 kHz. Since this harmonic component appears in a form superimposed on the signals of other subcarriers (2 kHz, 3 kHz,...), Interference between subcarriers is caused.

  If the received signal when the Doppler shift occurs is divided into the desired symbol component, the above-described intersubcarrier interference component, and the noise component, the intersubcarrier interference component is a major factor in signal quality degradation. (For example, Okada et al., "Analysis of Doppler shift effect in OFDM and examination of its mitigation method", ITE Technical Report, Technical Report Vol.21, No.52, pp .1-5, see BCS '97). According to this document, the received signal of the mth subcarrier of the lth symbol when the Doppler shift occurs is

It is represented by here,

It was. C (m, n) represents a correlation coefficient between the mth subcarrier and the nth subcarrier among N subcarriers included in the lth symbol. d (l, m) represents the transmission signal of the mth subcarrier of the lth symbol.

  Of the received signals represented by Equation (1), the first term corresponds to the desired symbol component when n = m, the second term corresponds to the intersubcarrier interference component when n ≠ m, and the third term The term corresponds to the noise component. As described above, the inter-subcarrier interference component is caused by the Doppler shift, and the value increases as the Doppler frequency increases.

  Here, when the symbol number (time direction) 1 is different, the magnitude of the inter-subcarrier interference component of the second term of Expression (1) changes because the transmission signal d (l, n) changes. For this reason, even if the received signal has the same subcarrier frequency (equivalent to the same m) on the IQ constellation, a slight deviation occurs.

  Therefore, if the correlation between two pilot signals having the same subcarrier frequency among the received signals having different symbol numbers is calculated, the constellation shift is large when the intersubcarrier interference component is large, that is, when the Doppler frequency is large. , The correlation value becomes low. On the other hand, when the inter-subcarrier interference component is small, that is, when the Doppler frequency is small, the constellation shift is small and a high correlation value is obtained.

  Therefore, if the correlation value is determined based on a predetermined threshold value, the magnitude of the influence of intersubcarrier interference due to Doppler shift can be known. Furthermore, if the configuration is such that the characteristics of the interpolation filter are changed in accordance with the obtained degree of inter-subcarrier interference, the accuracy of transmission path estimation can be improved and stable reception performance can be ensured.

  Next, a method for calculating the correlation value between pilot signals will be described. The calculation of the correlation value is performed in the Doppler frequency estimation circuit 602 shown in FIG.

  FIG. 7 is a block diagram showing a configuration of the Doppler frequency estimation circuit 602 in FIG.

  Referring to FIG. 7, Doppler frequency estimation circuit 602 correlates memory circuit 700a storing a pilot signal of a predetermined number of symbols, and the stored pilot signal and a pilot signal of the same subcarrier frequency included in another symbol. A correlation circuit 702a that calculates the correlation value, an addition circuit 704a that adds the obtained correlation value for one symbol, a filter circuit 706a that removes high-frequency components from the addition result, and a determination that determines the magnitude of the Doppler frequency from the filter output Circuit 708a.

  As shown in FIG. 7, the pilot signal extracted in pilot signal extraction circuit 600 is applied to memory circuit 700a and correlation circuit 702a.

  Memory circuit 700a stores pilot signals for four symbols (for example, symbol numbers 0, 1, 2, and 3) of pilot signals provided from pilot signal extraction circuit 600. As shown in FIG. 8, the pilot signal of each symbol is arranged at a rate of one for 12 subcarriers in the frequency direction. In the time direction, the pilot signal insertion position is shifted by three subcarriers for each symbol. Each pilot signal is composed of an I signal and a Q signal. For example, the pilot signal of the i-th (i = 0, 12, 24,...) Subcarrier of symbol number 0 is (I (0, i) , Q (0, i)).

  Here, it is further assumed that pilot signal (I (4, i), Q (4, i)) of symbol number 4 is given.

  In the correlation circuit 702a, the pilot signal (I (4, i), Q (4, i)) of symbol number 4 and the pilot signal (I of symbol number 0) having a pilot signal of the same subcarrier frequency from the memory circuit 700a are provided. (0, i), Q (0, i)).

  Correlation circuit 702a calculates the correlation of pilot signals of the same subcarrier frequency existing every 12 subcarriers between the pilot signal of symbol number 0 and the pilot signal of symbol number 4. The correlation value DOP_CORR (i) at the i-th subcarrier frequency is

Given in.

Here, as described above, when the Doppler frequency is small, I (0, i) and I (4, i) and Q (0, i) and Q (4, i) are both small in deviation and almost the same value. Therefore, the correlation value has a certain size. On the other hand, when the Doppler frequency is large, since the deviation is large, the correlation value is low. In the equation ( 4 ), the correlation value is expressed as an absolute value, but the sign may be removed by squaring DOP_CORR (i).

  When the correlation value DOP_CORR (i) (i = 0, 12, 24,...) Calculated for the 12 subcarriers calculated by the correlation circuit 702a is supplied to the addition circuit 704a, the correlation value for one symbol is added. The

  The added value DOP_CORR_SUM is input to the filter circuit 706a.

  The filter circuit 706a is composed of a low-pass filter, and removes a high-frequency component included in the added value DOP_CORR_SUM. Since the addition value is determined for each symbol, the filter circuit 706a operates for each symbol.

  Subsequently, the filter output is given to the determination circuit 708a. The determination circuit 708a compares the added value DOP_CORR_SUM after passing through the filter circuit with a predetermined threshold value. As the threshold value, a Doppler frequency that improves reception performance by switching the filter characteristics of the transmission path estimation circuit is set. When the added value DOP_CORR_SUM is larger than the threshold value, it is estimated that the correlation between pilot signals is large, that is, the Doppler frequency is small. On the other hand, when the added value DOP_CORR_SUM is smaller than the threshold value, it is estimated that the correlation between pilot signals is small, that is, the Doppler frequency is large. The estimation result of the determination circuit 708a is transmitted to an interpolation filter selection circuit (not shown).

  As described above, the Doppler frequency estimation circuit 602 estimates the magnitude of the Doppler frequency from the magnitude of the correlation between pilot signals of the same subcarrier frequency of different symbols. Based on this estimation result, the interpolation filter described below is optimized.

  Next, the optimization of the interpolation filter in the equalization circuit 214a will be described.

  Referring to FIG. 5 again, the interpolation filter selection circuit 604 is given the estimation result from the Doppler frequency estimation circuit 602 and the mode / guard interval determined by the mode / guard interval determination circuit 300 (not shown). Given. The interpolation filter selection circuit 604 selects an optimum filter coefficient from a plurality of filter coefficients stored in the filter coefficient ROM 606a based on the Doppler frequency estimation result and the mode / guard interval information of the received signal.

  Here, the filter coefficient ROM 606a will be described in detail.

  The filter coefficient ROM 606a holds the filter coefficients of the interpolation filter included in the transmission path estimation circuit 608 as a table. The number of tables required for any one subcarrier is a number corresponding to the product of the total number of patterns of the combination of mode and guard interval and the two patterns having large and small Doppler frequencies. For example, in the case of digital terrestrial broadcasting, the number of tables in the filter coefficient ROM 606a is 5 × 2 = 10 since five combinations of mode and guard interval are employed in operation.

  The interpolation filter of the transmission path estimation circuit 608 may be configured to interpolate other subcarrier data using a pilot signal, and may be realized using any number of pilot signals. Therefore, the general interpolation filter configuration is

It is expressed as Here, SP1, SP2,..., SPn indicate pilot signals. A1, A2,..., An are filter coefficients for a plurality of tap outputs of the interpolation filter, and correspond to the weighting coefficient of the corresponding pilot signal SP.

  Therefore, if an interpolation filter corresponding to one mode / guard interval pattern is generated by a combination of j filter coefficients, the number of tables in the filter coefficient ROM 606a is 5 × 2 × j = 10 × j. Become.

  For example, as shown in FIG. 8, when the transmission path estimation of the data portion is performed using the pilot signals SPA to SPF of 6 points (A to F), the interpolation filter is configured with filter coefficients for these 6 points. . When the Doppler frequency is large, the fluctuation in the time direction is large, and therefore, the filter coefficients corresponding to the pilot signals SPB, SPC, SPD, and SPE are set smaller than the filter coefficients of the pilot signals SPA and SPF. . As a result, the weighting of the pilot signals SPB, SPC, SPD, and SPE is reduced, and is not considered much in transmission path estimation.

  On the other hand, when the Doppler frequency is small, since the amount of fluctuation in the time direction is small, the filter coefficients corresponding to the pilot signals SPB, SPC, SPD, and SPE are set to be equivalent to the pilot signals SPA and SPF. As a result, all of the pilot signals SPA to SPF are considered in the transmission path estimation. Thus, by adopting a configuration in which the magnitude of the filter coefficient is changed according to the magnitude of the Doppler frequency, the transmission path estimation accuracy of the data carrier can be further increased.

  Referring to FIG. 5 again, the interpolation filter selection circuit 604 selects a suitable filter coefficient from these tables based on the estimation result of the mode / guard interval and the magnitude of the Doppler frequency, and estimates the transmission path. Output to the circuit 608.

  The transmission path estimation circuit 608 inputs the pilot signal extracted by the pilot signal extraction circuit 600 to the optimized interpolation filter. Thereby, the transmission path of the data signal is estimated. The estimated transmission line characteristics are given to the divider circuit 610.

  The division circuit 610 equalizes the data signal by dividing the FFT output by the transmission path characteristic, and outputs the equalized data signal to the subsequent frequency deinterleave circuit 216.

  As described above, according to the first embodiment of the present invention, the filter coefficient of the interpolation filter in the transmission path estimation can be selected according to the Doppler frequency, so that equalization accuracy is maintained even during mobile reception. And stable reception quality can be obtained.

[Embodiment 2]
In the previous embodiment, a configuration has been proposed in which the filter coefficient of the interpolation filter in the transmission path estimation can be selected according to the amount of fluctuation in the time direction (Doppler frequency).

  By the way, there are multipaths in the actual transmission path environment, and different delay times occur for each path with respect to the direct wave. The delay time of each path varies depending on the mobile reception, and the amount of variation varies depending on the moving speed.

  Therefore, in order to improve the transmission path estimation accuracy at the time of mobile reception, it is desirable to realize filter characteristics adapted to the amount of delay time variation. In view of this, the present embodiment proposes a digital broadcast receiving apparatus that can select filter characteristics in accordance with fluctuations in delay time.

  First, the influence of the delay time on the received signal will be described.

  The influence of the delay time on the received signal appears in the phase of the output signal of the FFT circuit when the delay time is within the guard interval. Note that the case where the delay time exceeds the guard interval period hardly occurs in the system and is not considered here. In the above formulas (1) to (3), if only the delay time is taken into consideration and the Doppler frequency fd = Δf = 0, the formula (3) is

And becomes a function unrelated to k. Moreover, since the expression (1) may ignore the second term if the delay time is within the guard interval period,

It becomes. Here, ignoring the noise component of the third term, the received signal is given as a point rotating on a circle having a radius γ centered on the position of the transmission signal d (l, m) on the IQ constellation. Furthermore, since the rotating phase depends only on the frequency fm of the m-th subcarrier, the magnitude of the delay time is determined by comparing the positional relationship on the IQ constellation of subcarriers included in the same symbol. can do.

  Therefore, if the correlation of pilot signals of different subcarrier frequencies in the same symbol is calculated, the tendency of the delay time can be confirmed. That is, it is estimated that if the correlation value is large, the delay time is small, and if the correlation value is small, the delay time is large.

  Therefore, in this embodiment, a delay time estimation circuit is arranged in the equalization circuit, and the filter characteristic of the transmission path estimation circuit is selected according to the estimated delay time. Hereinafter, the equalization circuit will be described in detail. Since circuit parts other than the equalization circuit mounted on the digital broadcast receiving apparatus according to the present embodiment are the same as those in the first embodiment, detailed description thereof will not be repeated.

  FIG. 9 is a block diagram showing a configuration of equalization circuit 214b in the digital broadcast receiving apparatus according to the second embodiment of the present invention.

  Referring to FIG. 9, equalization circuit 214b has basically the same configuration as equalization circuit 214a in FIG. 5, and differs only in that the Doppler frequency estimation circuit is replaced with a delay time estimation circuit. Therefore, detailed description of overlapping circuit parts is omitted.

  According to the present embodiment, the interpolation filter selection circuit 604 selects an optimum filter coefficient from the filter coefficient ROM 606b based on the estimated delay time and mode guard interval information. Here, a delay time estimation method in the delay time estimation circuit will be described.

  FIG. 10 is a block diagram showing a configuration of delay time estimation circuit 612 in FIG.

  Referring to FIG. 10, delay time estimation circuit 612 correlates memory circuit 700b for storing a pilot signal of a predetermined symbol and a correlation between pilot signals of different subcarrier frequencies that are included in the same symbol as the stored pilot signal. Correlation circuit 702b to be calculated, addition circuit 704b for adding the obtained correlation value for one symbol, filter circuit 706b for removing high frequency components from the addition result, and determination circuit for determining the magnitude of the delay time from the filter output 708b.

  The pilot signal extracted by pilot signal extraction circuit 600 is applied to memory circuit 700b and correlation circuit 702b as shown in FIG.

  Memory circuit 700b stores a pilot signal for one symbol (for example, symbol number 4) of the pilot signals provided from pilot signal extraction circuit 600. As shown in FIG. 8, one pilot signal is arranged for one symbol with respect to 12 subcarriers. Each pilot signal is made up of an I signal and a Q signal. For example, the pilot signal of the i-th subcarrier of symbol number 4 is represented by (I (4, i), Q (4, i)).

  Correlation circuit 702b receives the same symbol from pilot signal extraction circuit 600 as well as the pilot signal (I (4, i), Q (4, i)) of i-th subcarrier of symbol number 4 from memory circuit 700b. , A pilot signal of symbol number 4 (I (4, i + 12), Q (4, i + 12)) having a pilot signal of the (i + 12) th subcarrier frequency adjacent to each other with the i-th pilot signal and the data signal interposed therebetween is given It is done.

Correlation circuit 702b calculates the correlation between the pilot signal of the i-th subcarrier and the pilot signal of the (i + 12) -th subcarrier . The correlation value DEL_CORR (i) at the i- th subcarrier frequency is

It becomes.

  Here, when the delay time is small, I (4, i) and I (4, i + 12) and Q (4, i) and Q (4, i + 12) are almost the same value. With the size of On the other hand, when the delay time is large, the correlation value is low because the signal shift is large. In Equation (4), the correlation value is expressed as an absolute value, but the sign may be removed by squaring DEL_CORR (i).

  When the correlation value DEL_CORR (i) (i = 0, 12, 24,...) Calculated by the correlation circuit 702b is given to the addition circuit 704b, the correlation value for one symbol is added. .

  The added value DOP_CORR_SUM is input to the filter circuit 706b.

  The filter circuit 706b is composed of a low-pass filter, and removes a high-frequency component included in the addition value DEL_CORR_SUM. Since the addition value is determined for each symbol, the filter circuit 706b operates for each symbol.

  Subsequently, the filter output is given to the determination circuit 708b. The determination circuit 708b compares the added value DEL_CORR_SUM after passing through the filter circuit with a predetermined threshold value. Note that it is desirable to set the threshold value so as to be a delay time that improves the reception performance by switching the filter characteristics of the transmission path estimation circuit.

  In the comparison result, when the added value DEL_CORR_SUM is larger than the threshold value, it is estimated that the correlation between pilot signals is large, that is, the delay time is small. On the other hand, when the added value DEL_CORR_SUM is smaller than the threshold value, it is estimated that the correlation between pilot signals is small, that is, the delay time is large. The estimation result of the determination circuit 708b is transmitted to an interpolation filter selection circuit (not shown).

  As described above, the delay time estimation circuit 612 estimates the magnitude of the delay time from the magnitude of the correlation between pilot signals adjacent to each other at 12 subcarriers of one symbol. Based on this estimation result, the interpolation filter described below is optimized.

  Next, the optimization of the interpolation filter in the equalization circuit 214b will be described.

  Referring to FIG. 9 again, the interpolation filter selection circuit 604 is given the estimation result from the delay time estimation circuit 612, and the mode / guard interval determined by the mode / guard interval determination circuit (not shown), Filter coefficients stored in the filter coefficient ROM 606b are given. The interpolation filter selection circuit 604 selects an optimum filter coefficient from a plurality of filter coefficients stored in the filter coefficient ROM 606b based on the delay time estimation result and the mode / guard interval information of the received signal.

  Here, the filter coefficient ROM 606b will be described.

  Similarly to the first embodiment, the filter coefficient ROM 606b holds the filter coefficients of the interpolation filter included in the transmission path estimation circuit 608 as a table. The number of tables required for a certain subcarrier is a number corresponding to the product of the total number of patterns of the combination of mode and guard interval and the two patterns with large and small delay times. For example, in the case of digital terrestrial broadcasting, the number of tables in the filter coefficient ROM 606b is 5 × 2 = 10 since five combinations of mode and guard interval are employed in operation.

  Further, if an interpolation filter corresponding to one mode / guard interval pattern is generated by a combination of j filter coefficients, the number of tables in the filter coefficient ROM 606b is 5 × 2 × j = 10 × j. Become.

  For example, as shown in FIG. 8, when the transmission path estimation of the data portion is performed using the pilot signals SPA to SPF of 6 points (A to F), the interpolation filter is configured with filter coefficients for these 6 points. . When the delay time is large, the fluctuation in the frequency direction is large. Therefore, the filter coefficients corresponding to the pilot signals SPA and SPF are set to be smaller than the filter coefficients of the pilot signals SPB, SPC, SPD and SPE. . As a result, the weighting of the pilot signals SPA and SPF is reduced, and is not considered much in transmission path estimation.

  On the other hand, when the delay time is small, the amount of fluctuation in the frequency direction is small, so that the filter coefficients corresponding to the pilot signals SPA and SPF are set to be equivalent to the pilot signals SPB, SPC, SPD and SPE. As a result, all of the pilot signals SPA to SPF are considered in the transmission path estimation. As described above, by adopting a configuration in which the size of the filter coefficient is changed according to the delay time, the transmission path estimation accuracy of the data carrier can be further increased.

  Referring to FIG. 9 again, the interpolation filter selection circuit 604 selects a suitable filter coefficient from these tables based on the estimation result of the mode / guard interval and the delay time, and estimates the transmission path. Output to the circuit 608.

  The transmission path estimation circuit 608 inputs the pilot signal extracted by the pilot signal extraction circuit 600 to the optimized interpolation filter. Thereby, the transmission path of the data carrier is estimated. The estimated transmission line characteristics are given to the divider circuit 610.

  In the division circuit 610, the data carrier is equalized by dividing the FFT output by the transmission path characteristic, and output to the frequency deinterleave circuit 216 in the subsequent stage.

[Embodiment 3]
In the above embodiment, a configuration has been proposed in which the characteristics of the interpolation filter of the equalization circuit are optimized based on the Doppler frequency estimation result or the delay time estimation result.

  The present embodiment further proposes a configuration for selecting a filter characteristic from estimation results of both the Doppler frequency and the delay time. Since Doppler frequency and delay time can be treated as independent from each other, more accurate transmission path characteristic estimation can be realized by selecting the optimum filter characteristics by combining the estimation results. Plan.

  FIG. 13 is a block diagram showing a configuration of an equalization circuit in the digital broadcast receiving apparatus according to the third embodiment of the present invention.

  Referring to FIG. 13, equalization circuit 214c includes pilot signal extraction circuit 600, Doppler frequency estimation circuit 602, delay time estimation circuit 612, interpolation filter selection circuit 604, filter coefficient ROM 606c, and transmission path estimation circuit. 608 and a divider circuit 610.

  As is apparent from FIG. 13, the equalization circuit 214c according to the present embodiment has a configuration in which a delay time estimation circuit 612 is added to the equalization circuit 214a according to the first embodiment shown in FIG. The delay time estimation circuit 612 is the same as that described in the second embodiment. Therefore, a detailed description of each part of the equalization circuit 214c is omitted.

  In FIG. 13, the pilot signal output from pilot signal extraction circuit 600 is provided to transmission path estimation circuit 608 and is also provided to Doppler frequency estimation circuit 602 and delay time estimation circuit 612.

  As described in Embodiment 1, the Doppler frequency estimation circuit 602 calculates the correlation between pilot signals of the same subcarrier frequency included in different symbols, and estimates the magnitude of the Doppler frequency from the magnitude of the correlation value. The estimation result is output to the interpolation filter selection circuit 604.

  As described in the second embodiment, the delay time estimation circuit 612 calculates a correlation between pilot signals separated by 12 subcarrier frequencies included in the same symbol, and estimates the magnitude of the delay time from the magnitude of the correlation value. Then, the estimation result is output to the interpolation filter selection circuit 604.

  In addition to these two estimation results, the interpolation filter selection circuit 604 is given the estimation result of the mode / guard interval from a mode / guard interval circuit (not shown). The interpolation filter selection circuit 604 selects an optimum filter coefficient from the filter coefficient ROM 606c based on these three estimation results, and provides it to the transmission path estimation circuit 608.

  Here, the filter coefficient ROM 606c will be described.

  The filter coefficient ROM 606c holds the filter coefficient of the interpolation filter included in the transmission path estimation circuit 608 as a table, as in the previous embodiment. The number of tables required for a certain subcarrier is the number corresponding to the product of the total number of patterns of mode / guard interval combinations, two patterns of large and small Doppler frequencies, and two patterns of large and small delay times. Become. In the case of digital terrestrial broadcasting, the number of filter coefficient ROM tables is 5 × 2 × 2 = 20 because a combination of 5 patterns of mode and guard interval is employed in operation.

  Further, if an interpolation filter corresponding to one mode / guard interval pattern is generated by a combination of j filter coefficients, the number of tables in the filter coefficient ROM 606c is 5 × 2 × 2 × j = 20 × j. It becomes a piece.

  Therefore, when the pattern of one mode / guard interval is determined, the interpolation filter selection circuit 604 determines one pattern based on the estimation result from four patterns of filter coefficients composed of combinations of the magnitude of the Doppler frequency and the magnitude of the delay time. A pattern is selected and output to the transmission path estimation circuit 608.

  The transmission path estimation circuit 608 inputs the pilot signal extracted by the pilot signal extraction circuit 600 to the optimized interpolation filter. Thereby, the transmission path of the data carrier is estimated. The estimated transmission line characteristics are given to the divider circuit 610.

  In the division circuit 610, the data carrier is equalized by dividing the FFT output by the transmission path characteristic, and output to the frequency deinterleave circuit 216 in the subsequent stage.

As described above, according to the third embodiment of the present invention, the filter coefficient of the interpolation filter in the transmission path estimation can be selected according to the Doppler frequency and the delay time. Transmission path estimation accuracy can be maintained, and reception quality can be further stabilized.

  The embodiment disclosed this time should be considered as illustrative in all points and not restrictive. The scope of the present invention is defined by the terms of the claims, rather than the description above, and is intended to include any modifications within the scope and meaning equivalent to the terms of the claims.

It is a schematic block diagram which shows the whole structure of the digital broadcast receiver according to Embodiment 1 of this invention. FIG. 2 is a block diagram showing a configuration of an OFDM demodulator 102 in FIG. 1. FIG. 3 is a block diagram showing a configuration of a guard interval correlation circuit 206 in FIG. 2. FIG. 4 is an operation waveform diagram for explaining the operation of the guard interval correlation circuit 206 of FIG. 3. It is a block diagram which shows the structure of the equalization circuit 214a in FIG. 6 is a schematic diagram for explaining the relationship between input / output signals and Doppler frequency in FFT circuit 210. FIG. FIG. 6 is a block diagram showing a configuration of a Doppler frequency estimation circuit 602 in FIG. 5. It is a figure which shows an example of the insertion pattern of the pilot signal proposed in the terrestrial digital television broadcast. It is a block diagram which shows the structure of the equalization circuit in the digital broadcast receiver according to Embodiment 2 of this invention. FIG. 10 is a block diagram illustrating a configuration of a delay time estimation circuit 612 in FIG. 9. It is a block diagram which shows the 2nd structure of the delay time estimation circuit in FIG. 12 is a delay profile output from the FFT circuit 710 in FIG. It is a block diagram which shows the structure of the equalization circuit in the digital broadcast receiver according to Embodiment 3 of this invention. It is a wave form diagram for demonstrating an OFDM modulation signal. It is a figure which shows an example of the insertion pattern of the pilot signal proposed in the terrestrial digital television broadcast. For example, it is a block diagram which shows the structure of the conventional digital broadcast receiver described in patent document 1. FIG.

Explanation of symbols

  100,802 tuner, 102 OFDM demodulator, 104 TS decoder, 110 MPEG decoder, 120 additional sound generator, 122 PCM decoder, 130 on-screen display processor, 140 ROM, 142 RAM, 144 arithmetic processor, 146 high-speed digital Interface, 148 Built-in storage device, 150 Modem, 152 IC card interface, 160.1, 160.2 Synthesizer, 162 Audio output terminal, 164 Video output terminal, 180 External storage device, 182 External input device, 201 A / D conversion 202 I / Q separation unit 203, 211 input terminal 204 first carrier synchronization unit 206 guard interval correlation circuit 209 215 output terminal 210 710 810 FFT circuit 212 2-carrier synchronization unit, 214a to 214c, 814 equalization circuit, 216 frequency deinterleave circuit, 218 time deinterleave circuit, 220, 818 demapping circuit, 222 bit deinterleave circuit, 224 Viterbi decoding circuit, 226 byte deinterleave circuit, 228 TS reproduction circuit, 230 RS decoding circuit, 300 mode / guard interval determination circuit, 302 symbol synchronization unit, 304 clock synchronization unit, 303, 305, 508 output terminal, 400 guard interval correlation detection unit, 402 delay memory, 404 correlator 406 Moving average circuit, 408 Absolute value addition circuit, 500 mode / guard interval determination / setting unit, 502 Triangular wave peak value detection circuit, 504 Peak value comparison circuit, 506 Mode / guard interface Global setting circuit, 600,900 pilot signal extraction circuit, 602 Doppler frequency estimation circuit, 606a to 606c filter coefficient ROM, 604 interpolation filter selection circuit, 608 transmission path estimation circuit, 610,904 division circuit, 612, 614 delay time estimation circuit 700a, 700b memory circuit, 702a, 702b correlation circuit, 704a, 704b addition circuit, 706a-706c filter circuit, 708a-708c decision circuit, 800 antenna, 804, 806 multiplier, 808 carrier generation circuit, 812 FFT window circuit, 816 TPS detection circuit, 820 control circuit, 902 interpolation filter.

Claims (7)

A digital broadcast receiver capable of receiving a reception signal of an orthogonal frequency division multiplex transmission system specified by a combination of an effective symbol period length and a guard interval length,
A mode / guard interval determination means for receiving an in-phase signal and a quadrature axis signal after quadrature detection and determining an effective symbol period length and a guard interval length of the received signal;
Fourier transform means for converting the in-phase signal and the orthogonal axis signal from a time domain to a frequency domain signal;
Equalizing means for equalizing the frequency domain signal based on the estimated channel characteristics;
The equalizing means includes
Extracting means for extracting a known pilot signal from the frequency domain signal;
A channel characteristic estimating unit for estimating the channel characteristic using an interpolation filter unit for interpolating the channel characteristic from the pilot signal;
Dividing means for dividing the signal in the frequency domain by the estimated transmission path characteristics,
The transmission path characteristic estimation means includes
Doppler frequency estimation means for estimating a Doppler frequency of a transmission line from the extracted pilot signal;
Interpolation filter selection means for selecting a filter coefficient corresponding to a predetermined number of tap outputs of the interpolation filter means based on the estimation result of the Doppler frequency and the determination result of the effective symbol period length and the guard interval length. ,
The Doppler frequency estimation means includes
A plurality of first pilot signals included in the first symbol and a second symbol included in each of the plurality of first pilot signals, and each of the plurality of first pilot signals matches a subcarrier frequency; and a correlation value between the plurality of second pilot signals adjacent along the time direction, and means for calculating, respectively Re their,
Adding means for adding the plurality of calculated correlation values;
A digital broadcast receiving apparatus comprising: determination means for determining the magnitude of the Doppler frequency depending on whether or not the addition result of the correlation value exceeds a predetermined threshold value.   The transmission path characteristic estimation unit stores a plurality of sets of the filter coefficients specified by a combination of a plurality of effective symbol period lengths and a plurality of guard interval lengths and a magnitude of the Doppler frequency, and the interpolation filter selection unit The digital broadcast receiving apparatus according to claim 1, further comprising a storage unit that selects and outputs one set from the plurality of sets of filter coefficients in response to an instruction. A digital broadcast receiver capable of receiving a reception signal of an orthogonal frequency division multiplex transmission system specified by a combination of an effective symbol period length and a guard interval length,
A mode / guard interval determination means for receiving an in-phase signal and a quadrature axis signal after quadrature detection and determining an effective symbol period length and a guard interval length of the received signal;
Fourier transform means for converting the in-phase signal and the orthogonal axis signal from a time domain to a frequency domain signal;
Equalizing means for equalizing the frequency domain signal based on the estimated channel characteristics;
The equalizing means includes
Extracting means for extracting a known pilot signal from the frequency domain signal;
A channel characteristic estimating unit for estimating the channel characteristic using an interpolation filter unit for interpolating the channel characteristic from the pilot signal;
Dividing means for dividing the signal in the frequency domain by the estimated transmission path characteristics,
The transmission path characteristic estimation means includes
Delay time estimating means for estimating a delay time of a transmission line from the extracted pilot signal;
Interpolation filter selection means for selecting filter coefficients corresponding to a predetermined number of tap outputs of the interpolation filter means based on the estimation result of the delay time and the determination results of the effective symbol period length and the guard interval length. ,
The delay time estimating means includes:
A plurality of first pilot signals included in a predetermined symbol and a plurality of first pilot signals included in the predetermined symbol, each of which is adjacent to each of the plurality of first pilot signals with a data signal therebetween. a correlation value between the second pilot signal, means for calculating, respectively Re their,
Adding means for adding the plurality of calculated correlation values;
And a determination unit that determines whether the delay time is large or not based on whether or not the addition result of the correlation value exceeds a predetermined threshold value.   The transmission path characteristic estimation means holds a plurality of sets of the filter coefficients specified by combinations of a plurality of effective symbol period lengths and a plurality of guard interval lengths and the delay time as stored data, and the interpolation filter 4. The digital broadcast receiving apparatus according to claim 3, further comprising storage means for selecting and outputting one set from the plurality of sets of filter coefficients in accordance with an instruction from the selection means. A digital broadcast receiver capable of receiving a reception signal of an orthogonal frequency division multiplex transmission system specified by a combination of an effective symbol period length and a guard interval length,
A mode / guard interval determination means for receiving an in-phase signal and a quadrature axis signal after quadrature detection and determining an effective symbol period length and a guard interval length of the received signal;
Fourier transform means for converting the in-phase signal and the orthogonal axis signal from a time domain to a frequency domain signal;
Equalizing means for equalizing the frequency domain signal based on the estimated channel characteristics;
The equalizing means includes
Extracting means for extracting a known pilot signal from the frequency domain signal;
A channel characteristic estimating unit for estimating the channel characteristic using an interpolation filter unit for interpolating the channel characteristic from the pilot signal;
Dividing means for dividing the signal in the frequency domain by the estimated transmission path characteristics,
The transmission path characteristic estimation means includes
Doppler frequency estimation means for estimating a Doppler frequency of a transmission line from the extracted pilot signal;
Delay time estimating means for estimating a delay time of a transmission line from the extracted pilot signal;
Interpolation filter selection for selecting filter coefficients corresponding to a predetermined number of tap outputs of the interpolation filter means based on the estimation result of the Doppler frequency and the delay time and the determination result of the effective symbol period length and the guard interval length Means,
The Doppler frequency estimation means includes
A plurality of first pilot signals included in the first symbol and a second symbol included in each of the plurality of first pilot signals, and each of the plurality of first pilot signals matches a subcarrier frequency; and time first correlation value between the plurality of second pilot signals adjacent to each other along the direction, and means for calculating, respectively Re their,
Adding means for adding a plurality of calculated first correlation values;
Determining means for determining the magnitude of the Doppler frequency according to whether or not the addition result of the first correlation value exceeds a predetermined threshold;
The delay time estimating means includes:
A plurality of first pilot signals included in a predetermined symbol and a plurality of first pilot signals included in the predetermined symbol, each of which is adjacent to each of the plurality of first pilot signals with a data signal therebetween. a second correlation value between the second pilot signal, means for calculating, respectively Re their,
Adding means for adding the plurality of calculated second correlation values;
A digital broadcast receiving apparatus comprising: a determination unit configured to determine the magnitude of the delay time based on whether or not the addition result of the second correlation value exceeds a predetermined threshold value.   The transmission path characteristic estimation means stores a plurality of sets of filter coefficients specified by a combination of a plurality of effective symbol period lengths and a plurality of guard interval lengths, a magnitude of the Doppler frequency, and a magnitude of the delay time. The digital broadcast receiving apparatus according to claim 5, further comprising a storage unit that holds and outputs one set from the plurality of sets of filter coefficients in accordance with an instruction from the interpolation filter selection unit. The mode / guard interval determination means includes:
Delay means for receiving the in-phase signal and the quadrature axis signal after quadrature detection and delaying a period corresponding to one of the plurality of effective symbol period lengths;
Correlation detecting means for detecting a correlation between the in-phase signal and the quadrature axis signal and the delayed in-phase axis signal and the quadrature axis signal;
Moving average means for receiving the output of the correlation detecting means and outputting moving average processing for a period corresponding to any of the plurality of guard interval lengths;
Absolute value adding means for outputting the sum of absolute values of the output of the moving average means;
Peak value detecting means for detecting the maximum peak value of the output value of the absolute value adding means;
Depending on the output of the peak value detecting means, the delay amount of the delay means and the setting for designating the period for performing the moving average processing of the moving average means are set to the effective symbol period length and the guard interval length of the received signal. 7. The digital broadcast receiving apparatus according to claim 1, further comprising: a determination unit that determines whether or not they match and updates the setting according to a result of the determination.

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