JP3399017B2 - Phase synchronizer - Google Patents

Description

Description: BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a phase synchronizer used in a direct conversion FM receiver and the like. 2. Description of the Related Art A technique disclosed in US Pat. No. 4,653,117 is known as a technique for performing phase synchronization control in a direct conversion FM receiver. In the technique disclosed herein, a phase comparison signal for phase synchronization is extracted by comparing the phase of the up-converted signal with the local oscillator (local oscillation) signal used here. But,
Such a phase synchronizing means requires a frequency divider especially when the frequency of the local oscillation signal is high, which causes a problem that the phase comparison sensitivity is lowered. SUMMARY OF THE INVENTION The present invention has been made in view of the above points, and is intended to execute phase synchronization control without comparing the phase between an up-conversion signal and a local oscillation signal. Even when using a frequency divider,
An object of the present invention is to provide a phase synchronizer which can maintain a sufficient phase comparison sensitivity and can be applied to a direct conversion FM receiver or the like. A phase synchronizer according to the present invention is characterized in that an input signal is firstly shifted by first and second frequency signals obtained by shifting an oscillation signal from a controlled oscillation means by 90 °. And a quadrature base for converting to a second baseband signal
Band converting means, and the converted orthogonal baseband signal and
A quadrature signal of a predetermined frequency with a 90 ° phase difference
And the multiplied first and second bases
Image rejection for adding and subtracting band signals
Frequency conversion using an up-conversion mixer
Means and the position of the added and subtracted signals relative to each other.
Comparison means for phase or frequency comparison and this comparison means
The controlled oscillator is controlled by the output corresponding to the comparison result.
A synchronous control means for performing synchronous control.
is there. [0005] That is, in the phase synchronizer configured as described above, the first and second baseband signals are shifted by 90 °.
By multiplying by a quadrature signal of a predetermined frequency in which the phase difference is set, an up-converted signal is obtained.The phase difference between the sum of the up-converted signals and the difference obtained by subtracting the up-converted signals is , The phase difference is twice as large as the phase difference with the local oscillation signal. Therefore, a phase comparison signal for synchronizing the controlled oscillating means with the received signal is obtained from the phase comparison result of the two up-conversion signals. The phase comparison sensitivity at this time is twice as high as that obtained when comparing the conventional local oscillation output with the up-conversion signal. An embodiment of the present invention will be described below with reference to the drawings. FIG. 1 shows an embodiment in which the present invention is applied to a direct-conversion FM receiver.
Entered in 2. And this mixer 111 and 112
, A 90 ° phase shifter 13 supplied with an oscillation signal from a voltage controlled oscillator (VCO) 12 constituting a controlled oscillator.
And a signal having a phase difference of 90 ° is provided. That is, the received signal is supplied to the mixer 111 and the mixer 111.
In 112, the baseband quadrature signal from each of the mixers 111 and 112 is converted into a baseband quadrature signal by a low-pass filter (LPF) 14.
The LPFs 141 and 142 remove adjacent undesired wave signal components. Output signals from the LPFs 141 and 142 are input to mixers 151 and 152, respectively. The mixers 151 and 152 have 90 °
A signal from a local oscillator (local oscillator) 17 having a 90 ° phase difference set by the phase shifter 16 is supplied to the mixer 15.
The output signals from 1 and 152 are added in an adder 18 and subtracted in a subtractor 19. The mixers 151 and 152 have a 90 ° phase shifter 16,
7. Together with the adder 18 and the subtractor 19, an image rejection type up-conversion mixer is constituted. That is, the baseband quadrature signals output from the LPFs 141 and 142 are mixed with the mixers 151 and 15 respectively.
Upconverted by 2,
The input from the local oscillator 17 of the mixers 151 and 152 is 9
Since they are orthogonal by the 0 ° phase shifter 16,
The outputs from mixers 151 and 152 are added to adder 18
, And subtracted by subtractor 19, the received signal is
The frequency can be converted to 17 frequencies. At this time, the frequency-converted signal is
Since the undesired wave components have been removed in the LPFs 141 and 142, the demodulation circuit 20 FM-demodulates the output signal from the subtracter 19 to obtain a received demodulated signal. Output signals from the adder 18 and the subtractor 19 are input to limiter amplifiers 211 and 212, respectively.
Amplify each with the limiter applied at the required level. The output signals from the limiter amplifiers 211 and 212 are compared in phase by the phase comparator 22.
The signal obtained corresponding to the phase comparison result is supplied to the loop filter 23, integrated, and converted to DC. The DC signal corresponding to the phase comparison result from the loop filter 23 is VC
This is supplied to O12 so that the oscillation frequency of the VCO 12 is controlled in accordance with the result of the phase comparison. Here, if the oscillation frequency of the VCO 12 is the same as that of the received signal, the outputs from the adder 18 and the subtractor 19 each have an opposite spectrum configuration around the oscillation frequency of the local oscillator 17. . Even if the oscillation frequency of the VCO 12 and the received signal are different, the adder
Since the outputs of 18 and the subtracter 19 are vertically shifted with respect to the oscillation frequency of the local oscillator 17, the frequency difference between them is twice as large as the difference from the oscillation frequency of the VCO 12. The same can be said for the phase. Accordingly, the phase difference or the frequency difference between the output signals from the adder 18 and the subtractor 19 is detected by the comparator 22, and the VCO 12 is detected in the same manner as in the ordinary PLL synthesizer.
, The oscillation signal of the VCO 12 can be synchronized with the received signal. The above operation is performed by changing the received signal center frequency ω _s ,
The oscillation frequency ω _u of the VCO 12 will be described below. Cos ｛ω where the received signal is a modulation component i (t)
_{s t + i (t)}} , 90 ° 2 two outputs respectively cos .omega _vco t of the phase shifter 13, it is expressed as sin .omega _vco t, mixer
The outputs from 111 and 112 are given by the following equations (1) and (2), respectively, assuming that the gain is "1". [0016] (mixer 111 output) = (1/2) [cos { ω s t + i (t) -ω vco t} + COS {ω s t + i (t) + ω vco t}] ............ (1) ( mixer 112 output) = (1/2) [sin { -ω s t-i (t) + ω vco t} + sin {ω s t + i (t) + ω vco t}] ............ (2) the (1 ) And (2), the second term is LP
Since it is removed in F141 and 142,
Each is as follows. (The output of the LPF141) = (1/2) [cos {ω s t + i
(t) −ω _vco t｝ (output of LPF 142) = (1/2) [sin ｛−ω _s t−
Expressed _{i (t) + ω vco t} } 90 ° phase shifter 16 of the two outputs respectively cos .omega _u t and sin .omega _u t, each output of the mixer 151 and 152 is as follows. (Mixer 151 output) = (1/4) [cos ｛ω _s t + i
_{(t) -ω vco t-ω} u t} + cos {ω s t + i (t) -ω
_{vco t + ω u t}]} ( mixer 152 output) = (1/4) [cos { -ω s t-
_{i (t) + ω vco t} -ω u t} - cos {-ω s t-i (t)
+ Ω _vco t + ω _u t}] Therefore, the outputs from the adder 18 and the subtracter 19 are as shown by the equations (3) and (4). [0017] (adder 18 output) = (1/2) cos {ω s t + i (t) -ω vco t + ω u t} ......... (3) ( subtractor 19 output) = (1/2) cos { _{ω s t + i (t)} -ω vco t-ω u t} ......... (4) , where the angular frequency difference between the received signal and vCO 12 omega _e (=
ω _s −ω _vco ), the above equations (3) and (4) are respectively (output of the adder 18) = (1/2) cos ｛ω _ut + i (t) +
ω _e t} (subtractor 19 output) = (1/2) cos {ω u t-i (t) -
ω｝ 、 、 し た_がっ て 、 、 、 、 、 、 、 、 、 、 、, 、 _した が っ て 、 、 し_たが っ て し_たが っ て 、 、 、 、 、 、 、 、. Since the phase comparator 22 supplies a control voltage corresponding to the phase difference to the VCO 12 through the loop filter 23, the oscillating output of the VCO 12 is the same as that of a normal PLL loop. , And the sensitivity is doubled. As shown in this embodiment, when this phase synchronizer is used as an FM receiver, i (t)
The VCO 12 can be frequency-locked to the received signal frequency by removing the components related to the VCO 12 with the loop filter 23. FIG. 2 shows an example in which the phase synchronizer as shown in FIG. 1 is applied to a high-frequency multi-channel receiver such as a receiver of a car telephone. Band-pass filter (B
PF) 30. The received signal that has passed through the BPF 30 is amplified by a high-frequency low-noise amplifier 31 having sufficient gain and noise figure characteristics in a desired receiving frequency band, and an output received signal from the amplifier 31 is supplied to a mixer 32. The mixer 32 has a PLL synthesizer.
The PLL synthesizer 33 is used as a local oscillator for converting an input signal from the amplifier 31 into a required intermediate frequency. Therefore, the PLL synthesizer 33 includes a variable frequency divider, a VCO,
It is configured by a phase comparator, a loop filter, and the like, and oscillates on multiple channels based on an input reference frequency signal. The output signal of the intermediate frequency from the mixer 32 is amplified by the amplifier 34 and then supplied to the BPF 35 so that the output signal from the BPF 35 is input to the mixers 111 and 112 constituting the phase locked loop. . The mixers 151 and 152 in the phase synchronization circuit are supplied with signals from the 1/2 frequency dividers 361 and 362, respectively.
In 1, the input reference frequency signal is divided by a 1/64 frequency divider 37.
, And the signal inverted by the inverter 38 is input. The output from the 1/64 frequency divider 37 is supplied to the 1/2 frequency divider 362 as it is.
In 361 and 362, a quadrature signal based on the signal from the 1/64 frequency divider 37 is generated. In this case, particularly, in order to fix the output phase relationship, the frequency dividers 361 and 3623 are connected to each other. The output from the adder 18 is supplied to a demodulation circuit.
39, limiter amplifier 40, monostable multivibrator 41,
An LPF 42 and a low-frequency amplifier 43 are provided, and a demodulated signal is output from the low-frequency amplifier 43. In this demodulation circuit, a monostable multivibrator 41 replaces the FM modulation signal with a pulse number change,
The demodulated signal is obtained by integration by the LPF. That is, in the receiver configured as described above, the input received signal is converted into a constant intermediate frequency signal in the mixer 32 based on the signal from the PLL synthesizer 33. Then, the intermediate frequency signal is amplified in the amplifier 34 and the BPF 35, and the interference wave is removed, and is input to the mixers 111 and 112. Thereafter, the signals are converted into baseband quadrature signals and supplied to mixers 151 and 152 in the same manner as described in the embodiment of FIG.
Since the signal input to 2 is a quadrature signal, the VCO 12 can be locked at a predetermined intermediate frequency based on the output from the adder 18 and the subtractor 19. In this example, the received signal is converted to a frequency of 1/128 of the input reference frequency and FM-demodulated. FIG. 3 shows a third embodiment. In this embodiment, the FM in the second embodiment shown in FIG.
The limiter amplifier 40 for detection and the limiter amplifier 212 for phase comparison are shared. The output from the limiter amplifier 211 to which the output from the subtractor 19 is supplied to the phase comparator 22 and the output from the limiter amplifier 40 for FM detection are supplied to the phase comparator 22, and the output from the limiter amplifier 211 and the limiter The phase comparison of the output from the amplifier 40 is performed. Therefore, the number of components is reduced as compared with the embodiment of FIG. 2, and the same effect can be obtained. In the fourth embodiment shown in FIG.
The LPFs 141 and 142 in the embodiment shown in FIG. 4 are constituted by two sets of LPFs 451 and 452 and 461 and 462, respectively. The output from the mixer 111 is supplied to one of the LPFs 451 and 452 via the changeover switch 471, and the output from one of the selected LPFs 451 and 452 is supplied to the mixer 151 via the changeover switch 472. . The output from the mixer 112 is supplied to the LP through a switch 481.
The output from the selected LPF 461 or 462 is input to the mixer 152 via the changeover switch 482. The LPFs 451 and 461 have higher cutoff frequencies than the LPFs 452 and 462, respectively. With this configuration, it is possible to select an optimum receiving band for the occupied band of the received FM signal. Further, since the input bandwidth to the phase comparator 22 is widened, the lock pull-in range of the VCO 12 is widened. The configuration shown in this embodiment can be combined with the embodiment shown in FIGS. 2 and 3. In the embodiment described above, an example in which the present invention is applied particularly to a direct conversion type FM receiver has been described. However, the present invention is not limited to such an example as a circuit for synchronizing an oscillation signal with respect to an input signal. Other applications are possible. Also, regarding the application to the receiver, F
It can be applied to not only the M receiver but also each modulation system such as an amplitude modulation system, a phase modulation system, and a pulse modulation system, and as a receiver for a combination signal thereof, similarly to a normal direct conversion receiver. . In the embodiment, the mixer, the LPF, and the like are described as being configured by hardware. However, these components can appropriately realize similar functions by digital signal processing. As described above, in the phase synchronization device according to the present invention, the phase synchronization control is performed without comparing the phase of the up-conversion signal with the local oscillation signal. In addition, even when a frequency divider or the like is used, a sufficient phase comparison sensitivity can be obtained, so that it can be effectively applied to a direct conversion FM receiver or the like.

BRIEF DESCRIPTION OF THE DRAWINGS FIG. 1 is a circuit configuration diagram for explaining a phase synchronizer according to an embodiment of the present invention. FIG. 2 is a circuit diagram illustrating a second embodiment in which the phase synchronizer is applied to a high-frequency multi-channel telephone receiver. FIG. 3 is a circuit diagram illustrating a third embodiment. FIG. 4 is a circuit configuration diagram illustrating a fourth embodiment of the synchronous phase device. [Description of References] 111, 112, 151, 152 ... mixer, 12 ... voltage-controlled oscillator (VCO), 13, 16 ... 90-degree phase shifter, 141, 142 ...
Low pass filter (LPF), 17: local oscillator (local oscillation), 18: adder, 19: subtractor, 20: FM demodulation circuit.

──────────────────────────────────────────────────続 き Continuation of the front page (56) References JP-A-4-79546 (JP, A) JP-A-6-77737 (JP, A) JP-A-64-72626 (JP, A) 83038 (JP, B2) JP-B-53-5549 (JP, B2) JP-B-63-47313 (JP, B1) Patent 2964573 (JP, B2) (58) Fields investigated (Int. Cl. ⁷ , DB name) H04L 27/00 H03D 7/00 H04B 14/00

Claims (1)

(57) The first and second baseband signals are converted from an input signal by first and second frequency signals obtained by shifting the oscillation signal from the controlled oscillation means by 90 °. Orthogonal baseband converting means for converting the orthogonal orthogonal baseband signals into 90 °
Multiply by a quadrature signal of a predetermined frequency with the phase difference set ,
The multiplied first and second baseband signals
Image rejection type
Frequency conversion means comprising a pre-conversion mixer ; comparison means for comparing the signals of the addition and subtraction results with each other in phase or frequency; and an output corresponding to the comparison result from the comparison means. A phase synchronization device, comprising: synchronization control means for performing synchronization control.