JP3353781B2 - Motor control device - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

The present invention relates to a motor control device for controlling a permanent magnet synchronous motor and the like. In particular, the present invention relates to a motor control device that stably controls a motor even at the time of voltage saturation in current control.

[0002]

2. Description of the Related Art Induction motors
tor. (Hereinafter referred to as IM) is connected to a commercial power supply, is rotated at a substantially constant speed, and controls the air flow by the damper. On the other hand, the variable speed control using the inverter to control the rotation speed of the IM is efficient from a low air volume to a high air volume. In recent years, demand for energy saving has been increasing, and replacement of damper control with variable speed control using an inverter has begun.

Further, a permanent magnet synchronous motor (Permanent Ma
gnet Sychronous Motor. Hereinafter, abbreviated as PMSM)
Are highly efficient because a permanent magnet arranged in the rotor generates a field. However, since it is necessary to change the current of the stator winding in synchronization with the magnetic pole position, an inverter is indispensable.

[0004] Synchronous Reluctance Motor (hereinafter SynRM)
) Is low in cost because it does not have a permanent magnet. In this SynRM, like the PMSM, it is necessary to change the current of the stator winding in synchronization with the rotor angle, and thus an inverter is essential.

As described above, the high-efficiency drive control of the IM, the PM
An inverter is required for SM drive control and SynRM drive control. This inverter uses a DC power supply obtained by rectifying and smoothing a commercial power supply when applying AC power to the stator winding using PWM control or the like. Therefore, a voltage higher than the voltage of the DC power supply cannot be applied. Then, when controlling the motor using current control, a sufficient voltage cannot be applied at the time of high load and high speed rotation, and the voltage is saturated. In such a region where the voltage is saturated, responsiveness is deteriorated due to operation amount saturation, and noise and vibration are generated due to current waveform distortion.

[0006] A motor control device which realizes good control even when the voltage is saturated is disclosed in Japanese Patent Application Laid-Open No. H11-308900.
The one disclosed in the gazette (hereinafter referred to as a conventional example) is known. Hereinafter, this conventional motor control device will be described.

FIG. 14 is a block diagram showing a configuration of a motor control device in a conventional example. Here, a Surface Permanent Magnet Synchronous M
otor. Hereinafter, an example of controlling the SPMSM 110 will be described. The conventional motor control device includes a q-axis current command value iq *, a d-axis voltage command value vd *, and a q-axis voltage command value vq
Perform anti-wind-up on *.

That is, the q-axis current command value iq * 'is created by using the proportional-integral control 135 so that the speed ω of the rotor 112 becomes equal to the speed command value ω *. Then, a limit 136 is applied to the q-axis current command value iq * '. Here, when the limit 136 operates, the speed command value ω * is reduced by an amount Δeω proportional to the difference Δiq before and after this. Then, the saturation of the q-axis current command value iq * is avoided.

The d-axis current value id is equal to the d-axis current command value i.
The d-axis voltage command value vd * 'is created using the proportional-integral control 153 so as to satisfy d *. Then, a limit 154 is applied to the d-axis voltage command value vd * '. Here, when the limit 154 operates, the difference Δ
The d-axis current command value id * is reduced by an amount eid proportional to vd. Then, the saturation of the d-axis voltage command value vd * is avoided.

The q-axis current value iq is equal to the q-axis current command value i.
The q-axis voltage command value vq * 'is created using the proportional-integral control 163 so that q * is satisfied. Then, a limit 164 is applied to the q-axis voltage command value vq * '. Here, when the limit 164 operates, the difference Δ
The q-axis current command value iq * after the limit is reduced by an amount eiq proportional to vq. Then, the saturation of the q-axis voltage command value vq * is avoided.

As described above, the conventional motor control device is
Performs anti-wind-up on q-axis current command value iq *, d-axis voltage command value vd *, and q-axis voltage command value vq *, deteriorating responsiveness due to operation amount saturation , and generating noise and vibration due to current waveform distortion. Was suppressed.

[0012]

The prior art motor control apparatus includes a proportional integral control 153 and a proportional integral control 1 respectively.
The d-axis voltage command value vd * and the q-axis voltage command value v obtained in 63
q * is used directly for saturation detection. Therefore, the induced voltage
Interior permanent magnet synchronous motor distortion of a large (InteriorPerman
ent Magnet Synchronous Motor: IPMSM), the d-axis voltage command value vd * and the q-axis voltage command value vq *
Vibrates, the amounts Δeid and Δeiq for decreasing the current command value vibrate, and the operation becomes unstable.

The present invention stably performs anti-wind-up even in a motor having a large induced voltage distortion,
It is an object to suppress deterioration of responsiveness due to operation amount saturation and generation of noise and vibration due to current waveform distortion.

[0014]

[0015]

In order to achieve the above object, a motor control device according to the present invention comprises a speed command value creating means for creating a pre-change speed command value indicating a command value of a rotor speed; Speed command value changing means for changing the speed command value before change to create a speed command value, speed detecting means for detecting the speed of the rotor, and a speed command value based on the speed command value and the speed.
Current limit indicating the command value of the current flowing through the stator winding
Current command value creating means for creating a command value;
Current command value limiter that limits the command value and creates a current command value
A stage, a current detecting means for detecting a phase current value indicating a current flowing through the stator winding, and a limit value indicating a command value of a voltage applied to the stator winding based on the current command value and the phase current value. Voltage command value creating means for creating a voltage command value; voltage command value limiting means for creating a voltage command value by limiting the voltage command value before limitation; and applying power to the stator winding based on the voltage command value. A driving unit, wherein the saturation control unit generates a saturation indicating how much the voltage command value is saturated based on the pre-limit voltage command value or the voltage command value; and Reference value creating means for creating a reference value indicating the reference of the saturation, wherein the speed command value changing means changes the speed command value when the saturation becomes larger than the reference value.
Change the previous speed command value so that it decreases
It is assumed that. With this configuration, it is possible to realize a motor control device that suppresses deterioration of responsiveness due to operation amount saturation and generation of noise and vibration due to current waveform distortion.

Further, in the motor control device of the present invention, the saturation creating means may create the saturation based on the pre-limit voltage command value or a value obtained by applying a filter to the voltage command value. It is. With this configuration, even in the motor high strain of the induced voltage, stably perform an anti-wind-up, the deterioration of responsiveness by the operation amount impregnating, suppresses motor controller generation of noise and vibration due to current waveform distortion realized I do.

Further, in the motor control device according to the present invention, the voltage command value creating means creates a voltage command value proportional component by a proportional operation, creates a voltage command value integral component by an integration operation, and Create the pre-limit voltage command value based on the value proportional component and the voltage command value integral component,
The saturation degree creation means creates the saturation degree based on the voltage command value integral component. With this configuration,
A motor control device that stably performs anti-wind-up even in a motor having a large induced voltage distortion and suppresses deterioration of responsiveness due to operation amount saturation and noise / vibration due to current waveform distortion.

Further, in the motor control device according to the present invention, the reference value generating means sets the reference value to be larger than a maximum voltage value which can be applied by the driving means. With this configuration, a motor control device that increases the effective value of the applied voltage and increases the output of the motor is realized.

[0019]

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS Hereinafter, specific embodiments of a motor control device according to the present invention will be described with reference to the accompanying drawings.

Embodiment 1 Hereinafter, a motor control device according to Embodiment 1 of the present invention will be described. The motor control device according to the first embodiment applies a low-pass filter to the voltage command value obtained by the proportional-integral control, creates saturation, and performs anti-windup. By doing so, anti-wind-up is stably performed even in a motor having a large induced voltage distortion, and the deterioration of the response due to the saturation of the operation amount and the generation of noise and vibration due to the current waveform distortion are suppressed.

Further, the motor control device according to the first embodiment
A reference value is provided separately from the limit applied to the voltage command value, and the saturation is compared with the reference value to perform anti-windup. By doing so, even when a low-pass filter is applied to the voltage command value, anti-wind-up is performed without reducing the output, the response is degraded due to the saturation of the manipulated variable, and noise and noise due to current waveform distortion are reduced.
Suppress the occurrence of vibration.

Further, the motor control device according to the first embodiment
The reference value is made larger than the maximum voltage value that can be applied by the driving means. By doing so, the effective value of the applied voltage is increased, and the output of the motor is increased.

[Structure of Motor] First, an interior permanent magnet synchronous motor (Interior motor) controlled by the motor controller of the first embodiment
or Permanent Magnet Synchronous Motor. After that, IP
(Abbreviated as MSM) will be described. FIG. 1 is a block diagram illustrating a configuration of the motor control device according to the first embodiment.

The IPMSM 10 is a laminated electromagnetic steel sheet product and has a stator (not shown) having teeth, and a stator winding 11u which is a coated copper wire and is wound around the teeth of a stator (not shown) through which a phase current flows. 11v, 11w, and a rotor 12 arranged close to and facing the stator, and an optical encoder 13 arranged to have the same rotation center as the rotor 12 and outputting encoder signals Z, A, B. . Here, the stator windings 11u, 11v, 11w
Are connected in a Y-connection (connection in which one end of each of the stator windings 11u, 11v, 11w is connected at one point).

The rotor 12 is rotatably supported, and has a shaft 14, a rotor yoke 15 having the same rotation center as the shaft 14 and a laminated magnetic steel sheet product, and a rotor yoke 1.
5 and a permanent magnet 16 arranged inside.
The magnetic permeability of the permanent magnet 16 is significantly smaller than that of the rotor yoke 15, and the rotor yoke 15 is not cylindrical. Therefore, in the rotor 12, the ease of passing the magnetic flux is not uniform and the saliency is large.

The torque (magnet torque) is generated by the interaction between the rotating magnetic field generated by the phase current flowing through the stator windings 11u, 11v, and 11w and the field generated by the permanent magnet 16. In addition, a torque (reluctance torque) is generated by using a difference in ease according to the magnetic flux and generating a rotating magnetic flux by a phase current.

[Configuration of Motor Control Device of First Embodiment]
Next, the configuration of the motor control device according to the first embodiment will be described. The motor control device 20 according to the first embodiment includes current sensors 21u and 21v, each of which includes a DC current transformer and outputs an analog u-phase current value ua and an analog v-phase current value iva, and encoder signals Z, A, and B, respectively. A microcomputer 22 that receives an analog u-phase current value ua, an analog v-phase current value iva, and an analog speed command value ω * and outputs switching command signals guh, gul, gvh, gvl, gwh, and gwl; and a switching command signal guh,
and gul, gvh, gvl, gwh, and gwl, and a drive unit 23 that applies electric power to the stator windings 11u, 11v, and 11w.

FIG. 2 shows a driving unit 23 according to the first embodiment.
FIG. 2 is a circuit diagram showing the configuration of FIG. The driving unit 23 includes a power supply 23a
And an upper IGBT (Insulated Gate Bipolar Transistor) having a collector connected to the positive electrode of the power supply 23a and an emitter connected to the stator windings 11u, 11v, 11w, respectively.
r: insulated gate bipolar transistor) 23b
u, 23bv, 23bw, upper IGBT 23bu,
Upper flywheel diodes 23cu, 23cv, 23cv respectively connected in anti-parallel to 3bw, 23bw.
And lower IGBTs 23du, 23dv, and 23dw whose collectors are connected to the stator windings 11u, 11v, and 11w, respectively, and whose emitters are connected to the negative electrode of the power supply 23a. Side flywheel diode 23e
u, 23ev, 23ew and the switching command signal gu
h, gul, gvh, gvl, gwh, and gwl are input, and the gate voltages of the upper IGBTs 23bu, 23bv, and 23bw and the lower IGBTs 23du, 23dv, and 23d, respectively.
gate driver 23f for controlling the gate voltage of w
It is composed of

The microcomputer 22 includes a CPU, an R
It comprises an AM, a ROM, a timer, an ADC, a port, and a bus connecting them. Also, the microcomputer 22
Functionally, an angle creation unit 31 that inputs the encoder signals Z, A, and B and outputs the angle θ, a speed creation unit 32 that inputs the angle θ and outputs the speed ω, and inputs an analog speed command value ω * a Analog-to-digital converter (Analog Digital Converter.
C) 33, a speed subtraction unit 34 for inputting the speed command value ω * and the speed ω and outputting a speed error eω, and inputting the speed error eω and the current command maximum value I * max to obtain the d-axis current It includes a current command value creation unit 35 that outputs a command value id * and a q-axis current command value iq *.

The microcomputer 22 receives an analog u-phase current value uaa and outputs an u-phase current value iu.
And the analog v-phase current value iva and input the v-phase current value iv
, And the angle θ and the u-phase current values iu and v
The phase current value iv is input, and the d-axis current value id and the q-axis current value i
and a three-phase to two-phase conversion unit 43 that outputs q and q.

Further, the microcomputer 22 calculates the d-axis current command value i
A d-axis subtraction unit 52 that inputs d * and d-axis current value id and outputs a d-axis current error eid, and a d-axis current command value that inputs d-axis current error eid and outputs a d-axis voltage command value vd * And a creating unit 53. Also, the microcomputer 22
A q-axis subtraction unit 62 that inputs a q-axis current command value iq * and a q-axis current value iq and outputs a q-axis current error eiq, and inputs a q-axis current error eiq and outputs a q-axis voltage command value vq * q
And a shaft current command value creating unit 63.

Further, the microcomputer 22 calculates a d-axis voltage command value v
Input d * and input a low-pass filter (Low-Pass Filter.
(Hereinafter abbreviated as LPF) d-axis LPF unit 71 that outputs post-d-axis voltage command value vd * l, q that inputs q-axis voltage command value vq * and outputs q-axis voltage command value vq * l after LPF Axis LPF section 72, d-axis current value vd * l after LPF and LPF
After inputting the q-axis current value vq * l and outputting the saturation v2, a saturation generation unit 73 that outputs the reference value v0, a reference value generation unit 74 that outputs the reference value v0, and inputting the saturation v2 and the reference value v0 to input the current Current command maximum value creation unit 75 that outputs the command maximum value I * max
It is comprised including.

The microcomputer 22 receives the angle θ, the d-axis voltage command value vd *, and the q-axis voltage command value vq *, and inputs the u-phase voltage command value vu *, the v-phase voltage command value vv *, and the w-phase voltage. A two-phase / three-phase conversion unit 81 that outputs a command value vw *, a u-phase voltage command value vu *, a v-phase voltage command value vv *, and a w-phase voltage command value vw *, and switching command signals guh, gu
PWM that outputs l, gvh, gvl, gwh, gwl
The control unit 82 is included.

[Principle of Operation of Motor Control Device of First Embodiment] Next, the operation of the motor control device of the first embodiment will be described. The motor control device according to the first embodiment generates the degree of saturation from the voltage command value applied with the LPF,
Do the up. Here, by using the LPF,
Anti-windup is performed stably even in motors with large induced voltage distortion. In addition, the motor control device according to the first embodiment provides a reference value separately from the limit applied to the voltage command value, so that even when the low-pass filter is applied to the voltage command value, the output is not reduced. Performs anti-wind-up, degrades responsiveness due to operation amount saturation, and reduces noise and noise due to current waveform distortion.
Suppress the occurrence of vibration. Further, the motor control device according to the first embodiment increases the effective value of the applied voltage by increasing the reference value to be greater than the maximum voltage value that can be applied by the driving means, thereby increasing the motor output.

First, the coordinate system will be described. FIG. 3 is an explanatory diagram of a coordinate system. FIG. 3 shows an IPMSM having two magnetic poles for the sake of simplicity. The d axis and the q axis are axes based on the actual angle θ of the rotor 12. d
The axis is oriented in the same direction as the magnetic flux generated by the permanent magnet 16 disposed on the rotor 12, and the q axis is oriented 90 degrees ahead of the d axis. The angle between the stator winding 11u and the d axis is defined as an angle θ. Here, the positive direction is the order of the u phase, the v phase, and the w phase.

Next, a description will be given of how to prevent voltage saturation by reducing the current amplitude when the voltage increases. The motor control device according to the first embodiment controls the d-axis current command value id * and the q-axis current command value iq * such that the speed ω * of the rotor 12 rotates according to the speed command value ω *. Perform speed control. The d-axis current value id and the q-axis current value iq, which are the d- and q-axis components of the current flowing through the stator windings 11u, 11v, and 11w, are as follows: d-axis current command value id * and q-axis current command value iq *, respectively. Is performed so as to control the d-axis voltage command value vd * and the q-axis voltage command value vq *.

FIG. 4 shows output torque, d-axis current value id, q-axis current value iq, current amplitude I, d-axis voltage value vd, q-axis voltage value vq, and voltage amplitude V with respect to speed ω in the first embodiment. FIG. 9 is a relationship diagram showing a change. When controlling a fan or the like, as shown in FIG. 4 and the following equation (1), the required output torque T increases with an increase in the speed ω. Where ω
＾ 2 indicates the square value of the speed ω, and Ktw indicates the ratio between the torque T and ω ＾ 2. As shown in the following equation (2), the d-axis current value id and the q-axis current value iq are increased in order to increase the output torque T. Here, P is the number of pole pairs, ψ is the effective value of the dq-axis winding interlinkage magnetic flux ψ, Lq is the q-axis inductance, and Ld is the d-axis inductance. In the equation (2), the first term indicates a magnet torque, and the second term indicates a reluctance torque. Here, in order to generate a positive reluctance torque, the d-axis current value id is made negative. Therefore, the expression that the d-axis current value id increases is expressed by the d-axis current value id.
Indicates that the absolute value of becomes larger. The speed control increases the d-axis current value id * and the q-axis current command value iq *, thereby increasing the d-axis current value id and the q-axis current value iq and increasing the output torque T. Then, as the d-axis current value id and the q-axis current value iq increase, the following equation (3)
The current amplitude I represented by Where √ is ()
Indicates the positive square root of.

T = Ktw · ω ＾ 2 (1) T = P · ｛ψ · iq− (Lq−Ld) · id · iq｝ (2) I = √ (id ＾ 2 + iq ＾ 2) (3) The d-axis voltage value vd and the q-axis voltage value vq, which are the d-axis component and the q-axis component of the voltage in the stator winding, are represented by the following equations (4) and (5). Here, R represents a phase resistance value, and p represents a differential operator. Therefore, the d-axis voltage value vd and the q-axis voltage value vq increase as the speed ω increases. here,
In order to generate a positive reluctance torque, there are many ranges where the d-axis voltage value vd is negative. Accordingly, like the d-axis current value id, the expression that the d-axis voltage value vd increases indicates that the absolute value increases. Then, as the d-axis voltage value vd and the q-axis voltage value vq increase, the following equation (6) is obtained.
Is increased.

Vd = (R + p) · id−ω · Lq · iq (4) vq = (R + p) · iq + ω · Ld · id + ω · ψ (5) V = √ (vd ＾ 2 + vq ＾ 2) (6) As described above, as the speed ω increases, the required voltage amplitude V increases. However, since the voltage of the power supply 23a of the driving unit 23 has a limit, a sufficient voltage cannot be applied in a high-speed and high-load region, and the voltage is saturated. In such a region where the voltage is saturated, the responsiveness is deteriorated due to the operation amount saturation,
Noise and vibration occur due to current waveform distortion.

Therefore, when the voltage amplitude V increases, the current amplitude I is limited to prevent voltage saturation.
That is, a reference value v0 slightly larger than the power supply voltage / 2 is set, and when the voltage amplitude V is larger than the reference value v0 (in FIG. 4, when the current amplitude I and the voltage amplitude V are in the state indicated by ●), The current amplitude I is reduced so that the amplitude V converges to the reference value v0 (in FIG. 4, the current amplitude I and the voltage amplitude V are changed from a state indicated by ● to a state indicated by ○). In this way, anti-wind-up is performed to suppress deterioration of responsiveness due to operation amount saturation and generation of noise and vibration due to current waveform distortion. In the actual operation, the saturation v2 described later is used instead of the voltage amplitude V, and the current command value amplitude I * is limited instead of the current command I.

Next, when creating the saturation degree v2, L
The use of the voltage command value with the PF applied will be described. FIG. 5 shows a d-axis voltage command value vd according to the first embodiment.
*, Q-axis voltage command value vq *, sum of squares of voltage command value (vd
* ＾ 2 + vq * ＾ 2), d-axis voltage command value vd * after LPF
1 is a waveform diagram of a q-axis voltage command value vq * l after LPF and a saturation degree v2. In IPMSM where the induced voltage is distorted,
Even if the d-axis current command value id * and the q-axis current command value iq * are constant, as shown in FIGS. 5A and 5B, the d-axis voltage command value vd * created by the current control. And the q-axis voltage command value vq * oscillates. With this vibration, FIG.
As shown in (c), these sums of squares (vd * ＾ 2 + vq *
＾ 2) also vibrates. Therefore, this sum of squares (vd * ＾ 2
If anti-wind-up is performed based on + vq * ＾ 2), control will not be stable.

On the other hand, the d-axis voltage command value vd * and the q-axis voltage command value vq * are each obtained by applying an LPF to the post-LPF d
The shaft voltage command value vd * l and the q-axis voltage command value vq * l are
As shown in FIGS. 5D and 5E, the vibration is suppressed. Therefore, as shown in FIG. 5F, the oscillation of the saturation v2 (the following equation (7)), which is the sum of the squares, is also suppressed. Therefore, if anti-wind-up is performed based on the degree of saturation v2, the control is stabilized.

V2 = vd * l ＾ 2 + vq * l ＾ 2 (7) Next, the independent setting of the voltage command value limit and the reference value v0 will be described. The d-axis voltage command value creation section 53 and the q-axis voltage command value creation section 63 respectively perform proportional integration control and act as limits to create a d-axis voltage command value vd * and a q-axis voltage command value vq *. Here, if voltage saturation is detected, it may be determined from the difference between the d-axis voltage command values before and after the limit and the difference between the q-axis voltage command values before and after the limit. However, as shown in FIGS. 5A and 5B, d
The shaft voltage command value vd * and the q-axis voltage command value vq * vibrate.

Therefore, as described above, the post-LPF d-axis voltage command value vd * l and the LPF shown in FIGS. 5D and 5E are used.
The saturation degree v2 using the rear q-axis voltage command value vq * l (FIG. 5)
(F)) is prepared, and anti-wind-up is performed based on the saturation v2 and the reference value v0. That is, the saturation is determined from the low frequency component of the voltage command and the anti-wind
Do the up. On the other hand, in order to cope with distortion of the induced voltage, it is necessary to apply the high frequency component of the voltage command as it is. Therefore, it is necessary to independently set the voltage command value limit and the anti-wind-up saturation judgment.

Next, how the reference value v0 is made larger than the maximum voltage value that can be applied by the drive unit 23 will be described. Even if the voltage is somewhat saturated, a sinusoidal current can continue to flow. The output of the motor can be increased as the effective value of the applied voltage increases. Therefore, the output of the motor can be increased by making the reference value v0 larger than the maximum voltage value that can be applied by the drive unit 23.

[Details of Operation of Motor Control Device of First Embodiment] The motor control device of the first embodiment controls the IPMSM 10 based on the above principle. Hereinafter, details of the operation of the motor control device according to the first embodiment will be described.

The optical encoder 13 has a rectangular wave-shaped A and B signals including an NP cycle per one rotation of the mechanical angle of the rotor 12 having a phase difference of 90 ° from each other, and a pulse only once per one rotation of the mechanical angle of the rotor 12. And outputs a Z signal that changes in the form. Here, the optical encoder 13 is attached in advance so that the angle when the Z signal changes in a pulse shape becomes 0 °.

The current sensors 21u and 21v detect the current flowing through the stator windings 11u and 11v, respectively, and create an analog u-phase current value ua and an analog v-phase current value iva.

The drive unit 23 outputs the switching command signal gu
The voltage of the stator windings 11u, 11v, 11w is controlled based on h, gul, gvh, gvl, gwh, gwl. The power supply 23a supplies power to the drive unit 23. The gate driver 23f outputs the switching command signal guh.
Is high, the upper IGBT 23bu is energized, and the gate voltage of the upper IGBT 23bu is controlled such that the upper IGBT 23bu is non-energized when the switching signal guh is L. On the other hand, the gate voltage of the lower IGBT 23du is controlled such that the lower IGBT 23du is energized when the switching command signal gu is H, and de-energized when the switching command signal gu is L. Similarly, for the v-phase and the w-phase, the upper IGBTs 23bv, 23bw, and the lower IGBT 23d based on the switching command signals gvh, gvl, gwh, gwl.
v, 23dw.

The microcomputer 22 is controlled by a certain period (hereinafter, referred to as a certain period).
This cycle is referred to as a control cycle T).
When the operation of these ADCs is completed, the angle creating unit 31, the speed creating unit 32, the speed subtracting unit 34, the current command value creating unit 35, the three-phase two-phase converting unit 43, d-axis subtraction unit 52, d-axis voltage command value creation unit 53, q-axis subtraction unit 62, q-axis voltage command value creation unit 63, d
Axis LPF unit 71, q-axis LPF 72, saturation creation unit 73,
The reference value creation unit 74, the current command maximum value creation unit 75, and the two-phase / three-phase conversion unit 81 are arithmetically executed. Further, the PWM control unit 8
2 is realized by hard logic.

The angle generator 31 outputs the encoder signals Z,
The angle θ is created based on A and B. The angle creation unit 31 includes a counter. The counter value TM of this counter
Is set to 0 when the Z signal changes in a pulse shape. When the A signal and the B signal are in the normal order (when the signals change in the order of A and B), the counter value is incremented by 1 at each rising edge and falling edge. On the other hand, A signal, B
When the signal is in reverse order (when changing in the order of B and A), the counter value is decremented by 1 at each rising and falling edge. Then, the angle creating unit 31 creates θ using the counter value TM as in the following equation (8).

Θ = TM · 2π / (4 · NP) (8) The speed creating unit 32 differentiates the angle θ to create the speed ω.
The velocity ω is created from the difference between the angles θ as in the following equation (9). Here, θ (i) is the angle θ when the speed creating unit 32 operates this time, and θ (i−1) is the angle θ when the speed creating unit 32 operates last time.

Ω = {θ (i) −θ (i−1)} / T (9) The ADC 33 outputs the analog speed command value ω which is an analog value.
* A is converted to analog speed command value ω *
Perform digital conversion. The speed subtractor 34 uses the following equation (1)
0), a difference between the speed command value ω * and the speed ω is set as a speed error eω.

Eω = ω * −ω (10) The current command value creation unit 35 creates a current command using proportional integral control, and further limits the current command based on the speed command maximum value i * max. FIG. 6 is a block diagram illustrating the operation of the current command value creating unit 35 according to the first embodiment. Here, z represents a z-transform.

As shown in the following equation (11), a value obtained by multiplying the speed error eω by the proportional gain Kpw is used as a current command value amplitude proportional component I * p. Further, as shown in the following equation (12), a value obtained by multiplying and integrating the speed error eω by the integral gain Kiw is used as a current command value amplitude integral component I * i. Here, the following equation (1)
As shown in 3), this current command value amplitude integral component I * i is
Limit the range from I * min to I * max. And
As shown in the following equation (14), the current command value amplitude proportional component I *
The sum of p and the current command value amplitude integral component I * i is defined as the current command value amplitude I *. Here, the current command value amplitude I * is limited to a range from I * min to I * max as in the following equation (15). Note that the current command maximum value I * max is
It is created by a current command maximum value creating unit 75 described later. In addition, I * min is determined by the maximum regenerative capacity of the drive unit 23 and the like.

I * p = Kpw · eω (11) I * i = ΣKiw · eω (12) I * min ≦ I * i ≦ I * max (13) I * = I * p + I * i (14) I * min ≦ I * ≦ I * max (15) Then, as in the following equations (16) and (17),
A d-axis current command value id * and a q-axis current command value iq * are created. Here, β * is a current phase that achieves the maximum output torque or the maximum efficiency when the current command value amplitude I * is given.

Id * = − I * · sin (β *) (16) iq * = I * · cos (β *) (17) The ADC 41 has an analog u-phase current value i which is an analog value.
ua is converted to a digital value u-phase current value iu by analog /
Perform digital conversion. The ADC 42 also performs analog / digital conversion of the analog v-phase current value iva, which is an analog value, into a v-phase current value iv, which is a digital value.

The three-phase to two-phase conversion unit 43 includes a stator winding 11
u, 11v, and 11w are represented by angles θ
D-axis current values id and q on the dq-axis, which is a rotating coordinate system by
This is converted to the shaft current value iq. Further, a two-phase to three-phase converter 81 described later performs an inverse conversion of the conversion performed by the three-phase to two-phase converter 43 on the voltage applied to the stator windings 11u, 11v, and 11w. Specifically, the three-phase to two-phase conversion unit 43 creates a d-axis current value id and a q-axis current value iq as in the following equations (18) and (19).

Id = {(2)} · {iu · sin (θ + π / 3) + iv · sin (θ)} (18) iδ = {(2)} · ｛iu · cos (θ + π / 3) + Iv · cos (θ)｝ (19) The d-axis subtractor 52 calculates the difference between the d-axis current command value id * and the d-axis current value id as a d-axis current error e as in the following equation (20).
id.

Eid = id * −id (20) The d-axis voltage command value creation unit 53 uses the proportional-integral control to calculate d.
A shaft voltage command value vd * is created and further limited. FIG. 7 shows a d-axis voltage command value creation unit 53 according to the first embodiment.
It is a block diagram which shows operation | movement.

As shown in the following equation (21), the d-axis current error e
The value obtained by multiplying id by the proportional gain Kpd is used as a d-axis voltage command value proportional component vd * p. Further, as shown in the following equation (22), a value obtained by multiplying and integrating the d-axis current error eid by the integration gain Kid is set as a d-axis voltage command value integration component vd * i. Here, the d-axis voltage command value integral component vd * i is limited to the range from vd * min to vd * max as in the following equation (23).

Then, as in the following equation (24), the sum of the d-axis voltage command value proportional component vd * p and the d-axis voltage command value width integral component vd * i is used as the d-axis voltage command value vd *. Here, as in the following equation (25), the d-axis voltage command value vd
* Is restricted to the range from vd * min to vd * max. This range is set to about two to three times the voltage value that can be given by the power supply 23a. That is, vd * mi
n is -2 from power supply -3 · (√ (3/2)) · (E / 2)
・ (√ (3/2)) ・ (E / 2). Also, v
The power supply d * max is reduced from about 2 · (√ (3/2)) · (E / 2) to about 3 · (√ (3/2)) · (E / 2). Here, E is the voltage value of the power supply 23a.

Vd * p = Kpd · eid (21) vd * i = ΣKid · eid (22) vd * min ≦ vd * i ≦ vd * max (23) vd * = vd * p + vd * i (24) vd * min ≦ vd * ≦ vd * max (25) The q-axis voltage command value vq * is created in the same manner as the d-axis voltage command value vd *.

The q-axis subtractor 62 calculates the difference between the q-axis current command value iq * and the q-axis current value iq by the following equation (26).
A shaft current error eiq is set.

Eiq = iq * −iq (26) The q-axis voltage command value creation unit 63 uses the proportional-integral control to make q
A shaft voltage command value vq * is created and further limited. FIG. 8 shows a q-axis voltage command value creating unit 63 according to the first embodiment.
It is a block diagram which shows operation | movement.

As shown in the following equation (27), the q-axis current error e
The value obtained by multiplying iq by the proportional gain Kpq is set as a q-axis voltage command value proportional component vq * p. Further, as shown in the following equation (28), a value obtained by multiplying and integrating the q-axis current error eiq by the integration gain Kiq is set as a q-axis voltage command value integration component vq * i. Here, the q-axis voltage command value integral component vq * i is limited to the range from vq * min to vq * max as in the following equation (29). Then, as in the following equation (30), the sum of the q-axis voltage command value proportional component vq * p and the q-axis voltage command value width integral component vq * i is used as the q-axis voltage command value vq *. Here, as in the following equation (31), the q-axis voltage command value vq
* Is restricted to the range from vq * min to vq * max. This range is set to about two to three times the voltage value that can be given by the power supply 23a. That is, vq * mi
n is -2 from power supply -3 · (√ (3/2)) · (E / 2)
・ (√ (3/2)) ・ (E / 2). Also, v
q * max is changed from power supply 2 (√ (3/2)) · (E / 2) to about 3 · (√ (3/2)) · (E / 2). Here, E is the voltage value of the power supply 23a.

Vq * p = Kpq · eiq (27) vq * i = ΣKiq · eiq (28) vq * min ≦ vq * i ≦ vq * max (29) vq * = vq * p + vq * i (30) vq * min ≦ vq * ≦ vq * max (31) The d-axis LPF 71 applies a digital primary LPF to the d-axis voltage command value vd * as shown in the following equation (32) to perform LPF.
The rear d-axis voltage command value is set to vd * l. Here, Kld, which is a coefficient of the LPF, takes a value from 0 to 1, and the effect of the LPF increases as the value decreases. Note that vd * l
(I) is the LP when the d-axis LPF unit 71 operates this time.
The post-F d-axis voltage command value, and vd * l (i-1) is the post-LPF d-axis voltage command value when the d-axis LPF unit 71 was operated last time.

Vd * l (i) = Kld · vd * + (1−Kld) · vd * 1 (i−1) (32) The q-axis LPF unit 72 calculates q as shown in the following equation (33). A digital primary LPF is applied to the shaft voltage command value vq * to produce an LPF
The rear q-axis voltage command value is set to vq * l. Here, Klq, which is a coefficient of the LPF, takes a value from 0 to 1, and the effect of the LPF increases as the value decreases. Note that vq * l
(I) is the LP when the q-axis LPF unit 72 operates this time.
The post-F q-axis voltage command value, and vq * l (i-1) is the post-LPF q-axis voltage command value when the q-axis LPF unit 72 last operated.

Vq * l (i) = Klq · vq * + (1−Klq) · vq * 1 (i−1) (33) The saturation creation unit 73 calculates the LPF as shown in the following equation (34).
Rear d-axis voltage command value vd * l and post-LPF q-axis voltage command value v
The sum of squares with q * l is set to the saturation degree v2.

V2 = vd * l · vd * l + vq * l · vq * l (34) The reference value creation unit 74 creates the reference value v0. This v0
Is about 1.1 to 2 times the voltage value of the power supply 23a.

The current command maximum value creating unit 75 calculates the saturation v2
The current command maximum value I * max is created based on the reference value v0 and the reference value v0. FIG. 9 is a block diagram illustrating an operation of the current command maximum value creating unit according to the first embodiment.

First, as shown in the following equation (35), a result obtained by subtracting the saturation v2 from the square value v0 ＾ 2 of the reference value is set as a saturation error ev. Then, as shown in the following equation (36), a value obtained by multiplying the saturation error ev by the proportional gain Kpi is set as a current command maximum value proportional component I * maxp. Further, as shown in the following equation (37), the integration error Kii is added to the saturation error ev.
Is multiplied and integrated by the current command maximum value integration component I * ma
xi. Here, as shown in the following equation (38), the current command maximum value integral component I * maxi is limited to a range from 0 to I0. Then, as in the following equation (39), the sum of the current command maximum value proportional component I * maxp and the current command maximum value integral component I * maxi is set as the current command maximum value I * max. Here, as shown in the following equation (40), the current command maximum value I * max is limited to a range from 0 to I0. This I0 is the maximum output of the IPMSM 10 or the driving unit 2
3 depending on the IGBT withstand capability and the like.

Ev = v0 ＾ 2−v2 (35) I * maxp = Kpi · ev (36) I * maxi = ΣKii · ev (37) 0 ≦ I * maxi ≦ I0 (38) I * max = I * maxp + I * maxi (39) 0 ≦ I * max ≦ I0 (40) The three-phase / two-phase conversion unit 81 outputs a d-axis voltage command value on the dq axis which is a rotating coordinate system based on the estimated angle θ. vd * and the q-axis voltage command value vq * are converted into a stationary coordinate system, and the stator winding 11u,
A u-phase voltage command value vu *, a v-phase voltage command value vv *, and a w-phase voltage command value vw * to be applied to 11v and 11w are created. Specifically, the following equations (41), (42), and (43)
Like

Vu * = {(2/3)} · {vd * · cos θ−vq * · sin θ} (41) vv * = {(2/3)} · ｛vd * · cos (θ −2 · π / 3) −vq * · sin (θ−2 · π / 3)｝... (42) vw * = {(2/3)} · ｛vd * · cos (θ + 2 · π / 3) −vq * · sin (θ + 2 · π / 3)｝ (43) The PWM controller 82 determines the pulse width of the u-phase voltage command value vu *, the v-phase voltage command value vv *, and the w-phase voltage command value vw *. Modulation (PWM: Pulse Width Modulation) is performed. Specifically, a triangular wave having a set period and an amplitude of E / 2 is generated. Here, E is the voltage value of the power supply 23a.
Then, the triangular wave is compared with the u-phase voltage command value vu *, and when the u-phase voltage command value vu * is larger, the switching signal guh is set to H and the gul is set to L. On the other hand, when the u-phase voltage command value vu * is smaller, the switching signal g
uh is set to L and gu is set to H.

When the states of the switching signals guh and gu transition, the switching signals guh and gul change.
Are both set to L (this short time is called dead time). Similarly, v-phase voltage command values vv * and w-phase
The switching signal gvh based on the phase voltage command value vw *,
Create gvl and gwh, gwl.

[Effects of Motor Control Device of First Embodiment]
By operating as described above, the motor control device of the first embodiment has the following effects. As the speed ω increases, the required voltage amplitude increases. However, since the voltage of the power supply 23a has a limit, a sufficient voltage cannot be applied in a high-speed and high-load region, and the voltage is saturated. In such a region where the voltage is saturated, responsiveness is deteriorated due to the operation amount saturation, and noise and vibration are generated due to current waveform distortion.

Therefore, the motor control device of the first embodiment sets the reference value v0, and when the saturation v2 is larger than the square of the reference value v0, the saturation v2 converges to the square of the reference value v0. The current command value is limited to a small value by reducing the current command maximum value I * max using proportional integral control. As described above, by limiting the current command value based on the saturation degree v2 and the reference value v0, a motor control device that suppresses deterioration of responsiveness due to operation amount saturation and noise / vibration due to current waveform distortion is realized. .

Further, in the IPMSM in which the induced voltage is distorted, even if the d-axis current command value id * and the q-axis current command value iq * are constant, as shown in FIGS. 5 (a) and 5 (b).
D-axis voltage command values vd * and q created by current control
The shaft voltage command value vq * vibrates. Therefore, these d
When anti-wind-up was performed using the shaft voltage command value vd * and the q-axis voltage command value vq *, the operation became unstable.

The motor control device according to the first embodiment includes a post-LPF d-axis voltage command value vd * l and a q-axis voltage command value obtained by applying an LPF to the d-axis voltage command value vd * and the q-axis voltage command value vq *, respectively. Use the value vq * l. As shown in FIG. 5D and FIG. 5E, the vibration of the post-LPF d-axis voltage command value vd * l and the q-axis voltage command value vq * l is suppressed by the action of the LPF. Therefore, as shown in FIG. 5F, the oscillation of the saturation degree v2, which is the sum of the squares, is also suppressed. Then, anti-wind-up is performed using the saturation v2.

As described above, by using the saturation degree v2 created from the voltage command value applied with the LPF, anti-wind-up can be stably performed even in a motor having a large induced voltage distortion, and the operation amount is saturated. A motor control device that suppresses noise and vibration due to deterioration of responsiveness and current waveform distortion is realized.

Further, the motor control device according to the first embodiment
As described above, d after the LPF in FIGS. 5D and 5E
Shaft voltage command value vd * l and q-axis voltage command value vq after LPF
* 1 is used to create a saturation v2 (FIG. 5 (f)), and an anti-winding factor is calculated based on the saturation v2 and the reference value v0.
Do the up. That is, saturation is determined from the low frequency component of the voltage command, and anti-windup is performed. On the other hand, in order to cope with distortion of the induced voltage, it is necessary to apply the high frequency component of the voltage command as it is. Therefore, it is necessary to independently set the voltage command value limit and the anti-wind-up saturation judgment.

As described above, by independently setting the voltage command value limit and the anti-wind-up saturation judgment, even when a low-pass filter is applied to the voltage command value, the output can be reduced. A motor control device that performs anti-wind-up to suppress responsiveness deterioration due to operation amount saturation and noise / vibration due to current waveform distortion.

Further, the motor control device according to the first embodiment
The reference value v0 is set to be larger than the maximum voltage value that can be applied by the drive unit 23. Here, a sinusoidal current can continue to flow even if the voltage is somewhat saturated. The output of the motor can be increased as the effective value of the applied voltage increases. As described above, by making the reference value v0 larger than the maximum voltage value that can be applied by the drive unit 23, a motor control device that increases the output of the motor is realized.

(Second Embodiment) Next, a motor control device according to a second embodiment will be described. The motor control device according to the first embodiment is configured such that the voltage command value applied with the LPF (d-axis voltage command value after LPF vd * l and q after LPF q
The saturation v2 was created based on the shaft voltage command value vq * l). The motor control apparatus of the second embodiment, to create a saturation v2 based on the integral component of the voltage command (d-axis voltage command value integral component vd * i and q-axis voltage command value integral component vq * i). By doing so, even in a motor having a large induced voltage distortion, the oscillation of the saturation degree v2 is suppressed, anti-wind-up is performed stably, the response is deteriorated due to the saturation of the operation amount, and the noise and vibration due to the current waveform distortion. The occurrence of is suppressed. Further, by omitting the operation of the LPF, the operation time is reduced.

[Configuration and Operation of Motor Control Device of Second Embodiment] First, the configuration and operation of the motor control device of the second embodiment will be described. FIG. 10 is a block diagram illustrating a configuration of a motor control device according to the second embodiment. In the configuration included in the motor control device 220, the microcomputer 222 is different from the first embodiment. Further, among the configurations included in the microcomputer 222, the d-axis voltage command value creation unit 253, the q-axis voltage command value creation unit 263, and the saturation creation unit 273 are different from the first embodiment.
And different. The d-axis LPF 71 and the q-axis LPF 72 included in the first embodiment are not included in the second embodiment. Other configurations are the same as those in the first embodiment, and are denoted by the same reference numerals as those in the first embodiment, and description thereof will be omitted.

The d-axis voltage command value creation section 253 receives the d-axis current error eid and outputs a d-axis voltage command value vd * and a d-axis voltage command value integral component vd * i. The operations other than the input / output of the d-axis voltage command value creation unit 253 are the same as those of the first embodiment, and the description is omitted.

The q-axis voltage command value creation unit 263 receives the q-axis current error eiq, and outputs a q-axis voltage command value vq * and a q-axis voltage command value integral component vq * i. The operation other than the input / output of the q-axis voltage command value creation unit 263 is the same as that of the first embodiment, and the description is omitted.

The saturation creation section 273 receives the d-axis voltage command value integral component vd * i and the q-axis voltage command value integral component vq * i, and outputs a saturation v2. As in the following equation (44), the saturation sum v2 of the d-axis voltage command value integral component vd * i and the q-axis voltage command value integral component vq * i is set to the saturation degree v2.

V2 = vd * i * vd * i + vq * i * vq * i (44) [Effect of Motor Control Device of Second Embodiment] Next, the effect of the motor control device of the second embodiment will be described. explain. FIG.
The d-axis voltage command value vd *, the q-axis voltage command value vq *, and the sum of squares of the voltage command value (vd * ＾ 2 + vq) in the second embodiment.
* ＾ 2) is a waveform diagram of a d-axis voltage command value integral component vd * i, a q-axis voltage command value integral component vq * i, and a saturation degree v2. In the IPMSM in which the induced voltage is distorted, even if the d-axis current command value id * and the q-axis current command value iq * are constant, they are created by current control as shown in FIGS. 11 (a) and 11 (b). The d-axis voltage command value vd * and the q-axis voltage command value vq * vibrate. With this vibration, FIG.
The sum of these squares (vd * ＾ 2 + vq * ＾ 2)
Also vibrate. Therefore, the sum of squares (vd * ＾ 2 + vq
* When anti-wind-up is performed based on 2),
Control is not stable.

On the other hand, as for the d-axis voltage command value integral component vd * i and the q-axis voltage command value integral component vq * i, vibration is suppressed as shown in FIGS. Therefore,
As shown in FIG. 11 (f), the saturation sum v
2 (Equation (44)) is also suppressed. Therefore, when anti-wind-up is performed using this degree of saturation v2, control is stabilized.

As described above, the induced voltage is obtained by using the saturation degree v2 created from the integral components of the voltage command value (d-axis voltage command value integral component vd * i and q-axis voltage command value integral component vq * i). also in motor distortion is large, stable perform anti-wind-up, the deterioration of responsiveness by the operation amount impregnating realizes suppressing motor controller generation of noise and vibration due to current waveform distortion.

Further, the motor control device according to the second embodiment
Since it has the same operation as the first embodiment, the first embodiment
Has the same effect as.

Further, the motor control device according to the first embodiment
Since the saturation degree v2 is created based on the voltage command values (the d-axis voltage command value after LPF vd * l and the q-axis voltage command value after LPF vq * l) applied with the LPF, the calculation time required for the LPF is required. Was. On the other hand, in the second embodiment, L
Without using the PF, the integral components of the voltage command value (d-axis voltage command value integral component vd * i and q-axis voltage command value integral component vq *
By creating the saturation degree v2 based on i), a motor control device with reduced calculation time is realized.

(Embodiment 3) Next, a motor control device according to Embodiment 3 will be described. When detecting the voltage saturation, the motor control device according to the first embodiment detects the current command maximum value I * ma
By changing x, anti-wind-up was performed to reduce the current command value. Embodiment 3
When the motor controller detects voltage saturation, it performs anti-wind-up to reduce the speed command value ω *, thereby suppressing responsiveness deterioration due to operation amount saturation and suppressing noise and vibration caused by current waveform distortion. I do.

[Configuration and Operation of Motor Control Device of Third Embodiment] First, the configuration and operation of the motor control device of the third embodiment will be described. FIG. 12 is a block diagram illustrating a configuration of a motor control device according to the third embodiment. The microcomputer 322 of the configuration included in the motor control device 320 is different from that of the first embodiment. Further, among the configurations included in the microcomputer 322, a speed subtraction unit 334, a current command value creation unit 335, and a speed command value subtraction amount creation unit 376 are different from the first embodiment. Further, the current command maximum value creation unit 75 included in the first embodiment is not included in the third embodiment. Other configurations are the same as those in the first embodiment, and are denoted by the same reference numerals as those in the first embodiment, and description thereof will be omitted.

The speed subtraction section 334 receives the speed command value ω *, the speed command value subtraction amount Δω, and the speed ω, and generates a speed error eω.
Is output. As shown in the following equation (45), the speed command value ω *
, The speed command value subtraction amount Δω is reduced, and the result obtained by further reducing the speed ω is set as a speed error eω.

Eω = ω * −Δω−ω (45) The current command value creation unit 375 calculates the current command maximum value I * max.
Use a certain set value. Other operations are the same as those in the first embodiment, and a description thereof will not be repeated.

The speed subtraction amount creating section 376 receives the saturation v2 and the reference value v0, and outputs a speed command value subtraction amount Δω. FIG. 13 is a block diagram illustrating the operation of the speed command value subtraction amount creating unit according to the third embodiment.

First, as shown in the following equation (46), a result obtained by subtracting the saturation v2 from the square value v0 ＾ 2 of the reference value is set as a saturation error ev. Then, as shown in the following equation (47), a value obtained by multiplying the saturation error ev by the proportional gain Kpdw is used as a speed command value subtraction amount proportional component Δωp. In addition, the following equation (4)
As shown in 8), the value obtained by multiplying and integrating the saturation error ev by the integration gain Kidw is used as a speed command value subtraction amount integration component Δωi. Here, as shown in the following equation (49), this speed command value subtraction amount integral component Δωi is limited to a range from 0 to ω0. Then, as in the following equation (50), the sum of the speed command value subtraction amount proportional component Δωp and the speed command value subtraction amount integral component Δωi is used as the speed command value subtraction amount Δω. Here, as shown in the following equation (51), the speed command value subtraction amount Δω is limited to a range from 0 to ω0. Note that this ω0 is the IPMS
It is determined from the maximum speed of M10 and the like.

Ev = v0 ＾ 2-v2 (46) Δωp = Kpdw · ev (47) Δωi = ΣKidw · ev (48) 0 ≦ Δωi ≦ ω0 (49) Δω = Δωp + Δωi (50) 0 ≦ Δω ≦ ω0 (51) [Effect of Motor Control Device of Third Embodiment] As described above with reference to FIG. 4, when controlling a fan or the like, a required voltage amplitude V increases with an increase in speed ω. Becomes larger. However, since the voltage of the power supply 23a of the driving unit 23 has a limit, a sufficient voltage cannot be applied in a high-speed and high-load region, and the voltage is saturated. In such a region where the voltage is saturated, responsiveness is deteriorated due to operation amount saturation, and noise and vibration are generated due to current waveform distortion.

Therefore, when the voltage amplitude V increases, the speed ω is reduced to prevent voltage saturation. That is, a reference value v0 slightly larger than the power supply voltage / 2 is set, and when the voltage amplitude V is larger than the reference value v0 (in FIG. 4, when the speed ω and the voltage amplitude V are in the state indicated by ●), V so that V converges to the reference value v0
(In FIG. 4, the speed ω and the voltage amplitude V
Is changed from the state indicated by ● to the state indicated by ○). In this way, anti-wind up,
Deterioration of responsiveness due to operation amount saturation and generation of noise and vibration due to current waveform distortion are suppressed. In the actual operation, the saturation v2 is used instead of the voltage amplitude V, and the speed command value ω * is reduced instead of the speed ω.

As described above, the motor control device according to the third embodiment detects voltage saturation based on the saturation degree v2 and the reference value v0, and reduces the speed command value ω * when the voltage is saturated. This has the same effect as the first embodiment. In the third embodiment, the saturation v2 used in the second embodiment may be used.

Although the first to third embodiments have been described in detail, in the first and third embodiments, the digital primary LPF is added to the d-axis voltage command value vd * and the q-axis voltage command value vq *. However, the present invention is not limited to this mode. For example, the d-axis voltage command values vd * and q
Sum of squares with shaft voltage command value vq * (vd * {2 + vq *}
LPF may act on 2). Further, the present invention includes the case where the LPF is applied to the value before the limit instead of the voltage command value after the limit (the d-axis voltage command value vd * and the q-axis voltage command value vq *) and used. The essential point of the present invention is to determine the voltage saturation by applying an LPF to the voltage command value.
To stabilize anti-wind-up,
It is not limited by the place where the LPF acts.

In the first and third embodiments, the digital primary LPF is operated.
Other types of LPF such as PF may be used. Further, a high-frequency component may be removed by using a band-pass filter. Further, the present invention includes a case where only a vibrating component is removed by using a notch filter. For example, in the case of a motor having 4 poles and 6 slots (the number of magnetic poles is 4 and the number of teeth is 6), the d-axis voltage command values vd * and q
Since the shaft voltage command value vq * oscillates at a cycle of an electrical angle of 60 °, using a digital notch filter that removes this component has a great effect.

Further, in the first and second embodiments, the current is reduced by reducing the current command maximum value I * max, but the current command value may be changed. Further, in the third embodiment, the speed command value ω * is reduced by the speed command value subtraction amount Δω, but the speed command value ω *
The speed may be reduced by setting a limit on * and changing the range of the limit. The essence of the present invention is
This is to detect the saturation of the voltage from the degree of saturation and the reference value to reduce the current or the speed, and the method is not limited to the method described in the embodiment.

Further, in the first and second embodiments, the current command maximum value I * max was determined by proportional integral control. On the other hand, in the third embodiment, the speed command value subtraction amount Δω was obtained by proportional integral control. The use of these alternative forms is also included in the present invention. For example, feed forward control or differential control may be added.

In the first to third embodiments, the speed control uses the proportional-integral control. However, the speed control may be performed in other forms. For example, feed forward control or differential control may be added.

In the first to third embodiments, the current control uses the proportional-integral control. However, the current control may be performed in other forms. For example, non-interference control is added, and the d-axis voltage command value vd * is calculated as in the following equations (52) and (53).
And the q-axis voltage command value vq * may be created. At this time, in the second embodiment, as in the following equation (54),
The saturation v2 is created in consideration of the term added in the non-interference control (non-interference term). Although the current command is used for the non-interference term, an actual current may be used. Also, the LPF may act only on the non-interference terms.

Vd * = vd * p + vd * i + R · id * + ω · Lq · iq * (52) vq * = vq * p + vq * i + R · iq * + ω · Ld · id * + ω · ψ (53) v2 = (vd * i + R · id * + ω · Lq · iq *) ＾ 2+ (vq * i + R · iq * + ω · Ld · id * + ω · ψ) ＾ 2 (54) In the third embodiment, the operation related to the speed control and the operation related to the current control are synchronized in the same cycle, but may be separated. In particular, the overall operation time can be reduced by increasing the period of the operation related to the speed control.

Further, in Embodiments 1 to 3, the angle is detected by using the optical encoder 13, but another position sensor may be used. For example, a magnetic encoder or a resolver may be used. Alternatively, a CS signal (commutation signal) may be created using a Hall IC or the like, and the angle may be extrapolated from the CS signal. Further, the angle may be estimated without using the position sensor.

The present invention is not limited to the configurations of the first to third embodiments. For example, the microcomputer 22, 2
22 and 322 may be composed of a plurality of microcomputers. Further, an IC including the driving unit 23 may be used.

Further, in the first to third embodiments, the case of controlling the fan has been described, but the present invention is not limited to this. Even if the output torque required by the rotation speed does not change, the voltage increases with the rotation speed.
Embodiment 3 can be applied.

Further, Embodiments 1 to 3 are not limited to IPMSM. For example, SPMS
M, IM, and SynRM may be controlled. These motors generally have a higher induced voltage (including the induced voltage by the magnet and the armature reaction) than the IPMSM, but there are some motors in which the induced voltage is distorted, and the above effects can be obtained by using the present invention. Can be

[0114]

[0115]

As described above, according to the present invention, the saturation
Becomes larger than the reference value, decrease the speed command value before change.
By making such a change, it is possible to realize a motor control device that suppresses deterioration of responsiveness due to operation amount saturation and noise / vibration due to current waveform distortion.

Further, according to the present invention, the degree of saturation is created from the voltage command value applied with the LPF, so that anti-wind-up is performed stably even in a motor having a large induced voltage distortion, and the operation amount is saturated. Worse responsiveness due to
The generation of noise and vibration due to current waveform distortion can be realized to suppress the motor controller.

Further, according to the present invention, the degree of saturation is created from the integral component of the voltage command value, so that even in a motor having a large induced voltage distortion, the anti-winding is stably performed.
An up, deterioration of responsiveness by the operation amount impregnating, suppresses motor controller generation of noise and vibration due to current waveform distortion can be realized.

Further, according to the present invention, it is possible to realize a motor control device that increases the output of the motor by setting the reference value to be larger than the maximum voltage value that can be applied by the drive unit.

[Brief description of the drawings]

FIG. 1 is a block diagram illustrating a configuration of a motor control device according to a first embodiment.

FIG. 2 is a circuit diagram illustrating a configuration of a driving unit in Embodiment 1.

FIG. 3 is an explanatory diagram of a coordinate system according to the first embodiment.

FIG. 4 shows output torque, d-axis current value, q-axis current value, current amplitude, d-axis voltage value,
Relationship diagram showing changes in q-axis voltage value and voltage amplitude

FIG. 5 is a waveform diagram of a d-axis voltage command value, a q-axis voltage command value, a sum of squares of the voltage command value, a d-axis voltage command value after LPF, a q-axis voltage command value after LPF, and a saturation degree in the first embodiment.

FIG. 6 is a block diagram illustrating an operation of a current command value creation unit according to the first embodiment.

FIG. 7 is a block diagram illustrating an operation of a d-axis voltage command value creating unit according to the first embodiment.

FIG. 8 is a block diagram illustrating an operation of a q-axis voltage command value creating unit according to the first embodiment.

FIG. 9 is a block diagram showing an operation of a current command maximum value creating unit according to the first embodiment.

FIG. 10 is a block diagram showing a configuration of a motor control device according to a second embodiment.

FIG. 11 is a waveform diagram of a d-axis voltage command value, a q-axis voltage command value, a sum of squares of the voltage command value, a d-axis voltage command value integral component, a q-axis voltage command value integral component, and a degree of saturation in the second embodiment.

FIG. 12 is a block diagram showing a configuration of a motor control device according to a third embodiment.

FIG. 13 is a block diagram showing an operation of a speed command value subtraction amount creating unit according to the third embodiment;

FIG. 14 is a block diagram showing a configuration of a motor control device in a conventional example.

[Explanation of symbols]

 Reference Signs List 10 IPMSM 11 Stator winding 20, 220, 320 Motor control device 21 u, 21 v Current sensor 22, 222, 322 Microcomputer 23 Drive unit 34, 334 Speed subtraction unit 35, 335 Current command value creation unit 52 d-axis subtraction unit 62 q-axis Subtraction unit 53,253 d-axis voltage command value creation unit 63,263 q-axis voltage command value creation unit 71,273 d-axis LPF unit 72 q-axis LPF unit 73 saturation degree creation unit 74 reference value creation unit 75 current command maximum value creation Unit 376 Speed subtraction amount creation unit

──────────────────────────────────────────────────の Continuing on the front page (72) Inventor Ichiro Oyama 1006 Kazuma Kadoma, Osaka Prefecture Inside Matsushita Electric Industrial Co., Ltd. (56) References JP-A-2000-245199 (JP, A) JP-A-8-9698 (JP, A) (58) Fields investigated (Int. Cl. ⁷ , DB name) H02P 5/408-5 / 412 H02P 7/628-7/632 H02P 21/00 H02P 6/00-6/24

Claims (5)

(57) [Claims] 1. A speed command value creating means for creating a speed command value before change indicating a command value of a speed of a rotor; a speed command value changing means for changing the speed command value before change to create a speed command value; A speed detecting means for detecting a speed of the rotor; and a current flowing through a stator winding based on the speed command value and the speed.
Current for creating the pre-limit current command value indicating the current command value
Command value creating means for limiting the current command value before limitation to create a current command value;
Flow command value limiting means, current detection means for detecting a phase current value indicating a current flowing through the stator winding, a command value of a voltage applied to the stator winding based on the current command value and the phase current value Voltage command value creating means for creating a pre-limit voltage command value, and voltage command value limiting means for creating the voltage command value by limiting the pre-limit voltage command value; and for the stator winding based on the voltage command value. A driving means for applying electric power, wherein a saturation level creation section for creating a saturation level indicating how much the voltage command value is saturated based on the pre-limit voltage command value or the voltage command value. Means, and a reference value creating means for creating a reference value indicating the reference of the saturation, wherein the speed command value changing means is configured such that the saturation is less than the reference value.
The speed command value before change is reduced when
Wherein the speed command value is changed to the speed command value . 2. The method according to claim 1, wherein the speed command value changing unit is configured to determine the saturation level.
There motor controller according to claim 1, characterized in that a change to the speed command value so as to reduce the pre-change speed command value by using proportional-plus-integral control to be larger than the reference value. 3. The apparatus according to claim 1, wherein the saturation creating unit creates the saturation based on the pre-limit voltage command value or a value obtained by applying a filter to the voltage command value. Motor control device. 4. The voltage command value creating means creates a voltage command value proportional component by a proportional operation, creates a voltage command value integral component by an integration operation, and outputs the voltage command value proportional component and the voltage command value integral component. 2. The motor control device according to claim 1, wherein the pre-limit voltage command value is created based on the following, and the saturation creation unit creates the saturation based on the voltage command value integral component. 3. 5. The motor control device according to claim 1, wherein the reference value creating unit sets the reference value to be larger than a maximum voltage value that can be applied by the driving unit.