JP2583457B2 - Switching power supply - Google Patents

Description

Description: BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a resonance type switching power supply device configured to reduce the loss of a switching element using resonance.

2. Description of the Related Art A resonance type switching regulator configured to reduce a switching loss of a switching element by utilizing a resonance phenomenon between an inductance of a transformer winding and a resonance capacitor is known.

[Problems to be Solved by the Invention] By the way, the off period of the switching element in the resonance type switching regulator is determined by the circuit constant of the LC resonance circuit, and is constant. Therefore, the voltage adjustment is performed by adjusting the ON time width. Therefore, the degree of freedom in circuit design is low.

Then, an object of the present invention is to provide a resonance type switching power supply device whose off period can be changed.

Means for Solving the Problems To achieve the above object, the present invention provides a DC power supply, a transformer having a winding connected between one end and the other end of the DC power supply, 2 main terminals and a control terminal, wherein the first main terminal is connected to one end of the DC power supply via the winding, and the second main terminal is connected to the other end of the DC power supply. A first switch, a capacitor comprising an independent capacitor or stray capacitance associated with the winding so as to resonate with the inductance of the winding, an output circuit coupled to the transformer, and a control voltage source. A second switch for directly or equivalently connecting the control voltage source to the winding in parallel, and an on / off control of the first switch, and an off period of the first switch Of the winding and the capacitor at Turning on the second switch within a range after a maximum amplitude value of the voltage of the capacitor caused by resonance and before the voltage becomes zero, and an ON time width of the second switch; And a control circuit formed so as to be able to change the switching power supply.

As described in claim 2, the ON period of the second switch can be set before the maximum amplitude value of the voltage of the capacitor.

Further, an additional capacitor and a third switch can be added.

[Operation] The voltage source and the second switch according to the first, second and third aspects of the invention are for interrupting resonance. When the voltage source is connected via the second switch, the resonance is interrupted, the voltage on the capacitor is kept substantially constant, and when the second switch is turned off, the resonance that continued before the interruption occurs.

The additional capacitor or voltage source according to claim 3 and the third switch serve to suppress the maximum amplitude value of the resonance waveform. When the on time of the third switch is changed, the off time width of the first switch changes. Therefore, it is also possible to change the off-time width of the first switch in both the second and third switches.

First Embodiment Next, a voltage resonance type switching regulator according to a first embodiment of the present invention will be described with reference to FIGS.

The DC power supply 1 of this switching regulator includes a full-wave rectifier 3 connected to an AC power supply terminal 2 and a smoothing capacitor 4. A series circuit of a primary winding (main winding) 6 of a transformer 5 and a first switching element 7 composed of a field effect transistor is connected between one end and the other end (ground terminal) of the DC power supply 1. ing. A control terminal (gate electrode) of the first switching element 7 is connected to a control circuit 8 for turning on / off the first switching element 7. Transformer 5-2
The secondary winding 9 is connected to DC output terminals 15 and 16 via an output rectifying / smoothing circuit 14 including diodes 10 and 11, a reactor 12 and a capacitor 13. A diode 17 is connected in anti-parallel with the first switching element 7 in order to flow a current in the opposite direction. In this example, the first switching element 7 is an N
Since it is a channel insulated gate field effect transistor, a diode is built in. Therefore, diode 17
It is not necessary to externally connect, but it is shown separately for ease of understanding.

The capacitor 18 connected in parallel with the first switching element 7 is for forming a resonance circuit with the inductance L of the transformer 5. In this example, since the resonance capacitor 18 is connected in parallel to the first switching element 7, the voltage between both terminals of the first switching element 7 and the voltage of the capacitor 18 become the same.

In order to control the off-time width of the first switching element 7 according to the present invention, a second switching element 19 comprising an N-channel insulated gate field effect transistor having a structure in which the substrate is connected to the source is provided. The second switching element 19 is connected in parallel to a tertiary winding 20 of the transformer 5 via a diode 21 and a control voltage source 22 to interrupt the LC resonance operation.
The polarity of the tertiary winding 20 is determined such that a downward voltage, that is, a voltage in the same direction as the control voltage source 22 is induced when the first switching element 7 is turned on. The direction of the diode 21 and the second switching element 19 is the voltage of the control voltage source 22.
It is determined to be forward biased at E2. Tertiary winding
Turn ratio between 20 and primary winding 6 and voltage E2 of control voltage source 24
Is the tertiary winding 20 during the off period of the first switching element 7.
The diode 21 is set to be forward-biased first, then reverse-biased, and then forward-biased again due to the relationship between the voltage V3 induced in the reverse direction and the control voltage E2.

When the second switching element 19 and the diode 21 are turned on and the voltage source 22 is connected to the tertiary winding 20 during the off period of the first switching element 7, the resonance operation is interrupted. When the width of the interruption time of the resonance operation is changed, the off period of the first switching element 7 changes, and as a result, the output voltage changes.

A voltage detection circuit 23 is connected to the output terminal 15 to control the ON time width of the second switching element 19 according to the output voltage. The voltage detection circuit 23 supplies a detection signal corresponding to the output voltage to the control circuit 8.

As shown in FIG. 2, the control circuit 8 in FIG. 1 includes a first control signal forming circuit 31 for forming a control signal for the second switching element 19 based on the detection of the voltage of the first switching element 7, The first control signal forming circuit 31
And a second control signal forming circuit 32 that forms a second control signal having a fixed time relationship with the first control element and supplies the second control signal to the first switching element 7. First control signal forming circuit 31
Is a voltage detection circuit 33 connected in parallel to the first switching element 7, a reference voltage source 34 for providing a reference voltage Vr lower than the maximum amplitude of the voltage applied to the first switching element 7 by LC resonance, and Is compared with a voltage V1 (or a divided output) of the first switching element 7 obtained from the voltage detection circuit 33, and a high-level output is output during a period when the voltage V1 of the first switching element 7 is higher than the reference voltage Vr. Generating a comparator
35, a trigger circuit 36 for generating a trigger pulse in response to the falling edge of the output pulse of the comparator 35, and a pulse generated in synchronization with the trigger pulse of the trigger circuit 36. A variable pulse generation circuit (variable timer circuit) 37 formed so as to change according to the detection voltage in the voltage detection circuit 22 of the first embodiment, and a first control signal corresponding to a pulse obtained from the variable pulse generation circuit 37 2 switching elements 19
And a drive circuit 38 for supplying the

The second control signal forming circuit 32 includes a variable pulse generating circuit 37.
, A delay circuit 41 connected to the trailing edge detection circuit 40, and a clock pulse generator.
42, a flip-flop 43 set by the output of the delay circuit 41 and reset by the clock pulse, and a drive circuit 44 connected between the output terminal of the flip-flop 43 and the first switching element 7.

[Operation] Before the time point t1 in FIG. 3, the first switching element 7 is turned on to supply power to the load. 3
As shown in FIG. 3D, the power supply voltage E1 is extremely low and is applied to the primary winding 6. When the clock pulse shown in FIG. 3A is generated from the clock pulse generator 42 shown in FIG. 2 at time t1, the flip-flop 43 is reset and the first switching element 7 is turned off. When the first switching element 7 is turned off, a flyback voltage is generated in the primary winding 6 of the transformer 5, and the flyback voltage follows the resonance frequency of the LC resonance circuit including the inductance of the primary winding 6 and the capacitor 18. The energy of the primary winding 6 is transferred to the capacitor 18 by the resonance operation in which the current Ic shown in FIG. 7F flows through the closed circuit including the primary winding 6, the capacitor 18 and the DC power supply 1, so that the capacitor 18 is charged. , The charging voltage gradually increases. Since the voltage of the capacitor 18 is the same as the voltage between both terminals of the first switching element 7, the voltage of the first switching element 7 also gradually increases. Thereby, turn-off of the first switching element 7 at substantially zero volt can be realized.

In the ON period of the first switching element 7, 3
A downward voltage is generated in the next winding 20 and forward biases the second switching element 19 and the diode 21 together with the control voltage E2. However, since the second switching element 19 is off, the voltage source 22 It has nothing to do with the next winding 6. Capacitor
The voltage of 18 and the voltage V1 of the first switching element 7 fall after reaching a maximum at t3. The comparator 35 in FIG. 2 compares the voltage V1 of the first switching element 7 with the reference voltage Vr,
A high-level output is generated during a period from t2 to t4 when the detection voltage V1 is higher than the reference voltage Vr. The time when the trailing edge (falling edge) of the output pulse of the comparator 35 is the time when the reverse bias of the diode 21 is released
The value of the reference voltage Vr is set to substantially match t4. The value of the reference voltage Vr is not limited to a value that satisfies the above condition, and the voltage of the first switching element 7
It can be set between the maximum amplitude value at time t3 of V1 and a value satisfying the above condition.

The trigger circuit 36 generates a trigger pulse at time t4 in FIG. 3 in response to the trailing edge (falling edge) of the output pulse of the comparator 35, and supplies this to the variable pulse generating circuit 37. The variable pulse generating circuit 37 is composed of, for example, a variable monomultivibrator, generates a control pulse (control signal) shown in FIG. 3B in response to a trigger pulse, and generates a voltage detecting circuit shown in FIG.
Control pulses of various pulse widths are generated in response to the 22 output. This control pulse is applied to the gate (control terminal) of the second switching element 19 via the drive circuit 38.

Since the tertiary winding 20 is electromagnetically coupled to the primary winding 6,
Both voltages change in relation to each other. The voltage V3 of the tertiary winding 20 shown in FIG. 3 (G) is equal to or less than the time t1 when the first switching element 7 is turned on and the power supply voltage E1 is applied to the primary winding 6. Although it is a constant voltage corresponding to the voltage E1, it becomes a value corresponding to the voltage of the primary winding 6 based on the LC resonance operation when the first switching element 7 is turned off. Thereafter, when the second switching element 19 and the diode 21 are turned on between t4 and t5, the voltage V3 of the tertiary winding 20 and the primary winding are connected because the control voltage source 22 is connected to the tertiary winding 20. 6 is fixed. During this fixed period, there is no change in the magnetic flux in the magnetic core of the transformer 5,
The LC resonance operation is interrupted. As a result, during the period from t4 to t5, the voltage V1 of the capacitor 18 and the first switching element 7 is kept almost constant.

FIG. 3G shows a change in the voltage V2 of the second switching element 19 and a change in the voltage Vd of the diode 21 corresponding to a change in the voltage V3 of the tertiary winding 20. When the sinusoidal voltage shown in FIG. 3D is generated by resonance in the primary winding 6 during the period from t1 to t4, an upward voltage having a waveform corresponding thereto is generated in the tertiary winding 20. The difference between the control voltage E2 and the upward voltage of the tertiary winding 20 is applied to the diode 21. The diode 21 is forward biased during the period from t1 to t2 when the upward voltage of the tertiary winding 20 is lower than the control voltage E2, but is during the period from t2 to t4 when the upward voltage of the tertiary winding 20 is higher than the control voltage E2. Then, the diode 21 enters a reverse bias state. Note that the second switching element 19 is a field-effect transistor and has a built-in diode equivalent to being connected in antiparallel.
In the period t4, the voltage V2 between both ends is substantially zero volt. In this embodiment, as shown in FIG. 3 (G), the comparator 35 almost coincides with the time point t4 when the reverse bias of the diode 21 is released.
3 falls, and in synchronization with this, the pulse shown in FIG. At time t4, the second switching element 19
And the reverse bias of the diode 21 is released.
When the ON control pulse is given to the switching element 19, the control voltage source 22 is connected in parallel to the tertiary winding 20, and the LC resonance is interrupted as described above. During the period when the control voltage source 22 is connected to the tertiary winding 20, the control voltage source 22
Supplies energy to the transformer 5. After a while
When the second switching element 19 is turned off at time t5, the control voltage source 22 becomes independent of the primary winding 6, and the resonance operation by the primary winding 6 and the capacitor 18 continues at t4, and the capacitor 18 and the voltage of the first switching element 7
V1 gradually decreases to zero volts at t6.

The trailing edge detection circuit 40 in FIG. 2 detects the trailing edge time t5 of the control pulse shown in FIG. 3B and delays this by the time Td by the delay circuit 41 to obtain a set signal of the flip-flop 43. At time t6 when the first switching element 7 is turned on, the voltage V1 across the first switching element 7 is zero, so that the switching loss at the time of turn-on is reduced.
Further, the energy loss of the capacitor 18 is reduced. t6
In the ON period, the stored energy in the primary winding 6 is released through the diode 17, so that a reverse current flows as shown in FIG. 3 (E), and thereafter, the first switching element 7, a forward current flows, and power is supplied to the load. The voltage V1 across the ON period t6 to t7 of the diode 17 and the first switching element 7 is substantially zero as shown in FIG. When the clock pulse shown in FIG. 3A is generated again at time t7, the same operation as in the period from t1 to t7 is repeated.

When the output voltage becomes lower than a predetermined value, for example, the trailing edge time of the pulse shown in FIG. 3B changes, and the width of this pulse becomes narrower, while the width of the pulse shown in FIG. 3C becomes wider. ,
The output voltage returns.

Since the clock pulse shown in FIG. 3A is generated at a constant cycle, if the OFF period of the first switching element 7 changes, the ON period necessarily changes. This makes it possible to change the duty while keeping the frequency constant. If the frequency is constant, noise countermeasures become easier.

Second Embodiment FIG. 4 shows only the control circuit 8 of the switching regulator according to the second embodiment. Parts other than the control circuit 8 are the same as those in FIG. Since many parts of the control circuit in FIG. 4 are the same as those in the control circuit 8 in FIG. 2, the common parts are denoted by the same reference numerals and description thereof is omitted. The circuit shown in FIG. 4 has a timer 43a for generating a pulse instead of the clock pulse generator 42 and the flip-flop 43 shown in FIG. The timer 43a is composed of, for example, a mono-multi vibrator, and generates a pulse having a constant pulse width in response to the output of the delay circuit 41 obtained at time t6 in FIG. This pulse functions as an ON control signal for the first switching element 7. As described above, if the ON time width of the first switching element 7 is fixed, it becomes easy to satisfy the resonance condition.

Third Embodiment Next, a switching regulator according to a third embodiment shown in FIG. 5 will be described. In FIGS. 5 to 17, parts common to FIGS. 1 to 3 are denoted by the same reference numerals, and description thereof is omitted. As is clear from the comparison between FIG. 1 and FIG. 5, in the circuit of FIG.
It is the opposite of the figure. The control circuit 8 shown in FIG. 5 is configured as shown in FIG. Variable pulse generation circuit
The trigger of 37 is that the clock pulse of FIG. 7A generated from the clock pulse generator 42 is timed by the first delay circuit 50.
It is determined by delaying by T1. The delay time T1 is set to be shorter than 1/2 of the half wave of the LC resonance waveform. Since the resonance frequency is determined by the circuit constant, it can be known in advance. The delay time T2 of the second delay circuit 41 is set to t3 to t6 as shown in FIG. 7 (C).

The operating principle of the switching regulator shown in FIG.
It is substantially the same as the figure. First, when a clock pulse is generated at t1, the flip-flop 43 is reset and the first flip-flop 43 is reset.
Of the primary winding 6 is switched off.
And resonance of the resonance circuit by C of the capacitor 18 occurs,
As shown in FIG. 7 (D), the voltage V1 of the first switching element 7, that is, the capacitor 18, gradually increases from t1 to t2. A trigger is given from the first delay circuit 50 to the variable pulse generation circuit 37, and the output pulse turns on the second switching element 19 and puts the diode 21 in a forward bias state.
When the control voltage source 22 is connected to the tertiary winding 20 at time t2, the resonance is interrupted as in the first embodiment. At this time, energy is supplied from the control voltage source 22 to the transformer 5. t2
The time point is the time point when the upward voltage of the tertiary winding 20 becomes higher than the control voltage E2. Since the diode 21 is reverse-biased during the period when the upward voltage of the tertiary winding 20 is lower than the control voltage E2, even if the second switching element 19 is turned on in this state, the control voltage source 22 becomes Not connected to the next winding 20. Therefore, the on-control pulse shown in FIG. 7B may be generated before the time t2.

When the second switching element 19 is turned off in response to the pulse of FIG. 7B going low at time t3, the interruption of the LC resonance is released, and the resonance operation continued at time t2 occurs. . When the potential of the capacitor 18 reaches the same potential as the interval between t2 and t3 after the voltage of the capacitor 18 reaches the maximum at t4, the diode 21 becomes reverse-biased and the built-in diode of the second switching element 19 becomes forward-biased. At time t6, when the voltage V1 of the first switching element 7 and the capacitor 18 becomes zero volt, an ON control signal is given to the first switching element 7. This results in an operation state similar to that of the circuit of FIG. Note that the output voltage is controlled by changing the trailing edge (falling edge) of the output pulse of the variable pulse generation circuit 37.

According to this embodiment, the same operation and effect as those of the first embodiment can be obtained.

Fourth Embodiment FIG. 8 shows a control circuit 8 of a switching regulator according to a fourth embodiment. The components other than the control circuit 8 are formed in the same manner as in FIG. The control circuit 8 in FIG. 8 is obtained by replacing the flip-flop 43 in FIG. 8 with a timer 43a. Therefore, the ON time width of the first switching element 7 is a fixed time determined by the timer. This fourth embodiment is similar to the second embodiment.
This has the same effect as the embodiment.

Fifth Embodiment In the switching regulator shown in FIG. 9, a control voltage source 22 is connected to a secondary winding 9 by a second switching element 19.
And a diode 21. This case is also equivalent to the circuit of FIG. 1, and a similar effect can be obtained. In this embodiment, the variable capacitance diode Dc having a higher rising voltage in the forward direction than the diode 10 is connected to the diode 10.
Are connected in parallel. The variable capacitance diode Dc has a large capacitance during a period in which the diode 10 generates noise and absorbs noise well. This variable capacitance diode
Dc can also be connected in parallel to the diodes 10 of the first to fourth and sixth to eleventh embodiments described later.

Sixth Embodiment A switching regulator shown in FIG. 10 is a circuit in which a capacitor 60, a third switching element 61, and a diode
62. The capacitor 60 has a third switching element 61 and a diode 62
Are selectively connected by switching means. The period during which the third switching element 61 is turned on is the third period.
From the time after t2 to the time before t4 in the figure,
This is the period from t3a to t3b in FIG. When the capacitor 60 is connected during the off period of the first switching element 7 as described above,
The resonance frequency in the LC resonance circuit decreases, and t3 in FIG.
As shown in the section from a to t3b, the peak of the sine wave indicated by the dotted line and a waveform in which the vicinity thereof is clamped are obtained, and the peak of the resonance voltage can be reduced. The ON period of the second switching element 19 is from t4 to t5 as in the case of FIG.

In addition, not only the first switching element 7 but also the on-time width of the third switching element 61 can be changed to control the off-period of the first switching element 7 to adjust the output voltage.

Seventh Embodiment The switching regulator shown in FIG. 12 is a modification of the circuit shown in FIG. Here, a quaternary winding 63 electromagnetically coupled to the primary winding 6 is provided, and a second capacitor 60 is connected in parallel with the quaternary winding 63. A parallel circuit of a diode 62 and a third switching element 61 is connected between the tertiary winding 63 and the second capacitor 60. Even if the capacitor 60 is connected via the quaternary winding 63 as shown in FIG. 12, it is equivalently connected in parallel with the primary winding 6. The ON / OFF control of the third switching element 61 is performed in the same manner as in the embodiment of FIG. According to the seventh embodiment, the same operation and effect as those of the sixth embodiment can be obtained.

Eighth Embodiment In the switching regulator according to the eighth embodiment shown in FIG. 13, a capacitor 60 is connected in parallel with the primary winding 6. The connection of the parallel circuit of the third switching element 61 and the diode 62 to the capacitor 60 in series is the same as in FIG. In the circuit of FIG. 13, a reactor L1 is connected in series with the primary winding 6 to assist resonance. This type of reactor can be inserted in other embodiments.

The circuit of FIG. 13 is substantially equivalent to the circuits of FIGS. 10 and 12, and has the same operation and effect.

Ninth Embodiment In a switching regulator according to a ninth embodiment shown in FIG. 14, a capacitor 60 is connected in parallel to the secondary winding 9. The parallel circuit of the diode 62 and the third switching element 61 is connected to the capacitor 60 in the same manner as in FIG. In this embodiment, the secondary winding 9 has the function of coupling the capacitor 60 to the primary winding 6 in addition to the function as the output winding. Also in the circuit of FIG. 14, the capacitor 60 is equivalently connected in parallel to the primary winding 6 during the off period, and the same operation and effect as those of the circuits of FIGS. 10, 12, and 13 can be obtained.

Tenth Embodiment In a switching regulator according to a tenth embodiment shown in FIG. 15, a constant voltage source 60 is used instead of the capacitor 60 shown in FIG.
a is connected. This voltage source 60a functions in the same manner as the capacitor 60, and has the function of clamping the section from t3a to t3b in FIG. When the voltage of the capacitor 18 is clamped,
The resonance operation is interrupted by the capacitor 18 and the primary winding 6. The voltage source 60a is equivalent to the one obtained by increasing the capacity of the capacitor 60 in FIG. Therefore, the same operation and effect as those of the embodiment shown in FIGS. 10 to 14 are obtained.

[Eleventh Embodiment] In the switching regulator shown in FIG.
The control voltage source 22 includes the second switching element 19 and the diode
21 and is directly connected in parallel to the primary winding 6.
Even if such connection is made, it is equivalent to FIG. 1, and the same operation and effect can be obtained.

[Modifications] The present invention is not limited to the above-described embodiment, and for example, the following modifications are possible.

(1) In the circuits shown in FIGS. 10, 12, 13, 14, and 15, the second switching element 19 and the diode 21
The polarity shown in FIG. 17 may be reversed, and the second switching element 19 may be turned on at the same timing as in FIG. 7 to obtain the waveform shown in FIG. In this case, the second switching element 19 is turned on from t2 to t3,
The third switching element 61 is turned on between t4a and t4b.

(2) Connect the capacitor 60 shown in FIGS. 12, 13 and 14 to
As shown in the figure, it can be replaced with a constant voltage source 60a.

(3) It is possible to connect a diode to the first switching element 7 in series.

(4) Like the second capacitor 60, the first capacitor 18 can be directly connected in parallel to the primary winding 6.
Further, the first capacitor 18 may be a stray capacitance of the primary winding 6.

(5) Some or all of the switch composed of the combination of the second switching element 19 and the diode 21 can be replaced with another analog switch such as a bipolar transistor. The first switching element 7 and the third switching element 61 can be replaced with a bipolar transistor or the like.

(6) The output rectifying / smoothing circuit 14 can be omitted to provide a DC-AC converter (inverter).

(7) The output can be taken out by using the transformer 5 as a single-turn transformer.

[Effects of the Invention] As is clear from the above, according to the invention of each claim,
Control of the off time width can be easily achieved. According to the third aspect, the maximum amplitude value of the voltage at the time of resonance can be suppressed.

[Brief description of the drawings]

1 is a circuit diagram showing a switching regulator according to a first embodiment of the present invention, FIG. 2 is a block diagram showing a control circuit in FIG. 1 in detail, and FIG. 3 is a waveform showing the state of each part in FIG. FIG. 4, FIG. 4 is a block diagram showing a control circuit of the switching regulator of the second embodiment, FIG. 5 is a circuit diagram showing a switching regulator of the third embodiment, and FIG. FIG. 7 is a waveform diagram showing the state of each part of FIG. 5, FIG. 8 is a block diagram showing a control circuit of the switching regulator of the fourth embodiment, and FIG. 9 is a diagram of the fifth embodiment. FIG. 10 is a circuit diagram showing a switching regulator according to a sixth embodiment. FIG. 11 is a circuit diagram showing the voltage across the first switching element in FIG. 10 in the same manner as FIG. 3 (C). Waveform diagram shown, Fig. 12, Fig. 13 FIGS. 14, 15, and 16 are circuit diagrams showing switching regulators of the seventh, eighth, ninth, tenth, and eleventh embodiments, respectively. FIG. 3 is a voltage waveform diagram of the switching element of FIG. 1 ... power supply, 5 ... transformer, 6 ... primary winding, 7 ...
1st switching element, 8 ... control circuit, 9 ... secondary winding, 18 ... capacitor, 19 ... second switching element, 20 ... tertiary winding, 21 ... diode, 22 ... Control voltage source.

Claims (3)

(57) [Claims] A DC power supply; a transformer having a winding connected between one end and the other end of the DC power supply; first and second main terminals and a control terminal; A first switch having a main terminal connected to one end of the DC power supply through the winding, and a second switch having the second main terminal connected to the other end of the DC power supply; A capacitor consisting of an independent capacitor or stray capacitance associated with the winding, an output circuit coupled to the transformer, a control voltage source, and connecting the control voltage source directly or to the winding. A second switch for equivalently connecting in parallel; an on / off control of the first switch;
The second switch within a range after the maximum amplitude value of the voltage of the capacitor caused by resonance between the winding and the capacitor during the off period of the switch and before the voltage becomes zero. A control circuit formed so as to be turned on and to change the on-time width of the second switch. 2. A power supply comprising: a DC power supply; a transformer having a winding connected between one end and the other end of the DC power supply; first and second main terminals and a control terminal; A first switch having a main terminal connected to one end of the DC power supply through the winding, and a second switch having the second main terminal connected to the other end of the DC power supply; A capacitor consisting of an independent capacitor or stray capacitance associated with the winding, an output circuit coupled to the transformer, a control voltage source, and connecting the control voltage source directly or to the winding. A second switch for equivalently connecting in parallel; an on / off control of the first switch;
The second switch within a range before the maximum amplitude value of the voltage of the capacitor caused by resonance between the winding and the capacitor during the off period of the switch and after the voltage rises from zero. On and the second
And a control circuit formed so that the on-time width of the switch can be changed. 3. An additional capacitor or voltage source associated with a resonant circuit of the winding and the capacitor, and the resonance of the winding and the capacitor causes the voltage of the capacitor to rise from zero. A third switch for associating the additional capacitor or voltage source with the winding within a period of time until the maximum amplitude value becomes zero again and within a range including the time point of the maximum amplitude value. 3. The switching power supply device according to claim 1, wherein