JP2019162011A - Motor control device and motor control method - Google Patents

Abstract

To provide a motor control device capable of reducing a rotational variation of a motor and a noise at a rotational number range wider than that of a conventional technique, and maintaining stability of a control.SOLUTION: A motor control device is a motor control device that control a motor driving part that converts a pole phase signal expressing a pole phase of a motor, and outputs a signal expressing a rotational amount and a rotational direction of an output axis of the motor and a rotational position detection signal with high resolution for the pole phase signal. The motor control device comprises a control part that control a power which has to be supplied to the motor by the motor driving part on the basis of the rotational position detection signal and a target signal. The control part includes a control gain setting part that sets a control gain so that a responsibility of a specific frequency is reduced in accordance with the rotational number of a pole number expressing a pole of the motor and the rotational number of the motor. The control gain setting part progressively increases or progressively decreases a drop amount of the responsibility in accordance with the specific frequency.SELECTED DRAWING: Figure 1

Description

  The present invention relates to a motor control device and a motor control method.

  A feedback control technique for detecting the speed and position of a rotary motor and controlling it to match a target value has been known for a long time. For this purpose, an encoder is generally attached to the motor shaft.

  However, if an encoder is attached, the cost increases and the size increases, and there is a demand for detecting the position and speed without using position detecting means such as an encoder.

  On the other hand, Patent Document 1 does not use an encoder, and originally uses a plurality of Hall elements provided for sequentially commutating the drive voltage of a brushless motor to each coil according to the motor angle, A technique for generating a high-resolution rotational position detection signal equivalent to the above is disclosed. For drive commutation, angle information with a relatively coarse resolution is used for the zero cross of the hall signal, but as a rotational position detection signal for position / speed control, the zero cross is achieved by multi-level slicing the sine wave before the hall signal zero cross. Position information with higher resolution is obtained.

  However, the generated rotational position detection signal (pseudo-encoder signal) has a periodic phase shift due to angular arrangement deviations, sensitivity variations, Hall signal offsets, and the like of a plurality of Hall elements. It is known that the cycle is once per n pole pairs of the motor rotor (rotor) or n (n> = 1). The number of pole pairs around the motor is often about 3 to 6 pole pairs. In the case of 6 pole pairs (that is, 12 poles), the motor speed is 6 times (n = 1) or 12 times (n = 2) ), A periodic detection position shift having a frequency component occurs.

  When speed control is performed using this position detection signal (pseudo encoder signal), control is performed so that the detection speed extracted from the position detection signal matches the target value. Will follow, causing a periodic shift in motor speed, that is, uneven rotation.

  Patent Document 2 discloses a notch filter (a control gain setting unit that sets a control gain so that the responsiveness of a specific frequency is reduced) that cancels a periodic shift component included in the pseudo encoder signal in the control loop. ) Is inserted to reduce the rotational unevenness caused by the follow-up control of the periodic deviation.

  However, if the center frequency of this notch filter (the center frequency at which the gain is to be reduced) is close to the control cross frequency (the frequency at which the gain of the control loop loop transfer function becomes 0 dB), the phase delay of the filter will cause a phase margin at the cross frequency. The speed or position control becomes unstable or oscillates. In order to avoid this, in Patent Document 2, it is preferable that the gain is not reduced by the filter (gain is set to 1) below the crossing frequency, preferably below 1.5 times the crossing frequency.

  For this reason, in the technique of Patent Document 2, the filter does not work at 1.5 times or less of the control cross frequency, and the rotation unevenness due to the pseudo encoder unevenness cannot be reduced. Also, if the filter is applied in this band, the control becomes unstable due to insufficient phase margin.

  As described above, in the conventional motor control device, the reduction in the rotation unevenness of the motor is sufficient, and when trying to apply the filter in the desired frequency band, the phase margin is insufficient, thereby There is a problem that the control becomes unstable.

  The present invention has been made in view of the above, and an object of the present invention is to use motor drive means that can generate a motor position detection signal without using expensive position detection means such as an encoder. To provide a motor control device that can reduce rotational unevenness and noise when controlled using this even in the presence of periodic errors in a wider rotational speed range than conventional technology, and maintain control stability. is there.

  In order to solve the above-described problems and achieve the object, a motor control device according to the present invention converts a magnetic pole phase signal representing a magnetic pole phase of a motor to represent a rotation amount and a rotation direction of the output shaft of the motor. A motor driving unit that outputs a rotational position detection signal having a high resolution with respect to the magnetic pole phase signal. The motor control device includes a control unit that controls electric power that the motor driving unit should supply to the motor based on the rotational position detection signal and the target signal. The control unit includes a control gain setting unit that sets a control gain so that the responsiveness of a specific frequency is lowered according to the number of poles representing the number of magnetic poles of the motor and the number of rotations of the motor, The control gain setting unit gradually increases or decreases the amount of decrease in the response in accordance with the specific frequency.

  According to the present invention, it is possible to reduce the rotation unevenness and noise of the motor in a wide rotation speed range and to maintain control stability.

FIG. 1 is a block diagram illustrating a functional configuration of the motor control device according to the first embodiment. FIG. 2 is a diagram for explaining the relationship between the Hall elements 22, 23 and 24 and the motor 2. FIG. 3 is a diagram illustrating an example of a hall signal and a signal generated therefrom. FIG. 4 is a block diagram illustrating a configuration example of the filter coefficient calculation unit. FIG. 5 is a diagram illustrating calculation algorithms of the first attenuation factor calculation unit and the second attenuation factor calculation unit in a notation according to the C language. FIG. 6 is an explanatory diagram of calculation algorithms of the first attenuation factor calculation unit and the second attenuation factor calculation unit. FIG. 7 is a diagram showing a configuration example of the filter means. FIG. 8 is a flowchart showing an example of a signal processing flow (signal flow) of two notch filters. FIG. 9 is a diagram illustrating an example of frequency characteristics of two notch filters. FIG. 10 is a diagram showing an example of frequency characteristics of the filter means when the filter characteristics set by the filter coefficient calculation means are operated at a relatively low rotational speed. FIG. 11 is a diagram showing an example of frequency characteristics of the filter means when the filter characteristics set by the filter coefficient calculation means are operated at a relatively high rotational speed. FIG. 12 is a block diagram illustrating a functional configuration of the motor control device according to the second embodiment. FIG. 13 is a block diagram illustrating a functional configuration of the motor control device according to the third embodiment. FIG. 14 is a block diagram illustrating a functional configuration of the motor control device according to the fourth embodiment. FIG. 15 is a diagram illustrating a schematic configuration of an image processing apparatus including the motor control device according to any one of the first to fourth embodiments. FIG. 16 is a principal perspective view for explaining a mechanism for transporting the transfer paper in the image processing apparatus.

  Embodiments of the present invention will be described below with reference to the drawings.

(Embodiment 1)
FIG. 1 is a block diagram illustrating a functional configuration of the motor control device according to the first embodiment. The motor control device 1 according to the present embodiment includes a control unit 4 that controls a motor driving unit 3 that drives a motor 2. In the motor control device 1, the control unit 4 controls the driving of the motor 2 so that the number and frequency of the clock CLK are proportional to the rotation angle (motor angle) and angular velocity of the output shaft of the motor 2.

  CLK means a command clock pulse, the number of CLK is a position command in motor drive control, and the frequency of CLK is a speed command in motor drive control.

  In FIG. 1, reference numeral 5 denotes a frequency detection means for detecting the frequency of the command clock CLK and outputting a speed command ftgt. Specifically, the frequency detection means 5 detects the frequency of the CLK by measuring the period of the CLK and calculating the reciprocal thereof, or counting the number of CLKs and taking the difference in the count value at regular intervals, etc. This can be realized by a known method.

  In FIG. 1, reference numeral 6 denotes a counting means, which counts the number of CLK edges and outputs a position command xtgt. Reference numerals A and B in the figure indicate two-phase pulses that are the result of detecting the position of the motor 2. In the two-phase pulses A and B, a pulse edge arrives every time the output shaft of the motor 2 moves by a predetermined angle. Further, it is preferable that the phase relationship between the two-phase pulses A and B is reversed depending on the rotation direction of the output shaft of the motor 2. For example, if the number of pulses of the two-phase pulses A and B is designed so that the output shaft of the motor 2 emits 90 at equal intervals for each revolution, the number of the two-phase pulses A and B is four times that at both edges. 360 times, and every time the output shaft of the motor 2 moves (rotates), one edge comes. Further, for example, when the rotation direction is normal rotation, the pulse A phase becomes a leading phase than the pulse B phase, and when the rotation direction is reverse rotation, the pulse B phase becomes a leading phase than the pulse A phase.

  The frequency detection means 7 detects the arrival frequency of the edge of the position detection pulse A or B, or both, and outputs a detection speed fdet corresponding to the rotation speed of the output shaft of the motor 2. For example, the frequency detecting means 7 detects both edge frequency equivalent values by counting both edges of the pulses A and B and subtracting the count value every predetermined time. Alternatively, the frequency detecting means 7 may measure the rising edge period of the pulse A and take the reciprocal. These are all known methods, but if the difference between the count values is used, detection is possible even at low frequencies including zero speed, and the current position of the output shaft of the motor 2 is maintained with position hold, that is, target speed zero. Can be controlled.

  The counting means 8 counts the edges of the two-phase position detection pulses A and B, and outputs a detection position xdet that is a value corresponding to the angle of the output shaft of the motor 2. When the phase relationship between the pulses A and B is reversed depending on the rotation direction of the output shaft of the motor 2, the count can also be increased or decreased by a known method depending on the phase relationship, and the current angle equivalent value can be obtained correctly.

The position control means 9 outputs a position control value xctl proportional to the difference between the position command xtgt and the detected position xdet. A calculation formula for obtaining the position control value xctl from the difference between the position command xtgt and the detected position xdet is, for example, the following formula (1).
xctl = Gpp * (xtgt-xdet) (1)
In the equation, Gpp indicates a position control gain (a constant determined by design).

The speed control means 10 outputs the speed control value fctl so that the difference (speed deviation) between the added value of the speed command ftgt and the position control value xctl and the detected speed fdet becomes small. Preferably, in order to set the steady speed deviation to 0, a proportional-integral control calculation is performed in which a value proportional to the deviation and a value obtained by integrating the deviation are added to obtain a control value. When the integral is expressed by the Laplace operator “1 / s”, the calculation formula is, for example, the following formula (2).
verr = ftgt + xctl-fdet (verr is speed deviation)
fctl = Gvp * verr + Gvi * verr * (1 / s) (2)
In equation (2), Gvp is a speed control proportional gain (a constant determined by design), Gvi is a speed control integral gain (a constant determined by design), and 1 / s represents an integral (known expression). .

  In this way, the speed control value fctl is controlled so that the speed deviation is 0, that is, ftgt = fdet, and the position deviation is also 0, that is, xtgt = xdet.

  The speed deviation may be created only by the position control value xctl and the detected speed fdet without adding the speed command ftgt to the speed deviation. However, in this case, xctl = fdet, and the position deviation is fctl / Gpp after xctl / Gpp, resulting in a position deviation proportional to the speed. By adding the speed command ftgt to the speed deviation, the position deviation can be made zero regardless of the speed.

  Further, by such a position control loop, when CLK does not arrive, hold control is performed so as to maintain the current position even if there is an external force.

  Next, the filter means 11 inputs the speed control value fctl, and outputs a control amount ctl obtained by attenuating a specific frequency component therefrom. In this example, the filter configuration attenuates two specific frequency components, and filter coefficient groups cn1 and cn2 that determine the characteristics are also input.

  The filter coefficient calculation means 12 calculates the center frequencies fn1 and fn2 of the two filters and the gains Gn1 and Gn2 at the center frequencies from the speed command ftgt and the motor pole pair number p, and from these, the respective filter coefficient groups cn1 , cn2 is calculated and given to the filter means 11.

  The gains Gn1 and Gn2 at the center frequency are 1.0 times or less because the filter is an attenuation filter. That is, the response of a specific frequency is reduced. In the present application, Gn1 and Gn2 may be referred to as attenuation rates or reduction amounts.

  As will be described later, the attenuation rate at the center frequency is calculated so that the higher the center frequency fn1 or fn2 is, the smaller the value is, that is, the smaller the amount of decrease, that is, the larger the decrease amount.

  The motor drive unit 3 rotationally drives the motor 2 with a voltage or current corresponding to the control amount ctl from the control unit 4 and outputs position detection signals A and B corresponding to the rotation angle of the output shaft of the motor.

  The drive means 20 of the motor drive unit 3 rotates the motor 2 by supplying a voltage or current proportional to the control amount ctl to the motor coil of the motor 2. In this example, a three-phase motor is used, and the voltage or current proportional to the controlled variable ctl is expressed as an electrical angle using the zero cross signal from the hall encoder 21 (an angular unit in which the magnet's magnetized one pole pair is 360 degrees). ) A rotational torque is generated by allocating to the three-phase coils U, V, W every 60 degrees, and the motor 2 is rotationally driven.

  In this example, the motor 2 includes a stator having a three-phase coil and a rotor (rotor) magnetized with multiple poles. The rotor is rotated by the electromagnetic action of the rotor magnet by the rotating magnetic field generated by the stator.

  The hall elements 22, 23, and 24 detect the magnetic field of the rotor magnet and output a sinusoidal hall signal along with the rotation of the magnetic pole pair. By arranging the three Hall elements 22, 23, 24 with an electrical angle of 120 degrees apart, it is possible to output Hall signals h1, h2, h3 whose phases are shifted by 120 degrees. Note that these signal output means are not limited to Hall elements, and other sensor devices may be used. Any means for generating a periodic signal corresponding to the magnetic pole of the rotor may be used.

  The relationship between the Hall elements 22, 23, 24 and the motor 2 is shown in FIG. In the figure, U, V, and W are three-phase coil ends, and the coils are not shown.

  Here, the rotor 30 of the motor 2 is magnetized to 8 poles (4 pole pairs). An angle unit in which an angle formed by a pair of N poles and S poles (here, a motor angle of 90 degrees) is 360 degrees is called an electrical angle. The Hall elements 22, 23, and 24 are arranged with an electrical angle shifted by 120 degrees.

  Returning to FIG. 1, the description of the motor driving unit 3 will be continued. The Hall Hall encoder 21 generates a zero cross signal from each of the Hall signals h1, h2, and h3 and supplies the generated zero cross signal to the driving unit 20. Furthermore, the Hall signals h1, h2, and h3 are sliced not only at the zero level but also at multiple levels to generate position detection pulse signals A and B having higher resolution than the zero cross.

  Next, Hall signals h1, h2, and h3 and signal examples generated therefrom are shown in FIG. Signals h1, h2, and h3 are sinusoidal hall signals obtained from the hall elements 22, 23, and 24, and make a round at an electrical angle of 360 degrees.

  Symbols h1z, h2z, and h3z in FIG. 3 are digital signals obtained by slicing the hall signals h1, h2, and h3 with zero crossing, and are used by the driving unit 20. Use of the digital signals h1z, h2z, and h3z has advantages that the circuit configuration is simplified and the motor driving unit 3 is inexpensive. Some Hall element outputs are differential "+" and "-" terminals. The differential voltage between terminals is h1, h2, h3, and the voltage comparison result between terminals is h1z, h2z. , h3z may be used.

  The pulse signals A and B are two-phase high-resolution position detection signals obtained from the result of slicing the hall signals h1, h2, and h3 not only at zero level but also at multiple levels. This can be generated, for example, by the technique disclosed in Japanese Unexamined Patent Application Publication No. 2015-019563.

  With such a configuration, the motor 2 is driven and controlled so that the number and frequency of the clocks CLK are proportional to the motor angle and the angular velocity.

  The edge phase and frequency of the position detection signals A and B obtained from the Hall signal due to the displacement of the Hall elements 22, 23, 24 (deviation from the 120-degree separation), the variation of the Hall signal amplitude, the offset, and the like. Causes a periodic shift according to the number of pole pairs. When the speed and position are controlled using the A and B signals, the control is performed following the periodic deviation, which may cause rotation unevenness and noise.

  Therefore, the periodic shift is caused by attenuating the specific frequency component corresponding to the number of pole pairs and the number of rotations by the filter unit 11 and the filter coefficient calculating unit 12 with an attenuation factor varied according to the frequency. By removing the components, uneven rotation and noise can be reduced.

  At this time, the attenuation rate of the specific frequency is gradually weakened to the cross frequency of the speed control (the frequency at which the gain of the loop transfer function becomes 0 dB), in other words, the attenuation rate of the high frequency is gradually increased from the cross frequency of the speed control. As a result, the effect of reducing rotational unevenness by the filter can be obtained even in the vicinity of the crossing frequency. At this time, since the attenuation rate becomes weak in the vicinity of the crossing frequency, the phase margin can be secured and the control can be kept stable.

  A configuration example of the filter coefficient calculation means 12 is shown in FIG. The first fixed value multiplication unit 40 calculates the motor rotation speed fm (Hz) from the speed command ftgt. As a specific example, the position detection pulses A and B are multiplied by the number of edges of one revolution of the motor, that is, the reciprocal of the encoder count number Nenc of one revolution.

  Next, the second fixed value multiplication unit 41 multiplies the motor rotation speed fm by the number p of the magnetized magnetic pole pairs of the rotor (4 in the case of 8-pole magnetization in FIG. 2). The result is the attenuation center frequency fn1 (Hz) of the first filter.

  Next, the third fixed value multiplier 42 multiplies the motor rotation speed fm by twice the number p of the magnetic pole pairs of the rotor (8 in the case of 8-pole magnetization in FIG. 2). The result is the attenuation center frequency fn2 (Hz) of the second filter.

These frequencies fn1 and fn2 are characteristic frequencies corresponding to the motor rotation speed and the number of pole pairs p.
fm = ftgt / Nenc (motor speed)
fn1 = fm * p (pole frequency multiple frequency component)
fn2 = fm * 2p (frequency component twice the number of pole pairs)
It becomes.

  Next, the first attenuation rate calculation unit 43 calculates a gain (attenuation rate) Gn1 at the center frequency fn1 of the first filter. Gn1 is decreased according to the frequency (the higher the frequency, the smaller the gain, that is, the greater the attenuation factor).

  Further, the second attenuation rate calculation unit 44 calculates a gain (attenuation rate) Gn2 at the center frequency fn2 of the second filter. Gn2 is decreased according to the frequency (the gain is decreased as the frequency is higher, that is, the attenuation rate is increased).

  FIG. 5 shows a calculation algorithm of the first attenuation rate calculation unit 43 and the second attenuation rate calculation unit 44 in a notation according to the C language. The calculation by the first attenuation rate calculation unit 43 and the second attenuation rate calculation unit 44 may be the same algorithm.

Let the input frequency be f.
When f is smaller than the predetermined frequency f1, the attenuation factor is a fixed value g1.
When f is from the predetermined frequency f1 to f2, the attenuation rate follows the straight line P12 * f + Q12.
When f is from a predetermined frequency f2 to f3, the attenuation rate follows the straight line P23 * f + Q23.
When f is equal to or higher than the predetermined frequency f3, the attenuation rate is a fixed value g3.

  In FIG. 6, the calculation algorithm of the 1st attenuation factor calculation part 43 and the 2nd attenuation factor calculation part 44 is shown in figure.

The frequency f1 or less is g1, the frequency f2 is g2, the frequency f3 is g3, and the frequency f3 or more is g3. Therefore, the coefficients of the straight lines P12 * f + Q12 and P23 * f + Q23 are as follows.
P12 = (g2-g1) / (f2-f1) (f1-f2 section gain linear slope)
Q12 = g1-P12 * f1 (f1-f2 interval gain linear intercept)
P23 = (g3-g2) / (f3-f2) (f2-f3 section gain linear slope)
Q23 = g2-P23 * f2 (f2-f3 section gain linear intercept)

  Preferably, the frequency f1 is in the vicinity of the speed control crossing frequency, f2 is 1.5 times f1, and f3 is about twice f1. Further, the gain g1 is set to 1, the gain g2 is set to 0.2 times, for example, and the gain g3 is set to about 0.1 times, for example.

  By doing this, the stability of the control is maintained without attenuation of the filter below the vicinity of the crossing frequency, and the filter attenuation factor increases proportionally (becomes low gain) from the crossing frequency to 1.5 times that. Thus, it is possible to obtain a filter effect and secure a control phase margin. Furthermore, the attenuation rate changes along a different straight line in the high range.

  By compensating for the change in the attenuation rate at two or more break points with a straight line, it is possible to realize a change in the attenuation rate with a high degree of freedom while using a simple calculation method called a straight line. Therefore, the attenuation rate change characteristic can be optimized according to the phase margin of the system and other conditions.

  Returning to FIG. 4, the description of the filter coefficient calculation means 12 will be continued. The first coefficient calculation unit 45 is a coefficient calculation unit for the first filter, and collects each coefficient (a1, a2, b0, b1, b2) of the digital filter from the center frequency fn1, the attenuation factor Gn1, and the sharpness Qn1. cn1) and the calculated value is given to the first filter of the filter means 11. The symbol Qn1 is a parameter that determines the steepness of the attenuation characteristic, and the attenuation factor and phase around the center frequency change. Therefore, it is determined by a trade-off between the rotational unevenness reduction characteristic and the control stability. This may also be varied according to the center frequency, but here it is simply a fixed value.

  The second coefficient calculation unit 46 is a coefficient calculation unit for the second filter. The second coefficient calculation unit 46 collects each coefficient (a1, a2, b0, b1, b2) of the digital filter from the center frequency fn2, the attenuation factor Gn2, and the sharpness Qn2. cn2) is calculated and given to the second filter of the filter means 11. The symbol Qn2 is a parameter that determines the steepness of the attenuation characteristic, and the attenuation factor and phase around the center frequency change. Therefore, it is determined by a trade-off between the rotational unevenness reduction characteristic and the control stability. This may also be varied according to the center frequency, but here it is simply a fixed value.

The coefficient calculation performed by the first coefficient calculation unit 45 and the second coefficient calculation unit 46 may be performed by, for example, the following equation for a second-order IIR digital filter having a sampling frequency fs (Hz). Here, the filter number (1, 2, etc.) is omitted, and the center frequency fn, the attenuation rate Gn, and the sharpness Qn are used.
wn = 2pi * fn; (pi is the pi)
w0 = wn / fs; (fs is the sampling frequency)
An = sqrt (Gn);
alfa = sin (w0) / (2 * Qn);
a0 = 1 + alfa / An;
a1 = -2 * cos (w0) / a0;
a2 = (1-alfa / An) / a0;
b0 = (1 + alfa * An) / a0;
b1 = -2 * cos (w0) / a0;
b2 = (1-alfa * An) / a0;

  FIG. 7 shows a configuration example of the filter unit 11. As shown in the figure, the filter means 11 includes a notch filter 51 and a notch filter 52 that attenuate a specific frequency. The notch filter 51 corresponds to the first filter described above, and the notch filter 52 corresponds to the second filter described above.

  When the well-known second-order IIR digital filter is used for the notch filter 1 and the notch filter 2, the coefficients a1, a2, b0, b1, and b2 are input. The filter coefficient is denoted by cn1, and the second filter coefficient is denoted by cn2.

  In FIG. 7, the input of each filter is indicated by x and the output is indicated by y. The notch filter 1 and the notch filter 2 are connected in cascade (cascade), and the frequency characteristics are a combination of both. That is, the specific frequency components of these filters 51 and 52 can be attenuated. The number of filters constituting the filter means 11 may be one or multiple. Even when the number of filters is one, the basic effect of the present application can be obtained. However, if a multi-stage filter configuration is used, a higher effect can be obtained.

FIG. 8 is a flowchart showing an example of a signal processing flow (signal flow) of the notch filter 51 and the notch filter 52. In the example of FIG. 8, the notch filters 51 and 52 are digital filters called IIR (Infinite Impulse Response) type and operate at a sampling frequency fs (Hz). In the figure, Z ⁻¹ indicates a one-sample delay, a1, a2, b0, b1, b2 indicate coefficient multiplication, and coefficient values are given from the outside. In the figure, x indicates input and y indicates output.

The latest x value is x [n], the x value before one sample is x [n-1], and the x value before two samples is x [n-2]. The latest y value is y [n], the y value before one sample is y [n-1], and the y value before two samples is y [n-2]. This signal processing is realized by the calculation of the following expression (3).
y [n] = b0 * x [n] + b1 * x [n-1] + b2 * x [n-2]-a1 * y [n-1]-a2 * y [n-2] (3)

  FIG. 9 shows frequency characteristic examples of the notch filter 1 and the notch filter 2. Here, the gain (solid line: left axis) when the center frequency fn is 100 Hz, the sharpness Qn is 10, and the gain (attenuation rate) is 0.1 times (-20 dB) and 0.5 times (-6 dB). ) And phase (broken line: right axis). It can be seen that when the attenuation factor is 0.5 times, the phase lag in the lower region is less than the center frequency. Therefore, when the center frequency is close to the speed control band, decreasing the attenuation factor (increasing the gain) results in less phase delay and makes it easier to ensure a control phase margin.

  FIG. 10 shows an example of the frequency characteristic of the filter unit 11 when the filter characteristic set by the filter coefficient calculation unit 12 is operated at a relatively low rotational speed.

The setting example and the rotation speed are as follows.
fs = 5000 Hz (sampling frequency)
Nenc = 720 (Edge / motor 1 round)
ftgt = 10800 Hz
p = 6 (pole pair)
f1 = 60 Hz
f2 = 100 Hz
f3 = 200 Hz
g1 = 1
g2 = 0.2
g3 = 0.1
fm = 10800/720 = 15 Hz (rotation speed)
fn1 = 15 * 6 = 90 Hz (Notch filter 51 center frequency)
fn2 = 15 * 12 = 180 Hz (Niche filter 52 center frequency)
Gn1 = 10, Qn2 = 10

Therefore, the gain at the center frequency of each filter is
Gn1 = 0.4
Gn2 = 0.12
Is calculated.

  If f1 is in the vicinity of the control band, fn1 is lower than 1.5 times, but the attenuation factor is not as high as 0.4. Therefore, the phase delay is small and the control is not unstable. Nevertheless, the effect of reducing rotation unevenness can also be obtained.

  In this case, the phase delay at f1 (= 60 Hz) of the filter means 11 (two synthesis filters) is about 12 degrees, and this phase delay is generally acceptable in terms of control stability.

  FIG. 11 shows an example of frequency characteristics of the filter means 11 when the filter characteristics set by the filter coefficient calculation means 12 are operated at a relatively high rotational speed.

The setting example and the rotation speed are as follows.
fs = 5000 Hz (sampling frequency)
Nenc = 720 (Edge / motor 1 round)
ftgt = 25200 Hz
p = 6 (pole pair)
f1 = 60 Hz
f2 = 100 Hz
f3 = 200 Hz
g1 = 1
g2 = 0.2
g3 = 0.1
fm = 25200/720 = 35 Hz (rotation speed)
fn1 = 35 * 6 = 210 Hz (center frequency of notch filter 51)
fn2 = 35 * 12 = 420 Hz (center frequency of notch filter 52)
Gn1 = 10, Qn2 = 10

Therefore, the gain at the center frequency of each filter is
Gn1 = 0.1
Gn2 = 0.1
Is calculated.

  In this case, both fn1 and fn2 are sufficiently higher than f1 (near the control band) and the attenuation factor is as high as 0.1, but the phase delay at f1 is small. Therefore, there is no problem because a margin for the control phase is ensured.

  Incidentally, the phase delay of the filter means 11 at 60 Hz under this condition is as small as about 7 degrees, and there is no problem in control stability.

  In FIG. 1, the filter unit 11 is arranged downstream of the speed control unit 10, but is not limited to this arrangement, and is downstream of the speed deviation calculation (verr = ftgt + xctl-fdet). I just need it.

  In order to directly reduce the periodic deviation of the position detection signal, not only a filter is inserted into the speed detection output fdet of the frequency detection means 7, but also the position detection output xdet of the counting means 8 is inserted. There must be. This is because both the positions fdet and xdet are used in the calculations of the position control means 9 and the speed control means 10. For this purpose, two filter means are required, which increases the cost. On the other hand, by inserting the filter means 11 after the speed deviation calculation, the filter means 11 can reduce the periodic detection deviation of the position and the speed with only one system, so that a low-cost device can be obtained.

  Further, as an optimal configuration, it is preferable to insert the filter means 11 immediately after the speed deviation calculation and before the proportional-integral control calculation. This is because, when an inexpensive fixed decimal arithmetic unit is used, when multiplication is performed, the output of the fraction increases to a long word length (bit length), but multiplication is performed before proportional integral control calculation. This is because the word length can be shortened because there are few or fewer times. The shorter the word length, the lower cost processing means can be used for the filter operation.

  Further, as described with reference to FIGS. 4 and 7, the filter means 11 is composed of two filters (notch filters 51 and 52), and the attenuation rate change characteristic thereof is an attenuation rate calculating means (first attenuation rate). The calculation means 45 and the second attenuation factor calculation means 46) are calculated by the same algorithm (FIG. 5).

  In the first embodiment, f1 described with reference to FIGS. 5 and 6 corresponds to the first predetermined frequency described in the claims, and f2 corresponds to the second predetermined frequency described in the claims. Further, f2 and f2 correspond to a plurality of predetermined frequencies described in the claims. The filter unit 11 corresponds to the control gain setting unit described in the claims.

(Embodiment 2)
Next, a second embodiment of the motor control device will be described with reference to FIG. The motor control device 1A according to the second embodiment is the same as the motor control device 1 according to the first embodiment except that the selection unit 60 is added. Therefore, in FIG. 12 and the following description, the same components as those in FIG.

  The selection unit 60 selects a control output fctl that does not pass through the filter unit 11 or a control output ctl0 that passes through the filter unit 11 according to the speed command ftgt, and outputs a control amount ctl to the drive unit 20.

  The above selection will be described in more detail. The control output fctl that does not pass through the filter unit 11 is selected when ftgt, that is, the value corresponding to the motor rotation speed is equal to or smaller than the predetermined value, and the control that passes through the filter unit 11 when ftgt is equal to or larger than the predetermined value. Select output ctl0.

  If the gain (attenuation rate) at the center frequency is multiplied by 1.0, the filter means 11 is expected to pass the entire frequency range and be the same as when no filter is inserted. Therefore, in the setting example of the gains Gn1 and Gn2 of the filter coefficient calculation means 12, the gain is 1.0 at low frequencies.

  However, when the filter means 11 is implemented with an inexpensive fixed decimal digital filter, if the word length of the coefficient is not sufficiently long, the center frequency setting is generally low (which is considerably lower than the sampling frequency). It is known that accuracy will deteriorate. This also affects the gain, and at a very low center frequency setting, even if the gain is set to 1.0, the gain may be completely different or an unintended frequency characteristic may be obtained.

  Therefore, when the motor rotational speed is low, that is, when the center frequency setting value of the filter means 11 is very low, the filter itself is not allowed to pass, so that the control can be performed without unintended filter characteristics. It can be made unstable.

  Note that the motor control device 1B according to the second embodiment can be modified in the same manner as the motor control device 1 described in the first embodiment.

(Embodiment 3)
Next, a third embodiment of the motor control device will be described with reference to FIG. The motor control device 1B according to the third embodiment uses a target generation unit 70 instead of the frequency detection unit 5 and the counting unit 6 in the motor control device 1 according to the first embodiment shown in FIG. Except this point, the motor control device 1 is the same as that of the first embodiment. Therefore, in FIG. 13 and the following description, the same components as those in FIG.

  The target generation means 70 outputs a target speed command ftgt and a target position command xtgt based on a predetermined target trajectory. For example, when the motor is rotated at a constant target speed, ftgt is a constant value, and xtgt may be a value corresponding to the integral value of ftgt. When the motor is accelerated at a constant acceleration, ftgt is a value that increases linearly, and xtgt may be a value corresponding to the integral value of ftgt. Alternatively, the trajectory of the target position with respect to time may be set in advance to be xtgt, and ftgt may be a value corresponding to the differential value of xtgt.

  In this way, the interface can be omitted by generating the target speed and position command inside the control unit 4 instead of the clock input from the outside, thereby reducing the cost and higher flexibility than the clock input. A target trajectory can be generated, and various controls are possible.

  Note that the motor control device 1B according to the third embodiment can be modified in the same manner as the motor control device 1 described in the first embodiment.

(Embodiment 4)
Next, a fourth embodiment of the motor control device will be described with reference to FIG. The motor control device 1C of the fourth embodiment is the same as the motor control device 1 of the first embodiment except that the position control means 9 is omitted and the signal fctl is output only by the speed control means 10. Therefore, in FIG. 14 and the following description, the same components as those in FIG. 1 are denoted by the same reference numerals, and the description thereof is omitted.

The speed deviation calculation of the speed control means 10 is
verr = ftgt-fdet
It's okay.

  With this configuration, it is not necessary to accurately control the rotational position of the motor. This configuration is sufficient when only the rotational speed is important. Also in this configuration, the filter means 11 and the filter coefficient calculation means 12 can reduce rotation unevenness due to detection deviation components included in the position detection signals A and B.

  Further, in this configuration, the CLK input and the frequency detection means 5 are deleted, and replaced with a speed command generation function by the target generation means 70 in the control unit 4B of the motor control device 1B of the third embodiment shown in FIG. May be.

(Image processing device)
FIG. 15 is a diagram illustrating the configuration of an example of an image processing apparatus using any one of the motor control apparatuses 1 to 1C.

  First, a schematic configuration of the image processing apparatus 100 illustrated in FIG. 15 will be described. The image forming apparatus 100 includes a scanner unit 111, photoconductor units 112a to 112d, a fixing unit 113, an intermediate transfer belt 114, a secondary transfer roller 115, a registration roller 116, a paper feed roller 117, and paper conveyance. A roller 118, transfer paper 119, a paper feed unit 120, a repulsive roller 121, a paper discharge unit 122, and an intermediate transfer scale detection sensor 123 are provided.

  The scanner unit 111 reads an image of a document placed on the upper surface of the document table. Each of the photosensitive units 112a to 112d corresponds to four colors of YCMK, and includes a drum-shaped photosensitive drum as a latent image carrier, a photosensitive member cleaning roller, and the like. Hereinafter, when the color is not specified, the photoconductor unit 112 may be simply referred to.

  The fixing unit 113 fixes the transferred toner image on the transfer paper 119. The intermediate transfer belt 114 superimposes and transfers the YCMK color toner images formed by the photoconductor units 112 a to 112 d onto the transfer paper 119. The intermediate transfer belt 114 is an image forming medium on which an image is formed by the photoreceptor units 112a to 112d. The intermediate transfer belt 114 is driven by a driving roller 130. The driving roller 130 is driven by a motor 152 at a reduced rotational speed via gears 155 and 56 connected with different numbers of teeth.

  The secondary transfer roller 115 transfers the image on the intermediate transfer belt 114 onto the transfer paper 119. The registration roller 116 performs skew correction of the transfer paper 119 and transfer paper transfer. The paper feed roller 117 sends the transfer paper 119 from the paper feed unit 120 to the transport unit. The paper transport roller 118 transports the transfer paper 119 sent from the paper feed roller 117 to the registration roller 116.

  The paper feed unit 120 loads the transfer paper 119. The repulsive roller 121 is disposed at a portion facing the secondary transfer roller 115, and generates and maintains a nip between the intermediate transfer belt 114 and the secondary transfer roller 115. The paper discharge unit 122 discharges the transfer paper on which the image has been transferred and fixed. The intermediate transfer scale detection sensor 123 is an encoder for measuring the conveyance speed of the intermediate transfer belt 114, detects a scale formed on the intermediate transfer belt 114, and generates a pulse output.

  In the image processing apparatus 100 configured as described above, various rollers are used. By using the motor control device according to any one of the first to fourth embodiments as a motor for driving them, image transfer of the apparatus is performed. Accuracy can be improved. For example, by using the motor control device according to any one of the first to fourth embodiments to control the motor 152 that determines the conveyance accuracy of the transfer belt 114 that is directly related to the transfer accuracy, the image transfer accuracy of the device is improved. Can do.

  Further, any one of Embodiments 1 to 4 is used to control a motor that drives a conveyance roller provided in a conveyance path for conveying the transfer paper 119 to a position where an image is transferred from the intermediate transfer belt 114 to the transfer paper 119. Using a motor control device is also effective for improving the accuracy of image transfer of the device. An example of a transport mechanism provided in this transport path is shown in FIG.

  FIG. 16 is a schematic diagram mainly showing a schematic configuration of the transport mechanism 720. The transport mechanism 200 mainly has two transport rollers 201 and 202 on the transport path as transport members. Here, the transport roller 202 is disposed on the upstream side in the transport direction of the transfer paper 119, and the transport roller 201 is disposed on the downstream side in the transport direction of the transfer paper 119. The conveyance roller 201 is a measurement target roller when the sensor 300 detects a sign of occurrence of an abnormality such as a slip of the transfer paper 119.

  The motor 203 rotationally drives the transport roller 201. The motor 204 drives the conveyance roller 202 to rotate. By using the motor control device according to any one of the first to fourth embodiments to control the drive motor 203 of the conveyance roller 201 that is a measurement target, the transfer delay of the transfer paper 119 is prevented and the image transfer accuracy of the device is reduced. Can be improved.

  A program executed by the motor control device of the present embodiment is a file in an installable format or an executable format, and is a computer such as a CD-ROM, a flexible disk (FD), a CD-R, a DVD (Digital Versatile Disk). It is recorded on a readable recording medium and provided.

  Further, the program executed by the motor control apparatus of the present embodiment may be stored on a computer connected to a network such as the Internet and provided by being downloaded via the network. Further, the information processing program executed by the information processing apparatus of the present embodiment may be provided or distributed via a network such as the Internet.

  In addition, the information processing program of the present embodiment may be provided by being incorporated in advance in a ROM or the like.

DESCRIPTION OF SYMBOLS 1 Motor control apparatus 3 Motor drive part 4 Control part 11 Filter means (control gain setting part)

Japanese Patent Application Laid-Open No. 2015-019563 JP, 2006-178860, A

Claims (8)

A motor that converts a magnetic pole phase signal representing a magnetic pole phase of the motor and outputs a rotational position detection signal having a high resolution with respect to the magnetic pole phase signal, the signal representing a rotation amount and a rotation direction of the output shaft of the motor A motor control device for controlling a drive unit,
Based on the rotation position detection signal and the target signal, the motor drive unit includes a control unit that controls power to be supplied to the motor,
The controller is
In accordance with the number of poles representing the number of magnetic poles of the motor and the number of rotations of the motor, a control gain setting unit that sets a control gain so that the response of a specific frequency is reduced,
The control gain setting unit gradually increases or decreases the amount of decrease in responsiveness according to the specific frequency.
The motor control apparatus characterized by the above-mentioned. The control gain setting unit does not decrease the responsiveness below the first predetermined frequency, and changes the decrease in the responsiveness in proportion to the rotation speed from the first predetermined frequency to the second predetermined frequency. thing,
The motor control device according to claim 1. The control gain setting unit is configured to change the amount of decrease different in a plurality of predetermined frequencies at a frequency higher than the first predetermined frequency, and to complement and change the amount of decrease at a frequency in between.
The motor control device according to claim 2. The control gain setting unit includes filter means for reducing responsiveness to each of a plurality of specific frequencies,
The amount of decrease in responsiveness at the plurality of specific frequencies is changed by the same function with respect to the frequency,
The motor control device according to claim 1, wherein: The control gain setting unit is inserted in a stage subsequent to a deviation means for obtaining a deviation between the rotational position detection signal and the target signal;
The motor control device according to claim 1, wherein: The controller is configured not to use the filter means when the rotational speed of the motor is a predetermined value or less;
The motor control device according to claim 1, wherein: A motor that converts a magnetic pole phase signal representing a magnetic pole phase of the motor and outputs a rotational position detection signal having a high resolution with respect to the magnetic pole phase signal, the signal representing a rotation amount and a rotation direction of the output shaft of the motor A motor control method for controlling a drive unit,
Based on the rotational position detection signal and the target signal, the motor drive unit controls the power to be supplied to the motor.
In accordance with the number of poles representing the number of magnetic poles of the motor and the number of rotations of the motor, a control gain is set so that the response of a specific frequency is reduced,
The amount of decrease in the response is gradually increased or decreased according to the specific frequency.
The motor control method characterized by the above-mentioned. A plurality of rollers for conveying a sheet to which an image is transferred; one or more motors for driving the plurality of rollers; and at least one motor control device for controlling the driving of the motors;
At least one of the motor control devices is the motor control device according to any one of claims 1 to 6.
An image processing apparatus.

Images