JP2013062937A - Drive device for electric motor, and refrigeration cycle device - Google Patents

Abstract

PROBLEM TO BE SOLVED: To prevent a short-circuit current from flowing through opposing SiC-MOSFETs.SOLUTION: A drive device 11 for an electric motor has: a rectification circuit 17 connected with an AC power supply and outputting a DC power; a three-phase inverter circuit 21 converting the DC power of the rectification circuit 17 into a three-phase AC power to drive the electric motor; and a control circuit 25 switching an energization phase of the three-phase inverter circuit 21 and supplying a PWM signal for controlling drive of the electric motor 23 to the three-phase inverter circuit 21. The three-phase inverter circuit 21 has an upper arm and a lower arm each having a parasitic diode and comprising an Si-MOSFET element, for each of the three phases. The PWM signal is set to have a dead time period between an ON period for driving the upper arm and an ON period for driving the lower arm so that both ON periods are not overlapped with each other, for each phase.

Description

  The present invention relates to a motor drive device that drives a motor such as a DC brushless motor using a three-phase bridge inverter circuit, and a refrigeration cycle apparatus.

  In a motor drive device that drives a motor such as a DC brushless motor using a three-phase bridge inverter circuit, the three-phase bridge inverter circuit includes a semiconductor switching element. As the semiconductor switch element, an element made of Si (silicon) material is generally used. In applications where the voltage is relatively high, an IGBT (Insulate Gate Bipola Transisitor) structure is used as a semiconductor switching element.

  In recent years, in order to reduce the loss of semiconductor switching elements, wide band gap semiconductors such as SiC (silicon carbide) or GaN (gallium nitride) have been developed as materials exceeding the theoretical limit of silicon. Wide band gap semiconductors have excellent characteristics such as low loss, high speed, and high temperature operation, and research on MOSFET (Metal Oxide Semiconductor Field Effect Transistor) structure (hereinafter referred to as “SiC-MOSFET”) has been conducted. Progressing.

  For example, Patent Document 1 discloses that an “inverter circuit has six switching elements. The switching element is composed of a SiC-MOSFET that is a unipolar element using a wide bandgap semiconductor. The parasitic diode of the MOSFET is used as a freewheeling diode, and the inverter circuit performs synchronous rectification to turn on the SiC MOSFET at a predetermined timing when a reverse current flows through the parasitic diode of the SiC MOSFET. The effect is described.

JP 2011-36020 A (Claim 2 etc.)

  However, in the inverter circuit according to Patent Literature 1, the SiC-MOSFET is turned on during the period in which the reverse current flows even in the process in which the opposite SiC-MOSFET is turned on and the reverse current flowing through the parasitic diode ends and decays. Synchronous rectification is performed so as to turn on. For this reason, the subject that the short circuit current which penetrates SiC-MOSFET of an ON state will flow occurred.

  In the first place, in the inverter circuit according to Patent Document 1, the process in which the opposite SiC-MOSFET is turned on and the reverse current flowing through the parasitic diode ends and decays is considered in the same process. This is because the reverse recovery current inevitably flows when transitioning from to the cut-off state. When a reverse recovery current flows through the parasitic diode in the conductive state, the parasitic diode transitions to the cutoff state. For this reason, the current flowing through the SiC-MOSFET that is turned on by synchronous rectification is a short-circuit current that does not contribute to the reverse recovery of the parasitic diode, and is merely a useless current that causes power loss.

  The present invention has been made in view of the above circumstances, and in the process in which the opposite SiC-MOSFET is turned on and the reverse current flowing through the parasitic diode ends and decays, a short-circuit current that does not contribute to reverse recovery of the parasitic diode is generated. The purpose is to prevent it from flowing.

  An electric motor drive device according to the present invention includes a rectifier circuit that is connected to an AC power source and outputs DC power, a three-phase inverter circuit that drives the motor by converting DC power of the rectifier circuit into three-phase AC power, A control circuit for switching the energized phase of the three-phase inverter circuit and supplying a PWM signal for controlling the driving of the electric motor to the three-phase inverter circuit, the three-phase inverter circuit having a parasitic diode An upper arm and a lower arm made of MOSFET elements using SiC or GaN are provided for each of the three phases, and the PWM signal has an ON period for driving the upper arm and the lower arm for each phase. The main feature is that the on periods for driving the arms are set with a dead time period so as not to overlap each other.

  According to the present invention, it is possible to prevent a short-circuit current that does not contribute to reverse recovery of the parasitic diode from flowing in the process in which the opposite SiC-MOSFET is turned on and the reverse current flowing through the parasitic diode is terminated and attenuated.

1 is an overall configuration diagram of an electric motor drive device according to a first embodiment of the present invention. FIGS. 2A to 2D are diagrams for explaining the operation of the PWM signal generation circuit of the motor drive device according to the first embodiment. It is explanatory drawing showing the line voltage of the pseudo sine wave which is an output of the drive device of the electric motor which concerns on 1st Embodiment. It is explanatory drawing showing a mode that the PWM signal in a non-overlapping relationship is produced | generated. FIG. 5A is a diagram for explaining the operation of the electric motor drive device according to the comparative example, and FIG. 5B is a diagram for explaining the operation of the electric motor drive device according to the first embodiment. It is a whole block diagram of the drive device of the electric motor which concerns on 2nd Embodiment of this invention.

Hereinafter, a plurality of embodiments of the present invention will be described in detail with reference to the drawings.
[First Embodiment]
(Whole structure of the electric motor drive device 11 according to the first embodiment)
First, the overall configuration of the electric motor drive device 11 according to the first embodiment will be described with reference to FIG. FIG. 1 is an overall configuration diagram of an electric motor drive device 11 according to a first embodiment of the present invention. The electric motor drive device 11 according to the first embodiment of the present invention will be described with reference to an example in which an electric motor 23 composed of a DC brushless motor is driven by a three-phase bridge inverter circuit 21 using SiC (silicon carbide) -MOSFET. . Further, the electric motor 23 according to the first embodiment will be described below on the assumption that, for example, a compressor (not shown) provided in a refrigerating cycle device of an air conditioner (not shown) is driven.

  As shown in FIG. 1, the motor drive device 11 according to the first embodiment of the present invention includes an AC power supply 13, a reactor 15, a full-wave rectifier circuit 17, a smoothing circuit 19, and a three-phase bridge inverter circuit 21. And a PWM signal generation circuit 25, and is configured to drive the electric motor 23.

  The AC power supply 13 is, for example, a single-phase AC power supply. However, a three-phase AC power source may be adopted as the AC power source 13. A reactor 15 is connected to one side terminal of the AC power supply 13. The reactor 15 plays a role of improving the power factor of the AC power supply 13.

  As shown in FIG. 1, the full-wave rectifier circuit 17 includes first to fourth rectifier diodes D1 to D4 that are bridge-connected to each other. The full-wave rectifier circuit 17 has a function of full-wave rectifying the AC voltage waveform of the AC power supply 13 using the first to fourth rectifier diodes D1 to D4. As shown in FIG. 1, the first to fourth rectifier diodes D <b> 1 to D <b> 4 are connected to the positive DC bus PL on the anode side and to the negative DC bus NL on the cathode side, respectively. Between the first connection point Nd1 located between the first and second rectifier diodes D1 and D2 and the second connection point Nd2 located between the third and fourth rectifier diodes D3 and D4. The AC power supply 13 is connected via the reactor 15.

  The smoothing circuit 19 has a smoothing capacitor C1. The smoothing circuit 19 has a function of smoothing the voltage that has been full-wave rectified by the full-wave rectification circuit 17. As shown in FIG. 1, the smoothing capacitor C1 having a pair of terminals has one terminal connected to the positive DC bus PL and the other terminal connected to the negative DC bus NL. A drive voltage Ed (see FIG. 2D) for driving the electric motor 23 is generated between both terminals of the smoothing capacitor C1.

  As shown in FIG. 1, the three-phase bridge inverter circuit 21 functioning as the “three-phase inverter circuit” of the present invention has first to sixth switching elements Sup, Sun, Svp, Svn, Swp, and Swn. As the first to sixth switching elements Sup, Sun, Svp, Svn, Swp, and Swn, for example, wide band gap semiconductors such as SiC (silicon carbide) -MOSFET can be preferably used. The three-phase bridge inverter circuit 21 converts the DC power rectified by the full-wave rectifier circuit 17 and smoothed by the smoothing circuit 19 into pseudo three-phase AC power of u-phase, v-phase, and w-phase. This has a function of supplying the pseudo three-phase AC power to the electric motor 23.

  The first and second switching elements Sup and Sun are connected in series via a third connection point Nd3. First and second parasitic diodes Dup and Dun are connected in antiparallel to the first and second switching elements Sup and Sun, respectively. The third connection point Nd3 is connected to the u-phase power line of the electric motor 23. In the following description, the first switching element Sup may be referred to as a first upper arm UA1, and the second switching element Sun may be referred to as a first lower arm LA1.

  The third and fourth switching elements Svp and Svn are connected in series via the fourth connection point Nd4. Third and fourth parasitic diodes Dvp and Dvn are connected in antiparallel to the third and fourth switching elements Svp and Svn, respectively. The fourth connection point Nd4 is connected to the v-phase power line of the electric motor 23. In the following description, the third switching element Svp may be referred to as a second upper arm UA2, and the fourth switching element Svn may be referred to as a second lower arm LA2.

  The fifth and sixth switching elements Swp and Swn are connected in series via the fifth connection point Nd5. Fifth and sixth parasitic diodes Dwp and Dwn are connected in reverse parallel to the fifth and sixth switching elements Swp and Swn, respectively. The fifth connection point Nd5 is connected to the w-phase power line of the electric motor 23. In the following description, the fifth switching element Swp may be referred to as a third upper arm UA3, and the sixth switching element Swn may be referred to as a third lower arm LA3.

  A series connection circuit of first and second switching elements Sup, Sun, a series connection circuit of third and fourth switching elements Svp, Svn, and a series connection circuit of fifth and sixth switching elements Swp, Swn Each is connected in parallel with each other between a positive DC bus PL and a negative DC bus NL.

  The PWM signal generation circuit 25 functioning as the “control circuit” of the present invention is connected to the gates of the first to sixth switching elements Sup, Sun, Svp, Svn, Swp, and Swn. The individual switching operations of the first to sixth switching elements Sup, Sun, Svp, Svn, Swp, and Swn are collectively controlled by the PWM signal generation circuit 25.

  In other words, the PWM signal generating circuit 25 is based on a carrier signal composed of a triangular wave having a predetermined frequency, which is a reference when performing inverter control, and a voltage command signal for each phase (u phase, v phase, w phase). It has a function of switching the energized phase of the bridge inverter circuit 21 and generating a PWM (Pulse Width Modulation) signal for controlling the drive of the electric motor 23 and supplying the PWM signal thus generated to the three-phase bridge inverter circuit 21.

  The PWM signal generated in the PWM signal generation circuit 25 is the ON period for driving the first to third upper arms UA1 to UA3 and the first to third for each phase (u phase, v phase, w phase). The dead time period DT (see FIG. 3B) is set so that the ON periods for driving the lower arms LA1 to LA3 do not overlap each other. Such a relationship may be referred to as a non-overlapping relationship.

(Operation of PWM signal generation circuit 25 of electric motor drive device 11 according to the first embodiment)
Next, the operation of the PWM signal generation circuit 25 of the motor drive device 11 according to the first embodiment is exemplified by the u-phase target voltage characteristic, the v-phase target voltage characteristic, and the uv line target voltage characteristic. A description will be given with reference to FIG. FIG. 2A is an explanatory diagram showing a time-series relationship between a carrier signal waveform serving as a reference when performing inverter control and a voltage command signal waveform of each phase, and FIG. FIG. 2C is an explanatory diagram illustrating the target voltage characteristics of the v phase, and FIG. 2D is a target voltage characteristic between the uv lines between the u phase and the v phase. It is explanatory drawing showing.

  The PWM signal generation circuit 25 calculates the target voltage of each phase based on the carrier signal and the voltage command signal of each phase (u phase, v phase, w phase) and calculates the line-to-line target voltage between the phases. It works to ask for.

  Specifically, the PWM signal generation circuit 25 uses the triangular wave carrier signal waveform as a reference based on the carrier signal and u phase voltage command signal shown in FIG. While the waveform portion is drawn as a positive voltage half of the drive voltage Ed, the u-phase target voltage in which the u-phase voltage command signal waveform portion below it is drawn as a negative voltage half the drive voltage Ed is obtained by calculation. The u-phase target voltage thus obtained is shown in FIG.

  Further, the PWM signal generation circuit 25 uses the triangular carrier signal waveform as a reference based on the carrier signal and the v phase voltage command signal shown in FIG. Is calculated as a positive voltage that is half of the drive voltage Ed, and a v-phase target voltage that is drawn as a negative voltage that is half of the drive voltage Ed is a portion of the v-phase voltage command signal waveform that is lower than that. FIG. 2C shows the v-phase target voltage thus obtained.

Then, the PWM signal generation circuit 25 generates a potential difference between the u-phase target voltage and the v-phase target voltage based on the u-phase target voltage shown in FIG. 2B and the v-phase target voltage shown in FIG. Is a drive voltage Ed, and a portion where the u-phase target voltage is positive and the v-phase target voltage is negative is drawn as a positive drive voltage (Ed), and both the u-phase target voltage and the v-phase target voltage are common. The voltage difference between the u-phase target voltage and the v-phase target voltage is the drive voltage Ed, the u-phase target voltage is negative, and the v-phase target voltage is positive. The target voltage between the uv lines in which the part is drawn as a negative drive voltage (-Ed) is obtained by calculation. The target voltage between uv lines thus obtained is shown in FIG.
Note that the w-phase target voltage, the u-w line target voltage, and the v-w line target voltage can be obtained by calculation according to the above description, and thus the description and illustration thereof are omitted.

  The PWM signal generation circuit 25 switches the energized phase of the three-phase bridge inverter circuit 21 and controls the drive of the electric motor 23 so as to realize the line-to-line target voltage obtained as described above. A signal is generated, and the PWM signal thus generated is operated to be supplied to the three-phase bridge inverter circuit 21.

  Next, a procedure for generating a PWM signal in a non-overlapping relationship will be described with reference to FIGS. 3 and 4A to 4G. FIG. 3 is an explanatory diagram illustrating a line voltage of a pseudo sine wave output from the motor drive device 11 according to the first embodiment. FIG. 4A is an explanatory diagram showing a signal waveform of a double triangular wave used for generating a PWM signal. FIG. 4B is an explanatory diagram showing how the PWM signal related to the first upper arm UA1 is generated, and FIG. 4C shows how the PWM signal related to the first lower arm LA1 is generated. FIG. 4D is a diagram illustrating how a PWM signal related to the second upper arm UA2 is generated, and FIG. 4E illustrates a PWM signal related to the second lower arm LA2. FIG. 4 (f) is an explanatory diagram showing how the PWM signal related to the third upper arm UA3 is generated, and FIG. 4 (g) is related to the third lower arm LA3. It is explanatory drawing showing a mode that a PWM signal is produced | generated.

  The three-phase bridge inverter circuit 21 of the motor drive device 11 according to the first embodiment applies the pseudo sine wave line voltage shown in FIG. 3 to the motor 23 to drive the motor 23. For that purpose, the comparison values 1 to 3 shown in FIG. 4A are set appropriately changing with time so that the output of the three-phase bridge inverter circuit 21 traces the line voltage waveform of the pseudo sine wave. It is necessary to do. This setting procedure is called a modulation method.

  In order to generate a non-overlapping PWM signal, the PWM signal generation circuit 25 includes a comparison value generation unit 27 that generates comparison values 1 to 3 and a pair of counters 1 and 2 as shown in FIG. And comprising. The comparison values 1 to 3 and the pair of counters 1 and 2 play an important role in generating a PWM signal in a non-overlapping relationship.

  The comparison values 1 to 3 are variable values that change their values over time in order to reproduce the line voltage waveform of each phase (u phase, v phase, w phase). The comparison value 1 corresponds to the u phase, the comparison value 2 corresponds to the v phase, and the comparison value 3 corresponds to the w phase. The comparison value generation unit 27 generates comparison values 1 to 3 that are variable values that change over time with reference to the line voltage waveform of each phase (u phase, v phase, w phase).

The counter 1 has a function of creating a timing for defining an ON period during which the first to third upper arms UA1 to UA3 are driven. In FIG. 4A, a triangular waveform drawn by the count value of the counter 1 that rises and falls linearly is represented by a one-dot chain line. For example, when the current count value coincides with the comparison value 1 during the count-up of the counter 1 (increase process of the triangular waveform), the coincidence timing is set as the start period of the ON period related to the first upper arm UA1. (See FIGS. 4A and 4B). Further, when the current count value coincides with the comparison value 1 during the countdown of the counter 1 (descent process of the triangular waveform), the coincidence timing is set as the end of the ON period related to the first upper arm UA1. (See FIGS. 4A and 4B).
The start period and the end period of the on periods related to the second to third upper arms UA2 to UA3 are also set in accordance with the above (see FIGS. 4A, 4D, and 4F).

  The counter 2 has a function of creating a timing for defining an off period during which the first to third lower arms LA1 to LA3 are driven (however, if the logic is reversed, it is also an on period). In FIG. 4A, a triangular waveform drawn by the count value of the counter 2 that rises and falls linearly is represented by a solid line. The triangular waveform related to the counter 1 and the triangular waveform related to the counter 2 have a similar relationship, and the time related to the vertex is set to be the same. The size of the triangular waveform related to the counter 2 is set slightly larger than that of the triangular waveform related to the counter 1. This difference in size creates a dead time period DT.

For example, when the current count value coincides with the comparison value 1 during the count-up of the counter 2 (increase process of the triangular waveform), the coincidence timing is set as the start time of the off period related to the first lower arm LA1. (See FIGS. 4A and 4C). If the current count value coincides with the comparison value 1 during the countdown of the counter 2 (the triangular waveform descending process), the coincidence timing is set as the end of the off period related to the first lower arm LA1. (See FIGS. 4A and 4C).
The start period and the end period of the off period related to the second to third lower arms LA2 to LA3 are also set according to the above (see FIGS. 4A, 4E, and 4G).

  As described above, the PWM signal generation circuit 25 compares the current values of the counters 1 and 2 that sequentially and continuously count up and count down with the three comparison values 1 to 3 while comparing the three values. A PWM signal in a non-overlapping relationship for driving the bridge inverter circuit 21 is generated, and the PWM signal thus generated is supplied to the three-phase bridge inverter circuit 21.

As shown in FIGS. 4 (b) and 4 (c), the drive signal for the u phase generated by the above procedure is such that when the first upper arm UA1 is on, the first lower arm LA1 is off. Become. When the first upper arm UA1 is off, the first lower arm LA1 is on. In short, the drive signal for the u phase is a complementary PWM signal.
Note that a predetermined dead time period is provided between the ON period of the first upper arm UA1 and the OFF period of the first lower arm LA1 as will be described in detail later. .

Further, as shown in FIGS. 4D and 4E, when the second upper arm UA2 is turned on, the second lower arm LA2 is turned off in the drive signal related to the v phase. In short, the drive signal for the v phase is a complementary PWM signal.
Note that a predetermined dead time period is provided between the ON period of the second upper arm UA2 and the OFF period of the second lower arm LA2 as will be described in detail later. .

As shown in FIGS. 4 (f) and 4 (g), when the third upper arm UA3 is on, the third lower arm LA3 is turned off. In short, the drive signal for the w phase is a complementary PWM signal.
Note that a predetermined dead time period is provided between the ON period of the third upper arm UA3 and the OFF period of the third lower arm LA3, as will be described in detail later. .

  Of the first to third upper arms UA1 to UA3 and the first to third lower arms LA1 to LA3, the upper and lower arms (first upper arm UA1 and first lower arm LA1, The output voltage characteristics of the three-phase bridge inverter circuit 21 are determined by the ON ratio between the two upper arms UA2 and the second lower arm LA2, and the third upper arm UA3 and the third lower arm LA3).

(Operation of the electric motor drive device 11 according to the first embodiment)
Next, the operation of the electric motor drive device 11 according to the first embodiment will be described with reference to FIG. 5A and FIG. 5B while comparing with the operation of the electric motor drive device according to the comparative example. . FIG. 5A is a diagram for explaining the operation of the electric motor drive device according to the comparative example, and FIG. 5B is a diagram for explaining the operation of the electric motor drive device 11 according to the first embodiment.
In the first embodiment and the comparative example, each phase (u phase, v phase, w phase) performs the same operation. Therefore, out of each phase (u phase, v phase, w phase), the description of the operation of the u phase is replaced with the description of the operation of the other phases (v phase, w phase).

The difference between the motor drive device 11 according to the first embodiment and the motor drive device according to the comparative example is that the PWM signal generation circuit 25 of the motor drive device 11 according to the first embodiment has a PWM signal. For each phase (u phase, v phase, w phase) to drive the first to third upper arms UA1 to UA3 and to drive the first to third lower arms LA1 to LA3 On the other hand, in the PWM signal generation circuit of the motor drive device according to the comparative example, the PWM signal is set so as to sandwich the dead time period DT (see FIG. 5B) so that the ON periods do not overlap each other. The point is set without considering the dead time period DT as in the first embodiment.
The configuration of the three-phase bridge inverter circuit is common between the first embodiment and the comparative example. For this reason, in operation | movement description of a comparative example, suppose that the code | symbol attached | subjected to each component of the three-phase bridge inverter circuit 21 of the drive device 11 of the electric motor which concerns on 1st Embodiment is used.

  First, the operation of the three-phase bridge inverter circuit of the motor drive device according to the comparative example will be described.

  In the motor drive device according to the comparative example, as shown in FIG. 5A, the first upper arm UA1 is turned on, the first lower arm LA1 is turned off, and the first upper arm UA1 is turned off. The period and the ON period of the first lower arm LA1 are set without considering the dead time period DT.

  In the case of the comparative example, when it is assumed that the first upper arm UA1 is turned on and the first lower arm LA1 is turned off, the current is the smoothing capacitor C1 → first upper arm UA1 → third connection point. Nd3 → the motor 23 → the second and third lower arms LA2 and LA3 flow.

  In this state, when the first upper arm UA1 is turned off and the first lower arm LA1 is turned on (see times t1 and t3 in FIG. 5A), current flows through the first upper arm UA1. Can not be. Since the current in this state flows through an inductance (not shown) built in the electric motor 23, it has a property of continuously flowing. For this reason, when the first upper arm UA1 is turned off, the current from the smoothing capacitor C1 turns to the first lower arm LA1. The first lower arm LA1 has an internal circuit portion of the second switching element Sun and a parasitic diode Dun portion. For this reason, the current from the smoothing capacitor C1 flows in a distributed manner according to the ease of flow between these parts.

  Next, when the first upper arm UA1 is turned on and the first lower arm LA1 is turned off again (see time t2 in FIG. 5A), the smoothing capacitor C1 is switched to the first lower arm LA1. This time, the current that has flowed turns to the first upper arm UA1. At this time, the current change is not instantaneous, but requires a finite transition time. For this reason, the first upper arm UA1 and the first lower arm LA1 are turned on at the same time for a very short time. As a result, a short-circuit current that passes through the first upper arm UA1 and the first lower arm LA1 flows.

  On the other hand, in the portion of the parasitic diode Dun in the first lower arm LA1, the direction of the applied voltage changes from the forward direction to the reverse direction. In the parasitic diode Dun, when a reverse voltage is applied in a state where a forward current is flowing, a current flows in the reverse direction for a very short time during the transition from the conduction state to the cutoff state. This is called reverse recovery current.

  The reverse recovery current that flows through the parasitic diode Dun is unavoidable due to the nature of the diode, but the short-circuit current that passes through the first upper arm UA1 and the first lower arm LA1 only causes loss. Not too much. In addition, the flow of the short-circuit current may damage the first upper arm UA1 or the first lower arm LA1.

  In the above description of the operation according to the comparative example, the case where the first upper arm UA1 is turned on and the first lower arm LA1 is turned off again is illustrated. Even in the case where the first upper arm UA1 is turned off and the first lower arm LA1 is switched back on, the first upper arm UA1 and the first lower arm LA1 are penetrated in the same manner as described above. Short circuit current may flow.

  In contrast to this comparative example, in the motor drive device 11 according to the first embodiment, as shown in FIG. 5B, the on period of the first upper arm UA1, the off period of the first lower arm LA1, and The off period of the first upper arm UA1 and the on period of the first lower arm LA1 are set with a dead time period DT interposed therebetween so as not to overlap each other.

  In short, the PWM signal that drives the three-phase bridge inverter circuit 21 in the motor drive device 11 according to the first embodiment is a non-overlapping PWM signal as described above. By driving the three-phase bridge inverter circuit 21 using such a non-overlapping PWM signal, a short circuit that penetrates the first upper arm UA1 and the first lower arm LA1 as in the comparative example. A situation in which current flows can be prevented in advance.

  That is, in the electric motor drive device 11 according to the first embodiment, assuming that the first upper arm UA1 is turned on and the first lower arm LA1 is turned off, the current is the smoothing capacitor C1 → first It flows through the path of the upper arm UA1 → the third connection point Nd3 → the electric motor 23 → the second and third lower arms LA2 and LA3. This operation is the same as in the comparative example.

  In this state, the first upper arm UA1 is turned off (see times t11 and t15 in FIG. 5B). In the first embodiment, since the three-phase bridge inverter circuit 21 is driven using a PWM signal having a non-overlapping relationship, the first lower arm LA1 remains off for a predetermined dead time period DT. This operation is different from the comparative example.

  The current that can no longer flow through the first upper arm UA1 flows through the inductance built in the electric motor 23, and therefore has a property of continuing to flow. For this reason, when the first upper arm UA1 is turned off, the current from the smoothing capacitor C1 turns to the first lower arm LA1. In the dead time period DT, the current from the smoothing capacitor C1 flows through the parasitic diode Dun in the first lower arm LA1.

  When the dead time period DT has passed (see time t12 in FIG. 5B), the first lower arm LA1 is turned on. As a result, the current from the smoothing capacitor C1 is distributed and flows in accordance with the degree of ease of flow between the internal circuit portion of the second switching element Sun and the portion of the parasitic diode Dun. In general, the resistance value of the internal circuit portion of the second switching element Sun is much smaller than the forward resistance value of the parasitic diode Dun.

  Therefore, most of the current from the smoothing capacitor C1 flows to the internal circuit portion of the second switching element Sun. Thereby, the loss due to the voltage drop can be reduced as compared with the case where all of the current from the smoothing capacitor C1 flows through the parasitic diode Dun. This loss reduction effect is called a synchronous rectification effect. This operation is the same as in the comparative example.

  Next, an operation of switching to the state where the first upper arm UA1 is turned on and the first lower arm LA1 is turned off will be described. In the first embodiment, since the three-phase bridge inverter circuit 21 is driven using a PWM signal having a non-overlapping relationship, the timing when the first upper arm UA1 is turned on (time t14 in FIG. 5B). Prior to (see), the first lower arm LA1 is turned off first by the dead time period DT (see time t13 in FIG. 5B).

  Since the first lower arm LA1 is off, the current from the smoothing capacitor C1 is commutated from the internal circuit portion of the second switching element Sun to the parasitic diode Dun portion. In this state, the first upper arm UA1 is turned on (see time t14 in FIG. 5B). For this reason, a reverse recovery current flows through the parasitic diode Dun in the first lower arm LA1, and the parasitic diode Dun transitions from the conductive state to the cut-off state.

  Therefore, according to the electric motor drive device 11 according to the first embodiment, it is possible to prevent a situation in which a short-circuit current passing through the first upper arm UA1 and the first lower arm LA1 flows as in the comparative example. Can do. Further, there is no loss due to the flow of the short-circuit current.

In the electric motor drive device 11 according to the first embodiment, the electric motor 23 drives a compressor provided in the refrigeration cycle of the air conditioner.
Therefore, it is possible to provide an air conditioner with excellent energy saving effect in which loss due to short-circuit current distribution is suppressed.

[Second Embodiment]
(Whole structure of the drive device 31 of the electric motor which concerns on 2nd Embodiment)
Next, the overall configuration of the electric motor drive device 31 according to the second embodiment will be described with reference to FIG. FIG. 6 is an overall configuration diagram of an electric motor drive device 31 according to the second embodiment of the present invention.
The electric motor drive device 11 according to the first embodiment and the electric motor drive device 31 according to the second embodiment share most of the components. Therefore, in principle, members having the same function are denoted by the same reference numerals, the duplicate description thereof is omitted, and the description will be focused on the difference between the two.

There are two major differences between the first embodiment and the second embodiment.
The first difference is the internal configuration of the “rectifier circuit”. That is, the “full wave rectifier circuit 17” according to the first embodiment includes first to fourth rectifier diodes D1 to D4 that are bridge-connected to each other as shown in FIG. The structure which carries out full wave rectification of the alternating voltage waveform of the alternating current power supply 13 using 4 rectifier diodes D1-D4 is employ | adopted.

On the other hand, the “synchronous rectifier circuit 33” according to the second embodiment includes the first to fourteenth switching elements Sp1, Sn1, Sp2, and Sn2 connected to each other as shown in FIG. A configuration is employed in which the AC voltage waveform of the AC power supply 13 is full-wave rectified using the 11th to 14th switching elements Sp1, Sn1, Sp2, and Sn2.
The 11th to 14th parasitic diodes Dp1, Dn1, Dp2, Dn2 are connected in reverse parallel to the 11th to 14th switching elements Sp1, Sn1, Sp2, Sn2.

  The second difference is that a “synchronous rectification signal generation circuit 35” is added as shown in FIG. 6 in accordance with the adoption of the “synchronous rectification circuit 33”.

  The synchronous rectification signal generation circuit 35 performs the same operations as the operations of the first to fourth rectifier diodes D1 to D4 bridge-connected to each other, so that the first to fourteenth switching elements Sp1, Sn1, Sp2 are realized. , Sn2 has a function of supplying a drive signal.

  Here, in order to realize the same operation as the operation of the first to fourth rectifier diodes D1 to D4 bridge-connected to each other, the 11th to 14th switching are performed according to the voltage polarity of the AC power supply 13. Of the elements Sp1, Sn1, Sp2, and Sn2, there are two sets of switching elements that should be turned on in a brushed pattern. That is, the set of the eleventh switching element Sp1 and the fourteenth switching element Sn2 and the set of the twelfth switching element Sn1 and the thirteenth switching element Sp2 are the same.

  The synchronous rectification signal generation circuit 35 performs first to fourth rectifications that are bridge-connected to each other by performing a synchronous rectification operation that alternately turns on the above-described pair of switching elements according to the voltage polarity of the AC power supply 13. A full-wave rectification operation similar to the operation of the diodes D1 to D4 can be realized while reducing loss due to conduction loss. As a result, the AC power of the AC power supply 13 is rectified to DC power.

  According to the motor drive device 31 according to the second embodiment, it is possible to perform sinusoidal approximation of the power supply current waveform, power factor improvement, and boost operation of the drive voltage Ed generated at both ends of the smoothing capacitor C1.

  In the synchronous rectification signal generation circuit 35, when the full-wave rectification operation is not performed, a normal bridge rectification circuit is configured by the presence of the 11th to 14th parasitic diodes Dp1, Dn1, Dp2, and Dn2. .

  Further, as a configuration of the synchronous rectification signal generation circuit 35, for example, the twelfth switching element Sn1 is turned on so that the short-circuit current is caused to flow through the reactor 15 so that the AC power supply 13 is short-circuited via the reactor 15. Subsequently, the twelfth switching element Sn1 may be turned off to allow the smoothing capacitor C1 to release energy through the eleventh switching element Sp1.

  At this time, when the synchronous rectification signal generation circuit 35 supplies the drive signal to the eleventh switching element Sp1 to turn it on, the current from the reactor 15 is mainly not the part of the parasitic diode Dp1 in the eleventh switching element Sp1, but mainly. It flows through the internal circuit part. Thereby, the synchronous rectification (loss reduction) effect is acquired.

  Further, by adopting the non-overlapping PWM signals according to the first embodiment as drive signals of the first to fourteenth switching elements Sp1, Sn1, Sp2, and Sn2, the same as in the first embodiment. In addition, it is possible to prevent a short circuit current passing through the switching elements (first to fourteenth switching elements Sp1, Sn1, Sp2, and Sn2) connected in series with each other, and to eliminate a loss caused by the flow of the short circuit current. .

[Other Embodiments]
The plurality of embodiments described above show examples of implementation of the present invention. Therefore, the technical scope of the present invention should not be limitedly interpreted by these. This is because the present invention can be implemented in various forms without departing from the gist or main features thereof.

  For example, in the first and second embodiments, a single-phase AC power supply has been described as an example, but the present invention is not limited to this example. In the present invention, a three-phase AC power source may be used. In this case, the number and arrangement configuration of the diodes of the full-wave rectification circuit 17 and the number and arrangement configuration of the switching elements of the synchronous rectification circuit 33 may be appropriately changed in accordance with the three-phase AC power supply.

DESCRIPTION OF SYMBOLS 11 Motor drive device according to first embodiment 13 AC power source 15 Reactor 17 Full wave rectifier circuit (rectifier circuit)
19 Smoothing circuit 21 Three-phase bridge inverter circuit (Three-phase inverter circuit)
DESCRIPTION OF SYMBOLS 23 Electric motor 25 PWM signal generation circuit 27 Comparison value generation part 31 Electric motor drive device which concerns on 2nd Embodiment 33 Synchronous rectification circuit 35 Synchronous rectification signal generation circuit C1 Smoothing capacitor D1-D4 1st-4th rectifier diode Dup 1st Parasitic diode Dun second parasitic diode Dvp third parasitic diode Dvn fourth parasitic diode Dwp fifth parasitic diode Dwn sixth parasitic diode DT dead time period Ed drive voltage Sup (UA1) first switching element ( First upper arm)
Sun (LA1) Second switching element (first lower arm)
Svp (UA2) Third switching element (second upper arm)
Svn (LA2) Fourth switching element (second lower arm)
Swp (UA3) Fifth switching element (third upper arm)
Swn (LA3) Sixth switching element (third lower arm)
Sp1, Sn1, Sp2, Sn2 11th to 14th switching elements Dp1, Dn1, Dp2, Dn2 11th to 14th parasitic diodes

Claims (4)

A rectifier circuit connected to an AC power source and outputting DC power;
A three-phase inverter circuit for driving the motor by converting the DC power of the rectifier circuit into three-phase AC power;
A control circuit for switching the energized phase of the three-phase inverter circuit and supplying a PWM signal for controlling the driving of the electric motor to the three-phase inverter circuit,
The three-phase inverter circuit includes a parasitic diode, and includes an upper arm and a lower arm made of a MOSFET element using SiC or GaN for each of the three phases.
The PWM signal is set for each phase with a dead time period interposed so that an on period for driving the upper arm and an on period for driving the lower arm do not overlap each other.
An electric motor drive device. The electric motor drive device according to claim 1,
The rectifier circuit has a parasitic diode and is composed of a MOSFET element using SiC or GaN.
An electric motor drive device. The electric motor drive device according to claim 1 or 2,
The electric motor is mounted on an air conditioner,
An electric motor drive device. A compressor driven by an electric motor driven by the drive device according to claims 1 to 3,
A refrigeration cycle apparatus characterized by that.

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