JP2010124614A - Switching power supply unit - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a switching power supply unit including an overcurrent protecting circuit which can surely limit a switching current and an output current within a safe range with respect to various operation states and reduce a loss in current detection resistance. <P>SOLUTION: The switching power supply unit includes a DC voltage generating circuit 22 outputting a DC voltage for controlling the output current, a current detecting resistor R0 detecting the current flowing in a switching element TR1, and a current limit signal generating circuit 20 limiting the switching current by detecting a current detecting voltage. The current limit signal generating circuit 20 includes an output voltage monitoring circuit 20a generating a first DC voltage to be output in response to a change in output voltage based on a pulse voltage generated in an auxiliary winding T1c. The switching power supply unit further includes a second DC voltage generating means R32 to be changed based on the current controlling DC voltage of the DC voltage generating circuit 22, and a transistor TR32 which outputs a current limiting signal limiting the switching current when the added voltage of the first and second DC voltages is compared with the current detecting voltage and the detected current voltage increases over the added voltages. <P>COPYRIGHT: (C)2010,JPO&INPIT

Description

  The present invention relates to a switching power supply device that converts an input voltage into a desired DC voltage by performing on / off control of a switching element using a pulse width modulation signal, and more particularly to a switching power supply device including an overcurrent protection circuit.

  A general switching power supply device is provided with an overcurrent protection circuit that limits the output current supply capability so that an output current exceeding a predetermined value does not flow. This prevents the electronic device from being heated or burnt down due to the excessive output current flowing when the electronic device that is the load connected to the output has a low impedance failure, etc. Its purpose is to prevent its own failure, heat generation and burning.

  Conventionally, as a switching power supply device provided with an overcurrent protection circuit, for example, as disclosed in Patent Document 1, an output voltage of an element current detection circuit that detects a switching current flowing in a switching element, and an overcurrent protection reference voltage source An overcurrent protection circuit that limits the peak value of the switching current by comparing with the overcurrent protection reference voltage generated by the power supply, a second primary winding provided in the transformer, and a second primary winding voltage. Rectifying and smoothing the voltage of the second primary winding generated during the ON period of the switching element to rectify and smooth and output a voltage proportional to the input voltage, and to the output voltage of the input voltage detection unit And a variable signal generator that generates a variable signal that varies the overcurrent protection reference voltage based on the peak value of the switching current when the input voltage increases. To be limited, there is a switching power supply performs an operation to correct the overcurrent protection reference voltage. According to this configuration, it is possible to prevent an increase in the peak value of the switching current due to a delay time of overcurrent protection that occurs remarkably at a high input voltage.

Further, as disclosed in Patent Document 2, the detection resistor inserted in series in the flow path of the load current, the transistor forward-biased by the detection voltage, and the output voltage is lowered when the transistor is turned on. And a control means for performing overcurrent protection by adding at least a part of a forward voltage drop of a diode to which a forward current is supplied irrespective of a load current to a voltage across the detection resistor. There is a DC-DC converter for forward-biasing the transistor. According to this configuration, it is possible to reduce the loss of the detection resistor and to compensate for the fluctuation of the overcurrent protection performance due to the temperature fluctuation of the base-emitter voltage Vbe of the transistor.
JP 2004-343900 A JP-A-10-174427

  However, in the case of the switching power supply device disclosed in Patent Document 1, there is only one means for correcting the overcurrent protection reference voltage from the outside, which is insufficient to ensure the safety of the switching power supply device. . For example, in the case of an overcurrent protection method that limits the peak value of the switching current, normally, the on-time of the switch element is reduced by the overcurrent protection operation and the output voltage is lowered. It becomes easy to be affected by the delay time, and even at a low input voltage, the peak value of the switching current cannot be limited and the safe range may be exceeded. Therefore, in order to always ensure safety even when the input voltage fluctuates, it is often preferable to correct the overcurrent protection reference voltage based on the output voltage information. In addition, if the overcurrent operation continues, the temperature of the internal components rises due to the switching current, which may exceed the allowable range. Therefore, it may be preferable to have a function of correcting the overcurrent reference voltage based on the temperature information of the internal components. However, the switching power supply device of Patent Document 1 corrects the overcurrent protection reference voltage only based on the input voltage, and is not necessarily sufficient to ensure safety against various operating states of the switching power supply device. I can't say that.

  In the switching power supply device of Patent Document 1, an element current detection circuit, an overcurrent protection circuit, and the like are integrated on one chip, but the internal configuration of the element current detection circuit is not described in detail. In a general switching power supply device, in order to detect a switching current with high accuracy, for example, an element current detection circuit is used in which a current detection resistor is inserted in series with a switching element and the element current is converted into a voltage by the current detection resistor. . When the configuration of the overcurrent protection circuit of Patent Document 1 is applied to such a general switching power supply device, in order to detect the element current with high accuracy, a large voltage drop of a predetermined value or more is applied to the current detection resistor. Since it must be generated, there is a problem that the loss of the current detection resistor is increased and the detection resistor is increased in size.

  On the other hand, in the case of the DC-DC converter disclosed in Patent Document 2, while having the effect of reducing the loss of the current detection resistor, which is a problem of the switching power supply device of Patent Document 1, the threshold at which the transistor that lowers the output voltage is turned on In other words, the upper limit of the detection voltage of the current detection resistor only corrects the operating point of the transistor using the temperature variation of the forward voltage of the diode. Therefore, like the switching power supply device disclosed in Patent Document 1, it is not always sufficient to ensure safety against various operating states of the DC-DC converter.

  Further, since the DC-DC converter of Patent Document 2 is a method of detecting a load current that fluctuates relatively slowly as compared with the switching current flowing through the switching element and lowering the output voltage, for example, the peak value of the switching current Even if a steep current stress is applied to the switching element due to a rapid increase in the switching element, it is difficult to detect it at a high speed and with high accuracy, so that the switching element may not be damaged.

  The present invention has been made in view of the above-described background art, and includes means for correcting an overcurrent protection reference voltage based on at least output voltage information, and includes a switching current and various operating states of the switching power supply device. An object of the present invention is to provide a switching power supply device including an overcurrent protection circuit that can reliably limit the output current to a safe range and reduce the loss of the current detection resistor.

The present invention relates to a drive pulse generation circuit that outputs a drive pulse that is pulse-width modulated at a predetermined switching frequency, and is turned on / off by the drive pulse from the drive pulse generation circuit to intermittently connect a DC input voltage to an AC voltage A transformer having an input winding to which the AC voltage is applied, an output winding for outputting a voltage obtained by transforming the AC voltage, and rectifying and smoothing the AC voltage output from the output winding A rectifying / smoothing circuit that generates an output voltage and supplies an output current to a load,
A DC voltage generation circuit that outputs a changeable DC voltage for current control that is a predetermined target value relating to the control of the output current, and a current detection voltage that is inserted into a flow path of a switching current that flows through the switching element. A current detection resistor and a current limit signal generation circuit that outputs a current limit signal for limiting the switching current by detecting the current detection voltage, and the current limit signal generation circuit includes an auxiliary winding provided in the transformer. An output voltage monitor circuit that generates and outputs a first DC voltage corresponding to a change in the output voltage based on a pulse voltage generated in the auxiliary winding, and current control of the DC voltage generation circuit A second DC voltage generating means that is variable based on the DC voltage for use, and the current detection voltage is compared with the addition voltage of the first and second DC voltages and the current detection voltage. A comparator for outputting a current limit signal for limiting the switching current when the current limit signal is larger, and the drive pulse generation circuit outputs a drive pulse for driving the switching element when the current limit signal is output. It is a switching power supply device that operates to stop or narrow the on-duty.

  The output voltage monitor circuit has a positive potential side of the output terminal connected to one end of the current detection resistor, and the second DC voltage generating means and the comparing means have one end connected to the output of the DC voltage generating circuit. A first resistor, a second resistor having one end connected to the other end of the first resistor, a collector terminal connected to the other end of the second resistor, and a base terminal being the first resistor A first NPN transistor connected to the middle point of the resistor and the second resistor, an emitter terminal connected to the reference potential side of the output terminal of the output voltage monitor circuit, and a base terminal to the collector of the first NPN transistor And a second NPN transistor whose emitter terminal is connected to the other end of the current detection resistor and whose collector terminal is an output of the current limit signal generation circuit, and the current detection resistor includes the switch The current detection voltage may be generated in a direction in which the emitter terminal of the second NPN transistor is at a lower potential than the emitter terminal of the first NPN transistor when a current flows through the switching element. .

  Further, the transformer and the rectifying / smoothing circuit are configured to perform single forward type power conversion, and the output voltage monitor circuit includes a parallel circuit of a first monitor resistor and a monitor capacitor, and the parallel circuit. An integrating circuit composed of a second monitoring resistor connected in series, and the input of the integrating circuit is connected to both ends of the auxiliary winding via a rectifying diode, and the rectifying diode is connected to the switching element. The voltage generated during the ON period is connected in the direction of half-wave rectification, and the output of the output voltage monitor circuit may be a voltage generated at the output of the integrating circuit.

  According to the switching power supply device of the present invention, there is provided means for extracting the output voltage information through at least the output voltage monitor circuit, and correcting the threshold value for operating the current control signal generation circuit so as to decrease when the output voltage becomes low. Since the influence of the switching delay time when the output voltage decreases is corrected, the switching power supply device and the load can be reliably protected. Further, the DC voltage generation circuit generates a DC voltage for current control that becomes a predetermined target value related to output current control based on an arbitrary signal input such as input voltage and environmental temperature, and the current control signal generation circuit operates. The operation for correcting the threshold can be performed, and the overcurrent protection characteristic can be optimized for various operation states of the switching power supply device.

  Hereinafter, an embodiment of the switching power supply device of the present invention will be described with reference to FIGS. First, the schematic configuration of the entire circuit will be described based on the circuit diagram shown in FIG. In the switching power supply device 10, a series circuit of an input winding T1a of a transformer T1 and a switching element TR1 is connected in series with a DC input power supply Ein. An output winding T1b is provided on the secondary side of the transformer T1, and a rectifying / smoothing circuit 12 for rectifying and smoothing an AC voltage to generate an output voltage Vout is connected to both ends of the transformer T1, and an output voltage Vout terminal is connected to a load. 14. The output voltage Vout terminal is connected to an error amplification circuit 16 including an inverting amplifier that outputs an output voltage control signal V (vol) obtained by amplifying the difference between the output voltage Vout and a predetermined reference voltage Vref. Further, a drive pulse generation circuit 18 capable of performing pulse width modulation based on the output voltage control voltage V (vol) output from the error amplifier circuit 16 and outputting the drive pulse Vg toward the drive terminal of the switching element TR1 is provided. I have.

  A current detection resistor R0 is inserted between the source terminal of the switching element TR1 and the negative terminal of the input power supply Ein, and a voltage V (R0) corresponding to the switching current Isw flowing through the current detection resistor R0 is generated at both ends thereof. Both ends of the current detection resistor R0 are connected to a current limit signal generation circuit 20 that constitutes an overcurrent protection circuit described later.

  In addition, a DC voltage generation circuit 22 constituting an overcurrent protection circuit is provided with the source terminal of the switching element TR1 as a reference potential. The DC voltage generation circuit 22 generates a current control DC voltage Vc for limiting the switching current Isw and inputs the current control DC voltage Vc to the subsequent current limit signal generation circuit 20. The current control DC voltage Vc can be adjusted based on a variable signal received from outside.

  The current limit signal generation circuit 20 to which the current control DC voltage Vc is input further includes an output voltage monitor circuit 20a. The output voltage monitor circuit 20a outputs an output voltage Vout through an auxiliary winding T1c provided in the transformer T1 and an voltage generated in the auxiliary winding T1c via an rectifying diode D21, an integration circuit composed of a resistor and a capacitor. Is generated and output as a voltage V (C21) which is a first DC voltage.

  In addition to the output voltage monitor circuit 20a, the current limit signal generation circuit 20 includes a circuit composed of a plurality of transistors and resistors, and a voltage V that is a second DC voltage depending on the current control DC voltage Vc. (R32) is generated, and the added voltage of the voltage V (C21) and the voltage V (R32) is compared with the voltage V (R0) of the current detection resistor R0, and the voltage V (R0) exceeds the added voltage. If it is going to be, it has the function to output the current limiting signal V (cur) for limiting a switching current.

  When the current limit signal V (cur) is input, the drive pulse generation circuit 18 operates from pulse width modulation using the voltage control signal V (vol) described above to pulse width modulation using the current limit signal V (cur). The mode is switched and a function of outputting the drive pulse Vg of the switching element TR1 is provided.

  Next, the detailed configuration and operation of each circuit block of the switching power supply device 10 will be described. As shown in FIG. 1, the inverter circuit composed of the switching element TR1, the transformer T1, and the rectifying / smoothing circuit 12 has a primary winding T1a of the transformer T1 and the switching element TR1 connected in series with the input power source Ein. The AC voltage V2 is generated in the secondary winding T1b of the transformer T1 by the on / off operation of the element TR1. The polarity of each winding of the transformer T1 is configured in a well-known single forward system.

  A rectifying / smoothing circuit 12 is connected to the secondary winding T1b of the transformer T1. The rectifying / smoothing circuit 12 includes a forward-side rectifying element TR2 that conducts a pulse current when the switching element TR1 is on, a flywheel-side rectifying element TR3 that is turned on / off complementarily with the forward-side rectifying element TR2, The synchronous rectification driving circuit 12a that drives the rectifying elements TR2 and TR3 in synchronization with the on / off operation of the element TR1 has a rectifying function, and the choke coil Lo and the capacitor Co have a smoothing function. That is, the rectifying / smoothing circuit 12 rectifies and smoothes the voltage induced in the secondary winding T1b, and outputs the output voltage Vout across the capacitor Co. A load 14 is connected to both ends of the capacitor Co, and an output voltage Vout and an output current Iout are supplied.

  As shown in FIG. 1, the error amplifying circuit 16 includes an operational amplifier OP1 whose output voltage analog signal is input to its inverting input terminal, a predetermined reference voltage Vref connected to its non-inverting input, gain adjustment and phase compensation. And an inverting amplifier circuit that outputs an output voltage control signal V (vol). Therefore, the output voltage control signal V (vol) is obtained by amplifying the difference between the output voltage and the reference voltage Vref. When the output voltage becomes higher than the reference voltage Vref, the output voltage control signal V (vol) continuously decreases, and vice versa. Will rise continuously.

  The DC voltage generation circuit 22 can be configured using, for example, a general-purpose digital processor (microcomputer), and here, is sent from a temperature detection means (not shown), an input / output voltage detection means, or the like that detects the temperature of the switching element TR1. When the detection signal crosses a predetermined threshold, such as when the variable signal is received and the detection temperature exceeds a predetermined temperature, an operation of changing the current control DC voltage Vc is performed. When the temperature of the switching element TR1 is detected, the rate of change in the temperature is very slow compared with one cycle of switching of the switching element (for example, about several hundred kHz). High-speed response of the current control DC voltage Vc to the variable signal is not necessary, and can be configured inexpensively using a general-purpose digital processor that operates at a clock frequency as low as about 10 kHz. As shown in FIG. 1, the current detection resistor R0 is inserted in a path through which the switching current Isw flows. In the switching power supply device 10, the switching current Isw is a pulse current that repeats a substantially trapezoidal shape, and the relationship between the peak value Iswp and the output current Iout uses the primary winding number N1 and the secondary winding number N2 of the transformer T1. In this case, the peak value Iswp≈ (N2 / N1) × Iout. That is, the peak value of the current detection voltage V (R0) generated in the current detection resistor R0 is substantially proportional to the output current Iout if the influence of the switching delay time is ignored. As described above, the switching power supply device 10 detects the output current Iout by observing the switching current Isw via the current detection resistor R0. When the output current Iout flows, the current detection voltage V (R0) is generated in a direction in which the side connected to the source terminal of the switching element TR1 becomes a high potential.

  The output voltage monitor circuit 20a included in the current limit signal generation circuit 20 includes a resistor R21 that is a first monitor resistor, a capacitor C21 that is a monitor capacitor, a parallel circuit, and a second monitor that is connected in series to the parallel circuit. And an input of the integrating circuit is connected to both ends of the auxiliary winding T1c via a rectifying diode D21. The diode D21 is attached in a direction capable of rectifying the voltage of the auxiliary winding T1c generated while the switching element TR1 is on. The voltage V (C21) across the capacitor C21, which is the output of this integration circuit, becomes the output voltage of the voltage monitor circuit 20a.

The output voltage monitor circuit 20a operates as follows. First, the rectifying diode D21 performs half-wave rectification on the rectangular waveform AC waveform of the voltage V3 generated in the auxiliary winding T1c. The voltage generated in the auxiliary winding T1c during the period when the switching element TR1 is on is a voltage Vin · (N3 / N1) with the dot side of the auxiliary winding T1c in FIG. Thus, the switching element TR1 continues for the duration of the on-time Ton. Here, N1 is the number of turns of the input winding T1a of the transformer T1, and N3 is the number of turns of the output winding T1c. On the other hand, during the period other than the on-time Ton, a voltage is generated with the dot side at the low potential side. Accordingly, when the half-wave rectified waveform as described above is passed through the integrating circuit composed of R21, R22, and C21, a voltage V (C21) represented by the following expression (1) is substantially output.

  Further, the current limit signal generation circuit 20 includes a resistor R31 having one end connected to the output of the DC voltage generation circuit 22, a resistor R32 having one end connected to the other end of the resistor R31, and a collector terminal other than the resistor R32. Is connected to one end of the capacitor C21, which is the output end of the output voltage monitor circuit 20a, and the base terminal of the transistor TR31 is a resistor R31 and a resistor R32. The point is connected. Further, a transistor TR32 having a base terminal connected to the collector terminal of the transistor TR31 and an emitter terminal connected to the negative terminal side of the input power source Ein of the current detection resistor R0 is provided. The output of the generation circuit 20 is configured. Transistors TR31 and TR32 are NPN transistors.

  Next, the overall operation of the current limit signal generation circuit 20 will be described with reference to FIG. The transistor TR31 operates in an active state, and a voltage VBE1 generated by a base-emitter current is generated between the base and the emitter.

  Here, when the voltages VBE1 and VBE2 generated by the base-emitter currents of the transistors TR31 and TR32 exhibit the same characteristics, and the voltages at which the transistors TR31 and TR32 can be turned on are equal to VBE, the current detection voltage V (R0 ) Becomes equal to the voltage V (R32) + V (C21), Vbe2 = VBE, and the transistor TR32 can be turned on. For example, when the switching current Isw = 0, the voltage Vbe2 applied between the base and the emitter of the transistor TR32 cannot be turned on because it is lower than the voltage VBE between the base and the emitter. However, when the switching current Isw increases and the current detection voltage V (R0) reaches the sum of the voltage V (R32) and the voltage V (C21), the voltage Vbe2 applied between the base and emitter of the transistor TR32 is , TR32 reaches the base-emitter voltage VBE which can be turned on, and can be turned on.

  Therefore, the current limit signal generation circuit 20 adds the voltage V (R32) determined by the current control DC voltage Vc supplied from the DC voltage generation circuit 22 and the voltage V (C21) output from the output voltage monitor circuit 20a. Is compared with the current detection voltage V (R0) via the base-emitter voltage VBE when the transistors TR31 and TR32 are turned on. When the current detection voltage V (R0) is lower, that is, when the switching current Isw and the output current Iout are smaller, the collector terminal of the transistor TR32 outputs a high level. Conversely, when the current detection voltage V (R0) reaches the addition voltage, that is, when the switching current Isw and the output current Iout are about to exceed predetermined values, a low level is output.

  When the temperature of the switching element TR1 exceeds a predetermined temperature, the DC voltage generation circuit 22 performs an operation of changing the current control DC voltage Vc so as to decrease the added voltage of the voltage V (R32) and the voltage V (C21). . When the current control DC voltage Vc decreases, the voltage V (R32) decreases, so that the transistor TR32 can be turned on with a relatively small switching current Isw. Further, the output voltage monitor circuit 20a performs an operation of reducing the voltage V (C21) when the output voltage Vout is lowered. Accordingly, the transistor TR32 can be turned on with a relatively small switching current Isw. Hereinafter, for convenience of explanation, an added voltage of the voltage V (R32) and the voltage V (C21) is referred to as an overcurrent protection threshold Vr.

  As shown in FIG. 3, the drive pulse generation circuit 18 has a sawtooth wave generation circuit 18 b that generates a sawtooth wave voltage V (osc) and a sawtooth wave voltage V (osc) input to an inverting terminal. And an output voltage control signal V (vol) input from the terminal a1, which is an output, and a comparator CP11 input to the non-inverting terminal. Furthermore, the NAND circuit NAND11 to which the output signal of the comparator CP11 and the current control signal V (cur) input from the terminal a2 are input, and the output signal of the NAND circuit NAND11 are input to the reset terminal R, and the oscillator OS11. Is formed by a known set-reset-flip-flop FF11 (hereinafter referred to as RS-FF11) having an output terminal Q connected to a terminal a0 of the drive pulse generating circuit 18 The signal selection circuit 18a is provided. The signal output from the output terminal Q of the RS-FF 11 becomes a driving pulse Vg for driving the switching element TR1, and is input to the gate which is the driving terminal of the switching element TR1 via the terminal a0.

  The sawtooth wave generation circuit 18b includes a constant voltage DC power source Vcc11, a charging resistor R11 having one end connected to the DC power source Vcc11, a timer capacitor C11 connected between the other end of the charging resistor R11 and the ground, and a timer. A reset element S11 connected to both ends of the capacitor C11 and an oscillator OS11 for controlling the reset element S11 are provided, and a voltage V (osc) generated at both ends of the timer capacitor C11 is output.

  The oscillator OS11 generates an impulse trigger pulse. The trigger pulse is a repetitive pulse having a constant period, and determines the switching frequency of the switching element TR1 and the turn-on timing of the switching element TR1. Further, the reset element S11 short-circuits both ends of the timer capacitor C11 when a trigger pulse is input, and is instantaneously opened, and continues to be open until the next trigger pulse is input.

  The drive pulse generation circuit 18 configured as described above operates as follows. First, the sawtooth wave generation circuit 18b receives a trigger pulse from the oscillator OS11, the reset element S11 short-circuits both ends of the timer capacitor C11, and the electric charge is discharged to V (osc) ≈0. Further, the reset element S11 is instantaneously opened, the timer capacitor C11 is charged by the charging current supplied from the DC power supply Vcc11 via the charging resistor R11, and V (osc) increases. At this time, the time constant of the series circuit of the charging resistor R11 and the timer capacitor C11 is set to a sufficiently large value as compared with the period of the trigger pulse, so V (osc) increases linearly with a substantially constant slope. To do. Thereafter, when the next trigger pulse is input, the reset element S11 is short-circuited and the above operation is repeated. By such an operation, the sawtooth voltage V (osc) is generated at both ends of the timer capacitor C11.

  The comparator CP11 outputs a high level when the output voltage control signal V (vol) is higher than the sawtooth voltage V (osc), and outputs a low level in the opposite case. The current limiting signal V (cur) input from the terminal a2 is input to one input terminal of the NAND circuit NAND11, and the output signal of the comparator CP11 is input to the other input terminal. As a result, when the signal from the comparator CP11 is at a low level or one of the current limit signals V (cur) is at a low level, the output of the NAND circuit NAND11 becomes a high level. The output of the output terminal Q becomes high level by the impulse-like trigger pulse of the oscillator OS11 input to the set terminal S, and the RS-FF 11 is output when a high level signal is input from the NAND circuit NAND11 to the reset terminal R. The output of terminal Q is inverted to low level. A signal output from the output terminal Q of the RS-FF 11 is a drive pulse Vg input to the drive terminal of the switching element TR1. That is, the switching element TR1 is turned on while the output terminal Q is outputting a high level, and the switching element TR1 is turned off while the output terminal Q is outputting a low level.

  Next, a series of operations of the switching power supply 10 configured as described above will be described with reference to FIGS. Period 1 in the time chart of FIG. 4 is a period during which the output voltage Vout of the output voltage Vout-output current Iout characteristic shown in FIG. In this period 1, since the switching current Isw is a small value and the current detection voltage V (R0) has not reached the overcurrent protection threshold Vr, the current limit signal V (cur) output from the current limit signal generation circuit 20 is The high level is maintained, and a high level signal is continuously input to one input terminal of the NAND circuit NAND11 of the drive pulse generation circuit 18. Accordingly, the drive pulse Vg generated at the output terminal Q of the RS-FF 11 has the same logic as the output signal of the comparator CP11, and the drive pulse Vg is exclusively generated by the error amplification circuit 16 as shown in the time chart of FIG. The pulse width is controlled based on the output voltage control signal V (vol) to be output.

Thus, the switching power supply 10 determines the on-time Ton of the drive pulse Vg of the switching element TR1 so that the output voltage Vout becomes equal to the reference voltage Vref in the error amplifier circuit 16, and the output voltage Vout is controlled. The output voltage Vout can be approximately expressed as in Expression (2).

  Here, N1: the number of turns of the input winding T1a, N2: the number of turns of the output winding T1b, T: the switching period, and Ton: the ON time of the switching element TR1.

  Note that it can be seen from the equation (2) and the above equation (1) that the output voltage V (C21) of the output voltage monitor circuit 20a and the output voltage Vout have a substantially proportional relationship.

  Period 2 is a state in which, for example, the load electronic device has a low impedance failure, the output current Iout increases, the overcurrent protection circuit operates, and the switching current Isw is limited so as not to exceed a predetermined current value. is there.

  The error amplification circuit 16 raises the output voltage control signal V (vol) because the output voltage Vout is lower than the predetermined reference voltage Vref. Eventually, when the output voltage control signal V (vol) becomes higher than the sawtooth voltage V (osc), the output of the comparator CP11 always becomes low level, and pulse width modulation based on the output voltage control signal V (vol) is performed. It becomes a state that can not be.

  On the other hand, as the switching current Isw increases, the current detection voltage V (R0) generated in the current detection resistor R0 increases to reach the overcurrent protection threshold Vr, and the current limit signal V (cur) becomes low level. Then, the input of the NAND circuit NAND11 of the drive pulse generation circuit 18 becomes low level, and the output of the NAND circuit NAND11 changes from low level to high level. Accordingly, a reset signal is output from the NAND circuit NAND11 to the reset terminal R at the timing when the current limit signal V (cur) is inverted from the high level to the low level. The drive pulse Vg generated by the output terminal Q of the RS-FF 11 is inverted from the high level to the low level at the timing when the reset signal is input, thereby turning off the switching element TR1. Further, the RS-FF 11 holds the state of the output terminal Q until the next set signal is input to the set terminal S. Therefore, when the output terminal Q becomes low level and the switching element TR1 is turned off, the switching current Isw does not flow and the current limit signal V (cur) is inverted to high level, but the output is continued until the next period starts. The state (low level) of the terminal Q is maintained, and the switching TR1 maintains the off state. With the above operation, the output current Iout is limited so as not to exceed the overcurrent set value, and at the same time, the ON time Ton of the switching element TR1 is shortened, so that the output voltage Vout decreases according to the equation (2).

  When the output voltage Vout decreases, the error amplifier circuit 16 that is an inverting amplifier saturates to a high level, and the output voltage control signal V (vol) continues its high-level saturation voltage value. Accordingly, the drive pulse Vg generated at the output terminal Q of the RS-FF 11 is inverted to the low level in synchronization with the timing when the current limit signal V (cur) is inverted to the low level, as shown in the time chart of FIG. The drive pulse Vg is generated exclusively based on the output voltage control signal V (cur) output from the current limit signal generation circuit 20.

  Here, the relationship between the output voltage Vout and the output current iout in the period 2 will be described with reference to FIG. In the period 2, the operation is performed in a state where the output voltage Vout is slightly reduced in the graph of the output voltage Vout-output current Iout characteristic. If the output voltage V (C21) of the output voltage monitor circuit 20a is a fixed value that does not change in accordance with the output voltage Vout, the output voltage Vout-output current Iout characteristic draws a locus (a), and the output voltage Vout is As the voltage decreases, the influence of the delay time of the operation of the drive pulse generation circuit 18 becomes more significant, and the output current Iout increases. However, in the switching power supply circuit 10, when the output voltage Vout decreases, the voltage V (C21) also decreases at the same time, and the overcurrent protection threshold Vr also decreases accordingly, so that the output current Iout increases as shown in the locus (b). The operation | movement which suppresses is performed.

  In the period 3, since the switching element TR1 has exceeded a predetermined temperature as a result of being left in the state of the period 2, the DC voltage generating circuit 22 changes the DC voltage Vc for current control as shown in FIG. This is a period in which the overcurrent protection operation continues in the state. When the current control DC voltage Vc decreases, the voltage V (R32) of the resistor R32 decreases at the same time, and the overcurrent protection threshold Vr decreases accordingly, so that the output current Iout is reduced to a smaller value as shown in FIG. Therefore, the switching element TR1 can be prevented from being damaged due to heat generation.

  As described above, the switching power supply device 10 extracts the voltage V (C21) that is the output voltage information via the output voltage monitor circuit 20a, and the peak value of the switching current Isw decreases as the output voltage Vout decreases. Since the overcurrent protection threshold Vr is corrected, the switching power supply device 10 and the load 14 can be reliably protected regardless of the input voltage. Furthermore, the DC voltage generation circuit 22 can change the overcurrent protection threshold Vr by changing the current control DC voltage Vc in accordance with an arbitrary signal input such as the input voltage Vin and the environmental temperature, and can perform switching. The overcurrent protection characteristics can be optimized for various operating states of the power supply device.

  In general, in order to reliably protect the switching element TR1 from a steep current stress, the overcurrent protection threshold value Vr is changed within a short time within several pulses of the drive pulse Vg with respect to fluctuations in the output voltage Vout. However, in the switching power supply device 10, high-accuracy and high-speed response performance is realized by using a simple circuit that can be configured with a capacitor and a resistor, such as the output voltage monitor circuit 20 a. On the other hand, since the environmental temperature etc. fluctuate relatively slowly, satisfactory response performance can be obtained even with the DC voltage generation circuit 22 constituted by an inexpensive general-purpose microprocessor having a clock frequency of about 10 kHz, for example.

  In addition, since the above-mentioned reliable overcurrent protection characteristics can be obtained, safety can be ensured without changing parts that are susceptible to electrical stress due to switching current to large ones with an excess margin for the rated current and temperature. Therefore, the switching power supply device 10 itself can contribute to downsizing and cost reduction.

  Note that the present invention is not limited to the above embodiment, and a resistor may be inserted at the connection point between the emitter terminal of the transistor TR31 and the capacitor C21 to finely adjust the correction operation of the overcurrent protection threshold value Vr. In addition, a capacitor, snubber circuit, or filter that absorbs switching noise and external noise is added to a predetermined circuit part within a range that does not affect the operation related to overcurrent protection described above, thereby preventing malfunction of the circuit. May be.

  In addition to the single forward method, the power conversion method can also be applied to a switching power supply configured in a push-pull method, a bridge method, a flyback method, a polarity inversion chopper method, or the like.

  Furthermore, the current control DC voltage Vc generated by the DC voltage generation circuit 22 and the output voltage V (C21) of the output voltage monitor circuit 20a are not necessarily complete DC voltages. For example, by superimposing a ripple voltage having the same frequency as the switching frequency on the voltage V (C21), it is possible to cause a gain adjustment action equivalent to a known slope correction technique and to improve the stability of the overcurrent protection operation. It is.

It is a circuit part which shows one Embodiment of the switching power supply device of this invention. It is a circuit diagram explaining operation | movement of the DC voltage generation circuit with which this embodiment is provided, a voltage limiting signal generation circuit, and a current detection resistor. It is a functional block diagram which shows the structure of the drive pulse generation circuit with which this embodiment is provided. It is a time chart which shows operation | movement of this embodiment. It is a graph of the output voltage-output current characteristic which shows operation | movement of this embodiment.

Explanation of symbols

DESCRIPTION OF SYMBOLS 10 Switching power supply device 12 Rectification smoothing circuit 18 Drive pulse generation circuit 20 Current limiting signal generation circuit 20a Output voltage monitor circuit 22 DC voltage generation circuit C21 Monitor capacitor D21 Rectifier diode R0 Current detection resistors R21, R22 Monitor resistors R31, R32 Resistance T1 Transformer T1a Input winding T1b Output winding T1c Auxiliary winding TR1 Switching elements TR31, TR32 Transistors

Claims (3)

A drive pulse generation circuit that outputs a drive pulse that is pulse-width modulated at a predetermined switching frequency, and a switching that is turned on / off by the drive pulse from the drive pulse generation circuit to generate an AC voltage by intermittently applying a DC input voltage A transformer having an element, an input winding to which the AC voltage is applied, and an output winding that outputs a voltage obtained by transforming the AC voltage, and an output voltage obtained by rectifying and smoothing the AC voltage output from the output winding. And a rectifying / smoothing circuit that supplies an output current to a load.
A DC voltage generation circuit that outputs a changeable DC voltage for current control that is a predetermined target value relating to the control of the output current, and a current detection voltage that is inserted into a flow path of a switching current that flows through the switching element. A current detection resistor and a current limit signal generating circuit that outputs a current limit signal for limiting the switching current by detecting the current detection voltage;
The current limit signal generation circuit includes:
An auxiliary winding provided in the transformer, and an output voltage monitor circuit that generates and outputs a first DC voltage corresponding to a change in the output voltage based on a pulse voltage generated in the auxiliary winding;
Second DC voltage generating means that is varied based on a DC voltage for current control of the DC voltage generating circuit;
Comparing means for comparing the added voltage of the first and second DC voltages with the current detection voltage and outputting a current limit signal for limiting the switching current when the current detection voltage becomes larger. ,
The drive power generation circuit operates so as to stop or narrow the on-duty of the drive pulse for driving the switching element when the current limit signal is output. The output voltage monitor circuit has a positive potential side of its output terminal connected to one end of the current detection resistor,
The second DC voltage generating means and the comparing means are:
A first resistor having one end connected to the output of the DC voltage generating circuit, a second resistor having one end connected to the other end of the first resistor, and a collector terminal being the other end of the second resistor A first NPN transistor having a base terminal connected to a midpoint of the first resistor and the second resistor, and an emitter terminal connected to a reference potential side of the output voltage monitor circuit output terminal; A terminal connected to a collector of the first NPN transistor, an emitter terminal connected to the other end of the current detection resistor, and a collector terminal comprising a second NPN transistor that is an output of the current limiting signal generation circuit;
The current detection resistor includes the current detection voltage in a direction in which the emitter terminal of the second NPN transistor is at a lower potential than the emitter terminal of the first NPN transistor when a current flows through the switching element. The switching power supply device according to claim 1, wherein: The transformer and the rectifying / smoothing circuit are configured to perform a single forward type power conversion, and the output voltage monitor circuit is connected in parallel to a first monitor resistor and a monitor capacitor, and is connected in series to the parallel circuit. And an input of the integrating circuit is connected to both ends of the auxiliary winding via a rectifying diode, and the switching element is turned on by the rectifying diode. 3. The switching power supply device according to claim 1, wherein the voltage generated during the period is connected in a direction in which half-wave rectification is performed, and the output of the output voltage monitor circuit is a voltage generated at the output of the integrating circuit. .

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