JP2006157198A - Non-waveguide line/waveguide transformer - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To suppress a desired frequency component without causing increase in the volume or the area occupied by the circuit incident to provision of an extra filter and without having characteristic adverse effect on the main frequency, by providing a means for suppressing a desired frequency substantially two times or higher of the main frequency, in a transformer of non-waveguide line (e.g. distributed constant line or coaxial line) and waveguide. <P>SOLUTION: In the non-waveguide line-waveguide transformer, where a probe 3 is inserted into a waveguide 2 from a direction substantially perpendicular to the transmission direction of the waveguide 2 and one end side thereof becomes a short circuit surface 6a, a metal piece 5 operating to ground the tip of the probe 3 artificially at a desired suppression frequency is provided, in a plane substantially perpendicular to the transmission direction of the waveguide 2 at the position of the probe 3. <P>COPYRIGHT: (C)2006,JPO&NCIPI

Description

  The present invention relates to a converter between a non-waveguide line and a waveguide line such as a distributed constant line or a coaxial line used in a microwave circuit. More specifically, the present invention relates to a non-waveguide line-waveguide converter having a function of a filter that suppresses a desired unnecessary frequency component such as a harmonic that is twice or more the main frequency.

  For example, in microwave components such as LNB (Low Noise Block Down Converter) and BUC (Block Up Converter) for satellite communications, a distributed constant circuit using a microstrip line or the like is used for the internal circuit configuration. A waveguide with a low insertion loss is generally used for the interface. For this reason, a stripline-waveguide converter that mutually converts the microstrip line and the waveguide is used between them (see, for example, Patent Document 1). As shown in FIG. 8, the stripline-waveguide converter generally includes a dielectric substrate in a waveguide 2 formed as a rectangular cavity by a metal casing 6 or a metal plate. The tip of the microstrip line 1 formed on 4 is inserted as a probe 3. Further, the coaxial line-waveguide converter is generally configured such that the tip of the coaxial line 7 is inserted as the probe 3 into the waveguide 2 having the same configuration as shown in FIG. ing.

  On the other hand, in this microwave circuit having a mixer (mixer), when suppressing unnecessary frequency components generated by the mixer, or when suppressing gain and noise outside the desired band with an amplifier, (1) Filtering by a distributed constant circuit such as a microstrip line produced on a substrate and (2) filtering by a waveguide having one or more resonators are generally performed. When these two filterings suppress harmonics that are about twice the main frequency, they have the following advantages and disadvantages.

(1) Filtering by distributed constant circuit A distributed constant circuit represented by a microstrip line is usually manufactured on a substrate that is a dielectric, and forms a circuit equivalently by a conductor pattern, so that there is no need for parts. have. However, the line impedance is mainly determined by the relative permittivity, the substrate thickness, and the line width of the substrate used, but it is a distributed constant circuit that uses the optimal substrate material (relative permittivity, substrate thickness) at the main frequency. In the case of suppressing a frequency higher than the main frequency and relatively close to the main frequency or a frequency lower than the main frequency, there is no major problem in design, but when the frequency becomes higher than about twice the main frequency, In many cases, the line width of the strip line becomes a thickness that cannot be ignored with respect to the wavelength. For this reason, when designing up to a higher frequency than the main frequency, the optimization is achieved by reducing the board thickness and narrowing the line width in order to design the same impedance, but the frequency is very high. However, when the substrate is made of a resin that is easily available and inexpensive, the substrate thickness becomes too thin, making it difficult to handle in the manufacturing process. In addition, it becomes difficult to realize a high impedance line (narrow line width) with respect to a general transmission line impedance (50Ω), and the impedance that can be used for circuit design is greatly limited, and desired characteristics are obtained. Becomes difficult. In addition, the Q value of a distributed constant circuit such as a general microstrip line is lower than that of a waveguide and has a large loss. Rejection using side couples, edge couples, and λ / 4 resonators is easy to design. In the filter, when a suppression filter having a frequency close to the main frequency is designed, a suppression operation is not performed for almost twice the frequency. Therefore, a filter for suppressing the frequency almost twice is provided separately from these filters. It has the disadvantages of being necessary, leading to an increase in the occupied area and often affecting the characteristics of the main frequency.

(2) Filtering by a waveguide A waveguide circuit has an advantage that a loss is lower than that of a distributed constant circuit using a dielectric, and thus a circuit having a high Q value can be produced. The tube itself has a structure that requires machining accuracy, and its production is costly. In addition, since it is a three-dimensional circuit, there is a drawback that the physical occupation volume is generally increased. In general, the waveguide frequency is such that a transmission mode (TE ₁₀ mode) with a small transmission loss can be used at the main frequency, but transmission speeds such as TE ₂₀ and TE ₀₁ are used at a frequency almost twice as high. Because it is possible to transmit higher-order modes with different frequencies, it is difficult to design the filter. Furthermore, a general bandpass filter with a resonator (cavity) with the main frequency set in the passband is almost doubled. Since the suppression operation cannot be obtained at the frequency of, a frequency suppression filter that is almost twice as large as that is required, resulting in an increase in cost and an increase in occupied volume.
JP 11-41010 A

  As described above, in a distributed constant circuit or a waveguide, in order to obtain suppression characteristics at a frequency approximately twice or more than the main frequency, a filter must be prepared separately from the filter for suppressing the vicinity of the main frequency. There is a problem that the occupied area or the occupied volume of the circuit increases, the number of parts increases, and the cost increases.

  The present invention has been made in view of such a situation, and is a non-waveguide line (distributed constant line or coaxial line) such as a converter between a distributed constant circuit and a waveguide generally used for microwave components. Etc.) in the converter between the waveguide and the waveguide, a means for suppressing an arbitrary frequency that is almost twice or more of the main frequency causes an increase in the volume or area occupied by the circuit when a separate filter is provided. There is provided a non-waveguide line-waveguide converter having a filter function that suppresses a desired frequency component that is approximately twice or more than the main frequency with little adverse characteristic to the main frequency. For the purpose.

  A non-waveguide line-waveguide converter according to the present invention includes a waveguide and a probe inserted into the waveguide from a direction perpendicular to the transmission direction of the waveguide- In the waveguide converter, a metal piece that operates to quasi-ground the tip of the probe at a desired suppression frequency is provided in a plane substantially perpendicular to the transmission direction of the waveguide at the position of the probe. It is characterized by.

  Here, the transmission direction of the waveguide means the direction along the tube axis of the waveguide, and the non-waveguide line means other than the waveguide, such as a distributed constant line or a coaxial line. It means a transmission line.

  The probe may be a transducer between the distributed constant line and the waveguide at the tip of the distributed constant line formed on the surface of the dielectric substrate, or coaxial at the tip of the central conductor of the coaxial line. A converter between a line and a waveguide may be used.

  According to the non-waveguide line-waveguide converter of the present invention, when it is desired to suppress, for example, twice the harmonic of the main frequency, which is the frequency originally handled in the microwave component, the probe tip is at that frequency. Since the metal piece is provided in the vicinity of the probe so as to be a pseudo ground, the electric field radiated from the probe at the frequency to be suppressed is minimized, and the emission is suppressed. Therefore, in the past, when trying to suppress almost twice the harmonics, in addition to a filter near the main frequency and a filter below the main frequency, a filter that suppresses about twice the frequency must be provided. In the present invention, the non-waveguide line-waveguide converter has a function of suppressing a desired frequency of about twice or more of the main frequency, such as a harmonic of approximately twice and an odd multiple thereof. Therefore, there is no need to separately provide a filter that suppresses a desired frequency, such as a filter that suppresses twice the frequency in the distributed constant line and the waveguide, without increasing the occupied area and occupied volume, and increasing the cost. High-performance microwave components can be obtained with a very simple configuration.

  Next, the non-waveguide line-waveguide converter of the present invention will be described with reference to the drawings. A non-waveguide line-waveguide converter according to the present invention is shown in FIG. 1, which is a cross-sectional explanatory view of a principal part of a converter of a distributed constant line and a waveguide made of a microstrip line as one embodiment thereof, and FIG. The probe 3 is inserted into the waveguide 2 from a direction substantially perpendicular to the transmission direction of the waveguide 2 as shown in the plane explanatory views when the probe portion is viewed from the front in the transmission direction of the waveguide. In the non-waveguide line-waveguide converter in which one end of 2 is the short surface 6a, the probe 3 is shown in FIG. A metal piece 5 that operates so as to pseudo-ground the tip of the probe 3 at a desired suppression frequency is provided in a plane substantially perpendicular to the transmission direction of the waveguide 2 at the position.

  In the example shown in FIG. 1, a rectangular waveguide is used as the waveguide 2, and the probe 3 is inserted in parallel to the transmission surface and parallel to the H surface (narrow surface). The metal piece 5 is provided in the. However, as will be described later, the waveguide 2 is not limited to a rectangular waveguide, and may be a circular waveguide, and the metal piece 5 is not provided on both sides of the probe 3 and provided only on one side. May be. In the example shown in FIG. 1, the waveguide 2 is formed by forming a rectangular cavity with a housing 6 made of metal. However, the waveguide 2 is formed of a metal plate and formed into a normal waveguide. It may be. In the rectangular waveguide 2, the width of the wide surface and the height of the narrow surface are usually determined by standard dimensions, and are formed to the optimum standard dimensions according to the main frequency to be used. For realization, the main frequency should not be extremely higher than the cut-off frequency of the used waveguide (the condition is such that a higher-order mode of the waveguide appears near a frequency almost twice the main frequency). Is necessary).

  On the side opposite to the transmission direction of the waveguide 2, a short surface 6 a is formed at a position with an electrical length of ¼ wavelength of the main frequency from the probe 3. In the example shown in FIG. 1, the short surface 6 a is formed by the wall surface of the housing 6 constituting one wall surface of the waveguide 2, but is limited to one wall surface of such a waveguide 2. Instead, a metal rod such as a screw may be formed by being inserted into the waveguide 2, or may be formed by a movable wall surface.

  Further, in the example shown in FIG. 1, the probe 3 is a probe in which the tip of the distributed constant circuit formed by the microstrip line 1 is used as it is. The tip of the distributed constant circuit can be used, or as shown in FIG. 7 described later, a coaxial line, for example, a distributed constant circuit can be converted into a coaxial line, and the central conductor of the coaxial line can be used.

  The microstrip line 1 is formed by patterning a conductor film such as a copper coating provided on the surface of the dielectric substrate 4 to a thickness of about 10 to 50 μm, for example, so as to form a desired circuit. An inductor, a capacitor, and the like are formed by the microstrip line 1. The probe 3 is formed by forming the dielectric substrate 4 so that the tip of the microstrip line 1 can be inserted into the waveguide 2. However, even if the dielectric substrate 4 is not provided, it is sufficient that only the conductor portion is inserted with a predetermined impedance as in the case of a coaxial line described later. The insertion length and width of the tip of the probe 3 into the waveguide 2 are patterned in an optimized shape together with the short surface 6a in order to perform impedance conversion between the microstrip line 1 and the waveguide 2. However, if the width a of the probe 3 becomes too large with respect to the wavelength of the desired suppression frequency, as in the case of the width of the metal piece 5 to be described later, the operation according to the operation principle is hindered. , Should be sufficiently narrower than λ / 4, and is preferably about λ / 8 or less. When it is formed in a cylindrical shape as in the case of a coaxial line, the diameter is preferably set to the above dimensions.

  In the present invention, at the position of the probe 3, the metal piece 5 is disposed at least on one side of the probe 3 in a plane substantially perpendicular to the transmission direction of the waveguide 2 and at the desired suppression frequency. Is provided so as to operate in a pseudo grounding manner. That is, the probe 3 is the maximum portion of the electric field with respect to the main frequency because the tip B point is an open end. For example, for the frequency twice the main frequency to be suppressed, A metal piece 5 is provided so as to be a pseudo ground. In order to place the tip B of the probe 3 in a pseudo grounding state with respect to the frequency to be suppressed, for example, when the wavelength of the frequency to be suppressed is λ, the distance d laterally from the tip of the probe 3 on the surface of the dielectric substrate 4 described above. May be formed at a position (point A) separated by λ / 2, and if a metal piece 5 having a total length L of λ / 2 around the point A is formed, λ / 2 = λ / Since it is 4 × 2, both ends of the metal piece 5 are excited by the strongest electric field at the frequency to be suppressed, and the center point A forms a virtual ground point having the lowest electric field.

  In the example shown in FIG. 1, this metal piece 5 is formed on both sides of the surface of the dielectric substrate 4 (in a plane substantially perpendicular to the transmission direction of the waveguide 2) at positions symmetrical with respect to the tip of the probe 3. Has been. By being formed on both sides in this way, the pseudo grounding effect at the tip of the probe 3 can be further strengthened, and since it is symmetrical, it can be made difficult to generate unnecessary polarization, etc. There is no need to provide it. When provided on both sides, both metal pieces 5 are provided so as to satisfy the relationship as described above. If the width of the metal piece 5 is too wide, the operation according to the operation principle is hindered, as in the case of the probe 3 described above. Preferably, it is formed to be about λ / 8 or less. Therefore, the shape of the metal piece 5 is a rectangle that is long in the direction parallel to the probe 3 or a similar shape (see, for example, FIGS. 5 to 6). In addition, when forming the metal piece 5 by a column shape, it is preferable to form so that the diameter may become below the said dimension.

  FIG. 2 shows the dimensional relationship between the metal piece 5 and the probe 3 and the relationship between the position where the electric field is high and the position where the frequency is desired to be suppressed (wavelength λ). As shown in FIG. 2, both end portions of the metal piece 5 are electric field maximum points, the midpoint A is a pseudo grounding point, and the distance between the midpoint A and the tip B point of the probe 3 is λ / 2. As a result, the point B of the probe 3 also becomes a virtual ground point at the desired suppression frequency, the electric field distribution at the desired suppression frequency is reduced, and its emission is suppressed (insertion loss is increased), and the suppression characteristic is improved. can get.

If the λ / 4 portion (both ends) of the metal piece 5 at this time is expressed by a grounded LC series resonance circuit, an equivalent circuit as shown in FIG. 3 is obtained. That is, at the resonance frequency (desired suppression frequency) f = 1 / {2π (L · C) ^1/2 }, the impedance becomes 0 and the middle point A of the metal piece 5 is in the ground state. Since the midpoint A is separated from the point B of the probe 3 by λ / 2 as described above, the point B of the probe 3 is also in a grounded state. Therefore, according to the configuration of the present invention, it is shown that a certain suppression characteristic can be obtained at a desired suppression frequency. On the other hand, for the main frequency to be mainly passed, the addition of the metal piece 5 is considered to add only a capacitive component, so the shape of the probe 3 (dimension conditions such as length and width) is optimized. Therefore, necessary characteristics can be obtained.

  The microstrip line-waveguide converter structure shown in FIG. 1 has a main frequency of 30 GHz and a suppression frequency of 57 GHz. The converter of the present invention in which the above-described metal pieces 5 are provided on both sides and the metal piece 5 are provided. FIG. 4 shows the result of examining the insertion loss (I / L) and return loss (R / L) of the converter (structure shown in FIG. 8) formed with the same structure as in FIG. The solid line is the data of the converter according to the present invention, and the broken line is the data of the converter having a conventional structure in which no metal piece is provided. As is clear from FIG. 4, the insertion loss (I / L) at 57 GHz is significantly attenuated, whereas the conventional structure does not show attenuation at 57 GHz. As for the return loss (R / L), according to the present invention, the return loss is close to total reflection at 57 GHz. In the present invention, both the insertion loss and the return loss are degraded in characteristics at 55.6 GHz and 57.8 GHz. However, there is no problem because it is an influence of the higher-order mode and is out of the desired suppression frequency. .

  As described above, according to the present invention, noise in a desired frequency band can be suppressed without inserting a special filter and increasing the occupied area and volume. However, since the metal piece 5 becomes too large at a frequency close to the main frequency and the influence on the main frequency cannot be avoided, the frequency that can be suppressed should be approximately twice or more than the main frequency. is required. In addition, the relationship between the electric field intensity at the midpoint A of the metal piece 5 and the tip B of the probe 3 is the same for an odd multiple of the frequency to be suppressed. can do.

  5 and 6 are examples showing a modification of FIG. That is, the example shown in FIG. 5 is such that the central portion of the metal piece 5 is thin and thicker toward the tip, and if the metal piece 5 becomes too thick (the width becomes wider) as described above, Although it interferes with the principle, there is no problem in operation if the central part is formed thin, and if the tip part side is formed widely, the ground frequency becomes lower and the suppression frequency band (range) becomes wider. There is a merit. Also in this case, the length is λ / 4 with respect to the wavelength λ of the frequency to be suppressed from the center A point, and the total length is λ / 2. Other configurations are also shown in FIG. Same as example. In the example shown in FIG. 6, the metal piece 5 is formed so as to move away from the probe 3 as it goes from the center point A to the tip, and the length from the center point A to the tip is λ. The other configuration is the same as the example shown in FIG. By adopting this structure, since the parts other than the pseudo grounding point A are separated from the probe 3, the influence on the main frequency can be reduced.

  Each of the above examples is an example of a converter that converts a distributed constant line and a waveguide made of a microstrip line formed of a conductor film on the surface of a dielectric substrate. Not only the converter but also a converter of a coaxial line and a waveguide, for example, can be similarly configured. An example is shown in FIG. The example shown in FIG. 7 is similar to FIG. 1, but instead of a microstrip line, a central conductor 7a, a dielectric 7b, and a casing 6 that covers the outer periphery and serves as an external conductor. The constructed coaxial line 7 is used, and the tip of the middle conductor 7 a is inserted into the waveguide 2 as the probe 3. Then, at the position of the probe 3, metal pieces 8 a are provided on both sides of the probe 3 in a plane perpendicular to the transmission direction of the waveguide 2 with the same relationship between the dimensions d and L as in the example shown in FIG. 1. Yes.

  In this example, unlike the example shown in FIG. 1, since there is no dielectric substrate, one wall surface (a wall surface perpendicular to the probe 3) of the waveguide is partially metallized to form a metal piece. It is formed by erecting a dielectric rod 8 that is 8a. This metallization is performed so that the metal piece 8a has a length of λ / 4 (λ is a wavelength of a frequency to be suppressed) on both sides with the position of the tip of the probe 3 as the center. The dielectric rod 8 may be cylindrical or prismatic, but is preferably not square. Moreover, it does not have to be rod-shaped, but may be a plate-like body and may be a dielectric piece. Furthermore, as described above, if the metal piece 8a is too thick, the operation principle is hindered. Therefore, the diameter (passage) is preferably about λ / 8 or less. The other configuration is the same as the example shown in FIG. 1, and the operation is the same as the example of FIG. Therefore, the metal piece 8a may be only on one side. As a result, similarly to the above, the tip of the probe 3 can be set to the minimum position of the electric field with respect to a desired suppression frequency, and radiation into the waveguide 2 can be suppressed.

  In addition, each of the above examples is an example of a converter between a non-waveguide line and a rectangular waveguide, but the waveguide is not limited to a rectangular waveguide, and may be a circular waveguide. The same can be applied. In this case, since the cross-sectional shape of the waveguide is circular, if the waveguide is inserted from the direction perpendicular to the transmission direction, it is the same anywhere on the circumference.

It is a figure explaining the converter of the microstrip line and waveguide which are one Embodiment of the non-waveguide line-waveguide converter by this invention. It is a figure explaining distribution of the electric field strength of a suppression frequency with the structure of FIG. FIG. 2 is an equivalent circuit diagram in which the lumped constant circuit is replaced with respect to the suppression frequency in the structure of FIG. 1. It is a figure which shows the characteristic of I / L and R / L in the structure shown by FIG. 1, and the structure shown by the conventional FIG. It is a figure which shows the modification of FIG. It is a figure which shows the modification of FIG. It is a figure which shows the structural example of the coaxial line-waveguide converter which is other embodiment of the non-waveguide line-waveguide converter of this invention. It is a figure which shows the example of the conventional microstrip line-waveguide converter. It is a figure which shows the example of the conventional coaxial line-waveguide converter.

Explanation of symbols

DESCRIPTION OF SYMBOLS 1 Microstrip line 2 Waveguide 3 Probe 4 Dielectric board 5 Metal piece 6 Case 7 Coaxial line 8 Dielectric rod 8a Metal piece

Claims (3)

  In a non-waveguide line-waveguide converter having a waveguide and a probe inserted into the waveguide from a direction perpendicular to the transmission direction of the waveguide, the waveguide at the position of the probe A non-waveguide line-waveguide, characterized in that a metal piece that operates to quasi-ground the tip of the probe at a desired suppression frequency is provided in a plane substantially perpendicular to the transmission direction of converter.   2. The non-waveguide line-waveguide converter according to claim 1, wherein the probe is a converter between the distributed constant line and a waveguide at a tip of a distributed constant line formed on the surface of the dielectric substrate. .   The non-waveguide line-waveguide converter according to claim 1, wherein the probe is a converter between a coaxial line and a waveguide at a distal end portion of a central conductor of the coaxial line.

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