JP2005195490A - Radar system - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a radar system capable of restraining a grating lobe, while reducing the number of elements in a receiving array antenna. <P>SOLUTION: In this radar system provided with a transmission antenna having L<SB>t</SB>of beam width, and the receiving antenna of L<SB>r</SB>(L<SB>r</SB>>L<SB>t</SB>) of beam width having a plurality of receiving elements of element space d, the element space d is determined not to generate a main beam 17 in one reception pattern and the grating lobe 18 therein concurrently in a main beam in a transmission pattern. <P>COPYRIGHT: (C)2005,JPO&NCIPI

Description

  The present invention relates to a radar apparatus using an array antenna, and more particularly to a technique for determining the arrangement of array elements and reducing the number of elements.

  In order to improve the resolution in the azimuth direction of the radar, an array antenna having a large aperture is required. In addition, since it is necessary to arrange the element antennas at half-wavelength intervals in order to prevent the occurrence of grating lobes, an increase in the aperture diameter of the array antenna results in an increase in the number of element antennas constituting the array antenna. It was. As a conventional technique for avoiding such a problem, a method is known in which a transmitting array antenna and a receiving array antenna are separately provided to reduce the number of element antennas and simultaneously suppress the generation of grating lobes ( For example, see Patent Document 1).

  In this method, first, the number of elements is reduced by widening the element spacing between the transmitting array antenna and the receiving array antenna beyond half a wavelength. Then, by arranging the transmitting array antenna and the receiving array antenna so that the respective element intervals are different from each other, the grating lobe generated in the transmitting antenna pattern (hereinafter referred to as transmission pattern) and the receiving antenna pattern (hereinafter referred to as receiving) The intervals between the grating lobes generated in the pattern) do not coincide with each other, and the grating lobes generated in the pattern as a whole of transmission and reception (hereinafter referred to as transmission / reception pattern) are suppressed. Further, the region is scanned and observed by scanning the transmission pattern and the reception pattern in synchronization.

U.S. Pat. No. 3,825,928 "High Resolution Bistatic Rader System"

  The above-described method has a problem that the arrangement of the element antennas is restricted in order to keep the grating lobes of the transmission / reception pattern sufficiently low. As a result, when the element spacing of the transmitting array antenna and the element spacing of the receiving array antenna do not satisfy this restriction, the grating lobe of the transmission pattern matches the direction of the side lobe of the reception pattern. In addition, a relatively high grating lobe is produced.

  Also, with this method, when observing a wide area without overlapping the observation range, it is necessary to scan in angular increments corresponding to the angular resolution of the transmission / reception pattern, which requires observation time. It was.

  The present invention aims to solve such problems. In other words, by using a transmission antenna having a relatively small aperture diameter to thicken the main beam of the transmission pattern and forming a reception multi-beam in the main beam of the transmission pattern by digital beam forming (DBF). It is an object of the present invention to provide a radar apparatus that can reduce the scanning time while expanding the area that can be observed by transmission and reception of one pulse.

を満たす素子間隔ｄで前記複数の受信素子を配置したものである。 A radar apparatus according to the present invention includes a transmission antenna having a beam width of Lt, and a reception antenna having a plurality of receiving elements and a beam width of Lr (where Lr> Lt).
When the wavelength of the transmission wave radiated by the transmission antenna is λ and the effective beam width is θ _max ,

The plurality of receiving elements are arranged at an element interval d satisfying

  According to the present invention, it is possible to obtain a radar apparatus that can suppress grating lobes at the same time even though the number of receiving elements can be reduced by increasing the element spacing.

なお、送信アンテナ２は、電子的あるいは機械的に送信パターンの主ビームの中心方向θ_cを走査するようになっている。 An embodiment of the present invention will be described below with reference to the drawings.
Embodiment 1 FIG.
FIG. 1 is a block diagram showing a configuration of a radar apparatus according to Embodiment 1 of the present invention. In the figure, a transmitter 1 is an element or a circuit that generates a reference pulse signal. The wavelength of the pulse signal generated here is represented by λ. The transmitting antenna 2 is an antenna having an aperture length L _t for irradiating the reference pulse signal generated by the transmitter 1. As a result, the null to null width Δθ _t [rad] of the main beam of the transmitting antenna 2 is given by Equation (1).

The transmission antenna 2 is configured to scan the central direction θ _c of the main beam of the transmission pattern electronically or mechanically.

Receiving array antenna 3 is a array antenna comprising N array elements 4-1 to 4-N to the element spacing is d, is the aperture length and L _r. FIG. 2 is a diagram showing the relationship between the respective aperture lengths of the transmitting antenna 2 and the receiving array antenna 3 and the element spacing of the array elements 4-1 to 4-N. Here, since the number of elements is N, the aperture length L _r of the receiving array antenna 6 and the element spacing d satisfy the relationship of the following equation.

These array elements 4-1 to 4-N constitute a one-dimensional equally-spaced linear array, and are adapted to detect radio waves that are irradiated by the transmitting antenna 2 and reflected back to the target. . Therefore, the main beam width Δθ _r [rad] of the receiving array antenna 3 is given by Equation (2).

At this time, the gap θ _g [rad] between the grating lobes generated in the antenna pattern of the receiving array antenna 3 satisfies the equation (4). Therefore, when the element interval d is sufficiently larger than the wavelength λ, θ _g is calculated from the equation (5).

  Further, the element arrays 4-1 to 4-N of the reception array antenna 3 are arranged so that the main beam and the grating lobe of one reception pattern are not generated in the main beam of the transmission pattern at the same time. By doing so, it becomes possible to select the grating lobes that are not included in the main beam width of the transmission pattern, and to suppress the grating lobes.

FIG. 3 is a diagram for explaining the relationship between the beam width and the beam direction. In the figure, the reference numeral 16 is pointing is a main beam of the transmitting antenna 2, the main beam width is [Delta] [theta] _t. The direction of the main beam of the transmission pattern 16 is θ _c . Reference numeral 17 denotes a reception pattern of the main beam of any of the array elements 4-1 to 4-N. The beam width of these reception patterns is Δθ _{r as} described above. Here, the direction of the main beam of the reception pattern is θ _max . These reception patterns are multi-beams formed in a process (DBF: Digital Beam Forming) for synthesizing reception signals of respective element antennas described later to form a beam. Reference numeral 18 indicates a grating lobe corresponding to the main beam 17, and the interval between the grating lobe 18 and the main beam 17 is θ _{g as} described above.

を満たす必要があることが、図３より理解される。式（６）の右辺のθ_g、Δθ_r、Δθ_tに式（１）、式（３）、式（５）の関係を代入すると、式（７）が得られる。

Then, in order to prevent the main beam 17 of the reception pattern and the grating lobe 18 from occurring simultaneously in the main beam 16 of the transmission pattern, θ _max is set to

It can be understood from FIG. By substituting the relations of Equations (1), (3), and (5) into θ _g , Δθ _r , and Δθ _t on the right side of Equation (6), Equation (7) is obtained.

As described above, since θ _max is the direction of the main beam 17 of the reception pattern, Equation (7) is a condition that defines the directions that the main beam 17 of the reception pattern can take. Here, θ _max is referred to as an effective beam width. That is, the effective beam width is an amount representing the direction that the main beam 17 of the reception pattern can take in the main beam width Δθt of a certain transmission pattern. In order to increase the effective beam width, that is, to expand the coverage of the reception pattern, according to the equation (7), the aperture length of the transmission antenna 2, the aperture length of the reception antenna 3, and the array elements 4-1 to 4-N It is possible to determine what the element spacing should be.

  Further, when the aperture lengths of the transmitting antenna 2 and the receiving array antenna 3 are determined from the system requirements and θmax is given, by determining d so as to satisfy the equation (7), suppression of the grating lobe can be achieved. At the same time, it is possible to increase the element spacing and consequently reduce the number of elements.

このようにすることで、有効ビーム幅内において安定的に送信波を受信することができるのである。ただし、４ｄＢは例示であり、また十分な電力の大きさはＳＮＲ(Signal to Noise Power Ratio)などの条件によって変化するので、このことは必須ではない。 Note that a transmission wave with sufficient power needs to be transmitted in the direction of θ _max . For example, if it is assumed that sufficient power is transmitted in the direction within the 4 dB beam width of transmission, θ _max desirably satisfies the condition of equation (8).

By doing so, it is possible to stably receive the transmission wave within the effective beam width. However, 4 dB is merely an example, and the magnitude of sufficient power varies depending on conditions such as SNR (Signal to Noise Power Ratio), so this is not essential.

  Next, the configuration of the radar apparatus according to Embodiment 1 of the present invention shown in FIG. 1 will be described. Each of the array elements 4-1 to 4-n has a corresponding receiver 5-1 to 5-. n are connected, and the radio waves detected by the respective array elements are amplified and output as reception signals. Corresponding A / D converters 6-1 to 6-n are connected to the receivers 5-1 to 5-n, respectively, and each received signal is individually converted into a digital signal. .

  The storage means 7 is a storage element or circuit or storage medium that temporarily stores the received signal converted into a digital signal. The range compression means 8 is a part that obtains a signal (range profile) with improved range resolution by performing pulse compression processing on the digital received signal. In this description and the following description, the term “part” means an element or a circuit configured to perform a predetermined function, but a computer program is executed on a central processing unit (CPU). You may comprise so that a corresponding function may be implement | achieved. The beam synthesizing unit 9 is a part for synthesizing a beam in which the grating lobe is suppressed from the range profile obtained by the range compressing unit 8.

Next, beam forming processing in the radar apparatus according to Embodiment 1 of the present invention will be described. First, the transmission / reception process that is a premise of the grating lobe suppression process will be briefly described. The reference pulse signal generated by the transmitter 1 is irradiated and reflected by the transmission antenna 2 to the external target and arrives at the reception array antenna 3. . The array elements 4-1 to 4-N of the receiving array antenna 3 receive the incoming waves, and the receivers 5-1 to 5-N corresponding to the received signals amplify the received signals, and the A / D converter 6 −1 to N convert to digital received signals. Thereafter, the digital reception signal is temporarily stored in the storage means 7. The range compression means 8 outputs a signal with improved range resolution (range) by performing pulse compression processing on the digital reception signal output from the N array elements 4-1 to 4-N and stored in the storage means 7. Profile). Here, the number of range bins is M, n is a natural number less than or equal to N, m is a natural number less than or equal to M, and the range profile output by the range compression means 8 is X _{n, m} .

Next, the beam combining unit 9 forms a reception multi-beam by a method such as fast Fourier transform. Now, let K be the number of reception multi-beams formed by the beam combining means 9 and θ _k be the direction of the main beam of the reception pattern of the k-th reception beam formed by the beam combining means 9. Then, the beam combining means 9 calculates the range profile Y _{k, m} of _{the mth} range bin for the kth received beam based on the equation (9) and outputs it as an output signal.

Meanwhile, since the effective beam width of the receiving array antenna 3 is theta _max, in a single pulse transmission and reception, an angle range for receiving the receiving array antenna 3 has to be within theta _max. Therefore, the orientation θ _k of the center of the main beam of the reception pattern formed in one pulse transmission / reception satisfies Expression (10).

このようにすることで、一回のパルスの送受信により±θ_maxの範囲を角度分解能Δθrで観測することができるようになる。ただしこのことは必須ではない。 Further, _since the angular resolution of the range profile Y _{k, m} obtained by the equation (9) is Δθ _r , the range of ± θmax from the central direction θc of the main beam of the transmission pattern using the main beam of the plurality of reception patterns is set. In order to divide without overlap or gap, it is desirable to set the distance between the centers of two adjacent main beams in the reception pattern to be formed as shown in Equation (11). That is, the first null point of a certain reception pattern is made to coincide with the beam center direction of the main beam of the reception pattern adjacent to this reception pattern.

By doing so, it becomes possible to observe the range of ± θ _max with an angular resolution Δθr by transmitting and receiving a single pulse. However, this is not essential.

  In addition, a wider area can be observed by transmitting and receiving pulses a plurality of times while scanning the center direction of the main beam of the transmission pattern. At this time, it can be seen that in order to scan the transmission pattern so that there is no overlap of observation ranges or gaps, it is only necessary to scan in angular increments of 2 × θmax. According to the conventional technique, in order to eliminate overlapping observation ranges and gaps, it is necessary to scan in angular increments corresponding to the angular resolution of the transmission / reception pattern, so that the scanning time can be shortened compared to such a technique. It is.

  As is apparent from the above, according to the radar apparatus of the first embodiment of the present invention, the element spacing d can be determined so as to satisfy Expression (7), and as a result, the grating lobe is suppressed while reducing the number of elements. This is a very advantageous effect.

  In the above description, the receiving array antenna 3 is configured as a one-dimensional equally spaced linear array. However, it goes without saying that the method described here can be easily applied to a two-dimensional array or an unequally spaced array. Absent.

  Furthermore, there is no restriction | limiting in the antenna system of the transmission antenna 2, Any antenna systems, such as a phased array antenna and an aperture surface antenna, may be used.

Embodiment 2. FIG.
In the first embodiment, the radar apparatus that can suppress the grating lobe even though the element spacing is increased to reduce the number of elements has been described. In the first embodiment, as a method of suppressing the grating lobe, a method of selecting based on whether or not the main beam of the reception pattern is included in the main beam of the transmission pattern is used. However, in addition to this, for example, a plurality of pulses may be transmitted / received using a transmission / reception antenna having such an arrangement, and a reception signal obtained therefrom may be integrated coherently. The radar apparatus according to Embodiment 2 of the present invention has such a feature.

  FIG. 4 is a block diagram showing a configuration of a radar apparatus according to Embodiment 2 of the present invention. In the figure, coherent integration means 20 is a part for coherently integrating the formed reception beam. The other components having the same reference numerals as those in FIG. 1 are the same as those in the first embodiment, and thus the description thereof is omitted.

Next, the operation of the radar apparatus according to Embodiment 2 of the present invention will be described. Also in the radar apparatus according to the second embodiment of the present invention, the transmitter 1 to the beam synthesizing unit 9 operate in the same manner as in the first embodiment. The range profile Y _{k, m} of the m th range bin for the th receive beam is output as an output signal.

このようにすることで、雑音成分のレンジプロフィールに対応する<Ｙ_k,m>は大きくならない一方で、信号成分のレンジフィールに対応する<Ｙ_k,m>は大きな値となるので、信号対雑音電力比を向上させることとなるのである。 The coherent integration unit 20 coherently integrates the range profile obtained as a result of a plurality of times of pulse transmission / reception. Here, among the plurality of pulses, the range profile output by the beam combining means 9 for the p-th pulse is Y _{k, m} (p). First, in order to simplify the explanation, it is assumed that the center direction θc of the main beam of the transmission pattern is fixed without being scanned. In this case, the coherent integrator 20 coherently integrates the range profile based on the equation (12), and obtains <Y _{k, m} > as a result.

By doing so, <Y _{k, m} > corresponding to the range profile of the noise component does not increase, whereas <Y _{k, m} > corresponding to the range feel of the signal component becomes a large value. This will improve the noise power ratio.

  Next, processing of the coherent integration unit 20 when scanning the transmission pattern will be described. FIG. 5 is a diagram showing a plurality of transmission pulses and a reception beam for each transmission pulse. In the figure, reference numeral 21 denotes a transmission pattern at the time of p-th pulse transmission, and reference numeral 22 denotes a transmission pattern at the time of p + 1-th pulse transmission. Reference numerals 23 to 29 are reception patterns formed by the beam combining means 15. The inside of the transmission pattern 21 is divided by reception patterns 23 to 27. The region where the reception patterns 23 to 27 exist is a region of ± θmax centered on the direction θc of the transmission pattern main beam.

  The transmission pattern 22 is also divided by the reception patterns 25 to 29. Then, in the example of FIG. 5, the transmission pattern 21 and the transmission pattern 22 share the reception patterns 25 to 27, and the reception patterns 25 to 27 can be integrated coherently even when the transmission pattern is scanned. I understand.

なお、式（１３）において［］は［と］の間の式の値を超えない最大の整数である。コヒーレント積分手段２０は、同一の方向について式（１３）に基づいて得られた回数分のレンジプロフィールを式（１２）に基づいて積分して<Ｙ_k,m>を求めるのである。 Here, if the transmission pattern is scanned at the angular velocity ω, and the pulse is transmitted at the repetition period T _pri , the number of times of integration in a certain direction θ _k can be calculated from Equation (13).

In Expression (13), [] is the maximum integer that does not exceed the value of the expression between [and]. The coherent integrating means 20 integrates the number of range profiles obtained based on the equation (13) in the same direction based on the equation (12) to obtain <Y _{k, m} >.

  As is apparent from the above, according to the radar apparatus of the second embodiment of the present invention, the range profile obtained by beam synthesis is further coherently integrated, so that the signal-to-noise power ratio is improved.

  Note that the number of pulse integrations may be increased by scanning the same region a plurality of times, or the number of integrations may be increased by transmitting more pulses in a required direction by changing the scanning angular velocity of the transmission pattern.

Embodiment 3 FIG.
The radar apparatus according to the second embodiment improves the signal-to-noise power ratio by coherent integration. However, this radar apparatus does not consider the case where it is mounted on a moving body, although it considers scanning of the transmission pattern. Therefore, a method for integrating coherently when mounted on a moving body will be described next.

  FIG. 6 is a block diagram showing a configuration of a radar apparatus according to Embodiment 3 of the present invention. The radar apparatus shown in the figure is assumed to be mounted on a moving body. In the figure, the motion compensation means 30 is a part that shifts the range profile and associates different range bins including the same target. The inertial navigation means 31 is a part that outputs information relating to the displacement of the position of the moving body on which the radar apparatus is mounted during transmission of a plurality of pulses. The inertial navigation means 31 may acquire position information by a method such as GPS (Global Positioning System), for example, or a method that autonomously acquires a temporal change in position by combining a gyroscope or a speedometer. It may be adopted. In addition, since the component which attached | subjected the code | symbol same as FIG. 4 is the same as that of Embodiment 2, description is abbreviate | omitted. However, in the second embodiment, the coherent integrator 20 directly inputs the range profile output by the beam combiner 9, but in the radar apparatus according to the third embodiment of the present invention, the coherent integrator 20 is The motion compensation means 30 is configured to input a range profile subjected to motion compensation.

Next, the operation of the radar apparatus according to Embodiment 3 of the present invention will be described with reference to the drawings. Also in the radar apparatus according to the third embodiment of the present invention, the transmitter 1 to the beam synthesizing means 9 operate in the same manner as in the second embodiment. The range profile Y _{k, m} of the m th range bin for the th receive beam is output as an output signal. Now, let Y _{k, m} (p) be the range profile output by the beam combining means 9 for the p-th pulse among a plurality of transmitted and received pulses.

Subsequently, based on the position information obtained from the inertial navigation means 31, the motion compensation means 30 obtains the phase at the time t of the received signal of the pulse reflected by the target. FIG. 7 is a diagram for explaining how the motion compensation unit 30 calculates the phase of the received pulse. In the figure, reference numerals 32 to 34 indicate the positions of the transmitting antenna 2 and the receiving array antenna 3 mounted on the moving body at each time. As shown in the figure, it is assumed that the transmitting antenna 2 and the receiving array antenna 3 are moving linearly at a speed v. It is assumed that time T elapses while moving from the point 32 to the point 34. Then, the distance between the points 32 and 34 is v × T. Further, it is assumed that the point 33 is an intermediate point between the point 32 and the point 34. Here, the point 33 is set to the origin O of the xy coordinate system, and the moving directions of the transmitting antenna 2 and the receiving array antenna 3 are set to the positive y-axis. The direction. Furthermore, the target 35 exists at a position that forms an angle θ between the positive direction of the y-axis and the azimuth direction, and the distance between the origin O and the target 35 is R ₀ . In the following description, the angle θ is referred to as a squint angle.

As is clear from FIG. 7, the distance R (θ, t) from the moving object to the target 35 at time t is calculated by the equation (14) from the cosine theorem.

In Expression (14), when the moving distance vt of the moving body is smaller than R ₀ , the right side of Expression (14) can be approximated as Expression (15).

If Expression (15) is used, the radio wave transmitted and received at the wavelength λ reciprocates the distance of R (θ, t), so the phase is given by Expression (16). This is the phase value calculated by the motion compensation means 30.

Subsequently, the motion compensation unit 30 shifts the range profile of the received P received pulses Y _{k, m} (p) by the pulse transmission repetition period T _PRI so that the range bin including the target 35 has the same range bin number m. Adjust to. Then, the phase profile based on the range and the squint angle is performed with the range profile having the range bins as Y ′ _{k, m} (p).

Here, Q is the number of samples in the azimuth direction after coherent integration. However, in order to obtain a sufficient reception gain in the θ _q direction, it is desirable that θ _q and θ _k satisfy Expression (18).

The signal Z _{k, m} (p), whose range bins are aligned by the motion compensation unit 30 and phase-compensated by the processing of equation (17), is coherently integrated by the coherent integration unit 20 based on equation (19).

ただし式（２０）において分解能を４ｄＢ幅で定義した。一般に知られているように、ＳＡＲの処理によって進行方向正面付近における方位方向の分解能は向上しない。このことは、式（２０）においてθ＝０とすると右辺が発散することからも確かめられる。つまり観測領域全体にわたって式（２０）の分解能が達成されるわけではない。結局、全体としての方位方向の分解能は、受信アレーアンテナ３の開口長Ｌrによって決まる分解能Δθ_rと、式（２０）の分解能Δθ_dのうちの分解能の高い方によって求められる。すなわち、ｍｉｎ{Ａ，Ｂ}をＡとＢのいずれか小さい方とすれば、分解能Δθは、式（２１）で与えられる。

By the way, Expression (17) and Expression (19) represent SAR (Synthetic Aperture Rader) in the squint direction. Therefore, it is known that the azimuth resolution Δθ _d after the coherent integration by the equation (19) is calculated by the equation (20).

However, the resolution was defined as 4 dB width in the equation (20). As is generally known, the resolution in the azimuth direction near the front in the traveling direction is not improved by the SAR process. This can also be confirmed from the fact that the right side diverges when θ = 0 in equation (20). That is, the resolution of Expression (20) is not achieved over the entire observation region. Eventually, the overall azimuth direction resolution is determined by the higher one of the resolution Δθ _r determined by the aperture length Lr of the receiving array antenna 3 and the resolution Δθ _d of equation (20). That is, if min {A, B} is set to the smaller one of A and B, the resolution Δθ is given by equation (21).

On the other hand, in the region where the squint angle θ is large, the angular resolution in the azimuth direction obtained as a result of the coherent integration of Expression (19) may be higher than the angular resolution determined by the thickness of the main beam of the received pattern. From this, it can be seen that, when θ _j is determined so as to satisfy Expression (18), it is desirable to determine θ _j by chopping finer than θ _k .

  Further, in the coherent integration of Expression (19), the value increases only for the signal whose squint angle is in the θj direction, and the signal in the direction of the grating lobe generated in the reception pattern does not increase even when integrated. For this reason, the grating lobe can be suppressed relatively.

  As is apparent from the above, according to the radar apparatus of the third embodiment of the present invention, even when mounted on a moving body, coherent integration is performed by compensating for the range shift and phase change caused by the movement. Therefore, it is possible to improve the resolution with respect to the azimuth direction and achieve suppression of grating lobes.

  In the above description, the case where the opening surface of the receiving array antenna 3 faces the traveling direction of the moving body has been described. However, even if the moving surface faces the other direction, the equation (15) is adjusted. Needless to say, it can be easily applied if calculated.

  The present invention can be applied particularly to a radar apparatus using an array antenna.

It is a block diagram which shows the structure of the radar apparatus by Embodiment 1 of this invention. It is a figure which shows the arrangement state of the transmission antenna of the radar apparatus by Embodiment 1 of this invention, and a receiving antenna. It is a figure which shows the relationship of the beam width of the transmission pattern and receiving pattern by the radar apparatus by Embodiment 1 of this invention. It is a block diagram which shows the structure of the radar apparatus by Embodiment 2 of this invention. It is a figure which shows the relationship between the transmission beam pattern between several pulses in Embodiment 2 of this invention, and a reception beam pattern. It is a block diagram which shows the structure of the radar apparatus by Embodiment 3 of this invention. It is a figure explaining the method to calculate the phase in the received pulse of the radar apparatus by Embodiment 3 of this invention.

Explanation of symbols

1 transmitter,
2 Transmitting antenna,
3 receiving array antenna,
4-1 to 4-N receiving antenna,
5-1-5-N receiver,
6-1-6-NA A / D converter,
7 storage means,
8 Range compression means,
9 Beam combining means,
20 coherent integration means,
30 motion compensation means,
31 Inertial navigation means.

Claims (5)

A transmitting antenna beam width is L _t, beam width has a plurality of receiving elements in the radar device and a receiving antenna is L _{r (provided} that L _r> L _t),
When the wavelength of the transmission wave radiated by the transmission antenna is λ and the effective beam width is θ _max ,

A radar apparatus, wherein the plurality of receiving elements are arranged at an element interval d satisfying The effective beam width θ _max is sufficient when transmitting sufficient power in a direction within DdB (D is a positive number) of the transmission pattern.

The radar apparatus according to claim 1, wherein: A receiver that outputs a reception signal of a reception wave received by the reception antenna;
Beam combining means for performing beam combining based on the received signal output by the receiver and outputting a range profile;
The radar apparatus according to claim 1, further comprising: The receiver outputs received signals for a plurality of pulses received by the receiving antenna;
The beam combining means outputs a range profile for each of the pulses based on the received signals for the plurality of pulses,
The radar apparatus according to claim 3, further comprising coherent integration means for coherently integrating these range profiles. The transmitting antenna and the receiving antenna are mounted on a mobile platform,
5. The radar apparatus according to claim 4, further comprising motion compensation means for compensating for movement and phase change of a range bin of a range profile output by the beam synthesis means in accordance with the movement of the moving platform.

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