JP2004312612A

JP2004312612A - Diversity receiver

Info

Publication number: JP2004312612A
Application number: JP2003106571A
Authority: JP
Inventors: Hidenori Usuki; 秀範臼杵
Original assignee: Hitachi Kokusai Electric Inc
Current assignee: Kokusai Denki Electric Inc
Priority date: 2003-04-10
Filing date: 2003-04-10
Publication date: 2004-11-04

Abstract

【課題】最大比合成ダイバーシチ処理を行うダイバーシチ受信装置において、ゾーン境界での合成処理によるビット誤りの発生を軽減する手段を提供する。
【解決手段】中間周波数帯信号から復調された同相成分信号及び直交成分信号と、その同相成分信号及び直交成分信号に含まれる固定パターンとの相関値を演算する相関演算処理部１１３−１、１１３−２と、前記の演算により求められた相関値を基に、同相成分信号及び直交成分信号と固定パターンとの尤もらしさ（尤度）を計算する相関尤度計算処理部１１４−１、１１４−２と、受信した無線信号の電力に正比例した重み付け係数に、前記の計算により求められた尤度に比例した値を乗ずることにより、重み付け係数を補正する乗算係数補正部１１５−１、１１５−２とを具備する。
【選択図】図１A diversity receiving apparatus that performs maximum ratio combining diversity processing provides means for reducing the occurrence of bit errors due to combining processing at a zone boundary.
Kind Code: A1 A correlation operation processing unit for calculating a correlation value between an in-phase component signal and a quadrature component signal demodulated from an intermediate frequency band signal and a fixed pattern included in the in-phase component signal and the quadrature component signal. -2 and correlation likelihood calculation processing units 114-1 and 114- for calculating the likelihood (likelihood) between the in-phase component signal and the quadrature component signal and the fixed pattern based on the correlation value obtained by the above calculation. 2 and a weighting coefficient directly proportional to the power of the received radio signal, multiplied by a value proportional to the likelihood calculated by the above calculation, thereby multiplying coefficient correction units 115-1 and 115-2 for correcting the weighting coefficient. And
[Selection diagram] Fig. 1

Description

【０００１】
【発明の属する技術分野】
本発明は、デジタル移動無線通信の受信機において使用されるダイバーシチ受信装置に関し、特に、ＬＣＸ（ＬｅａｋｙＣｏａｘｉａｌＣａｂｌｅ：漏洩同軸ケーブル）などを利用して構築される列車無線システムにおける車上局での最大比合成ダイバーシチ受信性能の改善に関するものである。
【０００２】
【従来の技術】
従来のダイバーシチ受信装置における、検波後合成ダイバーシチ処理ブロック図を図２に示す。図２において、１０１−１，１０１−２は列車等の移動体の前後に配置された受信アンテナ（図３参照）、１０２−１，１０２−２は周波数変換回路、１０３−１，１０３−２，１１１−１，１１１−２はアナログ・デジタル変換器（ＡｎａｌｏｇＤｉｇｉｔａｌＣｏｎｖｅｒｔｅｒ、以下ＡＤＣと称する）、１０４−１，１０４−２は直交検波処理部、１０５−１，１０５−２は受信フィルタ処理部、１０６−１，１０６−２は遅延検波処理部、１０７はダイバーシチ合成処理部、１０８は復号処理部、１０９は復号データ出力端子、１１０−１，１１０−２は受信電力検出部、１１２−１，１１２−２は乗算係数計算部を示す。
【０００３】
以下、図２及び図３を用いて、従来技術における２つのアンテナを持つ最大比合成ダイバーシチ受信装置の動作について説明する。
【０００４】
まず、基地局無線機から送信された送信波は、受信アンテナ１０１−１及び１０１−２で受信される。受信アンテナ１０１−１は、受信した無線周波数帯の信号を周波数変換回路１０２−１及び受信電力検出部１１０−１へ出力する。受信アンテナ１０１−２も同様に、無線周波数帯の信号を周波数変換回路１０２−２及び受信電力検出部１１０−２へ信号を出力する。
【０００５】
ここで、周波数変換回路１０２−１に入力された無線周波数帯の信号は、周波数変換回路１０２−１により所要周波数の中間周波数帯信号に変換され、ＡＤＣ１０３−１へ出力される。周波数変換回路１０２−２に入力された無線周波数帯の信号も同様に、中間周波数帯信号に変換されＡＤＣ１０３−２へ出力される。ＡＤＣ１０３−１は、入力信号をサンプリング処理しデジタル信号とした後、直交検波処理部１０４−１へ出力する。ＡＤＣ１０３−２も同様に、入力信号に対しサンプリング処理を行い、デジタル信号を直交検波処理部１０４−２へ出力する。
【０００６】
直交検波処理部１０４−１は、入力されたデジタル信号に対し直交検波を行い、得られた同相成分及び直交成分信号を受信フィルタ処理部１０５−１へ出力する。直交検波処理部１０４−２も同様に、直交検波処理を行い、同相成分及び直交信号成分を受信フィルタ処理部１０５−２へ出力する。受信フィルタ処理部１０５−１は、入力信号に対し周波数帯域制限処理を行い、処理後の同相信号及び直交信号成分を遅延検波処理部１０６−１へ出力する。受信フィルタ処理部１０５−２も同様に、周波数帯域制限処理を行い、処理後の同相信号及び直交信号成分を遅延検波処理部１０６−２へ出力する。
【０００７】
遅延検波処理部１０６−１は、入力信号に対し遅延検波処理を行い、遅延検波された同相信号成分Ｉ_１及び直交信号成分Ｑ_１をダイバーシチ合成処理部１０７へ出力する。遅延検波処理部１０６−２も同様に、遅延検波処理を行い、同相信号成分Ｉ_２及び直交信号成分Ｑ_２をダイバーシチ合成処理部１０７へ出力する。
【０００８】
一方、受信電力検出部１１０−１に入力された無線周波数信号は、受信電力検出部１１０−１により受信帯域の信号成分の電力が検出され、電力の対数値に比例した電圧がＡＤＣ１１１−１へ出力される。受信電力検出部１１０−２に入力された無線周波数信号も同様に、電力の対数値に比例した電圧が検出されＡＤＣ１１１−２へ出力される。ＡＤＣ１１１−１は、入力電圧をサンプリング処理しデジタル信号とした後、乗算係数計算部１１２−１へ出力する。ＡＤＣ１１１−２も同様に、入力電圧をサンプリング処理し、デジタル値を乗算係数計算部１１２−２へ出力する。乗算係数計算部１１２−１は、入力された電力値に正比例した重み付け係数Ｃ_１を算出し、ダイバーシチ合成処理部１０７へ出力する。乗算係数計算部１１２−２も同様に、重み付け係数Ｃ_２を入力電力値から求め、ダイバーシチ合成処理部１０７へ出力する。
【０００９】
そして、ダイバーシチ合成処理部１０７は、最大比合成ダイバーシチ処理を行う。最大比合成は、受信電力が高くＣＮ（ＣａｒｒｉｅｒｔｏＮｏｉｓｅ）比の高い信号の寄与する割合を多くする合成方式であるため、各系の受信電力に比例した重み付け係数を求め、信号に乗じた後加算する。数式で表すと、同相信号成分合成値Ｉ_ｏ＝Ｉ_１×Ｃ_１＋Ｉ_２×Ｃ_２及び直交信号成分合成値Ｑ_ｏ＝Ｑ_１×Ｃ_１＋Ｑ_２×Ｃ_２を求め、復号処理部１０８へ出力する。復号処理部１０８は入力された同相成分Ｉ_ｏ及び直交成分Ｑ_ｏを用い符号判定を行い、復号データ出力端子１０９へ出力する。
【００１０】
上記のように、入力された電力値に正比例した重み付け係数を用いて最大比合成ダイバーシチ処理を行うダイバーシチ受信機として、例えば、特許文献１のようなものがある。
【００１１】
【特許文献１】
特開２０００−１１５０４３号公報
【００１２】
【発明が解決しようとする課題】
図４において、４０１−１，４０１−２は基地局無線機、４０２−１，４０２−２はＬＣＸ、４０３は車上局無線機及び１０１−１，１０１−２は従来例で説明した受信アンテナを示す。車上局４０３は当初、受信アンテナ１０１−１及び１０１−２で、ＬＣＸ４０２−１を介した基地局４０１−１からの送信波を受信し、前述のダイバーシチ処理を行う。今、車上局が進行し、基地局Ａ４０１−１の通信可能領域と基地局Ｂ４０１−２の通信可能領域との境界（以下、ゾーン境界と称する）に差し掛かった場合を考える。この際、進行方向側の受信アンテナ１０１−２は、基地局４０１−１の送信波と４０１−２からの送信波とが干渉しあう領域の信号を受信することになる。基地局４０１−１及び基地局４０１−２は、同一の信号内容を同一のタイミングで送出しているが、伝播距離などの関係からゾーン境界では完全にタイミングが一致せず、受信電界にも差が生じる可能性がある。この場合、受信アンテナ１０１−２で受信される干渉領域の信号は、受信アンテナ１０１−１で受信される信号とほぼ同等の受信電界を持つが、送信波が干渉しているため情報としては正確性に欠ける。この信号に対し最大比合成処理を行うと、例えば図５に示すように、干渉していない信号（Ｉ_１，Ｑ_１）のように、本来は合成後のベクトルが同相・直交成分とも正の値となるべき場合に、一方の信号（Ｉ_２，Ｑ_２）が干渉の影響でベクトルの方向が大きく異なっていると、合成後の信号（Ｉ_ｏ，Ｑ_ｏ）の同相成分Ｉ_ｏが負になってしまうような合成処理が行われる。図５中、各象限が復号するビット（例えば、第一象限が００、第二象限が０１、第三象限が１１、第四象限が１０）に対応しているため、合成処理によりベクトルの方向が異なってしまうと、ベクトルが属する象限も異なってしまい、結果としてビット誤りが生じることになる（上記の例では、００が０１になってしまう）。
【００１３】
本発明の目的は、このようなゾーン境界での合成処理によるビット誤りの発生を軽減する手段を提供することである。
【００１４】
【課題を解決するための手段】
上記目的を達成するため、本発明のダイバーシチ受信装置は、中間周波数帯信号から復調されたｋ組（ｋは２以上の整数）の同相成分信号及び直交成分信号と、その同相成分信号及び直交成分信号に含まれる固定パターンとの相関値をそれぞれ演算するｋ個の相関演算処理部と、前記の演算により求められたｋ個の相関値を基に、ｋ組の同相成分信号及び直交成分信号と固定パターンとの尤もらしさ（尤度）をそれぞれ計算するｋ個の相関尤度計算処理部と、受信した無線信号の電力に正比例したｋ個の重み付け係数に、前記の計算により求められたｋ個の尤度に比例した値を乗ずることにより、ｋ個の重み付け係数をそれぞれ補正するｋ個の乗算係数補正部とを具備するようにしたものである。
【００１５】
【発明の実施の形態】
デジタル無線通信において、送信波は図６に示すようなフレーム構造であることが一般的であり、フレーム構造中には同期ワードという固定パターンのデータが存在する。本発明では、干渉による信号の影響を検出するために、この同期ワードを利用する。
【００１６】
同期ワード受信が想定される前後の区間で、あらかじめ用意した理想状態の同期ワード遷移パターンと受信信号との相関自乗演算を行う。相関自乗演算処理は、時点ｔを演算開始点、受信信号同相信号成分をＲＩ（ｔ）、受信信号直交信号成分をＲＱ（ｔ）、理想状態の同期ワード同相信号成分をＳＩ（ｊ）（ｊは１からｎまでの整数、ｎは同期ワードシンボル数×オーバーサンプル数）、理想状態の同期ワード直交信号成分をＳＱ（ｊ）、及び演算実施区間をｍとすると、求める相関自乗値ＣＲ（ｉ）（ｉは０からｍ−１までの整数）は以下の式で求められる。
【００１７】
【数１】

今、同期ワードシンボル数を１０シンボル、オーバーサンプル数を４とし、演算を同期ワードの前後１シンボル（演算区間長ｍ＝（１（前シンボル分）＋１（後シンボル分））×４（オーバーサンプル数）＋１（中心時）＝９）に渡って実行した場合、例えば非干渉領域では図７、干渉領域では図８に示すような結果を得ることができる。同期ワードが正確に受信されている場合は、図７に示すように明確なピークが存在し、この時点で同期ワードが受信したと判定できる。対して、干渉などにより信号成分が劣化している場合は、図８に示すようにピークが現れないか、表われても値は非常に小さなものとなる。この相関結果ＣＲ（ｉ）の尤もらしさ（以下、尤度と称する）を求めるため、図９に示すような理想状態での相関自乗演算結果ＣＩ（ｉ）（ｉは０からｍ−１までの整数）を用い、（式２）に示す演算により尤度Ｌを求める。
【００１８】
【数２】

尤度ＬはＣＲ（ｉ）が理想状態に近いほど値が大きくなり、干渉などが発生した場合、ＣＲ（ｉ）は理想状態と異なる値をとるため、尤度Ｌは小さくなる。よって、最大比合成を行う際の乗算係数をこの尤度に比例して変動させることにより、尤度が低い系の信号成分を減少させ、合成による劣化を軽減することが可能となる。
【００１９】
図１に本発明に基づくダイバーシチ受信装置の一実施例を示す。図１において、１１３−１，１１３−２は同期ワード相関演算処理部、１１４−１，１１４−２は相関尤度計算処理部及び１１５−１，１１５−２は乗算係数補正部を示し、他の従来例と同様の動作をするものに関しては、従来例と同一の番号を付した。
【００２０】
以下、図１を用いて、本実施例における最大比合成ダイバーシチ受信装置の動作について説明する。
【００２１】
まず、基地局無線機から送信された送信波は、受信アンテナ１０１−１及び１０１−２で受信される。受信アンテナ１０１−１は、受信した無線周波数帯の信号を周波数変換回路１０２−１及び受信電力検出部１１０−１へ出力する。受信アンテナ１０１−２も同様に、無線周波数帯の信号を周波数変換回路１０２−２及び受信電力検出部１１０−２へ信号を出力する。
【００２２】
ここで、周波数変換回路１０２−１に入力された無線周波数帯の信号は、周波数変換回路１０２−１により所要周波数の中間周波数帯信号に変換され、ＡＤＣ１０３−１へ出力される。周波数変換回路１０２−２に入力された無線周波数帯の信号も同様に、中間周波数帯信号に変換されＡＤＣ１０３−２へ出力される。ＡＤＣ１０３−１は、入力信号をサンプリング処理しデジタル信号とした後、直交検波処理部１０４−１へ出力する。ＡＤＣ１０３−２も同様に、入力信号に対しサンプリング処理を行い、デジタル信号を直交検波処理部１０４−２へ出力する。
【００２３】
直交検波処理部１０４−１は、入力されたデジタル信号に対し直交検波を行い、得られた同相成分及び直交成分信号を受信フィルタ処理部１０５−１へ出力する。直交検波処理部１０４−２も同様に、直交検波処理を行い、同相成分及び直交信号成分を受信フィルタ処理部１０５−２へ出力する。受信フィルタ処理部１０５−１は、入力信号に対し周波数帯域制限処理を行い、処理後の同相信号及び直交信号成分を遅延検波処理部１０６−１及び同期ワード相関演算処理部１１３−１へ出力する。受信フィルタ処理部１０５−２も同様に、フィルタ処理を行い、処理後の同相信号及び直交信号成分を遅延検波処理部１０６−２及び同期ワード相関演算処理部１１３−２へ出力する。
【００２４】
一方、受信電力検出部１１０−１に入力された無線周波数信号は、受信電力検出部１１０−１により受信帯域の信号成分の電力が検出され、電力の対数値に比例した電圧がＡＤＣ１１１−１へ出力される。受信電力検出部１１０−２に入力された無線周波数信号も同様に、電力の対数値に比例した電圧が検出されＡＤＣ１１１−２へ出力される。ＡＤＣ１１１−１は、入力電圧をサンプリング処理しデジタル信号とした後、乗算係数計算部１１２−１へ出力する。ＡＤＣ１１１−２も同様に、入力電圧をサンプリング処理し、デジタル値を乗算係数計算部１１２−２へ出力する。乗算係数計算部１１２−１は、入力された電力値に正比例した重み付け係数Ｃ_１を算出し、乗算係数補正部１１５−１へ出力する。乗算係数計算部１１２−２も同様に、重み付け係数Ｃ_２を入力電力値から求め、乗算係数補正部１１５−２へ出力する。
【００２５】
ところで、遅延検波処理部１０６−１に入力された同相信号及び直交信号成分は、遅延検波処理部１０６−１により遅延検波処理が行われ、遅延検波された同相信号成分Ｉ_１及び直交信号成分Ｑ_１はダイバーシチ合成処理部１０７へ出力される。遅延検波処理部１０６−２に入力された同相信号及び直交信号成分も同様に、遅延検波処理が行われ、同相信号成分Ｉ_２及び直交信号成分Ｑ_２がダイバーシチ合成処理部１０７へ出力される。
【００２６】
一方、同期ワード相関演算処理部１１３−１に入力された同相信号及び直交信号成分は、あらかじめ用意した理想状態の同期ワード遷移パターンとの（式１）に示す相関自乗演算処理が行われ、その結果が相関尤度計算処理部１１４−１へ出力される。同期ワード相関演算処理部１１３−２に入力された同相信号及び直交信号成分も同様に、（式１）に示す相関自乗演算処理が行われ、その結果が相関尤度計算処理部１１４−２へ出力される。相関尤度計算処理部１１４−１では、入力された相関自乗演算結果とあらかじめ用意した理想状態の同期ワード相関自乗値の出力パターンを用い（式２）に示す演算を行い、その演算値を尤度Ｌ_１として乗算係数補正部１１５−１へ出力する。相関尤度計算処理部１１４−２も同様に、（式２）に示す演算を行い、演算値を尤度Ｌ２として乗算係数補正部１１５−２へ出力する。
【００２７】
乗算係数補正部１１５−１では、尤度Ｌ_１と基準値を比較し、乗算係数の補正量を決定する。補正量は尤度の大きさに比例するものとし、補正後の乗算係数Ｃ_１’を求め、ダイバーシチ合成処理部１０７へ出力する。乗算係数補正部１１５−２も同様に、乗算係数Ｃ_２を尤度Ｌ２に従い補正し、補正後乗算係数Ｃ_２’を求め、ダイバーシチ合成処理部１０７へ出力する。
【００２８】
そして、ダイバーシチ合成処理部１０７では、同相信号成分合成値Ｉ_ｏ＝Ｉ_１×Ｃ_１’＋Ｉ_２×Ｃ_２’及び直交信号成分合成値Ｑ_ｏ＝Ｑ_１×Ｃ_１’＋Ｑ_２×Ｃ_２’を求め、復号処理部１０８へ出力する。復号処理部１０８は、入力された同相成分Ｉ_ｏ及び直交成分Ｑ_ｏを用い符号判定を行い、復号データ出力端子１０９へ出力する。
【００２９】
上記のような処理を行うことにより、例えば図１０に示すように、干渉していない信号（Ｉ_１，Ｑ_１）の合成後のベクトルが同相・直交成分とも正の値となっていて、一方の信号（Ｉ_２，Ｑ_２）が干渉の影響でベクトルの方向が大きく異なっていても、干渉信号側の尤度Ｌ_２が低い値となるため、補正後の乗算係数Ｃ_２’も小さい値となる。よって、干渉信号の影響が減り、合成後の信号（Ｉ_ｏ，Ｑ_ｏ）についても同相・直交成分とも正の値となり、ビット誤りが発生しない正確な合成処理を行うことが可能となる。
【００３０】
【発明の効果】
本発明により、ゾーン境界において、干渉により情報が不正確になった送信信号の影響を軽減したダイバーシチ合成処理を実現することが可能となる。このことにより、ゾーン境界での合成処理によるビット誤りの発生を軽減することができる。
【図面の簡単な説明】
【図１】提案法による検波後合成ダイバーシチ処理ブロック図。
【図２】従来法による検波後合成ダイバーシチ処理ブロック図。
【図３】受信アンテナ配置例を示した図。
【図４】受信状態例を示した図。
【図５】従来法でのゾーン境界時信号合成の一例を示した図。
【図６】フレーム構成例を示した図。
【図７】非干渉領域での同期ワード相関自乗演算結果例を示した図。
【図８】干渉領域での同期ワード相関自乗演算結果例を示した図。
【図９】理想状態での同期ワード相関自乗演算結果例を示した図。
【図１０】提案法でのゾーン境界時信号合成の一例を示した図。
【符号の説明】１０１−１，１０１−２：受信アンテナ、１０２−１，１０２−２：周波数変換回路、１０３−１，１０３−２，１１１−１，１１１−２：アナログ・デジタル変換器（ＡＤＣ：ＡｎａｌｏｇＤｉｇｉｔａｌＣｏｎｖｅｒｔｅｒ）、１０４−１，１０４−２：直交検波処理部、１０５−１，１０５−２：受信フィルタ処理部、１０６−１，１０６−２：遅延検波処理部、１０７：ダイバーシチ合成処理部、１０８：復号処理部、１０９：復号データ出力端子、１１０−１，１１０−２：受信電力検出部、１１２−１，１１２−２：乗算係数計算部、１１３−１，１１３−２：同期ワード相関演算処理部、１１４−１，１１４−２：相関尤度計算処理部、１１５−１，１１５−２：乗算係数補正部、４０１−１，４０１−２：基地局無線機、４０２−１，４０２−２：ＬＣＸ、４０３：車上局無線機。[0001]
TECHNICAL FIELD OF THE INVENTION
The present invention relates to a diversity receiving apparatus used in a digital mobile radio communication receiver, and more particularly to a diversity receiving apparatus used in a train radio system constructed using an LCX (Leaky Coaxial Cable). The present invention relates to the improvement of the ratio combining diversity reception performance.
[0002]
[Prior art]
FIG. 2 is a block diagram showing a post-detection combining diversity process in a conventional diversity receiving apparatus. In FIG. 2, 101-1 and 101-2 are receiving antennas (see FIG. 3) arranged before and after a moving object such as a train, 102-1 and 102-2 are frequency conversion circuits, and 103-1 and 103-2. , 111-1 and 111-2 denote analog-to-digital converters (ADCs), 104-1 and 104-2 denote quadrature detection processing units, and 105-1 and 105-2 denote reception filter processing units. , 106-1 and 106-2 are delay detection processing units, 107 is a diversity combining processing unit, 108 is a decoding processing unit, 109 is a decoded data output terminal, 110-1 and 110-2 are received power detection units, 112-1 , 112-2 indicate a multiplication coefficient calculator.
[0003]
Hereinafter, the operation of the maximum ratio combining diversity receiver having two antennas in the related art will be described with reference to FIGS. 2 and 3.
[0004]
First, the transmission wave transmitted from the base station radio is received by the receiving antennas 101-1 and 101-2. The receiving antenna 101-1 outputs the received signal in the radio frequency band to the frequency conversion circuit 102-1 and the received power detection unit 110-1. Similarly, the reception antenna 101-2 outputs a signal in the radio frequency band to the frequency conversion circuit 102-2 and the reception power detection unit 110-2.
[0005]
Here, the radio frequency band signal input to the frequency conversion circuit 102-1 is converted into an intermediate frequency band signal of a required frequency by the frequency conversion circuit 102-1 and output to the ADC 103-1. Similarly, the radio frequency band signal input to the frequency conversion circuit 102-2 is converted into an intermediate frequency band signal and output to the ADC 103-2. The ADC 103-1 samples the input signal and converts it into a digital signal, and then outputs the digital signal to the quadrature detection processing unit 104-1. Similarly, the ADC 103-2 performs a sampling process on the input signal and outputs a digital signal to the quadrature detection processing unit 104-2.
[0006]
Quadrature detection processing section 104-1 performs quadrature detection on the input digital signal, and outputs the obtained in-phase component and quadrature component signals to reception filter processing section 105-1. Similarly, quadrature detection processing section 104-2 performs quadrature detection processing, and outputs in-phase components and quadrature signal components to reception filter processing section 105-2. Receiving filter processing section 105-1 performs frequency band limiting processing on the input signal, and outputs the processed in-phase signal and quadrature signal component to differential detection processing section 106-1. Similarly, reception filter processing section 105-2 performs frequency band limiting processing, and outputs the processed in-phase signal and quadrature signal component to differential detection processing section 106-2.
[0007]
Delay detection processing section 106-1 performs delay detection processing on the input signal, and outputs an in-phase signal component delayed detection I ₁ and the quadrature signal component Q ₁ to diversity combining section 107. Similarly, delay detection processing section 106-2 performs delay detection processing, and outputs an in-phase signal components I ₂ and the quadrature signal component Q ₂ to diversity combining section 107.
[0008]
On the other hand, in the radio frequency signal input to the reception power detection unit 110-1, the power of the signal component in the reception band is detected by the reception power detection unit 110-1 and a voltage proportional to the logarithmic value of the power is sent to the ADC 111-1. Is output. Similarly, a voltage proportional to the logarithmic value of the power is detected from the radio frequency signal input to the reception power detection unit 110-2 and output to the ADC 111-2. The ADC 111-1 performs a sampling process on the input voltage to generate a digital signal, and outputs the digital signal to the multiplication coefficient calculation unit 112-1. Similarly, the ADC 111-2 samples the input voltage and outputs a digital value to the multiplication coefficient calculator 112-2. Multiplication coefficient calculation unit 112-1 calculates the weighting factors C ₁ which is directly proportional to the input power value, and outputs it to diversity combining section 107. Similarly, the multiplier coefficient calculating section 112-2 calculates a weighting coefficient _{C 2} from the input power value, and outputs it to diversity combining section 107.
[0009]
Then, diversity combining processing section 107 performs maximum ratio combining diversity processing. Since the maximum ratio combining is a combining method that increases the contribution ratio of a signal having a high received power and a high CN (Carrier to Noise) ratio, a weighting coefficient proportional to the received power of each system is obtained, and after multiplying the signal, to add. Expressed in a formula to obtain the in-phase signal component synthesis value _{_{_{I o = I 1 × C 1}}} + I 2 × C 2 and the orthogonal signal component synthesis value _{_{_{Q o = Q 1 × C 1}}} + Q 2 × C 2, the decoding processing unit 108 Output to Decoding unit 108 performs the code judgment using the in-phase component I _o and a quadrature component Q _o inputted, and outputs the decoded data output terminal 109.
[0010]
As described above, as a diversity receiver that performs the maximum ratio combining diversity process using the weighting coefficient that is directly proportional to the input power value, for example, there is one as disclosed in Patent Document 1.
[0011]
[Patent Document 1]
JP 2000-115043 A
[Problems to be solved by the invention]
4, reference numerals 401-1 and 401-2 denote base station radios, reference numerals 402-1 and 402-2 denote LCXs, reference numeral 403 denotes an onboard radio, and reference numerals 101-1 and 101-2 denote receiving antennas described in the conventional example. Is shown. First, the on-board station 403 receives the transmission wave from the base station 401-1 via the LCX 402-1 with the receiving antennas 101-1 and 101-2, and performs the above-described diversity processing. Now, consider a case in which the on-vehicle station advances and approaches a boundary between the communicable area of the base station A 401-1 and the communicable area of the base station B 401-2 (hereinafter, referred to as a zone boundary). At this time, the receiving antenna 101-2 on the traveling direction side receives a signal in an area where the transmission wave from the base station 401-1 and the transmission wave from the 401-2 interfere with each other. Although the base station 401-1 and the base station 401-2 transmit the same signal content at the same timing, the timing does not completely match at the zone boundary due to the propagation distance and the like, and the reception electric field also differs. May occur. In this case, the signal in the interference area received by the reception antenna 101-2 has a reception electric field substantially equal to the signal received by the reception antenna 101-1. However, since the transmission wave interferes, the information is accurate. Lacks sex. When the maximum ratio combining process is performed on this signal, as shown in FIG. 5, for example, as in the case of a non-interfering signal (I ₁ , Q ₁ ), the vector after the combination is originally positive for both in-phase and quadrature components. If one of the signals (I ₂ , Q ₂ ) has significantly different vector directions due to the influence of interference, the in-phase component I _{o of} the combined signal (I _o , Q _o ) is negative. Is performed. In FIG. 5, since each quadrant corresponds to the bit to be decoded (for example, the first quadrant is 00, the second quadrant is 01, the third quadrant is 11, and the fourth quadrant is 10, the direction of the vector is determined by the synthesis processing. Is different, the quadrant to which the vector belongs is also different, resulting in a bit error (00 becomes 01 in the above example).
[0013]
An object of the present invention is to provide means for reducing the occurrence of bit errors due to such a synthesis process at a zone boundary.
[0014]
[Means for Solving the Problems]
In order to achieve the above object, a diversity receiving apparatus according to the present invention comprises: k sets (k is an integer of 2 or more) of in-phase component signals and quadrature component signals demodulated from an intermediate frequency band signal; K correlation calculation processing units for calculating correlation values with a fixed pattern included in the signal, and k sets of in-phase component signals and quadrature component signals based on the k correlation values obtained by the above calculation, K correlation likelihood calculation processing units for respectively calculating likelihood (likelihood) with a fixed pattern, and k weighting coefficients directly proportional to the power of a received radio signal, and k number of weighting coefficients obtained by the above calculation. And k multiplying coefficient correction units for respectively correcting the k weighting coefficients by multiplying by a value proportional to the likelihood.
[0015]
BEST MODE FOR CARRYING OUT THE INVENTION
In digital wireless communication, a transmission wave generally has a frame structure as shown in FIG. 6, and a fixed pattern of data called a synchronization word exists in the frame structure. In the present invention, this synchronization word is used to detect the influence of the signal due to the interference.
[0016]
In a section before and after a synchronization word is assumed to be received, a correlation square calculation between a synchronization signal transition pattern in an ideal state prepared in advance and a received signal is performed. In the correlation square calculation processing, a time point t is a calculation start point, a received signal in-phase signal component is RI (t), a received signal quadrature signal component is RQ (t), and an ideal synchronization word in-phase signal component is SI (j). (J is an integer from 1 to n, n is the number of synchronization word symbols × the number of oversamples), SQ (j) is a synchronization word orthogonal signal component in an ideal state, and m is a correlation execution value CR to be calculated. (I) (i is an integer from 0 to m-1) is obtained by the following equation.
[0017]
(Equation 1)

Now, assuming that the number of synchronization word symbols is 10 and the number of oversamples is 4, the calculation is performed for one symbol before and after the synchronization word (calculation section length m = (1 (for the previous symbol) +1 (for the next symbol)) × 4 (for oversampling). When the processing is performed over (number) +1 (at the center) = 9), for example, a result as shown in FIG. 7 can be obtained in a non-interference area, and a result as shown in FIG. 8 can be obtained in an interference area. When the synchronization word is correctly received, there is a clear peak as shown in FIG. 7, and it can be determined at this point that the synchronization word has been received. On the other hand, when the signal component is degraded due to interference or the like, the peak does not appear as shown in FIG. 8 or the value becomes very small even if it appears. In order to determine the likelihood (hereinafter, referred to as likelihood) of the correlation result CR (i), the correlation square calculation result CI (i) (i is 0 to m−1) in an ideal state as shown in FIG. (Integer) and the likelihood L is calculated by the operation shown in (Equation 2).
[0018]
(Equation 2)

The value of likelihood L increases as CR (i) approaches the ideal state, and when interference or the like occurs, CR (i) takes a value different from the ideal state, so that likelihood L decreases. Therefore, by varying the multiplication coefficient in performing the maximum ratio combining in proportion to the likelihood, it is possible to reduce the signal components of the system having a low likelihood and reduce the deterioration due to the combining.
[0019]
FIG. 1 shows an embodiment of the diversity receiving apparatus according to the present invention. In FIG. 1, reference numerals 113-1 and 113-2 denote synchronous word correlation operation processing units, 114-1 and 114-2 denote correlation likelihood calculation processing units, and 115-1 and 115-2 denote multiplication coefficient correction units. The same operations as those of the conventional example are denoted by the same reference numerals.
[0020]
Hereinafter, the operation of the maximum ratio combining diversity receiver according to this embodiment will be described with reference to FIG.
[0021]
First, the transmission wave transmitted from the base station radio is received by the receiving antennas 101-1 and 101-2. The receiving antenna 101-1 outputs the received signal in the radio frequency band to the frequency conversion circuit 102-1 and the received power detection unit 110-1. Similarly, the reception antenna 101-2 outputs a signal in the radio frequency band to the frequency conversion circuit 102-2 and the reception power detection unit 110-2.
[0022]
Here, the radio frequency band signal input to the frequency conversion circuit 102-1 is converted into an intermediate frequency band signal of a required frequency by the frequency conversion circuit 102-1 and output to the ADC 103-1. Similarly, the radio frequency band signal input to the frequency conversion circuit 102-2 is converted into an intermediate frequency band signal and output to the ADC 103-2. The ADC 103-1 samples the input signal and converts it into a digital signal, and then outputs the digital signal to the quadrature detection processing unit 104-1. Similarly, the ADC 103-2 performs a sampling process on the input signal and outputs a digital signal to the quadrature detection processing unit 104-2.
[0023]
Quadrature detection processing section 104-1 performs quadrature detection on the input digital signal, and outputs the obtained in-phase component and quadrature component signals to reception filter processing section 105-1. Similarly, quadrature detection processing section 104-2 performs quadrature detection processing, and outputs in-phase components and quadrature signal components to reception filter processing section 105-2. Receiving filter processing section 105-1 performs frequency band limiting processing on the input signal, and outputs the processed in-phase signal and quadrature signal component to differential detection processing section 106-1 and synchronous word correlation calculation processing section 113-1. I do. Similarly, reception filter processing section 105-2 performs filter processing, and outputs the processed in-phase signal and quadrature signal component to differential detection processing section 106-2 and synchronous word correlation calculation processing section 113-2.
[0024]
On the other hand, in the radio frequency signal input to the reception power detection unit 110-1, the power of the signal component in the reception band is detected by the reception power detection unit 110-1 and a voltage proportional to the logarithmic value of the power is sent to the ADC 111-1. Is output. Similarly, a voltage proportional to the logarithmic value of the power is detected from the radio frequency signal input to the reception power detection unit 110-2 and output to the ADC 111-2. The ADC 111-1 performs a sampling process on the input voltage to generate a digital signal, and outputs the digital signal to the multiplication coefficient calculation unit 112-1. Similarly, the ADC 111-2 samples the input voltage and outputs a digital value to the multiplication coefficient calculator 112-2. Multiplication coefficient calculation unit 112-1 calculates the weighting factors C ₁ which is directly proportional to the input power value and outputs to the multiplier coefficient correction unit 115-1. Similarly, the multiplier coefficient calculating section 112-2 calculates a weighting coefficient _{C 2} from the input power value, and outputs to the multiplier coefficient correction unit 115-2.
[0025]
Meanwhile, in-phase signal and quadrature signal components input to the delay detection processing section 106-1, delay detection processing by delay detection processing section 106-1 is performed, differential detection in-phase signal components I ₁ and the quadrature signal component _{Q 1} is output to diversity combining section 107. Similarly, in-phase signal and the orthogonal signal component input to the delay detection processing section 106-2, delay detection processing is performed, the in-phase signal component I ₂ and the quadrature signal component Q ₂ is output to diversity combining section 107 You.
[0026]
On the other hand, the in-phase signal and the quadrature signal component input to the synchronous word correlation operation processing unit 113-1 are subjected to a correlation square operation shown in (Equation 1) with an idealized synchronous word transition pattern prepared in advance, The result is output to correlation likelihood calculation processing section 114-1. Similarly, the in-phase signal and the quadrature signal component input to the synchronous word correlation operation processing unit 113-2 are subjected to the correlation square operation processing shown in (Equation 1), and the result is used as the correlation likelihood calculation processing unit 114-2. Output to The correlation likelihood calculation processing unit 114-1 performs the calculation shown in (Equation 2) using the input correlation square calculation result and the output pattern of the synchronous word correlation square value in the ideal state prepared in advance, and calculates the calculated value as the likelihood. output as degrees _{L 1} to the multiplier coefficient correction unit 115-1. Similarly, the correlation likelihood calculation processing unit 114-2 performs the calculation shown in (Equation 2) and outputs the calculated value to the multiplication coefficient correction unit 115-2 as the likelihood L2.
[0027]
The multiplier coefficient correction unit 115-1 compares the likelihoods L ₁ and the reference value, determines a correction amount of the multiplication factor. The correction amount is assumed to be proportional to the likelihood, and a corrected multiplication coefficient C ₁ ′ is obtained and output to the diversity combining processing unit 107. Similarly multiplier coefficient correction unit 115-2, the multiplier coefficient _{C 2} corrected in accordance likelihood L2, multiplication coefficients determine the _{C 2} 'after the correction, and outputs it to the diversity combining unit 107.
[0028]
Then, the diversity combining unit 107, in-phase signal component combined value _{_{_{I o = I 1 × C 1}}} '+ I 2 × C 2' and the orthogonal signal component synthesis value _{_{_{Q o = Q 1 × C 1}}} '+ Q 2 × C 2 'And outputs it to the decoding processing unit 108. Decoding unit 108 performs a code determination using an in-phase component I _o and a quadrature component Q _o inputted, and outputs the decoded data output terminal 109.
[0029]
By performing the processing as described above, as shown in FIG. 10, for example, the combined vector of the non-interfering signals (I ₁ , Q ₁ ) has positive values for both the in-phase and quadrature components. Even if the signals (I ₂ , Q ₂ ) have significantly different vector directions due to the influence of interference, the likelihood L ₂ on the interference signal side becomes a low value, and thus the corrected multiplication coefficient C ₂ ′ is also a small value. It becomes. Accordingly, the influence of the interference signal is reduced, and the in-phase and quadrature components of the combined signal (I _o , Q _o ) also have positive values, and it is possible to perform an accurate combining process that does not cause a bit error.
[0030]
【The invention's effect】
ADVANTAGE OF THE INVENTION According to this invention, it becomes possible to implement | achieve the diversity combining process which reduced the influence of the transmission signal whose information became inaccurate by interference at the zone boundary. This can reduce the occurrence of bit errors due to the combining process at the zone boundary.
[Brief description of the drawings]
FIG. 1 is a block diagram of a post-detection combining diversity process according to a proposed method.
FIG. 2 is a block diagram of a post-detection combining diversity processing according to a conventional method.
FIG. 3 is a diagram showing an example of a receiving antenna arrangement.
FIG. 4 is a diagram showing an example of a reception state.
FIG. 5 is a diagram showing an example of signal synthesis at the time of zone boundary in the conventional method.
FIG. 6 is a diagram showing an example of a frame configuration.
FIG. 7 is a diagram illustrating an example of a result of a synchronous word correlation square calculation in a non-interference area.
FIG. 8 is a diagram showing an example of a result of a synchronous word correlation square calculation in an interference area.
FIG. 9 is a view showing an example of a result of a synchronous word correlation square calculation in an ideal state;
FIG. 10 is a diagram showing an example of signal synthesis at the time of zone boundary in the proposed method.
[Description of References] 101-1 and 101-2: receiving antennas, 102-1 and 102-2: frequency conversion circuits, 103-1 and 103-2, 111-1, and 111-2: analog-to-digital converters ( ADC: Analog Digital Converter), 104-1, 104-2: Quadrature detection processing unit, 105-1, 105-2: Reception filter processing unit, 106-1, 106-2: Delay detection processing unit, 107: Diversity combining Processing unit, 108: decoding processing unit, 109: decoded data output terminal, 110-1, 110-2: reception power detection unit, 112-1, 112-2: multiplication coefficient calculation unit, 113-1, 113-2: Synchronous word correlation operation processing units, 114-1, 114-2: correlation likelihood calculation processing units, 115-1, 115-2: multiplication coefficient correction units, 401-1, 401-2: Chikyoku radio, 402-1,402-2: LCX, 403: car top station radio equipment.

Claims

K (k is an integer of 2 or more) receiving antennas for receiving radio signals,
K frequency conversion circuits for respectively converting the radio signals received by the k reception antennas to k intermediate frequency band signals,
K quadrature detection processing units for demodulating the k intermediate frequency band signals into k sets of in-phase component signals and quadrature component signals,
K reception power detection units that respectively detect the power of the radio signal received by the k reception antennas,
K multiplying coefficient calculators for calculating weighting coefficients in direct proportion to the k powers,
A diversity receiving apparatus comprising: a diversity combining processing unit that performs a maximum ratio combining diversity process by multiplying each of the k sets of in-phase component signals and the quadrature component signals by the k weighting coefficients and then adding them.
K correlation calculation processing units for calculating correlation values of the k sets of in-phase component signals and quadrature component signals and fixed patterns included in the in-phase component signals and quadrature component signals, respectively;
K correlation likelihood calculation processing units for respectively calculating likelihood (likelihood) between the k sets of in-phase component signals and quadrature component signals and the fixed pattern based on the k correlation values;
A diversity receiving apparatus comprising: k multiplying coefficient correction units for correcting the k weighting coefficients by multiplying the k weighting coefficients by a value proportional to the k likelihoods. .