JP2004129237A - Radio equipment - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To reduce power consumption of radio equipment, to improve the efficiency of output power and to remove a DC offset which becomes problems in the radio equipment of output power control and direct conversion system. <P>SOLUTION: A band of a loop filter of a synthesizer is made narrower during an idle channel search in comparison with during a call. Besides, a radio wave environment is observed, characteristics required for the radio equipment in accordance with the radio wave environment are judged by itself, power consumption is reduced in accordance with performance, current consumption of a power amplifier PA is observed and a matching circuit (LNA or MIX) of an antenna 101 is adjusted to reduce a loss of the antenna. Besides, the DC offset is removed from transmission power and reflected waves during transmission. When removing the DC offset by using an AC coupling capacitor, deterioration of frequency characteristics of a receiving system including the capacitor is corrected by digital processing. A transmission power detecting part is integrated in an IC, and a detected signal is defined as power to be leaked to a power source or the ground of the IC. <P>COPYRIGHT: (C)2004,JPO

Description

The present invention relates to a wireless device such as a portable wireless terminal.

In recent years, demand for portable wireless terminals (hereinafter referred to as portable terminals) has been increasing, and high-performance mobile terminals have been actively commercialized.

傾向 Trends in commercialization include miniaturization that is easy for people to carry, low power consumption that reduces current consumption and long talk time, and high linearity that is resistant to interference.

提供 Not all problems have been solved in providing such products to the market, and further research is currently underway to solve individual problems.

Hereinafter, a conventional wireless device will be described with reference to FIGS.

FIG. 35 shows a transmission / reception configuration of a conventional heterodyne wireless device.

In the description of this wireless device, as the receiving system, an example of a receiving system that is frequently used as a configuration of quadrature quasi-coherent detection using a quadrature demodulation method will be described as an example. This wireless device is based on the premise that a linear modulation method (π / 4 shift QPSK modulation method) used in, for example, PHS is used.

In the case of this radio, a high-frequency signal received by an antenna 101 is given a predetermined gain by a high-frequency amplifier 102 for the purpose of improving NF of a radio unit, and then passes through a high-frequency filter 103 for image suppression, The reference carrier signal supplied from the local oscillator 125 is multiplied by the frequency converter 104 and frequency-converted to an intermediate frequency. Thereafter, the frequency is converted to a baseband frequency by a quadrature demodulation unit including the frequency converters 105 and 106, the local oscillator 107, and the π / 2 phase shifter 108. That is, a reference signal having substantially the same frequency as the intermediate frequency transmitted from the local oscillator 107 and having a phase difference of π / 2, and a baseband signal of ΙQ are obtained by the two frequency converters 105 and 106. The ΙQ baseband signal is subjected to anti-aliasing processing for the A / D converters 113 and 114 at the subsequent stage by the low-frequency filters 109 and 110 (filtering). In some cases, channel selection is performed by the low frequency filters 109 and 110.

The filtered IQ signal is given a predetermined gain by the low frequency amplifiers 111 and 112. Thereafter, the signal is converted into a digital signal by the A / D converters 113 and 114, and is demodulated into a data signal by the detector 117. In FIG. 35, capacitors 119 to 124 at the subsequent stage of the frequency converters 105 and 106 are inserted for the purpose of removing DC (direct current) components.

Next, the transmission system of the conventional wireless device will be described.

The orthogonal digital signals Ich and Qch from the baseband signal generator are converted into analog signals by the respective digital / analog converters D / A 161 and 162. The respective signals converted here are input to the respective root roll-off filters 159 and 160, thereby removing unnecessary waves generated in the D / A 161 and 162 and performing a first operation for removing interference between symbols. Is performed. The second operation after the first operation is performed by a roll-off filter of the receiving system.

The output of the local oscillator is modulated by a baseband signal by a quadrature modulator including multipliers 157 and 158, a local oscillator 163 (synthesizer), a π / 2 phase shifter 163, and an adder 156 to generate an IF signal. .

ＩＦThis IF signal is input to the local oscillator 125 and the frequency converter 154 via the low-pass filter 155 in order to convert the frequency to the same frequency as the transmission frequency, and is converted to an RF signal. The RF signal is output from the antenna 101 via the band pass filter 153, the variable attenuator 152, the power amplifier 151, the directional coupler 172, the band pass filter 150, and the transmission / reception switch 170. The transmission power is sensed by a directional coupler 172 between the power amplifier 151 and the filter 150, and based on the power detected by the power detector 173, the transmission power control signal is transmitted from the control signal unit 171 to the variable attenuator 152. Is entered. The low-pass filters and band-pass filters 155, 153, and 150 are used to avoid unnecessary radiation.

Hereinafter, the problems of such a conventional wireless device will be described in the order of a receiving system, a synthesizer (meaning the local oscillator described above), a transmitting system, and an antenna.

First, the problems of the conventional receiving system will be described.

There are two problems with the conventional receiving system. The first problem is that the current consumption increases. The second problem is that a DC offset (hereinafter referred to as DC offset) occurs at the time of reception. Deterioration of characteristics.

First, the first problem will be described. Since the receiving characteristics of the wireless device must always satisfy the performance required for the system, the wireless device is designed to be able to cope with the worst radio wave environment in which the wireless device is placed. An example of the worst radio wave environment here is a case where an unnecessary wave defined by, for example, intermodulation characteristics or adjacent channel selectivity exists. That is, when an unnecessary signal other than the desired signal exists in the system band, the level of the unnecessary signal is a maximum value that satisfies a required bit error rate defined by the system.

Generally, in order to satisfy the standard value defined by the system even in the worst radio wave environment, the radio always operates under the worst conditions. Therefore, even in environments other than the worst radio wave environment, the radio device is operated with performance satisfying the worst condition. In order to satisfy the standards even in the worst radio wave environment, it is necessary to maintain the linearity of the receiving system of the radio. In other words, it is necessary to reduce the distortion of the receiving system so as to satisfy the standard. This is related to the operating currents of the circuit blocks constituting the receiving system, mainly the low-noise amplifier and the frequency converter.

Generally, to increase the linearity of a circuit, it is necessary to increase the operating current of the circuit. For this reason, a wireless device that always assumes the worst radio wave environment consumes more power than necessary. This is because the wireless device is not always in the worst radio wave environment, and most of the time when the radio device is operating is outside the worst radio wave environment. Next, a second problem of the receiving system in which a direct current offset (hereinafter, referred to as a DC offset) occurs at the time of reception to degrade the reception characteristics will be described.

Generally, in an active circuit such as the above-described frequency converter, low-frequency filter, low-frequency amplifier, etc., a DC component is generated by superimposing a desired signal on its output. This DC component is also generated by self-mixing.

Here, the phenomenon of self-mixing will be described with reference to FIG.

As shown in FIG. 36A, the output of the local oscillator 107 leaks and is reflected by the frequency converter 104, and the reflected component 401 is multiplied again by the local oscillator 107 by the frequency converter 105. Since the reflected component 401 and the signal oscillated from the local oscillator 107 have the same frequency, a DC component is generated at the output of the frequency converter 105. FIG. 37 shows an output (desired signal) 301 of the frequency converter 105 on which the DC output 305 is superimposed. The line indicated by reference numeral 304 indicates the level of thermal noise.

Since this DC offset output causes a remarkable deterioration of the reception error rate, a conventional AC coupling (AC coupling) capacitor (reference numeral in FIG. 119-124) are inserted into the baseband stage.

ＤＣIn FIG. 38A, the DC output 305 can be eliminated by inserting a capacitor so that the AC couple frequency characteristic becomes 302. However, a part 303 of the desired signal component in the desired signal 301 is deleted. That is, a notch 306 shown in FIG. This notch naturally occurs in the thermal noise 304 (307). For this reason, the C / N characteristic may be improved in the case of an FSK signal having a small modulation factor and a high modulation index near DC, for example.

方式 A method of effectively using the AC couple to remove DC offset has already been proposed, and this method can be effectively used particularly in a binary FSK having a high modulation index used in a conventional pager or the like. This is because the signal components near DC are small, so that the signal components are less attenuated by the AC couple.

However, FSK or quaternary FSK with a low modulation index used for high-speed data transmission in recent years has many signal components near DC, which poses a problem in practical use.

ＤＣ The DC offset generated in the receiving unit has been regarded as a problem in the above-described heterodyne system as described above, but is particularly problematic in the case of a direct conversion system recently used in the mobile communication field and the like. The problem of the DC offset in the direct conversion method has a different element from the heterodyne method, and will be described in detail below.

2. Description of the Related Art A conventional direct conversion wireless device (hereinafter referred to as a direct conversion wireless device) converts a radio frequency signal by providing a mixer with an external radio input signal (RF signal) and a local signal having the same frequency. , A signal conversion type wireless device that directly converts the signal into a baseband signal. FIG. 39 shows the configuration of this direct conversion radio.

As shown in FIG. 39, the direct conversion wireless device differs from the heterodyne method in that the local oscillator 130 oscillates at the same frequency as the center frequency of the RF signal, the frequency converters 2131 and 2132 as mixers, and the local oscillator. And a phase shifter 2133 that shifts the phase of the local signal by π / 2 (shifts the phase) by π / 2, whereby the RF signal can be directly converted to a baseband signal.

Next, the operation of the direct conversion radio will be described.

As shown in FIG. 39, at the time of reception, a high-frequency signal received by the antenna 101 is guided to the high-frequency amplifier 102 via the transmission / reception switch 170. The signal amplified here is added to the frequency converters 2131 and 2132 together with the local signal from the local oscillator 130 and the phase shifter 2133. The frequency converters 2131 and 2132 are basically three-port devices and have three terminals for RF, LO, and baseband signals (terminal names are not shown). The high frequency signal is applied to RF, and the local signal is applied to LO. As a result, a frequency-converted signal is generated at the baseband output terminal.

By the way, in a mathematically ideal mixer, the isolation between the terminals is infinite, and a signal applied to a specific terminal does not appear at other terminals.

However, since the actual mixer, that is, the frequency converters 2131 and 2132 cannot take infinite isolation, the local signal generated by the local oscillator 130 as shown in FIG. And is generated at the antenna 101. Therefore, the direct conversion radio radiates a local signal from the antenna 101 at the time of reception.

The local signal radiated from the antenna 101 is reflected by an external reflector, is again received by the antenna 101, and is again input to the frequency converters 2131 and 2132 following the path indicated by a broken line 402b in the figure. become. Since the signals re-input to the frequency converters 2131 and 2132 and the local signal are signals having exactly the same frequency, the signals of the respective frequency converters 2131 and 2132 are multiplied by a multiplication process which is a mixing function of the frequency converters 2131 and 2132. A DC component output is generated at the baseband output terminal. That is, at the same time that the desired high-frequency signal is frequency-converted, a DC component, that is, a DC offset is generated by multiplication of the local signals of itself.

In addition, when the reflection at the amplifier 102 and the reflection at the filter 103 occur, a DC offset occurs in the paths indicated by broken lines 403 and 404 in FIG. 36B, as in the case of the heterodyne method.

The DC offset generated in the path indicated by the broken line 402a in FIG. 36B changes depending on the amount of reflection of the local signal from the outside, that is, the reflection object around the antenna 101, and thus occurs in the paths indicated by the broken lines 403 and 404. The problem is greater than the calculated DC offset or the DC offset inherent in the active device.

Of the direct conversion radios, especially mobile phones, etc., which are targeted for miniaturization, can be used by hand or carried around in a bag or pocket, so that the status of the reflector outside the antenna 101 is constantly changing. And change. Therefore, the amount of reflection of the local signal is also time-varying, and the DC offset also changes every moment, which cannot be suppressed and causes a decrease in reception sensitivity.

キ ャ パ シ タ To compensate for this DC offset, a capacitor is provided on the subsequent circuit. However, since the capacitance of the capacitor is fixed, the reception error rate deteriorates due to the time-varying transient response of the DC offset.

As described above, in the conventional receiving system, at present, it is not possible to cope with the two problems of the reduction of the current consumption and the improvement of the receiving characteristics at all, and particularly in the case of the direct conversion system, the problem is larger than that of the heterodyne system. .

Next, a conventional radio synthesizer will be described.

周波 数 A conventional radio uses a frequency synthesizer, which is a signal generator that can easily obtain a stable and accurate frequency signal by giving a control signal from the outside.

FIG. 40 shows the configuration of a conventional frequency synthesizer.

In the figure, reference numeral 1101 denotes a highly stable reference oscillator such as a temperature-compensated crystal oscillator (hereinafter, TCXO), 1103 denotes a reference frequency divider, 1105 denotes a phase comparator, 1107 denotes a loop filter, and 1109 denotes a voltage controlled oscillator (hereinafter, VCO). Reference numeral 1111 denotes a comparison frequency divider.

Next, the operation of the frequency synthesizer will be described.

In the case of this frequency synthesizer, the output of the highly stable reference oscillator 1101 is divided by the reference divider 1103, and the divided signal is used as the reference signal. The phase of this reference signal is compared with the phase of the signal obtained by dividing the output of the VCO 1109 by the comparison divider 1111, and a voltage corresponding to the phase difference between the two input signals is output by the phase comparator 1105. The loop filter 1107 smoothes the output voltage of the phase comparator 1105 to generate a control voltage for the VCO 1109. By configuring a phase locked loop (hereinafter, PLL) in this manner, an output signal having a stable frequency can be obtained from the VCO 1109.

When the frequency is switched by changing the frequency division number of the comparison frequency divider 1111 from N1 to N2, a voltage corresponding to the phase difference between the two signals is output from the phase comparator 1105, and the loop filter 1107 is output. , Frequency synchronization and phase synchronization are performed. The time until the synchronization at this time is determined by the natural angular frequency ωn of the loop determined by the loop filter 1107 and the damping coefficient ζ. In general, if a natural angular frequency and a damping coefficient with stable oscillation frequency and low noise are selected, the frequency switching time becomes longer.

By the way, since this kind of frequency synthesizer is required to have low phase noise characteristics, it takes time to switch the frequency. For this reason, when the conventional frequency synthesizer is applied to a radio device of the TDMA system, there is a serious drawback that an empty channel search cannot be performed using an interval between empty slots during a call.

Next, the transmission system of the conventional wireless device will be described.

In FIG. 35, the frequency converter 154, the variable attenuator 152, the power amplifier 151, the transmission power control circuit 171, the transmission / reception switch 170, and the like can be easily integrated into an IC, and these parts can be downsized by the integration technology. I have been.

However, since it is difficult to integrate the bandpass filters 150 and 153, the directional coupler 172, and the power detector 173 into ICs, it is necessary to mount the components alone on the motherboard. For this reason, the mounting area has increased. For example, the directional coupler is a chip component of about 5 mm × 5 mm. As the power detector DET, a diode switch as shown in FIG. 41 is used. However, when the mounting area of the diode D1, the capacitors C1, C2, and the resistor RES is considered, the power detector DET is also 5 mm × 5 mm or more. For this reason, there has been a problem that the volume of a portable wireless device that needs to be reduced in size becomes large. Further, it is necessary to increase the output power of the power amplifier 151 so as to compensate for the loss of the output power due to the directional coupler 172, and there is a disadvantage that the power consumption of the transmission system increases.

Next, the antenna of the conventional wireless device will be described.

Among the components that make up radio equipment, the miniaturization of radio circuits, especially batteries and antennas, is rapidly progressing in terms of improving portability, but the speed of miniaturization of the circuit part itself is far away. Absent. Therefore, considering the whole wireless device, the antenna is one of the ones that must be further reduced in size and thickness.

On the other hand, the effect of the human body on the antenna has become a problem with portable radios. The human body causes absorption or scattering of radio frequency radio waves. In addition, the human body fluctuates the operating impedance of the nearby antenna. This is because the human body acts as a radio wave absorber having a high dielectric constant in terms of high frequency. As a result, the radiation characteristics of the antenna are deteriorated by the human body.

耳 In addition, ears tend to be closer to antennas due to the recent miniaturization and thinning of wireless devices. This proximity further worsens the antenna characteristics due to the human body.

Here, the results of measuring the antenna characteristics of the conventional PHS terminal are shown in FIGS. The experimental data shown here is the result of an experiment made with a model of a portable wireless device (PHS terminal) having a frequency of about 2 GHz.

42 and 43 both show the vertical polarization pattern in the horizontal plane during a call, but FIG. 43 shows a case where an antenna and a housing smaller than those in FIG. 42 are used. Comparing the vertical polarization patterns of FIGS. 43 and 42, when the small antenna and the housing are used (FIG. 43), the case where the antenna and the housing of the normal size are used (FIG. 42) is used. It can be seen that the gain deteriorates by about 8 dB.

As mentioned above, one of the causes of this deterioration is that the impedance of the antenna fluctuates depending on the human body. Here, the state of deterioration will be described in detail. In the description, it is assumed that the antenna is used for transmission.

電力 To radiate radio waves from the antenna, power must first be input to the antenna. The optimal condition for power input to the antenna is that the impedance of the feed line and the antenna have the same value. If the impedance of the antenna changes from the optimum value, the power transmitted through the feed line is reflected at the input end of the antenna and returns to the transmission amplifier. Therefore, the power output from the amplifier is not input to the antenna. In addition, this reflection has a bad influence such as oscillating an amplifier in some cases.

Hereinafter, methods that can easily be inferred as methods for solving the above-mentioned problems and problems involved in the solutions will be described.

The first idea to suppress the reflection at the input end of the antenna is to broaden the antenna. In other words, this method utilizes the fact that even if the input impedance fluctuates due to the proximity of a human body, the fluctuation is smaller in a wideband antenna than in a narrowband antenna. However, broadening the antenna bandwidth requires increasing the volume of the antenna. Therefore, the method of broadening the antenna has a contradictory result to the miniaturization of the antenna described above and the miniaturization of the whole radio.

As another method, it is conceivable to adjust the impedance of the antenna so as to be optimal when the antenna is brought close to the human body. However, this method is not generally a good method. This is because the portable wireless device is used not only when the human body is in proximity. Since a person carries the portable wireless device by hand or carries it in a bag, the usage state of the portable wireless device changes in various ways. The amount of change in the impedance of the antenna changes depending on the state of use. This is because the substances and distances approaching each other vary depending on the use conditions. When the amount of change varies in this way, it is very difficult to adjust the impedance of the antenna to an optimum state in all states.

耳 Furthermore, the ear is raised as the human body part closest to the antenna, but the size of the ear varies greatly between individuals. Antenna performance is also affected by individual differences in ear size. The effect of the ear greatly changes the impedance of the antenna. This is because the permittivity of the ear is very high, around 80, and when it is close to the antenna, the electrical length of the antenna changes greatly. The impedance of the antenna changes greatly depending on whether the ear is attached to the antenna or not, and whether the ear is near or away from the antenna. The relative position of the ear antenna depends on the size of the ear. As described above, even if an attempt is made to optimize the impedance at the time of approaching the human body, the performance varies from person to person, such that an antenna designed optimally for one person is not optimal for another person.

い く つ か Some other methods other than the above two methods are considered to optimally control the matching circuit of the antenna according to the state of use.

One method is to change the matching circuit of the antenna by turning on and off the talk button. It has a call button. In other words, it is established by the guess that the antenna should be close to the ear because the user is talking.

こ の However, in this case, there is an advantage that it can be realized with a simple configuration, but it is not possible to cope with the above-described individual difference in ear size.

The second method is to check the level of the reflected wave reflected from the antenna and change the matching circuit of the antenna according to the amount of reflection.

However, in this case, it is necessary to insert a probe between the antenna and the radio circuit in order to check the amount of reflection, and this probe causes reflection loss, conductor loss, and loss of transmitted and received high-frequency signals. cause.

As described above, in the conventional wireless device described above, in the receiving system, the current consumption of the wireless device always needs to flow at the maximum value of the use, and is consumed more than necessary. Further, in the DC offset removal by the AC couple, a desired signal component is also attenuated. Further, there is a time-varying DC offset due to a reflector outside the antenna which cannot be removed even by the AC couple, and there is a problem that deterioration of reception sensitivity cannot be suppressed.

Further, in the synthesizer, there is a problem that it is not possible to search for an empty channel by using an interval between empty channel slots during a call.

Furthermore, in the transmission system, the mounting area of circuit components other than the antenna, for example, a directional coupler or a power detector is large, and it is difficult to further reduce the size.

Another problem with antennas is that when carrying a wireless device, the antenna characteristics deteriorate due to the proximity of the antenna to the human body, and to improve this, it is necessary to increase the size of the antenna or select a user. there were.

The present invention has been made to solve such a problem, and a first object of the present invention is to reduce power consumption. A second object is to remove the time-varying DC offset and improve the receiving sensitivity. A third object is to maintain the original performance of the antenna without increasing the size of the antenna and without selecting a user.

To achieve the above object, a wireless device according to the first aspect of the present invention includes a wireless circuit that supplies a signal transmitted from an antenna via a transmission amplifier, and a power supply line to the wireless circuit and the transmission amplifier. A power supply circuit for supplying power; an ammeter connected to the power supply line; and an antenna characteristic varying unit for varying a matching characteristic of the antenna based on a current value detected by the ammeter.

In the case of the present invention, for the first problem of the receiving system, in addition to a function of detecting power in a desired wave band which is essential for a radio, a function of detecting power in a system band is added. Then, it is determined whether or not the wireless device is in the worst radio wave environment. Based on the result, the current consumption of the receiving system is controlled. Further, the above determination is made with reference to the storage device.
As for the second problem of the receiving system, by adding an AC capacitor, the digital signal processing unit is provided with amplification corresponding to the amount of attenuation at each frequency.

With respect to the third problem of the receiving system, the control signal detecting unit detects the reflection coefficient of the antenna at the time of transmission or the reflected power of the power amplifier, and according to the detected signal, the frequency converter, the low frequency amplifier, Or it has a control unit for controlling the DC offset of the analog / digital converter.

Further, the reflection coefficient of the antenna detected at the time of transmission or the reflected power signal of the power amplifier is stored in the storage device, and the frequency converter and the low-frequency converter according to the reflection coefficient or the reflected power signal stored at the time of reception are stored. Controls the DC offset of the amplifier or analog / digital converter.

(4) Further, a value corresponding to the reflection coefficient of the antenna detected by the control signal detection unit or the reflection power of the power amplifier is subtracted or added from the value detected by the analog / digital converter. Therefore, according to the present invention, even when the state of the reflector outside the antenna changes, it is possible to suppress a change in the DC offset and prevent a decrease in the receiving sensitivity.

(4) To solve the problem of the synthesizer, when detecting an empty channel in the system band, the loop band is broadened from the band at the time of communication.

In order to reduce the mounting area of the directional coupler and the power detector, the first problem of the transmission system is to reduce the mounting area of the directional coupler and the power detector. Alternatively, a power coupler and a power detector that do not always have a direction are manufactured in the transmission / reception switch IC. Thereby, miniaturization is achieved.
Further, a signal proportional to the transmission power generated in the IC is used as a signal for detecting the transmission power. With respect to the second problem of the transmission system, the fluctuation amount of the power supply system is detected. As for the antenna, the current flowing through the feed line of the transmission amplifier is measured, and the matching characteristic of the antenna is varied based on the measured current value.

As described above, according to the present invention, in the receiving system of a wireless device, power in a system band and power of a desired wave are detected, and current consumption control is performed based on the result. It can be powered. In the case where a capacitor for AC coupling is provided in the receiving unit, the deterioration of the frequency characteristic including the characteristics of the capacitor is corrected, so that the DC offset that deteriorates the reception error rate can be removed. Further, the reflected power returning from the reflector is detected by the directional coupler, and each unit that generates the DC offset is controlled based on the detected reflected power and the transmission power, so that the time-varying DC offset can be removed.

(4) In the synthesizer system, control is performed to expand the band of the loop filter for searching for an empty channel, so that an empty channel search can be performed at high speed. In addition, as a result of searching for an available channel, if the number of available channels is large, the wireless device is operated with reduced power consumption, so that the wireless device can be reduced in power consumption.

In the transmission system, by forming the power sensing means and the detection means in the IC, it is possible to achieve downsizing and to omit the directional coupler which causes the transmission power loss which has been conventionally used, The power consumption of the wireless device can be reduced.

The antenna detects the operating current of the power amplifier and varies the matching circuit of the antenna according to the operating current. Power consumption and performance can be improved.

Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings.

FIG. 1 is a diagram showing a configuration of a receiving system of a wireless device of a direct demodulation system according to one embodiment of the present invention. Here, as an embodiment of the present invention, a radio device of a direct demodulation system will be described as an example, but the present invention can also be applied to a heterodyne system or the like.

In the figure, LNA is a low noise amplifier, BPF is a bandpass filter, MIX is a frequency converter, BUFF1 and BUFF2 are buffer amplifiers, LPF1 or BPF1 and LPF2 are low-pass filters (or BPF2 is a bandpass filter), RSSI1 and RSSI2 is a power detector, 10 is a subtractor (or divider), 11 is a decision device (decision), 12 is a delay device (deley), and 13 is a current control means (current-contol). LPF1 or BPF1 is a filter having a characteristic of passing only a signal band, and RSSI1 detects power in the signal band. LPF2 or BPF2 is a filter having a characteristic of passing the band of the entire system, and RSSI2 detects power in the system band. Since the pass bands of the LPF 2 and the BPF 2 need to cover all frequencies of the system band, a filter that passes at least the system band is required. The delay device 12 delays and outputs the input signal by one frame or one slot from the time when the current control is determined, for example. In order to reduce the current of the circuit block of the receiving system, it is necessary to detect that the radio is not presently in the worst radio wave environment. In order to do this, RSSI1 and RSSI2 are provided. RSSI1 detects power in a desired wave band, and RSSI2 detects power in a system band. The determination device 11 determines whether or not the radio wave environment where the wireless device is present is in the worst state by calculating the difference or ratio between the power detected by RSSI1 and the power detected by RSSI2. Actually, if the radio wave environment is not the worst, there is no guarantee that the circuit will operate without any problem even if the current is reduced to about half, for example. Therefore, the numerical value obtained by taking the difference or ratio between RSSI1 and RSSI2 is obtained. Are allocated to several stages, and a current is set for each of the allocated stages so as to flow through the low-noise amplifier LNA and the frequency converter MIX. The current set from the difference or ratio between the RSSI1 and RSSI2 is obtained by previously inspecting the distortion characteristics of the circuit block and writing the obtained result in a table created on a memory (not shown) in the determination device 11. Shall be. Then, it is assumed that a current flows to the low noise amplifier LNA and the frequency converter MIX with reference to the set current value in the table. If the number of current stages to be allocated is small, it is not always necessary to refer to the table.

Next, the operation of the receiving system will be described with reference to the flowchart of FIG.

In the description of the operation of the receiving system, it is assumed that the number of reception slots is only once per frame, and control is performed using only the reception slots. Further, for the sake of simplicity, it is assumed that there are only two stages of the current to be allocated: the normal current (for the worst radio wave environment) and the low current mode.

In the case of this reception system, as shown in FIG. 2, from the start of the reception slot (slot start) to the end of the reception slot (slot end), that is, after the end of the slot (S201), the RSSI1 determines the frequency within the desired band. In addition to detecting the power, the power in the system band is detected by RSSI2, and the power detection result is subtracted (RSSI2-RSSI1) or divided (RSSI2 / RSSI1), and a set value having the subtraction result (or division result) It is determined whether it is equal to or more than A (S202).

Here, for example, when the subtraction result is equal to or greater than a certain set value A (YES in S202), the determination device determines that the radio wave environment is bad, and determines that a normal current flows (normal mode) (S203). The determination result is applied to the next reception slot (S204).

When the subtraction value (RSSI2-RSSI1) or the division value (RSSI2 / RSSI1) detected in the next reception slot is smaller than the set value Α (NO in S202), the determination device 11 sets the low current mode (low current mode). ) Is determined (S205), and the current of the low noise amplifier LNA and the frequency converter MIX is reduced (S206).

In order to reduce the current, for example, a resistance value or a specified voltage in a bias circuit (not shown) for setting a bias current of the low noise amplifier LNA and the frequency converter MIX is changed. Hereinafter, the current is adjusted similarly. In this embodiment, the case where the detection is performed one frame before is described, but the detection may be performed one slot before. In this case, the determination is made by taking the difference or ratio between the output level of the desired wave detected one frame before and the output in the current system band. In practicing the present invention, it is desirable to set the current value of the initial value to the normal current mode. However, if all the channels are vacant by the detection of the vacant channels described above, there is clearly no unnecessary wave in the system. Since it can be estimated, the initial value may be set to the low current mode.

Next, since the band of the LPF2 and the bandpass filter 2 is the band of the entire system, white noise (thermal noise) is increased by the band ratio compared to the band of the desired wave. When the white noise (thermal noise) increases, the determination device 11 may erroneously determine that a radio wave of a specified value or more exists in the system. In order to avoid this, it is necessary to correct the power detection value of RSSI2 by subtracting the power component of white noise corresponding to the widened band from the power detection value obtained from RSSI2. As a method of this correction, the power component of the white noise may be simply subtracted from the power value detected by the RSSI2, and the power value detected by the RSSI2 may be corrected using a table in which the power component of the white noise is written. You may.

FIG. 3 is a diagram showing another embodiment related to the receiving system of the wireless device of the present invention.

As shown in FIG. 3, this receiving system is obtained by adding an I-channel multiplier 115, a Q-channel multiplier 116 and a memory 118 as storage means to the configuration of the conventional receiving system of FIG. .

The memory 118 has data of the inverse characteristic of the total frequency characteristic of the AC couple in the baseband section from the frequency converter 105 to the A / D converter 113 and the data from the frequency converter 106 to the A / D converter 114. And data of the inverse characteristic of the total frequency characteristic of the AC couple in the baseband section up to the first frequency. Then, the data of each inverse characteristic is multiplied by the desired signal in each of the multipliers 115 and 116. The desired signal output from the A / D converters 113 and 114) has a notch 306 as shown in FIG. 38B, but the multipliers 115 and 116 multiply the contents of the memory 118. Thus, the original signal is reproduced.

Next, the frequency characteristics stored in the memory 118 will be described with reference to FIGS.

4A, reference numeral 801 denotes the frequency characteristic of the AC couple in the baseband section from the output of the frequency converter 105 to the output of the A / D converter 113. The frequency characteristic 801 of the AC couple is 0 for direct current (DC). The inverse characteristic of the frequency characteristic 801 becomes the characteristic 802 shown in FIG. 4B, and becomes infinite at the DC frequency. This frequency characteristic (reverse characteristic) 802 cannot be stored as data in the memory 118 as it is.

(4) Therefore, of the inverse characteristic 802, a DC component of a predetermined level or more is removed according to the required compensation accuracy and used as the inverse characteristic.

In the example of FIG. 4B, a part 803 of the DC component of the inverse characteristic 802 of the AC couple is removed, and the remaining part is set as the inverse characteristic 804. Therefore, in the desired wave after multiplying the output signals of the A / D converters 113 and 114 by the inverse characteristic 804, a compensation error corresponding to the termination of the DC component of the inverse characteristic 804 occurs. In other words, notches 701 and 702 occur in the desired wave 301 and the thermal noise 304, respectively, as shown in FIG.

However, it is clear that the unnecessary DC output is completely removed, and the notch 701 of the signal component is improved as compared with the notch 306 shown in FIG. 38B before compensation. Since the DC offset outputs of the I channel and the Q channel are basically different, it is necessary to perform this self-compensation operation for both the IQ channels. The advantage of the DC output self-compensation function is that the effect is not impaired at all even if the reference carrier frequency supplied from the local oscillator 107 in FIG. 3 is offset from a desired value.

If, for example, the frequency of the local oscillator 107 is offset from the desired value, the center frequency of the baseband desired signal output from the frequency converters 105 and 106 is offset from DC. This is shown in FIG.

所 望 As shown in FIG. 6, the desired signal 601 has a center frequency offset from the DC component by a width indicated by reference numeral 602. This corresponds to a frequency offset from the desired frequency of the local oscillator 107. However, as can be seen from FIG. 6, even if the center frequency of the signal whose frequency has been converted to baseband is offset from DC, the frequency of the DC output 305 does not in principle be offset from DC. Therefore, the DC output 305 is completely removed by the AC couple frequency characteristic 302 of the capacitors 119 to 124. Thereafter, the inverse characteristic 804 stored in the memory 118 may be multiplied. Therefore, the DC output 305 can be eliminated without causing a notch in the desired signal 301, as in the case where the local oscillator 107 has no frequency offset. As the data of the inverse characteristic 804, data measured in advance may be stored in the memory 118.

Next, another embodiment of the receiving system of the wireless device according to the present invention will be described with reference to FIG. The receiving system of this embodiment is obtained by adding a sweep oscillator 901, a changeover switch 902, and a computing unit 907 to the receiving system shown in FIG. The sweep oscillator 901 is connected to the respective circuit systems of the I channel and the Q channel via the changeover switch 902. The sweep oscillator 901 sweeps the frequency range from the output of the frequency converters 105 and 106 to the output of the A / D converters 113 and 114 to a frequency range where the frequency characteristic 801 of the AC couple becomes flat. By this sweep operation, the frequency characteristic 801 of the AC couple of the I channel and the Q channel can be obtained, and is sent to the arithmetic unit 907 which is provided after the A / D converters 113 and 114 by branching the circuit. The arithmetic unit 907 calculates the inverse characteristic 804 in FIG. 4 from the measured frequency characteristic 801 of the AC couple and stores it in the memory 118. The operation of measuring the frequency characteristics using the sweep oscillator 901 can be performed in a time period in which the reception operation is not performed.

As described above, according to this embodiment, even when the frequency characteristic 801 of the AC couple changes due to the temperature characteristic or the like, the DC offset compensation can be performed by more flexibly obtaining the inverse characteristic 804.

Although this receiving system has been described as being of the heterodyne type, it can also be used in the direct modulation type. However, it cannot cope with the transient response of the time-varying DC offset caused by the reflector. However, depending on the wireless system, this measure may provide sufficient characteristics.

Next, a method of removing a time-varying DC offset caused by a reflector which is a problem in the direct conversion method will be described.

FIG. 8 is a diagram showing a configuration of a direct conversion wireless device (hereinafter, referred to as a direct conversion wireless device) which is one embodiment of the wireless device of the present invention.

In the figure, 101 is an antenna, 170 is a transmission / reception switch, 102 is a high-frequency amplifier, and 105 and 106 are frequency converters. Each of the frequency converters 105 and 106 has a DC offset control terminal 132-1. 130 is a local oscillator, 131 is a π / 2 phase shifter, 109 is a baseband filter, and 111 and 112 are low frequency amplifiers. The low-frequency amplifiers 111 and 112 have a DC offset control terminal 132-2. Reference numerals 135 and 136 denote baseband signal processing circuits each having a DC offset control terminal 132-3 therein. Each of the baseband signal processing circuits 135 and 136 includes an analog / digital converter 113 and 114 and an addition / subtraction circuit 133 and 134, respectively.

# 137 is a transmission system of the direct conversion radio. The transmission system 137 includes a bandpass filter 150, a directional coupler 172, a power amplifier 151, a variable attenuator 152, a power detector 173, a power control circuit 171, an adder 156, frequency converters 157 and 158, a low-pass filter. 159, 160, a digital / analog converter, a transmission signal generator 138, and the like. A DC offset control circuit 139 is connected to the directional coupler 172 and the control terminals 132-2, 132-2, and 132-3.

Next, the operation of the direct conversion radio will be described. First, the basic operation of transmission will be described.

In this direct conversion radio, the transmission / reception switch 102 is switched to the transmission system 137 at the time of transmission. Then, a transmission wave from the transmission signal generator 138 is amplified by the power amplifier 151 and transmitted from the antenna 101 via the directional coupler 172 and the transmission / reception switch 170.

<< Here, the same frequency is used for transmission and reception >> The DD system will be described as an example. At the time of transmission, the directional coupler 172 measures the reflected power from the antenna 101. The directional coupler 172 also measures the traveling power to the antenna 101. These measurement results are output to the DC offset control circuit 139. Then, a reflection coefficient of the antenna 101 at the time of transmission is obtained from the reflected power and the forward power input in the DC offset control circuit 139. Since the same frequency is used for transmission and reception in the TDD system, it can be seen that when the reflected power at the time of transmission is large, the reflection of the local signal from the antenna 101 at the time of reception is large. Conversely, when the reflected power is small, the magnitude of the reflection of the local signal during reception is also small. Therefore, by performing the transmission operation, the reflected power at the time of transmission from the antenna 101 can be known, and thereby, the magnitude of the reflection amount of the local signal to the receiving system can be obtained.

When the reflected wave is large, the DC offset control circuit 139 outputs a control signal having a polarity in the direction of decreasing the DC offset and a magnitude proportional to the reflected power to the DC offset control terminal 132-of the frequency converters 105 and 106. Output to 1. Thereby, the DC offset in the receiving system can be reduced.

As described above, according to the direct conversion radio of this embodiment, when transmitting and receiving by the TDD method, the directional coupler 172 measures the traveling power to the antenna 101 and the reflected power from the antenna 101 during transmission, Since the reflection coefficient of the antenna 101 is obtained based on these and reflected in the reception system, the reception operation can be started from the time of reception immediately after transmission in a state where the DC offset by the external reflector is reduced. Compared with the case where no processing is performed, the reception sensitivity at the start of reception can be improved. In addition, since reception can be performed from a state where the DC offset is reduced, it is not necessary to provide a large number of capacitors for AC coupling on a circuit.

In the above description, the control of the frequency converters 105 and 106 has been described. However, among the DC offset control terminals 132-2 and 132-3 provided in the low-frequency amplifiers 111 and 112 and the analog / digital converters 113 and 114, By outputting the control signal to either one of them, the same effect as when controlling the frequency converters 105 and 106 can be obtained. This is because the DC offset may be compensated as necessary at the stage after the mixer, that is, the frequency converters 105 and 106.

Next, a modified example of the wireless device of the embodiment shown in FIG. 8 will be described with reference to FIG. FIG. 9 is a view showing a modification of the embodiment of FIG.

In this modification, the memory 141 is provided between the DC offset control circuit 139 and the directional coupler 172. The memory 141 generates a predetermined control voltage according to the reflection power or the reflection coefficient from the antenna 101. Therefore, if a control voltage for reducing the DC offset is stored in advance in accordance with the reflected power of the antenna 101, only the control voltage is input to the DC offset control circuit 139, and the DC offset control circuit 139 8, the calculation of the reflection coefficient is not performed, and the DC offset can be reduced in a shorter time than in the embodiment of FIG.

Next, another modified example of the wireless device of the embodiment shown in FIG. 8 will be described with reference to FIG. FIG. 10 is a diagram showing a modification of the embodiment of FIG.

This modification is an example in which the DC offset control circuit 139 is not provided and the baseband signal processing circuit 135 performs collective DC offset reduction processing. Although FIG. 10 illustrates only the baseband signal processing circuit 135, the same applies to the baseband signal processing circuit 136.

In this case, a value corresponding to the traveling power of the power amplifier 151 or the reflected power of the antenna 101 measured by the directional coupler 172 and the value output from the analog / digital converters 113 and 114 are added / subtracted by a circuit 133. , 134 to reduce the DC offset. This is because performing subtraction or addition from the output values of the analog / digital converters 113 and 114 is equivalent to performing a DC value offset, that is, performing an analog DC offset.

As described above, according to this modification, the same effect as described above can be obtained without connecting the special DC offset control circuit 139 to an analog circuit such as a frequency converter or a low-frequency amplifier. Although the TDD system has been described as an example of the communication system here, the same effect can be obtained by measuring the amount of reflection at the time of transmission even when the same frequency band is used for transmission and reception.

Next, a synthesizer for a wireless device will be described.

FIG. 11 is a configuration diagram showing a first embodiment of the synthesizer of the wireless device.

In the figure, 1101 is a reference oscillator, 1103 is a reference frequency divider, 1105 is a phase comparator, 1151 is a normal mode loop filter, 1152 is a high-speed mode loop filter, 1153 is a switch, 1109 is a VCO, and 1111 is a comparison filter. It is a circulator. Note that the basic loop operation is the same as that of the conventional synthesizer, and the description thereof is omitted.

As shown in FIG. 11, the synthesizer includes a normal mode loop filter 1, a high speed mode loop filter 2, and a switch 1153. When the switch 1153 is switched to the normal mode loop filter 1, the loop enters the normal mode, and the output of the VCO 1109 has low phase noise characteristics, but the frequency switching time is long.

On the other hand, when the switch 1153 is switched to the high-speed mode loop filter 2, the loop enters the high-speed mode (empty channel search mode), and the phase noise characteristics of the output of the VCO 1109 deteriorate, but the frequency switching time becomes faster.

Next, the operation of the synthesizer used in the wireless device of one embodiment of the present invention will be described. FIG. 12 is a conceptual diagram showing a slot configuration of the TDMA system.

に お い て In FIG. 12, reference numeral 1160 denotes a frame, and 1161 denotes a slot. Here, it is assumed that a call is made using the reception slot R1 and the transmission slot # 1. Also, the frequency at which a call is made is F1.

場合 In the case of this synthesizer, a search for an empty channel is performed as follows.

{First, during the reception slot R1, the operation is performed in the normal mode having good phase noise characteristics. At the same time as the end of the reception slot R1, the loop is switched to the high-speed mode by the switch 1153.

(4) An empty channel search is performed by setting the synthesizer to a frequency F2 different from the communication channel. At this time, since the loop is in the high-speed mode, the frequency is switched to a desired frequency at a high speed, and an empty channel search is quickly performed. When the empty channel search is completed at this frequency F2, the frequency is switched to another frequency F3 and the empty channel search is performed again. After repeating such an operation and searching for an empty channel at a plurality of frequencies, the frequency returns to the original frequency F1 of the communication channel before the transmission slot # 1 arrives. At the same time, the mode is switched from the high-speed mode to the normal mode, and the operation in the transmission slot # 1 is again performed in the normal mode having good phase noise characteristics. By repeating the above operation, an empty channel search can be performed during a call. At the time of empty channel search, since the loop is in the high-speed mode, the S / N of the synthesizer, that is, the phase noise characteristic is not good and the receiving sensitivity is reduced, but it is determined whether or not there is an empty channel. For this reason, the phase noise characteristic of the synthesizer required for the communication is alleviated as compared with the time of speech, so that it is safe to perform the idle channel search in the high-speed mode. In the above description, the loop characteristics are switched by switching the frequency characteristics of the loop filter. However, the loop characteristics may be switched by, for example, switching the sensitivity of a phase comparator. Further, the empty channel search may be performed, for example, all within one slot.

Also, if all channels are vacant by the high-speed vacant channel search, it can be determined that there is no interference wave in the vicinity of the system band. good. That is, as a result of the empty channel search, for example, if all the channels are empty, control for reducing the current consumption of the receiving system can be performed.

In this case, the result of the vacant channel search is input to the determination device 11 shown in FIG. 1, the determination device 11 determines the current mode, and controls the current of the LNA and MIX.

Next, the transmission system will be described with reference to FIGS.

FIG. 13 is a basic conceptual diagram in the case where the power amplifier or the transmission / reception changeover switch is integrated into an IC, and FIG. 14 is a diagram illustrating another example of the integration of the embodiment of FIG.

に お い て In FIG. 13, 1200 denotes an IC. When the IC 1200 is, for example, a power amplifier IC (hereinafter referred to as PΑ-IC), an RF signal from the frequency converters 105 and 106 is input to an input terminal IN of the IC 1200. An RF signal amplified by the PA-IC 1200 is output to the output terminal OUT. In the IC 1200, a sensing means 1201 for sensing the output power is provided. From the sensing means 1201, a signal proportional to the sensed power is output from a terminal different from the output terminal OUT. It is output to a power detector DET or the like, which is a signal processing means.

In the same figure, when the IC 1200 is, for example, a transmission / reception switch (hereinafter, a T / R switch), the RF signal output from the power amplifier P # is input to the input terminal IN of the IC 1200. An RF signal is output to the OUT terminal via the T / R switch. In the sensing means 1201, a signal proportional to the RF output power is sensed similarly to the PA-IC, and the sensed output is output from a terminal different from the output terminal OUT.

Another embodiment that is an extension of the embodiment of FIG. 13 will be described with reference to FIG. In FIG. 14, a sensing means 1201 is the same as in FIG. 13, but in this case, a sensed signal is input to an output power detector DET which is a signal processing means formed in the same IC. The output power detector DET 1202 converts the output power to a low frequency in response to the input of the sensing signal, and outputs a signal proportional to the output power to the power control circuit (CONT) 171. The reason why the output power detector DET 1202 is described as a signal processing means is that the output power detector DET also performs signal processing such as frequency conversion.

In the description of FIGS. 13 and 14, the PA-IC or the T / R switch IC does not assume a single IC, but may be any IC provided with at least one of them.

FIG. 15 is a diagram showing a circuit that is a more concrete example of the basic conceptual diagram related to the present proposal shown in FIG. 13, and shows an example using a PA-IC.

ＲＦ The RF signal outputs from the frequency converters 105 and 106 are input to the input terminal IN, and are input to the power amplifier P # 151. The output of the power amplifier PA151 is output to the next-stage band-pass filter BPF via the output terminal OUT.

The power supply terminal VDD1 in the IC 1200 is a terminal for supplying power to the power amplifier PA151, and the ground terminal GND1 is a ground terminal of the power amplifier PA151. The reason that the external power supply terminal VDD and the external ground terminal GND are separated as the terminal VDD1 and the terminal GND1 is that when an IC having such a configuration is actually mounted, parasitic inductors, resistors, and capacitor components are generated between the circuits. It is for showing. That is, the state is the same as the state where the impedance Zvdd1203 is connected between the terminal VDD and the terminal VDD1 due to the parasitic state, and the same as the state where the impedance Zgnd1204 is connected between the terminal GND and the terminal GND1 due to the parasitic state.

In the figure, a terminal s1 is a terminal for extracting a signal proportional to the RF output power of the power amplifier PA151, and is connected to the power supply terminal VDD1 via the sensing means 1201 described with reference to FIG. That is, in this example, a signal proportional to the multiplication of the parasitic impedance Zvdd1203 and the instantaneous current of the power amplifier P # is observed at the power supply terminal VDD1. In general, the change in the instantaneous current of the power amplifier PA151 is proportional to the output power of the power amplifier PA151. Therefore, the AC component observed at the terminal VDD1 is proportional to the output power of the power amplifier PA151.

The sensing means 1201 will be specifically described. Considering that a signal proportional to the output power of the power amplifier PA151 can be obtained by obtaining an AC component observed at the terminal VDD1, the simplest one is: For example, as shown in FIG. 16A, a capacitor C1 and the like may be provided between the terminal VDD1 and the terminal s1.
In this case, for example, the power detection circuit DET shown in the conventional example is connected to the next stage of the terminal s1. The capacitor C1 has no directivity, but can be used without any problem when detecting power, because a coupler having a directivity is not always necessary. In the circuit configuration of FIG. 16A, a signal proportional to the output power can be extracted by configuring the sensing means 1201 with a simple element configuration that only requires the provision of the capacitor C1. As the sensing means 1201, other elements such as a diode and a resistor may be used.

By the way, the values of the parasitic impedance Zvdd1203 and the parasitic impedance Zgnd1204 change depending on the mounting method of the PA-IC, so that the proportional coefficient depends on the mounting. For this reason, the detected power value output from the power detection circuit DET may change depending on the implementation. Further, the signal detected from the terminal VDD1 is about -50 dB of the RF signal output power, and the signal level is relatively small. Therefore, the low frequency detection signal output from the power detection circuit DET is converted to the power control circuit (CONT). Too small to receive at 171.

Therefore, as shown in FIG. 16 (b), a high-frequency amplifier (AMP) 1205 having a variable gain may be connected to the capacitor C1 and connected to the terminal s1 to form a sensing means 1201.

In this case, the reason why the variable gain high frequency amplifier (AMP) 1205 is connected to the next stage of the capacitor C1 is to match the output signal of the power detection circuit DET with the dynamic range of the power control circuit (CONT) 171. The variable gain high frequency amplifier (AMP) 1205 is provided with a gain control terminal (gain adjustment) 1206, and a control signal is input from the power control circuit (CONT) 171 to the gain control terminal (gain adjustment) terminal 1206. By doing so, an output signal corresponding to the control signal is obtained, and fluctuations in the amplitude of the output power detection signal due to mounting can be suppressed. As a result, a stable power detection signal having a high signal level is input to the power control circuit (CONT) 171, and the power control circuit (CONT) 171 can sufficiently detect the power detection signal.

As described above, in the circuit configuration of FIG. 16B, by using the variable gain high frequency amplifier (AMP) 1205 in cascade connection with the capacitor C1, the power control circuit (CONT) 171 can sufficiently detect the power detection signal, ICs that can be put to practical use can be performed.

Although a new power consumption is generated by the variable gain high frequency amplifier (AMP) 1205 formed in the IC and the total power consumption is somewhat increased, this power consumption is the power consumed by the power amplifier (PA) 151. Therefore, the power consumption of the transmission system does not matter.

Here, a specific internal configuration of the variable gain high frequency amplifier (AMP) 1205 will be described with reference to FIG.

Ｖ As shown in FIG. 17, VDD2 is a voltage source. The voltage source VDD2 has an anode connected to the base terminal of the transistor Q1 and a cathode connected to the ground terminal GND1 in the IC. The emitter terminal of the transistor Q1 is connected to the input terminal in and to the ground terminal GND1 via the variable current source I1. The gain control signal from the gain control terminal (gain adjustment) 1206 is input to the variable current source I1. The collector of the transistor Q1 is connected to the power supply terminal VDD1 in the IC via the load impedance Z1. The collector of the transistor Q1 is connected to the base terminal of the buffer transistor Q2. The collector terminal of the transistor Q2 is connected to the power supply terminal VDD1. The emitter terminal of the transistor Q2 is connected to the ground terminal GND1 via the constant current source I2 and to the output terminal out via the DC block capacitor C3. In this circuit configuration, the gain G can be obtained by approximation by the following equation.

G = gm (Q1) × Z1
= I1, dc x Z1 / Vt (Equation 1)
In this equation 1, i1 and dc are currents flowing through the variable current source I1, and Vt is a thermal voltage.

Therefore, by changing the current values i1 and dc of the variable current source I1 according to the gain control signal input from the gain control terminal (gain adjustment) 1206, the gain of the signal output from the variable gain high frequency amplifier (AMP) 1205 is adjusted. can do.

Next, some examples in which the T / R switch and the sensing means 1201 are manufactured as ICs based on the basic concept of FIG. 13 will be described.

One of the most common circuits for converting a T / R switch into an IC is a Single-Ploe-Dual-Throw (SPDT) switch. FIG. 18 shows a basic circuit of the SPDT switch IC.

In FIG. 18, Tin is a transmission system input terminal, and Rin is a reception system output terminal. The terminal ANT is an output terminal of the transmission system and an input terminal of the reception system, and is connected to the antenna 101. GND1 is a ground terminal of this IC. Complementary control signals are input to the terminals cont11 and cont2, and transmission and reception are switched by the control signals. When the terminal cont1 is high “Η” and the terminal cont2 is low “L”, the switching elements Q11 and Q12 are turned on, the switching elements Q10 and Q13 are opened, and the signal input from the terminal ANT is output from the reception output terminal Rin Is output to

On the other hand, when the terminal cont1 is low “L” and the terminal cont2 is high “Η”, the switching elements Q11 and Q12 are open, the switching elements Q10 and Q13 are conductive, and the signal input to the transmission system input terminal Tin is the terminal. Output to ANT.

ＳＰ A T / R switch IC is a SP / T switch IC in which a sensing circuit as a sensing means for sensing a signal proportional to the transmission output power is added to the SPDT switch IC, as shown in FIG.

よ う As shown in FIG. 19, the T / Rswitch IC 1200 has a sensing circuit 1201 connected between the source terminal of the switch element Q12 of FIG. 18 and a ground terminal GND1 in the IC. The specific configuration of the sensing circuit 1201 is shown in FIGS.

As an example of the sensing circuit 1201, for example, as shown in FIG. 20, the impedance circuit Z is inserted between the source terminal of the switch element Q12 and the ground terminal GND1. The impedance circuit Z includes a resistor, a capacitor, an inductor, a series circuit thereof, a parallel circuit thereof, and the like. Note that out is an output terminal.

Hereinafter, the operation of the IC 1200 will be described.

During transmission, cont1 is low “L”, cont2 is high “Η”, and the switching element Q12 is open. However, since the RF signal is input to the transmission system input terminal Tin, the RF signal is output to the terminal out by the capacitor between the source and the drain of the switch element Q12 or the series connection of the capacitor between the source and the gate and the capacitor between the gate and the drain. Leaks into. Since the leakage current flows through the impedance Z, a voltage proportional to the leakage current is generated. Since this leakage current is proportional to the power of the RF signal, a signal proportional to the RF power can be extracted from the terminal out in this manner.

Another example of the sensing circuit 1201 is a connection in which a variable gain high frequency amplifier (AMP) 1205 is inserted between the impedance circuit Z and the output terminal out as shown in FIG. The variable gain high frequency amplifier (AMP) 1205 amplifies a signal generated in the impedance circuit Z, and performs gain adjustment by inputting a gain control signal, for example, an applied voltage from a gain control terminal (gain adjustment) 1206. .

Next, a case where the power amplifier (PA) 151 is integrated into an IC, that is, a PA-IC will be described with reference to FIG. FIG. 22 shows a PA-IC in which a power amplifier (PA) 151, a sensing means 1201 and a detection means such as the output power detector DET 1202 shown in FIG. 14 are formed on a one-chip IC. FIG. In PA-IC 1200, a signal proportional to the power to be detected is output from terminal d1 and input to power control circuit (CONT) 171. By using the one-chip PA-IC 1200 including the sensing means 1201 and the output power detector DET 1202, the operation of sensing and detecting the power inside the IC can be performed, so that the transmission system can be downsized. be able to.

Next, a case where the T / R switch is formed into an IC, that is, a T / RswitchIC will be described with reference to FIG. FIG. 23 is a diagram showing a T / Rswitch IC in which a T / R switch circuit, a sensing means 1201 and an output power detector DET1202 are formed in a one-chip IC.

In this case, a signal proportional to the output power is output from the terminal d1 in the same manner as described above. The specific configuration of the sensing means 1201 is the same as that shown in FIGS. 16, 17, 20, and 21, for example. In addition, although not shown here, such an IC is not limited to making the power amplifier P #, the T / R switch, and the like alone, and may be any IC that includes one of them.

By using the one-chip T / Rswitch IC in which the sensing means 1201 and the output power detector DET 1202 are formed as described above, the operation of sensing and detecting the power inside the IC can be performed. The size can be reduced.

Further, since the power sensing means and the detection means shown in FIGS. 15, 19, 22, and 23 detect power leaking to, for example, a power supply or a ground without directly connecting to an RF signal line, directional coupling is performed. This eliminates the power loss in the device and the like, and the power consumption of the transmission system can be reduced accordingly.

FIG. 24 is a view showing one embodiment in which the power sensing and detecting elements shown in FIG. 16 are formed on an IC chip. FIG. 24 (a) is a plan view and FIG. 24 (b) is a view. It is AA sectional drawing of 24 (a).

In FIG. 24, reference numeral 1401 denotes a power supply wiring using a metal layer of a second metal layer to be supplied to the power amplifier (PA) 151, which is formed on the surface of the IC. Reference numeral 1402 denotes a metal wiring of a first layer metal, which is formed in an IC inner layer. The power supply wiring 1401 and the metal wiring 1402 constitute a power sensing element 1402 (sensing means). Reference numeral 1202 denotes an output power detector DET (detect means), and reference numeral 1404 denotes an insulating layer.

When the power sensing and detection means and the like are provided in the IC as described above, in a normal IC, a metal layer is formed on the power supply wiring on the surface layer through an insulating layer to form a capacitance component from the viewpoint of low loss. Often.

However, in the case of this embodiment, the capacitance coupling is performed by adding or subtracting the metal wiring 1402 of the first layer metal formed below the power supply wiring 1401 in the surface layer, that is, the inner layer.

Generally, the power supply wiring 1401 is formed with a wide area in order to reduce the impedance of the power supply. Therefore, the area of the power detection capacitor composed of the first and second layers can be increased, and leakage power can be easily picked up. This property is advantageous for power detection. It should be noted that the present invention can be applied without deteriorating the effect even when the assignment of the metal layers is reversed as in the related art. The capacitance to be coupled is adjusted by, for example, making the line width of the metal wiring 1402 of the first-layer metal formed in the inner layer smaller than that of the power supply wiring 1401, as shown in FIG. Further, as shown in FIG. 26, the line width of the first layer metal wiring 1402 may be adjusted to be wider than the power supply wiring 1401. Further, as shown in FIG. 27, the line length of the metal wiring 1402 of the first layer metal may be adjusted. Further, as shown in FIG. 28, the adjustment may be made by changing the forming direction of the metal wiring 1402 of the first layer metal.

Finally, another embodiment of the wireless device of the present invention, for example, an antenna portion of a portable wireless device will be described with reference to FIGS. FIG. 29 is a diagram showing a configuration of a portable wireless device according to one embodiment of the present invention.

In FIG. 29, 1500 is a housing, and 1501 is a wireless circuit, which includes, for example, the frequency converters 157 and 158 and the variable attenuator 152 shown in FIG. 1509 is a transmission amplifier, 101 is an antenna, 1504 is an ammeter, 1503 is a control circuit, 1502 is a matching circuit, 1505 is a power supply circuit, and 1506 is a current measurement probe.

(4) The power supply circuit 1505 supplies power to the wireless circuit 1501, the transmission amplifier 1509, and the control circuit 1503. The wireless circuit 1501 generates an information signal mixed with modulation and transmission frequency based on the supplied power, and sends the information signal to the transmission amplifier 1509. The transmission amplifier 1509 amplifies the transmitted signal and transmits it to the antenna 101. The antenna 101 sends out the amplified signal to the air, but a part of the signal is sent back to the transmission amplifier 1509 as a reflected wave, and the gain and efficiency of the transmission amplifier 1509 fluctuate. This fluctuation causes a fluctuation in current consumption. This variation is measured by the ammeter 1504, and the level of the current is sent to the control circuit 1503. The transmission amplifier 1509 may be varied in a manner such that the current increases or decreases. Here, one that changes monotonically will be used. The control circuit 1501 reads the value of the signal sent from the ammeter 1504 and electrically adjusts a variable portion on the matching circuit 1502. Note that as a variable portion on the matching circuit 1502, a variable capacitance such as a semiconductor switch or a semiconductor varicap may be used.

Hereinafter, it has been confirmed by experiments that the characteristics of the antenna 101 can be prevented from deteriorating due to the usage state of the portable wireless device.

FIG. 30 is a diagram showing a wireless device model created based on the circuit configuration of the portable wireless device of FIG. 29, and FIGS. 31 to 33 are graphs of the results of measurement using the wireless device model of FIG.

As shown in FIG. 30, the wireless device model includes a speaker 1511, a microphone 1512, an antenna cover 1514, and the like on the surface of the housing 1500 and a wireless circuit 1501 including a transmission amplifier 1509 in the housing 1500. Inside the antenna cover 1514, a coiled antenna (helical antenna) 101 is provided.

Power was supplied to the wireless circuit 1501 of this wireless device model by connecting a power supply line from the constant voltage source 1510 installed outside to the housing 1500. The current consumed by this wireless device model was measured by reading the swing of an ammeter 1513 included in the constant voltage source 1510. The matching circuit 1502 was simply simulated by directly changing the parameters of the antenna 101. The operating frequency of the wireless device model is around 2 GHz, and the size of the housing 1500 is about one wavelength in length, about one quarter wavelength in width, and about one-twentieth wavelength in thickness. The height of the antenna 101 is about one-tenth of a wavelength.

FIG. 31 shows the relationship between the usage state of the wireless device model and the current consumption. FIG. 32 shows the reflection coefficient at the input end of the antenna 101 for each use state. FIG. 33 is a diagram showing the relationship between the use state and the average power in the horizontal plane radiated from the wireless device model. 31 and 32, it can be seen that the reflection coefficient of the antenna 101 changes depending on the use condition, and the current consumption of the transmission amplifier 1509 increases depending on the use condition. Also, from FIG. 33, it can be seen that the power radiated from the antenna 101 decreases as the device is held by a single person and further enters a talking state. These phenomena can be explained by considering the antenna 101 as a load of the transmission amplifier 1509. That is, it is considered that the operation state of the power amplifier P # and the like changes due to the change in the load, and as a result, the current consumption increases. The load fluctuation at this time is clearly caused by the human body.

Next, we decided to optimize the antenna parameters while reading the current value. The current consumption increases from a single device to a handheld device and further to a call state. Therefore, in order to approach the original state of the simple substance, it was considered that the parameters should be reset so that the current consumption is reduced. The antenna parameter used for the adjustment was the antenna length. This is because the resonance frequency of the antenna 101 can be changed by changing the antenna length. When the antenna length was increased or decreased, it was found that the current consumption was reduced by reducing the antenna length. When the antenna radiated power was measured in this state, it was found that the radiated power also increased by about 2 dΒ as compared to before the antenna length was adjusted. Therefore, it has been experimentally confirmed that the proposed method reduces the characteristic deterioration of the antenna 101.

ア ン テ ナ In this experiment, the antenna length was adjusted, but as an equivalent method, it is conceivable to variably control the characteristics of the matching circuit 1502. FIG. 34 shows a specific configuration thereof.

In FIG. 34, 1521 is an antenna element shorter than a quarter wavelength, 1522 is a variable capacitor which is a part of a matching circuit, and 1523 is a path for preventing a control power supply 1526 from flowing directly into the high frequency source 1527. 1524, an inductance for preventing high frequency from flowing into the control circuit 1503, 1525, a variable resistor for controlling a voltage applied to the variable capacitor 1521, and 1528, a resistor.

(4) Changing the value of the variable resistor 1525 changes the value of the variable capacitor. As the value of the variable capacitance increases, the antenna 101 appears equivalently elongated, thereby lowering the resonance frequency. When the value of the variable capacitance decreases, the antenna 101 appears to contract, and the resonance frequency increases. In this way, by changing the resonance frequency of the antenna 101, the matching condition can be changed.

As described above, according to the portable wireless device of this embodiment, the characteristics of the antenna when the human body is in proximity can be improved.

According to each of the above embodiments, it is possible to reduce the current consumption and increase the efficiency of the portable wireless device. Also, time-invariant and time-varying DC offsets that increase the error rate of the receiving unit can be removed. Further, the size of the portable wireless device can be reduced. Further, the speed of the empty channel search can be increased.

FIG. 6 is a block diagram for detecting power in a system band and power of a desired wave for reducing power consumption of a receiving unit according to the present invention. The figure which shows the control flow for the receiving part low power consumption by this proposal. FIG. 2 is a diagram illustrating a configuration example of a receiver having a self-compensation function according to the present invention. The figure which shows the reverse characteristic of AC coupling used for the self-compensation of this invention. FIG. 4 is a diagram illustrating a desired wave after self-compensation according to the present invention. FIG. 4 is a diagram illustrating a DC offset output when a local oscillator frequency is detuned. The figure which shows the structure of the receiver of another embodiment provided with the self-compensation function. FIG. 1 is a diagram illustrating a configuration of a direct conversion wireless device according to one embodiment of the present invention. FIG. 9 is a diagram showing a modification of the wireless device of the embodiment shown in FIG. 8. FIG. 9 is a diagram showing another modification of the wireless device of the embodiment shown in FIG. 8. FIG. 1 is a diagram illustrating a configuration of a synthesizer of a wireless device according to one embodiment of the present invention. The figure explaining the high-speed free channel search operation | movement of this synthesizer. FIG. 3 is a basic conceptual diagram when a sensing means is provided in a front-end high-frequency IC. FIG. 3 is a basic conceptual diagram when a sensing means and a power detector are provided in a front-end high-frequency IC. FIG. 1 is a diagram showing a configuration of a PA-IC according to one embodiment of the present invention. (A) is a figure which shows an example of the sensing means (sensing means) of the PA-IC of FIG. (B) is a figure which shows another example of a sensing means (sensing means). FIG. 17 is a diagram illustrating an example of the variable gain control circuit illustrated in FIG. FIG. 2 is a diagram illustrating a configuration of a general T / R switch. FIG. 1 is a diagram showing a configuration of a T / RswitchIC according to an embodiment of the present invention. The figure which shows an example of a sensing means. The figure which shows the other example of a sensing means. FIG. 2 is a diagram illustrating a configuration of a PA-IC according to one embodiment of the present invention. FIG. 1 is a diagram showing a configuration of a T / RswitchIC according to an embodiment of the present invention. FIG. 17A is a plan view showing one embodiment in which the elements for power sensing and detection shown in FIG. 16 are formed on an IC chip. 24B is a sectional view taken along line AA ′ of FIG. The figure which shows the example which adjusted the coupling capacity by making the line width of the metal layer narrower than the power supply line. The figure which shows the example which adjusted the coupling capacitance by making the line width of the metal layer wider than the power supply line. The figure which shows the example which adjusted the coupling capacity by changing the line length of the metal layer. The figure which shows the example which adjusted the coupling capacity by changing the formation direction of the metal layer. FIG. 9 is a diagram illustrating a configuration of a portable wireless device according to another embodiment of the present invention. FIG. 2 is an external view of a wireless device model used in the experiment. FIG. 30 is a diagram illustrating a relationship between current consumption and an operation state of the portable wireless device in FIG. 29. FIG. 4 is a diagram illustrating a relationship between a reflection coefficient of an antenna input terminal and an operation state of a portable wireless device when viewed from a power supply line side. The figure which shows the relationship between the average radiation gain in a horizontal plane of this portable radio, and an operating state. FIG. 3 is a diagram illustrating a configuration of a matching circuit of an antenna. The figure which shows the structure of the conventional radio | wireless apparatus (heterodyne system). FIG. 3A is a diagram illustrating a DC offset generated by reflection of an LO signal output from a local oscillator. FIG. 3B is a diagram illustrating a DC offset generated in addition to that of FIG. It is a figure showing a DC offset output. FIG. 4 is a diagram illustrating the effect of AC coupling on a desired signal. The figure which shows the structure of the conventional radio | wireless apparatus (direct conversion method). FIG. 6 is a block diagram showing a synthesizer of a conventional wireless device. The figure which shows the conventional power detection circuit. The figure which shows the vertical polarization radiation pattern of a PHS terminal at the time of a human head wearing. FIG. 43 is a diagram showing a vertically polarized radiation pattern in the case of a PHS terminal smaller than the PHS terminal of FIG. 42.

Explanation of reference numerals

101: antenna, 102: high frequency amplifier, 103: high frequency filter, 104, 105, 106, 157, 158: frequency converter, 108, 131, 2133: π / 2 phase shifter, 109, 110, 159, 160: low Frequency filters, 111, 112 ... low frequency amplifiers, 113, 114 ... A / D converters, 115, 116, 2131, 2132 ... multipliers, 117 ... detectors, 118, 141 ... storage devices, 119, 120, 121, 122, 123, 124 ... capacitors, 107, 125, 130 ... local oscillators, 126 ... frequency converters, 132-1, 132-2, 132-3 ... DC offset control signal input terminals, 133, 134 ... adders or subtractors , 137: transmission block, 138: transmission signal generator, 139: DC offset control circuit, 151 ... Power amplifier, 152: variable attenuator, 156: adder for modulator, 161, 162: digital / analog converter, 170: transmission / reception switch, 171: transmission power control circuit (CONT), 172: directional coupler, 301 .. Desired signal, 302... AC couple frequency characteristic, 303... Signal deleted portion, 304... Thermal noise, 305... DC output, 306 and 307... Notch, 401, 402, 403 and 404. 702: Notch, 801: Frequency characteristic of AC couple, 802: Inverted characteristic of AC couple, 803: Part of the DC component of the inverse characteristic, 804: Inverted characteristic, 901: Sweep oscillator, 902: Switch, 903: I channel Test signal input, 904: Q channel test signal input, 905: I channel frequency characteristic, 906: Q channel Channel frequency characteristics, 907 arithmetic unit, 1101 reference oscillator, 1103 reference frequency divider, 1105 phase comparator, 1109 voltage controlled oscillator, 1111 comparison frequency divider, 1107, 1151 normal loop filter, 1152 ... High-speed loop filter, 1153... Changeover switch, 1160... 1 frame used in TDMA system, 1161... 1 slot used in TDMA system, 1200. PA-IC (or transmission / reception switch IC), 1201. means, 1202 ... power detector DET (detect means), 1203 ... impedance Zvdd parasitic on the power supply line, 1204 ... impedance Zgnd parasitic on the ground line, 1205 ... variable gain high frequency amplifier (AMP), 1401 ... second layer ( Power supply line of the surface layer) 1402. (Inner layer) metal layer, 1404: insulating layer, 1500: housing, 1501: wireless circuit, 1504: ammeter, 1503: control circuit, 1502: matching circuit, 1505: power circuit, 1506: probe for measuring current, 1509: transmission amplifier, 1520: antenna element, 1521: variable capacitance element, 1523: capacitance element, 1524: inductance element, 1525: variable resistance element, 1526: DC power supply, 1527: high frequency wave source, 1510: constant voltage source, 1511 ... Speaker, 1512: Microphone, 1513: Ammeter, 1514: Antenna cover, PA: Power amplifier, T / R: Transmission / reception switch, AMP: Variable gain high frequency amplifier, Z: Impedance circuit, MIX: Frequency converter, LO: Local Signal generator, BPF ... Band pass filter, CPL ... Coupler, ... Transmission power detection circuit (DET), ANT: antenna, D1: diode, RES: resistor, Cn (n = integer): capacitor, Qn (n = integer): transistor, Zn (n = integer): impedance circuit, In ( n = integer) ... current source.

Claims (1)

A wireless circuit that supplies a signal transmitted from an antenna via a transmission amplifier, a power supply circuit that supplies power to the wireless circuit and the transmission amplifier via a power supply line, an ammeter connected to the power supply line, An antenna characteristic varying means for varying a matching characteristic of the antenna based on a current value detected by an ammeter.

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