JP2001339258A - Voltage-current converting circuit - Google Patents

Abstract

PROBLEM TO BE SOLVED: To reduce fluctuation in output current Iout by widening the range of an input voltage Vin capable of controlling the output current Iout to 0V to a source voltage VDD. SOLUTION: A voltage developed by making a current flows through a resistance and the input voltage Vin are inputted to an operational amplifier 1 and a transistor is controlled with the output of the operational amplifier 1 to control the output current.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a circuit for inputting a voltage and outputting a current.

[0002]

2. Description of the Related Art FIG. 6 shows a conventional voltage-current converter, and FIG. 7 shows voltage-current characteristics of the conventional voltage-current converter. In the conventional voltage-current converter, when the input voltage Vin is higher than the threshold value Vtn of the transistor 2d, the output current Iout is determined with respect to the input voltage Vin in accordance with the following equation if the size of the transistor is sufficiently increased. . Iout = (Vin−Vtn) / Rd (1) Here, when the resistance value input voltage Vin of the resistor 3d is lower than the threshold value Vth of the transistor 2d, the transistor 2d does not turn on. Not flowing. Therefore, in the conventional voltage-current conversion circuit, the range of the input voltage Vin that can control the output current Iout is VDD-Vtn.

However, when the power supply voltage decreases, the input voltage range V that can control the output current Iout is reduced.
DD-Vtn becomes narrower than the voltage drop. Because, to meet the transistor leakage current tolerance,
This is because normally, the threshold value Vtn of the transistor cannot be reduced at the same rate as the power supply voltage.

When the range of the input voltage that can control the output current Iout becomes narrow, the gradient of the output current Iout / input voltage Vin becomes steep. The PLL shown in FIG.
When a conventional voltage-current conversion circuit is used in a feedback control system such as a DLL shown in (1), the gradient dIout / dVin of the output current Iout with respect to the input voltage Vin is steep, so that the output current Iout fluctuates greatly, The output of the control system fluctuated greatly.

[0005]

The problem to be solved is that the range of the input voltage that can control the output current Iout is narrow.

[0006]

The present invention provides an output current Io.
ut can be controlled by setting the input voltage Vin range to 0.
In order to extend the voltage from V to the power supply voltage VDD, the most main feature is to compare a voltage generated by flowing a current through the resistor with an input voltage and control the current so that the two voltages match.

[0007]

DESCRIPTION OF THE PREFERRED EMBODIMENTS The purpose of enabling the output current Iout to be controllable with respect to an input voltage in a range of 0 V to a power supply voltage VDD is to input a voltage generated by flowing a current through a resistor and an input voltage to an operational amplifier. This is achieved by controlling the current by controlling the transistor with the output.

[0008]

1 is a circuit diagram of an embodiment of the device of the present invention, wherein 1 is an operational amplifier, 2a is an NMOS transistor,
3a is a resistor.

The output N2 of the operational amplifier 1 is connected to the gate of the NMOS transistor 2a. The output current Iout increases as the output voltage of the operational amplifier increases, and the output current Iout decreases as the output voltage of the operational amplifier decreases. The operational amplifier 1 amplifies the potential difference between the node N1 and the input voltage Vin.
This circuit controls to be equal to

Therefore, the output current Iout is given by the following equation. Iout = Vin / Ra (2) Here, Ra indicates the relationship between the resistance value input voltage Vin of the resistor 3a and the output current Iout by a solid line in FIG.

In this embodiment, the input voltage Vin is 0 V to V
The output current Iout can be made proportional to Vin in the range of DD. Therefore, when the maximum value of the output current Iout is the same, in this embodiment, the range of the input voltage Vin can be made wider than in the conventional method, and the slope dIout / dVin of the output current with respect to the input voltage Vin can be made smaller. As a result, fluctuations in the output current caused by fluctuations in the input voltage can be further reduced. Therefore, if this circuit is used for a feedback system such as a PLL, the fluctuation can be reduced as compared with the related art.

In this embodiment, in order for the node N1 to reach the power supply voltage VDD, the output N2 of the operational amplifier 1 needs to drive a voltage equal to or higher than VDD + Vtn. Assuming that the voltage supplied to the operational amplifier 1 is VDD, the input voltage Vin is equal to VDD−Vtn as shown by a broken line in FIG.
In the above region, the output current becomes constant. Therefore, in order for the output N2 of the operational amplifier to drive a voltage equal to or higher than VDD + Vtn, the power supply voltage supplied to the operational amplifier must be equal to or higher than VDD + Vtn.

In the embodiment of FIG. 3, a booster circuit 13 is added to the circuit of the embodiment of FIG. 1, and power is supplied to the operational amplifier 1 at the output VCC of the booster circuit 13. In this configuration, only one type of power supply is required, VDD, so that the present embodiment is more suitable than the embodiment of FIG. 1 when only one type of power supply voltage is supplied. <BR>

In the embodiment of FIG. 4, the circuit of the embodiment of FIG. 1 and the conventional circuit of FIG. 5 are connected in parallel.

In this embodiment, the maximum output voltage of the operational amplifier 1 is up to the power supply voltage VDD. Output current Iou
t is represented by the following equation. Iout = Iout1 + Iout2 (3) Iout1 = Vin / Rb (0 ≦ Vin <VDD−Vtn) = (VDD−Vtn) / Rb (VDD−Vtn ≦ Vin ≦ VDD) (4) Iout2 = 0 (0 ≦ Vin <Vtn) = (Vin−Vtn) / Rc (Vtn ≦ Vin ≦ VDD) (5) where Rb is the resistance of the resistor 3b, and Rc is the resistance of the resistor 3c.

Also in this embodiment, the output current Iout can be controlled when the input voltage Vin is in the range of 0 V to VDD as in the embodiment of FIG. Further, in this embodiment, the voltage supplied to the operational amplifier 1 may be the power supply voltage VDD, so that it is not necessary to supply two kinds of power supply voltages from the outside or add a booster circuit.

Further, in the present invention, Iout / V in the region of 0 ≦ Vin <Vtn and VDD−Vtn ≦ Vin ≦ VDD.
The slope of “in” can be independently determined by the resistance values Rb and Rc, respectively. For this reason, the variation of the output current Iout can be changed according to the region of the input voltage Vin. For example, if the value of Rb is made larger than Rc, Iou
dIout / dVin when t is small can be made smaller than dIout / dVin when Iout is large.
As a result, current controllability in a region where Iout is small is improved.

As described above, the present invention has been described based on the embodiments. However, the present invention is not limited to the above-described embodiments, and can be freely modified and changed within the scope described in the claims. It is.

[0019]

As described above, the input voltage range of the voltage-current conversion circuit of the present invention can be widened, and the output current / input voltage gradient can be reduced. Can be smaller. Further, since the output current / input voltage gradient can be changed depending on the input voltage region, the fluctuation of the output current can be changed according to the input voltage region.

[Brief description of the drawings]

FIG. 1 is a circuit diagram showing a method for implementing a voltage-current conversion circuit. (Example 1)

FIG. 2 is a diagram showing voltage-current characteristics of Example 1.

FIG. 3 is a circuit diagram showing a method for implementing a voltage-current conversion circuit. (Example 2)

FIG. 4 is a circuit diagram showing a method for implementing a voltage-current conversion circuit. (Example 3)

FIG. 5 is a diagram showing voltage-current characteristics of Example 2.

FIG. 6 is a circuit diagram showing an implementation method of a conventional voltage-current conversion circuit.

FIG. 7 is a diagram showing a voltage-current characteristic of a conventional voltage-current converter.

FIG. 8 is a block diagram of a PLL.

FIG. 9 is a block diagram of a <BR> DLL.

[Explanation of symbols]

 DESCRIPTION OF SYMBOLS 1 Operational amplifier 2a, 2b, 2c NMOS transistor 3a, 3b, 3c Resistance 4 Phase frequency comparator 5 Charge pump 6 Loop filter 7 Voltage control oscillator 8 Current-voltage conversion circuit 9 Current control oscillator 10 Phase comparator 11 Voltage control delay line 12 Current Control delay line 13 Boost circuit

Continued on the front page F-term (reference) 5H430 BB01 BB09 BB12 EE06 FF08 FF13 GG11 HH03 5J090 AA03 AA42 CA04 CA34 CA81 CN04 FA05 FA10 FA17 FN06 FN10 HA10 HA16 HA25 HN07 KA01 KA02 KA12 KA17 KA18 MA03 MA03 MA04 CA81 FA05 FA10 FA17 HA10 HA16 HA25 KA01 KA02 KA12 KA17 KA18 KA47 MA08 MA13 MA21 TA01 TA02

Claims (5)

[Claims] A drain is an output, a source is connected to a first resistor and connected to a first potential, and a gate is a first insulated gate type of one conductivity type receiving an output of a first differential amplifier. A field-effect transistor, a first input signal having a positive input,
Voltage-current conversion circuit, comprising a first differential amplifier having a source of the insulated gate field effect transistor as a negative input. 2. The method according to claim 1, wherein the supply voltage to the first differential amplifier is higher than the amplitude of the first input signal by at least the threshold voltage of the first insulated gate field effect transistor. Voltage-current conversion circuit described 3. The power supply circuit according to claim 1, further comprising a first booster circuit for supplying power to the first differential amplifier.
Voltage-current conversion circuit described in 4. An insulated gate field effect transistor of one conductivity type having a drain as an output, a source connected to a second resistor and connected to a first potential, and a gate receiving a first input signal. The voltage-current conversion circuit according to claim 1, further comprising: 5. The voltage-current conversion circuit according to claim 4, wherein the value of the first resistor is larger than the value of the second resistor.