JP2000106577A - Method for compensating distortion of frequency converter and radio receiver - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a radio receiver with a distortion compensation function that ensures excellent linearity without causing increase in adjustment components and increase in a consumed current that leads to an increased cost of a frequency converter. SOLUTION: An IQ frequency converter 103 receives an RF signal from an antenna 101 and converts the frequency of the signal into a lower frequency by using two orthogonal reference signals generated by a local oscillator 105 and a π/2 phase shifter 106, A/D converter 109a, 109b convert the signal into a digital signal and a digital signal processing section 110 conducts demodulation of the signal through digital processing in a direct conversion receiver. The direct conversion receiver has a control section 114 that gives a distortion control signal 123 for changing secondary distortion to the IQ frequency converter 103 when an error detection circuit 113 detects a reception error of the RF signal consecutively for a prescribed number of times from an output of a demodulator 111.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a method for compensating distortion of a frequency converter suitable for a direct conversion receiving system used in a portable terminal in a digital mobile communication system and a radio receiving apparatus using the same.

[0002]

2. Description of the Related Art In recent years, with the development of mobile communication, demands for miniaturization and price reduction of portable terminals have been increasing. In such a mobile terminal, it is particularly important to reduce the size and cost of the wireless receiving unit. In order to reduce the size of the receiving radio unit, there are two methods, a method of mounting ultra-small circuit components at high density and a method of using a receiving method that does not require the use of large components. In the former case, the price of a micro component is generally higher than that of a normal component. Therefore, in consideration of the price, it is desirable to use the latter receiving method that does not require the use of large components.

A direct conversion receiving system is known as one of the receiving systems which can realize the miniaturization and low cost of the receiving radio section. The direct conversion reception system is a reception system in which a received RF signal is mixed with a local signal having the same frequency as this, and is directly frequency-converted into a baseband for demodulation. The basic configuration is shown in FIG.

[0004] An RF signal received by an antenna 1301 is amplified by a high-frequency amplifier 1302 and then divided into two channels to provide two frequency converters (mixers) 1303.
a, 1303b, two local signals (reference carrier signal) having the same frequency as the carrier frequency of the received RF signal and a phase difference of 90 ° generated by the local oscillator 1305 and the π / 2 phase shifter 1306. Is converted into two baseband signals having a phase difference of 90 °. These baseband signals pass through low-pass filters 1307a and 1307b, respectively, and then pass through baseband amplifiers 1308a and 1308a.
8b. Baseband amplifier 1308
a, 1308b are output from the A / D converter 1309
After being converted into a digital signal by a and 1309b, it is input to a digital signal processing unit 1310 and demodulated by digital signal processing.

As described above, the direct conversion receiving system directly converts the frequency of an RF signal into a base band. Therefore, the direct conversion receiving system has no intermediate frequency and has no image response in principle. There is an advantage that a steep filter for removing an image used is unnecessary, and a filter for selecting a baseband channel can be realized in an LSI, so that it can be realized at low cost.

However, when the direct conversion receiving method is applied to a wireless communication system using a modulation method in which an amplitude component exists in a signal envelope such as QPSK or π / 4-QPSK, there are the following problems. FIG.
FIGS. 4A and 4B are diagrams for explaining this problem. FIG. 4A shows a level relationship between a desired wave and an interference wave, and FIG. 4B shows a relationship between a desired wave after frequency conversion and secondary distortion. In the wireless communication system, in addition to the desired wave 1401 received as shown in FIG.
There are two. At this time, the desired wave 1401 and the interference wave 14
02 level difference, ie, desired wave to interference wave ratio (D / I)
As shown in FIG. 14A, it is necessary to realize a value of 50 to 60 dB or more.

As described above, the interference wave 1402 is
If the value is very large with respect to 1, nonlinear distortion occurs due to the interference wave 1402. Normally, third-order distortion is a problem in the superheterodyne receiving system, but in the direct-conversion receiving system, secondary distortion mainly occurs as nonlinear distortion.

FIG. 15 shows an outline of a frequency converter (mixer) that generates secondary distortion. Frequency converter 1303a (1
303b) mixes the RF signal input to the RF input port with the local signal supplied to the local input port. For direct conversion receivers, RF
The signal and the local signal have substantially the same frequency, and the mixed output is frequency-converted to a baseband (BB) band and output from a BB output port.

Here, when the desired wave 1401 and the interference wave 1402 as shown in FIG. 14A are inputted to the RF input port, the output of the BB output port becomes as shown in FIG. 14B. . That is, the interference wave 1402 and the second-order distortion 1404 generated by the nonlinearity of the frequency converter 1303a (1303b) are superimposed on the desired wave 1403 after the frequency conversion. When the desired wave 1403 is a QPSK signal, in order to secure a predetermined reception performance, the desired wave 1 after frequency conversion is used.
It is necessary to set the ratio between 403 and second-order distortion 1404, that is, the desired wave-to-distortion ratio D / U to 15 dB or more.

[0010] Such second-order distortion generally causes a problem in a system handling an amplitude modulation signal, and does not particularly cause a problem in a wireless communication system using a frequency modulation signal (for example, Ref. 1: RG Meyer, MJ Shensa,
R. Eschenbach: “Cross Modulation and Intermodulatio
n in Amplifiers at High Frequencies ”IEE Journalof
Solid-State Circuits, Vol.SC-7, No.1, pp.16,23 Febru
ary 1972).

In recent years, QPs used in wireless communication systems such as mobile telephone systems and cordless telephone systems have been developed.
In a linear modulation scheme with band limitation such as SK or π / 4-QPSK, since the signal component has an amplitude component, F
There is a problem that second-order distortion is more likely to occur as compared with the constant envelope modulation methods such as M and FSK. In a wireless communication system using a modulation method having such an amplitude component, it is very difficult for a frequency converter in a direct conversion receiver to satisfy particularly adjacent channel sensitivity suppression characteristics due to secondary distortion. .

Since the secondary distortion in the frequency converter is generated without depending on the frequency relationship between the frequency fD of the desired wave and the frequency fI of the interference wave, it cannot be solved only by adjusting this frequency relationship. It is essential to increase the linearity of the vessel itself, and various techniques have been devised. However, the desired wave to interference wave ratio D / I of 50 to 60 dB and 15 d
In order to realize a desired wave contrast D / U of B or more, it is necessary to realize a value of 65 to 75 dB as linearity (IM2) for second-order distortion. This linearity is a very large value,
Considering changes in temperature characteristics and other usage environments,
It is difficult to always be satisfied. Even if this value can be achieved, the current consumption of the circuit will increase, and the cost will increase due to the increase in the number of adjustment components, which will impair the low cost advantage of the direct conversion receiving method. Not realistic.

[0013]

As described above, in the conventional direct conversion receiver, a non-linear circuit in the receiver, in particular, a second-order non-linear component of the frequency converter causes a second order non-linear component of the interference wave to exist in the system band. Since the secondary distortion output is output in a desired frequency band after frequency conversion, reception becomes difficult. Therefore, when a direct conversion receiver is used in an actual radio wave environment where an interference wave exists, the nonlinear distortion characteristic of the frequency converter becomes a problem as compared with a conventional superheterodyne receiver, and especially, the direct conversion receiver is closest to a desired wave. It is very difficult to satisfy the specification of the adjacent channel sensitivity suppression that defines the interference characteristic of the interference wave having a different frequency, that is, the distortion characteristic when the adjacent channel interference wave is input.

Further, when the linearity of the frequency converter is to be improved, the current consumption of the circuit is increased, the cost is increased due to an increase in the number of adjustment components, and the advantage of the direct conversion receiver is impaired. there were.

SUMMARY OF THE INVENTION The present invention has been made to solve such a conventional problem, and does not cause an increase in current consumption and an increase in adjustment components which cause a cost increase in the frequency converter. It is an object of the present invention to provide a distortion compensation method capable of ensuring good linearity and a radio receiving apparatus using the distortion compensation method.

[0016]

According to the present invention, there is provided a radio receiving apparatus including a frequency converter for converting a received high-frequency signal to a low frequency. When detected a predetermined number of times,
Distortion compensation of the frequency converter is performed by controlling the frequency converter to change the amount of distortion.

A radio receiving apparatus according to the present invention to which the distortion compensation method is applied includes a receiving means for receiving a high-frequency signal modulated by an information signal, a reference signal source for generating a reference signal, and a receiving means for receiving the signal. A frequency converter configured so that the converted high frequency signal is converted into a lower frequency by the reference signal and output, and a distortion amount is changed by external control, and an output signal output from the frequency converter is received. Demodulation means for demodulating the information signal, an error detection means for detecting a reception error of a high-frequency signal from an output of the demodulation means, and a frequency conversion when the reception error is detected by the error detection means a predetermined number of times in succession. And control means for controlling the distortion so that the distortion amount changes.

In this way, the frequency converter is configured so that the amount of distortion can be changed by external control, and when a reception error is continuously detected a predetermined number of times, the amount of distortion is changed. If the control is stopped when the amount of distortion becomes equal to or less than the allowable value, good linearity is ensured and correct reception becomes possible.

Further, in this case, the requirement for the linearity of the frequency converter itself is relaxed, and an increase in current consumption and an increase in the number of adjustment components which cause an increase in cost are prevented.
Good linearity can be ensured.

According to the present invention, first and second orthogonal reference signals are generated by a reference signal source, and a high-frequency signal received as a frequency converter is converted into a lower frequency by the first and second reference signals. 1st and 2nd configured to output and change the amount of distortion by external control
, And can be applied to a radio receiving apparatus configured to receive output signals output from these first and second frequency converters and demodulate an information signal. It is only necessary to provide control means for controlling the frequency converter of (1) so that the distortion amount changes.

In this case, the control means may send a common control signal to the first and second frequency converters to perform common control, or may send separate control signals to each and separately. Control may be performed.

The control means for controlling the frequency converter compares the maximum value of the distortion generated by the frequency converter within the control range with a predetermined desired wave-to-distortion ratio to an input conversion value. When the measured reception electric field strength is small, the frequency converter may be controlled so that the distortion amount changes.

Further, the control means for controlling the frequency converter outputs a control signal consisting of a digital signal, and changes the control signal toward a point sequentially separated from both adjacent points of the initial value. You may.

[0024]

Embodiments of the present invention will be described below with reference to the drawings. (First Embodiment) FIG. 1 shows a configuration of a direct conversion receiver which is a radio receiving apparatus using a distortion compensation method according to the present invention. In FIG. 1, a high frequency signal (RF signal) modulated by an information signal (transmission data) such as a QPSK signal or a π / 4-QPSK is
A signal is received. The received RF signal output from the antenna 101 is amplified by the high frequency
The signal is input to the frequency converter 103.

The IQ frequency converter 103 is composed of two frequency converters 104a and 104b, and is generated by a local oscillator 105 and a π / 2 phase shifter 106 which oscillate at substantially the same frequency as the carrier frequency of the received RF signal. By mixing two local signals (reference carrier signal) having a phase difference of 90 ° and a received RF signal, two baseband signals (I and Q signals) having a phase difference of 90 ° are output. Note that the frequency converter 104
a and 104b are hereinafter referred to as mixers.

The two baseband signals output from the mixers 104 a and 104 b are
a and 107b, respectively, and the baseband amplifier 1
08a, 108b, and then A / D converter 1
The signals are respectively converted into digital signals by 09a and 109b and input to the digital signal processing unit 110. The digital signal processing unit 110 includes a demodulator 111, a received electric field strength (RSSI) measuring circuit 122, and an error detecting circuit 113.
And a control unit 114.

The demodulator 111 includes an A / D converter 109a,
Demodulate the digital baseband signal from 109b,
Play the original transmission data. For example, if the received RF signal is Q
In the case of a PSK signal, the demodulator 111 performs demodulation by delay detection, synchronous detection, or the like. The error detection circuit 113 detects a reception error, that is, an error of the received RF signal, from the demodulation result 121 of the demodulator 111. More specifically, the error detection circuit 113 measures a bit error rate (BER) or a frame error rate (FER) of a known signal portion included in the received RF signal, and when the BER value is equal to or more than a predetermined value. Sends the result 122 to the control unit 114.

The control section 114 receives the error detection result 122 from the error detection circuit 113, and when a reception error is detected a predetermined number of times continuously, outputs the distortion amount control signal 123 to the I.
It is transmitted to the Q frequency converter 103. Upon receiving the distortion amount control signal 123, the IQ frequency converter 103 outputs the distortion amount,
In particular, the second-order distortion is controlled so as to change.
The adjustment is performed so that the secondary distortion amount is equal to or less than the allowable value.

The reception electric field strength measuring circuit 112 measures the reception electric field strength of the RF signal at the input terminal of the antenna 101. FIG. 2 shows a detailed configuration example of the reception electric field strength measurement circuit 112. In this reception electric field strength measurement circuit 112,
The outputs (I, Q) of the A / D converters 109a and 109b in FIG.
Signal), the multipliers 201a and 201b calculate the squares of the I and Q signals, respectively, and then add these to the adder 2
02 and further added by the square root circuit 203 to (I ² +
Calculate the square root of Q ² ). And the square root circuit 203
Is compared with the specified value stored in the memory 205 by the comparator 204, and the result is sent to the control unit 114.

In order to calculate the received electric field strength at the input terminal of the antenna 101, for example, the received electric field strength measuring circuit 1
The output voltage level of the A / D converter 109a, 109
An operation of subtracting the total gain of the signal path up to 9b may be performed.

Next, the IQ frequency converter 10 will be described with reference to FIG.
3 will be described. The IQ frequency converter 103 is configured as shown in FIG.
Distortion control signal 123 transmitted from the control unit 114 of FIG.
3 shows a specific configuration example of the mixers 104a and 104b constituting the IQ frequency converter 103. FIG.

As an example of a specific circuit configuration for controlling the amount of second-order distortion, reference 2: T. Yamaji, H. Tanimoto, an
d H. Kokatsu: “An I / Q active balanced harmonic mixe
r with IM2 cancellers and a 45 ° phase shifter ”,
The circuit configuration disclosed in IEEE ISSCC '98 Dig.Tech.Papers, SP 23.3, 1998. is considered, and FIG. 3 shows this.

FIG. 3 shows one of the mixers 104a and 104b. Two sets of differential pair transistors Q1 and Q2 are shown.
And a balanced mixer mainly composed of Q3 and Q4. 1 is input between the bases of transistors Q1 and Q3 in the form of a differential signal, and a local (LO) signal is
2, Q4. Differential pair transistor Q
1 and Q2 have a common emitter of transistor Q5 and resistor R
1, and the common emitter of the differential pair transistors Q3 and Q4 is also connected to a current source by a transistor Q6 and a resistor R2. Transistor Q
5, the base of Q6 is provided with a constant bias voltage Vb. The common collector of the transistors Q1 and Q4 and the common collector of the transistors Q2 and Q3 are connected to load resistors R3 and R4, respectively, and connected to an output circuit including an amplifier A and resistors R5 and R6. The frequency-converted signal is output to OUT.

A variable current source CS for adjusting the amount of distortion is connected between the emitters of the transistors Q5 and Q6 constituting the two current sources. By adjusting the current value of the variable current source CS, two sets of differential pair transistors Q
The balance of the bias current to Q1, Q2, Q3, and Q4 is adjusted to change the amount of distortion of the mixer, particularly the amount of secondary distortion. The current value of the variable current source CS is called a mixer bias. This mixer bias is
The control is performed according to the distortion control signal 123 from the control unit 114 in FIG.

The configuration of the mixer to which the distortion compensation method according to the present invention can be applied is not limited to the example shown in FIG. 3 as long as the secondary distortion amount can be adjusted by an external control signal. However, any configuration may be used.

Next, the mixers 104a and 104a will be described with reference to FIG.
Next, a description will be given of a secondary distortion generated in 4b, a method of controlling the distortion amount, and a change in output due to the distortion amount control. FIG. 4 shows a desired wave output of the mixer having the secondary distortion amount adjusting function shown in FIG.
The measurement results of the interference wave output and the secondary distortion output are shown. The vertical axis represents the output level (dBμ), and the horizontal axis represents the level of the distortion control signal 123 (IM2 control signal code). In this example, the distortion amount control signal 123 is given as a code (IM2 control signal code) composed of a 3-bit digital signal, and 2 ³ = 8 indicated by “0” to “7” on the horizontal axis in FIG. Value. In the following description, for the sake of simplicity, it is assumed that the desired wave output, the interference wave output, and the secondary distortion output in FIG. 4 have not been subjected to gain compression due to saturation of the mixer circuit.

In FIG. 4, reference numeral 401 denotes an interference wave output;
2 is a second-order distortion output generated by the interference wave output 401;
03 is a desired wave output. Since the level of the desired wave output 403 in FIG. 4 is a value at the time of the reception sensitivity input (minimum signal input) level, it is not necessary to consider the case where the level of the desired wave output 403 is lower than this. D / I is a desired wave-to-interference wave ratio viewed from the output side of the mixer in the case of FIG. This D /
The value of I is required to be in the range of 50 to 60 dB as a system specification as described above.

At this time, the allowable value 40 of the secondary distortion output 402
4 is a value obtained by subtracting the required D / U 405 of about 15 dB from the desired wave output 403 as described above.
Suppressing the secondary distortion output 402 below 04 is the required specification for the secondary distortion amount of the mixer. Here, the secondary distortion suppression value IM2 representing the degree of distortion suppression of the mixer is the difference between the interference wave output 401 and the secondary distortion output allowable value 404.

In the example of FIG. 4, the IM2 control code is "6".
, The minimum value 409 of the secondary distortion output 402 is obtained. When the IM2 control code is set to “0”, the secondary distortion output 402 has a maximum value 406. According to the example of FIG. 4, it can be seen that as a whole tendency, the secondary distortion output 402 is deteriorated around the IM2 control code “6”. More specifically, when the IM2 control codes “4”, “5”, “6” and “7” are used, the secondary distortion output 40
2 is suppressed to the allowable value 404 or less, and it can be seen that the secondary distortion output 402 exceeds the allowable value 404 with the IM2 control codes “0”, “1”, “2”, and “3”.

In order to correctly receive a desired wave even when the secondary distortion output 402 is the maximum value 406, a level 407 obtained by adding a required D / U 408 to the allowable value 404 is used.
The above is required as the desired wave output 403. On the other hand, when the desired wave output 403 is lower than the level 407, the IM2 control code is changed to any one of “4”, “5”, “6”, and “7” in which the secondary distortion output 402 is equal to or less than the allowable value 404. Need to be changed. That is, the second-order distortion output 402
Is the maximum value 406, the desired wave output 403 is level 40
If it is lower than 7, the distortion amount control signal 123 may be sent to the mixer to adjust and reduce the secondary distortion amount. On the other hand, when the secondary distortion output 402 is not smaller than the allowable value 404, the IM2 control code is set to “4”, “5”, “6”.
What is necessary is just to move to any of "7".

Next, in order to explain a specific procedure of the distortion compensation method according to the present invention, various communication sequences including the distortion compensation sequence according to the present invention will be described with reference to FIGS. (Regarding the call sequence in FIG. 5) FIG.
In TDD system and TDMA / FDD system,
This is a form in which a distortion compensation sequence based on the distortion compensation method of the present invention is added to a communication sequence that can be used in a cordless telephone or the like in which a base station has a plurality of time slots at one frequency. Hereinafter, a communication sequence including the distortion compensation sequence of FIG. 5 will be described.

First, in step 501, the terminal is turned on, and in step 502, a control channel (CCH) transmitted from the base station (BS) is captured to establish a link (link channel) with the base station. I do. Then, in step 503, the traffic channel (TCH)
It becomes a transmission / reception and enters a call state. In a call state, the error detection circuit 113 in FIG. 1 uses a reception error, that is, a bit error rate (BE) using a known signal portion of a reception signal.
R) or the frame error rate (FER) is measured.
Here, error detection by CRC (Cyclic Redundancy Code) is assumed.

If a CRC error is detected in step 504, the next step 505 is to increment the CRC error cumulative number M so that M = M + 1. Also,
At step 505, simultaneously with the accumulated number M of CRC errors,
The CRC error detection count L used in the distortion compensation sequence 516 is also incremented.

As a cause of the CRC error and its countermeasures,
The following three are conceivable. (E1) When the interference wave is superimposed on the desired wave at the same time:
For this, it is necessary to change the transmission / reception slot (time). (E2) When the signal level is low and a CRC error occurs due to thermal noise: In this case, the base station (frequency) is changed. (E3) When secondary distortion occurs: This occurs when the secondary distortion output 402 exceeds the allowable value 404 in FIG. On the other hand, the second-order distortion compensation, that is, the mixer 10
4a and 104b are controlled.

In this embodiment, (E1) and (E2)
Is the same as the conventional method, and a distortion compensation sequence 516 shown in FIG. 5 is added as a sequence for (E3) based on the present invention. The distortion compensation sequence 516 will be described later in detail.

When a CRC error is detected in step 504, the received field strength (RSSI) is measured in step 506, and comparison with a predetermined level R1 is performed in step 507.
If the RSSI is smaller than the predetermined level R1, the value of N is incremented in step 508. Note that N is the number of times the RSSI has become smaller than the predetermined level R1. This operation is later performed when the cause of the CRC error is (E1),
It is necessary to determine which of (E2).

Next, in step 505, the value of the CRC error cumulative number M incremented when the CRC error is detected is compared with the specified number M1 in step 509, and M is set to the specified number M.
If it is larger than 1 (YES), the next step 51
According to the determination result of 0, the process proceeds to either “slot change request” in step 511 or “frequency change request” in step 513.

That is, if the number of times N at which the RSSI has become smaller than the predetermined level R1 is larger than the specified number N1 (YES) in step 510, the RSSI is smaller than the specified value, and (E2): the signal level is low. It is determined that a CRC error has occurred due to thermal noise.

On the other hand, if the number of times N at which the RSSI becomes smaller than the predetermined level R1 is smaller than the prescribed number N1 at step 510 (NO), (E1): the RSSI is larger than the prescribed value, and It is determined that the signal is superimposed on the desired wave at the time.

If the result of the determination in step 510 is (E2), that is, the signal level is low, and the CRC
If an error has occurred, the process proceeds to step 513 and issues a frequency change request to the BS.
Perform frequency change (handoff) in step 5
After capturing the CCH at 15 and establishing a communication link, the process returns to “TCH transmission / reception (communication)” of step 503.

On the other hand, the result of the determination at step 510 is (E
In the case of 1), that is, when the RSSI is larger than the specified value and the interference wave is superimposed on the desired wave at the same time, the process proceeds to step 511 to transmit a slot change request to the BS. Then, after the slot is changed in step 512 according to the instruction from the BS, the process returns to “TCH transmission / reception (communication)” in step 503.

It should be noted that the CRC error cumulative number M and the RSS
The value of the number N of times when I becomes smaller than the predetermined level R1 is cleared when a CRC error has not been detected for a predetermined time (not shown) in step 504, and the "slot change" in step 512 or the "change in slot" in step 515 is performed. Also cleared after "CCH capture".

Next, the distortion compensation sequence 516 in FIG. 5 will be described. In the conventional communication sequence, if the cumulative number of CRC errors M does not reach the specified number M1 in step 509, the process returns to “TCH transmission / reception (communication)” in step 503, and the steps from step 503 until the cumulative number of CRC errors M reaches M1. The processing up to 509 was simply repeated. In contrast, in the present embodiment, step 50 is performed.
Until it is determined in step 9 that the cumulative number of CRC errors M has reached the specified number M1, the process returns to step 503 through the distortion compensation sequence 516.

In the distortion compensation sequence 516, step 5
At 17, the number of CRC errors L is compared with a prescribed number L1. C
The number of RC errors L is calculated in step 5 every time a CRC error is detected.
At 16, the value is incremented together with the cumulative number M of CRC errors. Here, the following procedure is performed.

When the number L of CRC errors does not exceed the prescribed number L1 → The value (mixer bias) of the distortion amount control signal 123 transmitted from the control unit 114 to the IQ frequency converter 103 is kept as it is.

When the number L of CRC errors exceeds a prescribed number L1 → The value of the distortion control signal 123 (mixer bias) sent from the control unit 114 to the IQ frequency converter 103 is changed and generated by the mixers 104a and 104b. The amount of secondary distortion to be performed is adjusted (step 518). At this time, L
Is cleared.

That is, if a CRC error is detected, the mixer bias is changed by changing the value of the distortion control signal 123 each time the number L of CRC errors reaches the specified number L1 (step 518). Until it is determined in step 509 that the cumulative number M of CRC errors has reached the specified number M1, the amount of secondary distortion generated in the mixers 104a and 104b is changed, and TCH transmission / reception (step 503) and C
RC error detection (step 504) is continued. here,
If a CRC error is not detected for a predetermined time in the CRC error detection (step 504), as in the case of the conventional communication sequence, the cumulative number of CRC errors M, the number of times RSSI has become smaller than the predetermined level R1, and the number of CRC errors L Is cleared (not shown), and TCH transmission / reception (step 503) is continued.

Next, the specified number M of the cumulative number M of CRC errors
The relationship between 1 and the specified number L1 of the number of CRC errors L used in the distortion compensation sequence 516 will be described. Now, when the IM2 control code shown on the horizontal axis of FIG. 4 is considered as the distortion control signal 123, the IM2 control code (= mixer 104
It is preferable to equally allocate the biases a and 104b) within a specified number M1 that is the upper limit of the cumulative number M of CRC errors, and to perform CRC error detection under each mixer bias. That is, it is preferable that L1 = M1 ÷ (the number of settable mixer bias values = the number of IM2 control codes). For example, since the number of IM2 control codes is eight in the example of FIG.
Assuming that 1 = M1 ÷ 8, while performing CRC error detection,
It is preferable to adjust the secondary distortion amount.

During an actual call, even if the secondary distortion output 402 exceeds the allowable value 404, it does not become extremely large, and the IM2 control code indicates that the secondary distortion output 402 has a relatively small value. It is considered that it is set to be. For this reason, when the current IM2 control code (initial value) is, for example, in the state of the code “5” in FIG. 4, the second-order distortion output 402 can be smaller with the codes “4” and “6” on both sides. High in nature. Therefore, when changing the IM2 control code, more time is allocated to the codes “4” and “6” on both sides, in other words, the CRC error detection count L
It is effective to allocate the specified value L1 more. The time distribution, that is, the setting of the assignment of the specified value L1 may be determined according to the state of the secondary distortion output 402 from the mixers 104a and 104b actually used.

For the same reason, when changing the IM2 control code, it is effective to change the control code from the initial value of the control code to a point that is sequentially away from both adjacent points. For example, when the initial value of the IM2 control code is “4”, “4” → “3” → “5” → “2”
The control code may be changed in a zigzag manner in the order of “6” → “1” → “7” → “0”.

As described above, according to the communication sequence shown in FIG. 5, at least one point where the second-order distortion output 402 becomes the allowable value 404 or less within the control range of the control signal (OM2 control code) as shown in FIG. If there is one, (E
3): The error deterioration factor in the case "secondary distortion occurs" is automatically removed, and deterioration of the reception characteristic due to the second distortion can be avoided.

(Regarding the call sequence in FIG. 6)
6 will be described by assigning the same reference numerals to the same processing steps as in FIG. In the speech sequence of FIG. 6, the distortion compensation sequence 601 is different from the distortion compensation sequence 516 in FIG. 5, and before proceeding to the mixer bias change in step 518, the RSSI integral value ΔP is compared with the predetermined level R2 in step 602. Then, only when ΔP is smaller than R2, the process proceeds to step 518, where ΔP is equal to R2.
In the above case, the process returns to “TCH transmission / reception (communication)” in step 503. Here, the predetermined level R2 is R2
= Maximum secondary distortion output value 406 + Required D / U 408

As described above, in the call sequence of FIG.
Only when the RC error is detected and the RSSI integral value ΣP is less than the predetermined level R2, the control unit 114 sends the distortion amount control signal 123 to adjust the secondary distortion amount of the mixer bias. RSSI integral value ΣP is a predetermined level R
If 2 is greater than or equal to 2, it is considered that the required D / U 408 has been obtained, so it can be considered that at least a CRC error due to secondary distortion has not occurred. By this operation, it is possible to reduce erroneous adjustment of the second-order distortion caused by at least the high-level interference wave. At this time, ΔP may be the average value of the received field strength (RSSI) detected in step 506 up to that time.

When the distortion compensation method of the present invention is applied to the direct conversion receiver as shown in FIG. 1, it is more effective to perform the distortion compensation after the DC offset compensation. It is a target.

FIG. 7 shows an example of such a call sequence. In addition to the sequence of FIG. 5, after the power is turned on in step 501, the DC of the direct conversion receiver is turned on.
Step 701 of observing and storing the offset output value in advance and step 702 of acquiring the CCH while removing the DC offset thus observed are described in step 5.
03 is added before the TCH transmission / reception (call).

The DC offset observation step 701
For a specific method of the DC offset removal / CCH acquisition step 702, a method described in Japanese Patent Application No. 8-160368 “Receiver with DC offset removal function and communication system using the same” is used. It is possible.

(Regarding the Communication Sequence in FIG. 8) In the communication sequence in FIG.
During the processing of the H acquisition step 702, there is an interference wave having a high level, and 2
There is also a possibility that CCH acquisition cannot be performed due to the secondary distortion output. In such a case, a new distortion compensation sequence may be applied to the procedure of DC offset removal and CCH acquisition in step 702.

FIG. 8 shows an example of such a communication sequence. In step 702, it is determined in step 702 whether or not CCH has been captured. If CCH has not been captured, the process proceeds to a new distortion compensation sequence 802. In the distortion compensation sequence 802, t is incremented (step 803), and a comparison of t> T is performed in step 804, and the mixer bias is changed every specified time T (step 805).

Since there is always an IM2 control code in which the second-order distortion output 402 is smaller than the allowable value 404, if there is no interference wave exceeding the specification due to the change of the mixer bias, the CCH acquisition is always performed in step 801. It is determined that the transmission has been performed, and it is possible to proceed to TCH transmission / reception in step 503.

(Regarding the Communication Sequence of FIG. 9) In the communication sequences of FIGS. 5 to 8 described above, the distortion compensation sequence was executed while transmission / reception was performed in one slot. May be performed while performing the distortion compensation sequence.

FIG. 9 shows an example of such a call sequence, in which a DC offset removal / CCH acquisition step 702 is performed.
, The slot number S is set to 0 in a step 901, the value of the slot number S is incremented in a next step 902, and the process proceeds to “TCH transmission / reception (communication)” in a step 903. Error detection and reception field strength (RSSI) detection are performed. Then, the distortion compensation sequence 906 is executed until it is determined in step 905 that the slot number S has reached the predetermined value S1.

In the distortion compensation sequence 906, step 9
If a CRC error is detected at 07 and the RSSI is determined to be smaller than the predetermined level R1 at step 908, the mixer bias is changed at step 909 in the same manner as above, and the process returns to step 902.

Next, when the slot number S reaches the predetermined value S1 in step 905, the slot number S is set to 0 in step 911 until it is determined in step 910 that the cumulative number M of CRC errors has reached the specified number M1. Cleared, and the processing of steps 902 to 906 is repeated.

In step 905, the cumulative number of CRC errors M
Moves to step 912 when reaches the specified number M1, and the RSS
The average value I (mean (RSSI)) of I is compared with the specified value Ri. Step 912 is the same as step 602 in FIG. 6, and proceeds to either “slot change request” in step 511 or “frequency change request” in step 513 according to the result of this determination.

That is, at step 912, mean (R
If (SSI) is greater than Ri (NO), (E
1): It is determined that the RSSI is larger than the specified value and the interference wave is superimposed on the desired wave at the same time, and the process proceeds to step 511 to transmit a slot change request to the BS. Thereafter, the slot is changed in step 512 according to the instruction from the BS, and the process returns to step 901.

Conversely, at step 912, mean (RSS
If I) is smaller than Ri (YES), the RSSI is smaller than the specified value, and (E2): the signal level is small,
It is determined that a CRC error has occurred due to thermal noise, and the flow advances to step 513 to issue a frequency change request to the BS. In step 514, a frequency change (handoff) is performed. After establishing the link, the process returns to step 5901.

It is also possible to combine the above-described communication sequences of FIGS. 5 to 9 as appropriate. (Second Embodiment) FIG. 10 shows a configuration of a direct conversion receiver according to a second embodiment of the present invention.
In the first embodiment described above, the IQ frequency converter 1
Assuming that the second-order distortion characteristics (the characteristics of the second-order distortion output 402 in FIG. 4) of the mixers 104a and 104b in FIG.
14 from the mixers 104a, 104
The same control is applied by giving b in common. Mixers 104a and 104b for ICs of the same lot
Since the secondary distortion characteristics are similar, normally the same distortion amount control signal 123 can be used for the mixers 104a and 104b.

On the other hand, when the second-order distortion characteristics are different between the mixers 104a and 104b of the IQ frequency converter 103, as shown in FIG.
a and 104b are individually controlled by the distortion amount control signals 123a and 123a.
23b, and it is desirable to control the amount of secondary distortion independently for the I and Q channels.

In this case, in the example of the 8-valued IM2 control code shown in FIG. 4, the control code needs to be changed at the worst 8 × 8 = 64 times, and it takes time for control. However, as described above, during an actual call, the second-order distortion output 402 has an allowable value of 4
04, the second-order distortion output is 4
Since it is considered that 02 is set to a relatively small value, actually changing the control code starting from both adjacent points to the current value (initial value) reduces the number of times the control code changes. It is effective for. According to this method, for example, I and Q are each three points, that is, 3 × 3
Only by changing the IM2 control code by −1 (current minute) = 8 times, it is possible to drive the second-order distortion output 402 below the allowable value 404.

(Third Embodiment) The distortion compensation method according to the present invention can be applied to distortion compensation of any order generated by a frequency converter (mixer) used in, for example, a superheterodyne receiver other than a direct conversion receiver.
It is equally effective. FIG. 11 shows a configuration of a superheterodyne receiver to which the distortion compensation method according to the present invention is applied, as a third embodiment of the present invention.

In FIG. 11, a high-frequency signal (RF signal) modulated by an information signal (transmission data) is received by an antenna 301, and a received RF signal output from the antenna 301 is amplified by a high-frequency amplifier 302 and further After the image signal is removed by the filter 303,
The first frequency converter (first mixer) 304 mixes with the local signal (reference signal) supplied from the local oscillator 305 and frequency-converts it into a first intermediate frequency signal.

The first intermediate frequency signal from the first mixer 304 has its image signal removed by an intermediate frequency filter 306, and if necessary, is amplified by an intermediate frequency amplifier 307. In the mixer 308, the signal is mixed with a local signal (reference signal) supplied from the local oscillator 309 and frequency-converted into a second intermediate frequency signal.

The second intermediate frequency signal from the second mixer 308 is subjected to selection of a desired channel by an intermediate frequency filter 310 and further amplified by an intermediate frequency amplifier
The signal is converted into a digital signal by the D converter 312 and input to the digital signal processing unit 313. The digital signal processing unit 313 includes at least a demodulator 111, a received electric field strength (RSSI) measuring circuit 122, and an error detecting circuit 113, as in the direct conversion receiver shown in FIG.
And a control unit 114.

Here, in the superheterodyne receiver, the third-order distortion generated in the first mixer 304 usually becomes a problem. Therefore, if the mixer 304 is controlled by the distortion amount control signal 323 so that the third-order distortion output changes. Good. This control method is the same as in the case of the direct conversion receiver.

In FIG. 11, the first mixer 3 in the high frequency stage
Although the control of the distortion amount is performed only on the signal line 04, the control of the distortion amount may be performed on the second mixer 308 in the intermediate frequency stage. Further, although the control and the judgment become complicated, the first mixer 30
Control may be applied to both the fourth and second mixers 308.

As described above, according to the embodiment of the present invention, the minimum necessary characteristics for each circuit device are guaranteed by a single circuit of the frequency converter (mixer), and high-precision matching is systematic. By compensating for the frequency converter, high accuracy beyond the necessity at the frequency converter alone and device level, and good linearity can be obtained even though adjustment is unnecessary, miniaturization, cost reduction, and mass production A suitable circuit for a wireless device can be realized. Even when the distortion characteristic of the frequency converter changes due to a temperature change, a change in an actual use environment, or the like, by appropriately controlling the amount of distortion according to the distortion compensation method according to the present invention, a good distortion characteristic is always obtained. Can be realized.

[0087]

EXAMPLE Based on the above-described embodiment, for a frequency converter (mixer) for a direct conversion receiver for an actual PHS, an interference wave input level, a reception sensitivity input level, an input converted second-order distortion tolerance, an input conversion The second-order distortion level and RSSI specified value were examined. FIG.
FIG. 4 shows a result obtained by converting a measurement example of a desired wave output, an interference wave output, and a secondary distortion output of the frequency converter (mixer) to which the distortion compensation method according to the present invention shown in FIG.

Here, interference wave input level 1201: 69 dBμ reception sensitivity input level 1203: 19 dBμ desired wave to interference wave ratio D / I: 50 dB secondary distortion suppression value IM2: 65 dB input converted secondary distortion allowable value 1204: 4 dBμ Required D / U 1205: 15 dB Input-converted secondary distortion maximum value 1206: 11 dBμ Input-converted secondary distortion output minimum value 1209: -14 dBμ Input-converted secondary distortion level 1202: -14 to 11 dBμ Maximum required input signal level 1207: 26 dBμ (in the above numerical values, thermal noise degradation is omitted).

As shown in FIG. 12, the input-converted secondary distortion level 1202 has an IM2 control signal code of “4”.
In the case of “5”, “6”, “7”, the input converted allowable value 12
04, which indicates that the effect of the second-order distortion was actually removed by the distortion compensation method of the present invention. In addition,
It is clear from the above description that the distortion compensation method according to the present invention can be effectively used for distortion generated in circuits other than the frequency converter (mixer).

[0090]

As described above, in the present invention, the minimum necessary distortion characteristic for the circuit device of each frequency converter is guaranteed by the frequency converter circuit alone, and the distortion characteristics are accurately adjusted. Systematic compensation. Therefore, despite the frequency converter alone, higher accuracy than necessary at the device level and adjustment is unnecessary, good linearity can be obtained,
A wireless device suitable for miniaturization, low cost, and mass production can be realized.

Even when the distortion characteristics of the frequency converter change due to a change in temperature, a change in the actual use environment, or the like, good distortion characteristics are always obtained by appropriately performing distortion control by the distortion compensation method according to the present invention. There is an effect that can be realized.

Further, the distortion compensation method of the present invention is versatile, and compensates for distortion generated in circuits other than the frequency converter, and furthermore, for distortion that becomes a problem in various receiving systems such as a direct conversion system and a superheterodyne system. Applicable for compensation.

[Brief description of the drawings]

FIG. 1 is a block diagram showing a configuration of an embodiment of a direct conversion receiver using a distortion compensation method according to the present invention.

FIG. 2 is a block diagram showing a configuration of a reception electric field strength measurement circuit in FIG. 1;

FIG. 3 is a circuit diagram showing an example of a frequency converter having a secondary distortion amount control function.

FIG. 4 is a diagram showing a measurement example of a desired wave output, an interference wave output, and a second-order distortion output of the frequency converter to which the distortion compensation method according to the present invention is applied.

FIG. 5 is a flowchart showing a first example of a call sequence of the direct conversion receiver including the distortion compensation sequence according to the embodiment;

FIG. 6 is a flowchart showing a second example of the call sequence of the direct conversion receiver including the distortion compensation sequence according to the embodiment;

FIG. 7 is a flowchart showing a third example of the call sequence of the direct conversion receiver including the distortion compensation sequence according to the embodiment;

FIG. 8 is a flowchart showing a fourth example of the call sequence of the direct conversion receiver including the distortion compensation sequence according to the embodiment;

FIG. 9 is a flowchart showing a fifth example of the call sequence of the direct conversion receiver including the distortion compensation sequence according to the embodiment;

FIG. 10 is a block diagram showing the configuration of another embodiment of a direct conversion receiver using the distortion compensation method according to the present invention.

FIG. 11 is a block diagram showing a configuration of one embodiment of a superheterodyne receiver using the distortion compensation method according to the present invention.

FIG. 12 is a diagram showing a measurement example of a desired wave output, an interference wave output, and a secondary distortion output of a frequency converter using the distortion compensation method according to the present invention, in terms of RF input.

FIG. 13 is a block diagram showing a configuration of a conventional direct conversion receiver.

FIG. 14 is a diagram illustrating a level relationship between a desired wave and an interference wave and a relationship between a desired wave after frequency conversion and a second-order distortion for explaining a problem of the conventional direct conversion receiver.

FIG. 15 is a block diagram showing RF signal input / output and local signal input of a mixer;

[Explanation of symbols]

 DESCRIPTION OF SYMBOLS 101 ... Antenna 102 ... High frequency amplifier 103 ... IQ frequency converter 104a, 104b ... Mixer 105 ... Local oscillator 106 ... 90 degree phase shifter 107a, 107b ... Low pass filter 108a, 108b ... Base band amplifier 109a, 109b ... A / D conversion Unit 110 Digital signal processing unit 111 Demodulator 112 Received electric field strength measurement circuit 113 Error detection circuit 114 Control unit 123, 123a, 123b Distortion amount control signal 201a, 201b Multiplier 202 Adder 203 Square root circuit 204 Comparator 205 Memory 301 Antenna 302 High frequency amplifier 303 High frequency filter 304, 308 Mixer 305, 309 Local oscillator 306, 310 Intermediate frequency filter 307, 311 Intermediate frequency amplifier 312 / D converter 313 digital signal processing unit 323 distortion control signal D / I desired wave-to-interference wave ratio fD desired wave frequency fI interference wave frequency 401 interference wave output 402 secondary distortion output 403 ... desired wave output 404 ... allowable value of secondary distortion output 405 ... required D / U 406 ... secondary distortion output maximum value 407 ... desired wave output 408 ... required D / U 409 ... secondary distortion output minimum value 410 ... RSSI regulation Value IM2: Secondary distortion suppression value 1201: Interference wave input level 1202: Input converted secondary distortion level 1203: Reception sensitivity input level 1204: Input converted secondary distortion allowable value 1205: Required D / U 1206: Input converted Maximum secondary distortion 1207: Maximum required input signal level 1208: Required D / U 1210: RSSI specified value

 ────────────────────────────────────────────────── ─── Continuing from the front page (72) Inventor Miyuki Ogura 3-1-1, Asahigaoka, Hino-shi, Tokyo F-term (reference) in Toshiba Hino Plant 5K004 AA05 FA05 FA06 FD05 FG02 FH01 FH06 5K014 AA01 BA06 EA08 HA06 5K020 AA08 DD01 DD05 DD25 EE05 EE16 FF02 LL00

Claims (5)

[Claims] 1. A radio receiving apparatus including a frequency converter for converting a received high-frequency signal into a low frequency, wherein when a reception error of the high-frequency signal is continuously detected a predetermined number of times, the frequency converter is distorted. The distortion compensation method for a frequency converter, wherein the distortion is controlled so as to change. 2. A receiving means for receiving a high-frequency signal modulated by an information signal; a reference signal source for generating a reference signal; and converting the high-frequency signal received by the receiving means to a lower frequency by the reference signal. A frequency converter configured to output and change the amount of distortion by external control; a demodulator for receiving the output signal output from the frequency converter and demodulating the information signal; and the demodulator. Error detection means for detecting a reception error of the high-frequency signal from the output of the control signal, and control for controlling the frequency converter to change the amount of distortion when a predetermined number of consecutive reception errors are detected by the error detection means. Radio receiving apparatus comprising: 3. A receiving means for receiving a high-frequency signal modulated by an information signal; a reference signal source for generating orthogonal first and second reference signals; and a high-frequency signal received by the receiving means. First and second frequency converters configured to convert the first and second reference signals to lower frequencies and output the same, and to change the amount of distortion by external control; and the first and second frequency converters. Demodulation means for receiving the output signal output from the frequency converter and demodulating the information signal, error detection means for detecting a reception error of the high-frequency signal from the output of the demodulation means, and a predetermined number of times by the error detection means Control means for controlling the first and second frequency converters to change the amount of distortion when a reception error is continuously detected. Location. 4. A receiving electric field intensity measuring means for measuring a receiving electric field intensity of a high-frequency signal received from said receiving means, wherein said controlling means comprises a distortion generated by said frequency converter within a control range of said controlling means. When the reception electric field strength measured by the reception electric field strength measurement means is smaller than the input conversion value of the reception means of a value obtained by adding a predetermined desired wave-to-distortion ratio to the maximum value of the frequency converter, The radio receiving apparatus according to claim 2, wherein the control is performed such that the distortion amount changes. 5. The control means is configured to output a control signal composed of a digital signal for the control,
The radio receiving apparatus according to any one of claims 2 to 4, wherein the control signal is changed toward a point sequentially separated from points on both sides of the initial value.