JP2000172353A - Constant voltage generation circuit - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a constant voltage generation circuit capable of obtaining a sufficiently small reference voltage. SOLUTION: This constant voltage generation circuit has 1st and 2nd constant current sources, a 1st transistor MN1 whose drain is connected to the 1st constant current source and whose source is connected to the ground, a 2nd transistor MN2 whose drain is connected to the 2nd constant current source and whose gate is connected to the gate of the 1st transistor MN1 and a 1st resistor R1 connected between the source of the 2nd transistor MN2 and the ground. And, when the gate width of the 1st transistor MN1 is defined as W1, the gate length is defined as L1, the gate width of the 2nd transistor MN2 is defined as W2 and the gate length is defined as L2, the value represented by a formula (W1/L1) is different from the value represented by a formula (W2/L2). And, a reference voltage that is in proportion to the potential difference between both ends of the 1st resistor R1 is generated.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

The present invention relates to a constant voltage generator suitable for a semiconductor device having a plurality of MOS transistors having different threshold voltages, and more particularly to a constant voltage generator capable of generating a low reference voltage.

[0002]

2. Description of the Related Art Conventionally, various types of constant voltage generating circuits for generating a constant voltage with a small dependence on temperature and power supply voltage in a semiconductor device have been proposed. In particular, in recent dynamic random access memory (DRAM) products, a circuit for internally generating a constant reference voltage V _REF is essential in order to control various internal power supplies. Further, as an N-channel MOS transistor incorporated in a DRAM product, a transistor having at least three types of threshold voltages (Vt) is generally used. That is, a normal threshold transistor for a general circuit, a high threshold transistor for a memory cell, and a low threshold transistor for a sense amplifier and the like. As a conventional reference voltage V _REF generation circuit, there is a circuit called a band gap type. FIG. 8 is a circuit diagram showing a conventional reference voltage V _REF generation circuit.

The conventional reference voltage V _REF generating circuit is provided with P-channel MOS transistors MP11, MP12 and MP13 which are supplied with a power supply voltage V _cc and serve as constant current sources. Each P-channel MOS transistor MP
11, the gates of MP12 and MP13 are commonly connected,
The gate width and gate length are set equal. Therefore, each P-channel MOS transistor MP11, MP1
Current flows from the second and MP13 I _MP11, I _MP12 and I
_MP13 is equal to each other.

An N-channel MOS transistor having a drain connected to the drain of a P-channel MOS transistor MP11 is provided.
S transistor MN11 and drain are P-channel MO
An N-channel MOS transistor MN12 connected to the drain of the S transistor MP12 is provided. N
The threshold value of the channel MOS transistor MN11 is a normal threshold value, and its source is grounded. On the other hand, the threshold value of the N-channel MOS transistor MN12 is also a normal threshold value, and its source is connected to a resistor R1 whose other end is grounded.
1 is connected. The gates of the N-channel MOS transistors MN11 and MN12 are commonly connected.

Note that a P-channel MOS transistor MP
11 and MP12: ratio of gate width to gate width W _P / L _P
It is set sufficiently smaller than the ratio W _N12 / L _{N 12} of the gate width to gate length ratio W _N11 / L _N11 and MN12 of the gate width and gate length of the N-channel MOS transistor MN11. Therefore, current I _MP11 (= I _MP12 ) is sufficiently small, and N-channel MOS transistor MN
11 and MN12 operate in the weak inversion region. Also, the ratio W
The value of _N12 / _LN12 is about ten times the value of the ratio _WN11 / _LN11 .

Further, a P-channel MOS transistor MP
The drain of the resistor 13 is connected to a resistor R12, and the resistor 12 is connected to a diode D11 whose other end is grounded. And a P-channel MOS transistor M
The potential at the drain of P13 is output as the reference voltage _VREF .

The conventional reference voltage V configured as described above
_{In the REF} generation circuit, the following equations 1 to 4 hold.

[0008]

(Equation 1)

[0009]

(Equation 2)

[0010]

(Equation 3)

[0011]

(Equation 4)

Here, R _R11 and R _R12 are resistance values of the resistors R11 and R12, n is a process-dependent constant, k is a Boltzmann constant, q is an electron charge, and m is a value defined by the following equation (5). , Vf are forward voltages of the diode D11,
V _gs (MN11) and V _gs (MN12) are the gates of N-channel MOS transistors MN11 and MN12, respectively.
This is a source-to-source voltage.

[0013]

(Equation 5)

FIG. 9 shows the gate-source voltage V _gs on the horizontal axis and the drain-source current I _ds on the vertical axis, and shows the N-channel MOS transistor MN1 at various temperatures.
Is a graph showing the I _ds -V _gs characteristics of 1 and MN12. In FIG. 9, the solid line indicates the I _ds -V _gs characteristics of both transistors at 25 ° C., a two-dot chain line shows the I _ds -V _gs characteristics of both transistors at 85 ° C..

Is the MOS transistor characteristic in the weak inversion region?
From the N-channel MOS transistors MN11 and MN1
2 when the same current IMP11 = IMP12 is flowing
In this case, N-channel MOS transistors MN11 and MN1
2 gate-source voltage V _gs, That is, the resistance R11
Voltage applied to_R11Increases with increasing temperature. Toes
Voltage V_R11Has a positive temperature coefficient. Therefore, Equation 3
Is the voltage V_R11Is proportional to
The number is positive. Further, n, k and q in Equation 4 are all
High temperature (85 ° C) and normal temperature (25 ° C)
Voltage V between_R11Is larger as m is larger.
You.

On the other hand, as is well known, the temperature coefficient of the forward voltage Vf of the diode D11 is negative. That is, as the temperature rises, the voltage Vf shown in the second term of Equation 3 decreases. Therefore, the temperature coefficients of the first term and the second term in Expression 3 are opposite to each other. Since the first term is variable depending on the value of m, in the conventional reference voltage V _REF generation circuit, m
Is adjusted so that the temperature coefficients of the first and second terms cancel each other.

FIG. 10 shows the power supply voltage V _CC on the horizontal axis and the reference voltage V _REF , the power supply voltage V _CC and the resistance R1 on the vertical axis.
FIG. 4 is a graph showing these relationships by taking a potential difference V _R11 across the two terminals; Since the value of m is set appropriately, the temperature coefficient can be made substantially zero. As a result, FIG.
As shown in FIG. 0, the temperature dependency of the reference voltage V _REF is almost eliminated. The current flowing through the P-channel MOS transistors MP11, MP12 and MP13, since a substantially constant regardless of the supply voltage V _CC, as shown in FIG. 10, the power supply voltage V _CC dependency of the reference voltage V _REF is also small.

However, in the above-described reference voltage generation circuit, since the diode D11 is included, the number of elements is increased, and the reference voltage V _REF is lower than the forward voltage Vf (about 1.2 V) of the diode D11. Cannot be a value. Therefore, it is difficult to lower the reference voltage V _REF in a future reduction in power supply voltage.

A reference voltage generating circuit having a transistor operating in a saturation region is disclosed in Japanese Patent Laid-Open No. 9-1466.
No. 48 discloses this. In the conventional reference voltage generation circuit described in this publication, a current mirror circuit including a bipolar transistor operating in a saturation region is provided.

[0020]

However, there is a problem that a sufficiently small reference voltage cannot be obtained even by the conventional reference voltage generating circuit disclosed in the above-mentioned Japanese Patent Application Laid-Open No. 9-146648.

The present invention has been made in view of the above problems, and has as its object to provide a constant voltage generating circuit that can obtain a sufficiently small reference voltage.

[0022]

A constant voltage generating circuit according to the present invention comprises a first and a second constant current source, and a first constant current source having a drain connected to the first constant current source and a source connected to the ground. 1
And a second transistor having a drain connected to the second constant current source and a gate connected to the gate of the first transistor, and a transistor connected between the source of the second transistor and ground. A first resistor having a gate width of W1;
When the gate length is L1, the gate width W2 and the gate length of the second transistor are L2, the equation (W1 / L
When the value represented by 1) is different from the value represented by the formula (W2 / L2), or when the current flowing through the first transistor is I1 and the current flowing through the second transistor is I2, The current I1 is different from the current I2, and a constant voltage proportional to a potential difference between both ends of the first resistor is generated.

In the present invention, since the first and second transistors have different gate widths and gate lengths or currents, it is possible to obtain a reference voltage having extremely low temperature dependence and power supply voltage dependence. . At this time, since the value of the reference voltage is proportional to the potential difference between both ends of the first resistor, a desired sufficiently small reference voltage can be obtained by adjusting the resistance value of the first resistor. . Further, since no diode is required, the circuit configuration is simplified.

The absolute value of the threshold voltage of the first transistor may be higher than that of the second transistor.

The temperature dependence of the threshold voltage of the first transistor is higher than that of the second transistor, and is expressed by the formula (((W1 / L1) / (W2 / L2)) ×
The value represented by (I1 / I2) may be larger than 1, and the temperature dependence of the threshold voltage of the first transistor is lower than that of the second transistor, and the expression (((W1 / L1) / (W2 / L2)) × (I1 / I
The value represented by 2)) may be smaller than 1.

Further, the current flowing from the first constant current source and the current flowing from the second constant current source may be mutually proportional.

A third constant current source for supplying a current proportional to the current flowing from the first and second constant current sources;
A second resistor connected to the drain of the third constant current source.

Further, the first resistor and the second resistor may be formed on the same conductive layer.

Further, each of the first to third constant current sources may include a constant current source transistor whose gate is commonly connected.

Still further, at least one constant current source selected from the first to third constant current sources is a first constant current source that holds a drain-source voltage of the constant current source transistor at a predetermined voltage or less. A holding transistor may be provided, and a second holding transistor for holding a drain-source voltage of the first and second transistors at a predetermined voltage or less may be provided.

[0031]

DESCRIPTION OF THE PREFERRED EMBODIMENTS Hereinafter, a constant voltage generating circuit according to an embodiment of the present invention will be specifically described with reference to the accompanying drawings. FIG. 1 is a circuit diagram showing a constant voltage generating circuit according to a first embodiment of the present invention.

The first embodiment is provided with P-channel MOS transistors MP1, MP2 and MP3 which are supplied with a power supply voltage V _cc and serve as constant current sources. The gates of the P-channel MOS transistors MP1, MP2 and MP3 are commonly connected, and the gate width and the gate length are set equal. Therefore, the currents I _MP1 , I _MP2 and I _MP3 flowing from the respective P-channel MOS transistors MP1, MP2 and MP3 are equal to each other.

An N-channel MOS transistor having a drain connected to the drain of P-channel MOS transistor MP1
An N-channel transistor MN1 having a transistor MN1 and a drain connected to a drain of a P-channel MOS transistor MP2
An OS transistor MN2 is provided. The threshold value of the N-channel MOS transistor MN1 is a normal threshold value, and its source is grounded. On the other hand, N channel M
The threshold value of the OS transistor MN2 is a low threshold value, and the source thereof is connected to the resistor R1 whose other end is grounded. Also, N-channel MOS transistors MN1 and MN1
The gates of N2 are commonly connected.

The P-channel MOS transistor MP
The ratio W _P / L _P between the gate width and the gate length of 1 and MP2 is
The ratio is set to be sufficiently smaller than the ratio W _N1 / L _N1 of the gate width and the gate length of the N-channel MOS transistor MN1 and the ratio W _N2 / L _N2 of the gate width and the gate length of the MN2.
Therefore, current I _MP1 (= I _MP2 ) is sufficiently small, and N-channel MOS transistors MN1 and MN2 operate in the weak inversion region. The value of the ratio W _N2 / L _N2 is
It is about 10 times the value of _N1 / _LN1 .

Further, a P-channel MOS transistor MP
The resistor R2 whose other end is grounded is connected to the drain of
The potential difference between both ends of the resistor R2 is equal to the reference voltage (reference voltage).
Pressure) V _REFIs output as

In the constant voltage generating circuit according to the first embodiment thus configured, the following equations 6 to 9 hold.

[0037]

(Equation 6)

[0038]

(Equation 7)

[0039]

(Equation 8)

[0040]

(Equation 9)

Where R _R1 and R _R2 are resistors R1 and R
2, m is a value defined by the following equation 10, V
_gs (MN1) and V _gs (MN2) are N-channel MOs, respectively.
This is a gate-source voltage of the S transistors MN1 and MN2.

[0042]

(Equation 10)

Therefore, since the reference voltage V _REF is (R _R2 / R _R1 ) times the potential difference V _R1 , its magnitude can be set to an arbitrary value. Further, since the value of m is appropriately set, the temperature coefficient of the reference voltage V _REF can be made substantially zero.

FIG. 2 is a graph showing the relationship between the power supply voltage V _CC on the horizontal axis and the reference voltage V _REF , the power supply voltage V _CC and the potential difference V _R1 across the resistor R1 on the vertical axis. As shown in FIG. 2, in the present embodiment, the reference voltage V _REF
Of the power supply voltage V _CC is also small.

In the first embodiment, m is set to about 10. Next, an operation when m is inappropriately set to 1 will be described. FIG. 3 shows the gate-source voltage V _gs on the horizontal axis and the drain-source current I _ds on the vertical axis, and the I-channel MOS transistors MN1 and MN2 at various temperatures when m is 1. It is a graph which shows _ds - _Vgs characteristic. In FIG. 3, the solid line indicates the I _ds -V _gs characteristics of both transistors at 25 ° C., a two-dot chain line shows the I _ds -V _gs characteristics of both transistors at 85 ° C.. When m is 1, the voltage V _R1 is also represented by the following equation (11).

[0046]

[Equation 11]

Here, n is a process-dependent constant, k is Boltzmann's constant, and q is the amount of electron charge.

When m is 1, as shown in FIG.
The potential difference V _{R1 at} 5 ° C. is smaller than that at 25 ° C. That is, the temperature coefficient of the potential difference _VR1 is negative. on the other hand,
The first term in Equation 11 is zero regardless of the temperature.
Therefore, the temperature coefficient of the second term is negative.

Here, as the value of m is increased, the equation
The effect of the eleventh term becomes large. And the first
As shown in the embodiment, when m becomes about 10, the reference
Pressure V _REFIs substantially no longer temperature dependent. However,
When the value of m is increased, the shadow of the first term
Sound is too large and the reference voltage V_REFTemperature dependence of
It gets bigger.

FIG. 4 shows the gate-source voltage V _gs on the horizontal axis and the drain-source current I _ds on the vertical axis, and shows the N-channel MOS transistor MN1 at various temperatures.
FIG. 10 is a graph showing I _ds −V _gs characteristics of MN2 and MN2. In FIG. 4, a solid line indicates an I _ds -V _gs characteristics of both transistors at 25 ° C., a two-dot chain line shows the I _ds -V _gs characteristics of both transistors at 85 ° C..

Generally, the value of the drain-source current I _ds (I _ds −V _gs characteristic) with respect to the gate-source voltage V _gs in the weak inversion region is different between a normal threshold transistor and a low threshold transistor. As shown in FIG. 4, in this embodiment, the temperature dependence of the low threshold voltage transistor is smaller than that of the normal threshold voltage transistor. Therefore, the temperature coefficient of the second term of Expression 11 is negative. On the other hand, n, k, and q in the first term are all positive constants, and T is the absolute temperature, so the temperature coefficient of the first term is positive. Then, in the present embodiment, Expression 10
By setting the value of m defined by the above to about 10, the temperature coefficients of the first and second terms are set to cancel each other.

As described above, according to the present embodiment, the potential difference V
_Since the temperature coefficient of _R1 can be made substantially zero,
The temperature coefficient of the reference voltage V _REF , which is (R _R2 / R _R1 ) times, can be made substantially zero. Therefore, by appropriately setting the ratio of the resistance values of the resistors R1 and R2, the magnitude of the reference voltage V _REF can be set to an arbitrary value.

Further, since the diode element conventionally required is not required, the circuit is simplified. Furthermore, since the low threshold N-channel MOS transistor is also used for a sense amplifier in a DRAM, the present embodiment can be easily manufactured.

Next, a second embodiment of the present invention will be described. FIG. 5 is a circuit diagram showing a constant voltage generating circuit according to a second embodiment of the present invention. In the second embodiment shown in FIG. 5, the same components as those in the first embodiment shown in FIG. 1 are denoted by the same reference numerals, and detailed description thereof will be omitted.

In the second embodiment, the P-channel MO
A P-channel MOS transistor MP6 and an N-channel MOS transistor MN3 are connected between the S transistor MP1 and the N-channel MOS transistor MN1. An N-channel MOS transistor MP2 is connected between the P-channel MOS transistor MP2 and the N-channel MOS transistor MN2.
The OS transistor MN4 is connected. The gates of the N-channel MOS transistors MN3 and MN4 are commonly connected.

The P-channel MOS transistor MP
3 and a resistor R2, a P-channel MOS transistor M
P7 is connected. The gates of the P-channel MOS transistors MP6 and MP7 are commonly connected, and an N-channel MOS transistor MN5 and a P-channel MOS transistor MP5 each having a drain connected thereto are provided. The source of the P-channel MOS transistor MP5 is connected to a P-channel MOS transistor MP4 whose source is supplied with the power supply voltage V _CC .

Note that N-channel MOS transistor MN
5, the ratio W between the gate width and the gate length_N5/ L_N5Is P channel
Ratio and ratio in MOS transistors MP4 and MP5
It is set small enough. Therefore, P channel
Gate potential of MOS transistors MP6 and MP7
Is (V_CC-2V_tp) Degree. Note that the voltage V
_tpAre P-channel MOS transistors MP6 and MP7
Is the absolute value of the threshold voltage.

In the constant voltage generating circuit according to the second embodiment thus configured, the following equations 12 to 17 hold.

[0059]

(Equation 12)

[0060]

(Equation 13)

[0061]

[Equation 14]

[0062]

(Equation 15)

[0063]

(Equation 16)

[0064]

[Equation 17]

Here, W _N3 , W _N4 , W _N3 and W _N6 are N-channel MOS transistors MN3, MN4 and MN5, respectively.
And a gate width of the _{MN6, L N3, L N4,} L N3 and L
_N6 is an N-channel MOS transistor MN3, MN
4, the gate length of MN5 and MN6.

When m is 1, the following equation 18 is established.

[0067]

(Equation 18)

In the second embodiment, the P-channel MO
When the drain potentials of the S transistors MP1 and MP3 are P
Similar to the drain potential of the channel MOS transistor MP2, is held to a degree (V _CC -V _tp). That is, the drains of the P-channel MOS transistors MP1 and MP3
The source-to-source voltage V _ds becomes substantially constant. Therefore, the constant current characteristics of the drain currents of the P-channel MOS transistors MP1 and MP3 are improved as compared with the first embodiment.

The drain potentials of N-channel MOS transistors MN1 and MN2 are substantially equalized by N-channel MOS transistors MN3 and MN4 whose gates are commonly connected. That is, the voltage V _ds between the drain and the source of the N-channel MOS transistor MN2 is substantially constant, similarly to the N-channel MOS transistor MN1. As a result, N-channel MOS transistor M
The constant current characteristic of the drain current of N2 is improved as compared with the first embodiment.

As described above, according to the second embodiment, the power supply voltage dependency is further reduced as compared with the first embodiment by improving the constant current source circuit.

Next, a third embodiment of the present invention will be described. In the present embodiment, a normal threshold N-channel MOS transistor is provided instead of the low threshold N-channel MOS transistor in the first embodiment, and a normal threshold N-channel MOS transistor whose gate is commonly connected is provided. A high threshold N-channel MOS transistor is provided instead of the N-channel MOS transistor. FIG. 6 is a circuit diagram showing a constant voltage generating circuit according to a third embodiment of the present invention. In the third embodiment shown in FIG. 6, the same components as those in the first embodiment shown in FIG. 1 are denoted by the same reference numerals, and detailed description thereof will be omitted.

In the third embodiment, a high threshold N-channel MOS transistor MN1a is provided in place of the N-channel MOS transistor MN1 in the first embodiment, and a normal threshold is provided in place of the low threshold N-channel MOS transistor MN2. N-channel MOS transistor MN2a is provided. The ratio W _N1a between the gate width and the gate length of the N-channel MOS transistor MN1a
/ L _N1a is an N-channel MOS transistor MN2a
The ratio of the gate width to the gate length W _N2a / L _N2a is m times (m> 1).

In the constant voltage generating circuit according to the third embodiment having the above-described configuration, the following equations (19) to (20) are satisfied.

[0074]

[Equation 19]

[0075]

(Equation 20)

[0076]

(Equation 21)

[0077]

(Equation 22)

Therefore, as in the first embodiment, the reference voltage V _REF is (R _R2 / R _R1 ) times the potential difference V _R1 , and the magnitude can be set to an arbitrary value. Also, m
Is appropriately set, the temperature coefficient of the reference voltage V _REF can be made substantially zero.

In the third embodiment, m is larger than 1.
Next, what happens when m is set to improper 1?
The work will be described. FIG. 7 shows the gate-source voltage on the horizontal axis.
Pressure V _gsAnd the vertical axis represents the drain-source current I_dsAnd
Thus, N channels at various temperatures when m is 1
I of the MOS transistors MN1a and MN2a_ds-V
_gsIt is a graph which shows a characteristic. Note that in FIG.
The line shows the I of both transistors at 25 ° C._ds-V_gsCharacteristics
And the two-dot chain line shows the I_ds
-V_gsThe characteristics are shown. Also, when m is 1,
Is the voltage V_R1Is also represented by Equation 23 below.

[0080]

(Equation 23)

However, V _gs (MN1a) and V _gs (MN2
a) is a gate-source voltage in the transistors MN1a and MN2a, respectively.

When m is 1, as shown in FIG.
The potential difference V _{R1 at} 5 ° C. is smaller than that at 25 ° C. That is, the temperature coefficient of the potential difference _VR1 is negative. on the other hand,
The first term in Expression 23 is zero regardless of the temperature.
Therefore, the temperature coefficient of the second term is negative.

Generally, the I _ds -V _gs characteristics in the weak inversion region are different between the normal threshold voltage transistor and the high threshold voltage transistor. In this embodiment, the high threshold voltage transistor has the normal threshold voltage. Temperature dependence is greater than that of the transistor of FIG. Therefore, the second of equation 23
The temperature coefficient of the term is negative. On the other hand, n, k in the first term
And q are all positive constants, and T is the absolute temperature, so the temperature coefficient of the first term is positive. Then, in this embodiment, by setting the value of m defined by Expression 23 to an appropriate value (about 10) larger than 1, the temperature coefficients of the first and second terms are canceled. It is set.

As described above, also in this embodiment, the temperature coefficient of the potential difference V _R1 can be made substantially zero, so that the temperature of the reference voltage V _{REF which} is a value that is (R _R2 / R _R1 ) times as large. The coefficient can be made substantially zero. Therefore, by appropriately setting the ratio of the resistance values of the resistors R1 and R2,
The magnitude of the reference voltage V _REF can be set to an arbitrary value. At this time, as in the first embodiment, the temperature dependency and the power supply voltage dependency of the reference voltage V _REF are extremely small.

Further, since the diode element conventionally required is not required, the circuit is simplified. Further, as described above, the high threshold N-channel MOS transistor
In the DRAM, since this embodiment is also used for a sense amplifier and the like, this embodiment can be easily manufactured.

In the first to third embodiments,
P-channel MOS transistors MP1, MP2 and MP
3 are set to the same value, and the N-channel MOS
While properly setting the ratio m (m> 1) of the gate width of N-channel MOS transistor MN2 for the transistor MN1, the present invention is not limited thereto, an appropriate current I _MP1 current I _MP2 The same effect can be obtained even if it is set to be M times (M> 1). Further, the same effect can be obtained by a configuration in which both the gate size ratio m and the current ratio M are adjusted.

In each of the above embodiments, the gate length is made equal between the N-channel MOS transistors MN1 and MN2 so that only the gate width is different. However, the gate width is made equal and only the gate length is changed. The same effect can be obtained even if a difference is made or both the gate width and the gate length are made different.

Further, the resistors R1 and R
2 can be formed from a wiring layer or a diffusion layer,
In order to reduce the influence of manufacturing variations, it is desirable to form the wiring layers or diffusion layers of the same type.

Further, depending on the process conditions, it is conceivable that a transistor having a low threshold value has a greater temperature dependency in a weak inversion region than a transistor having a normal threshold value.
In such a case, the same effect can be obtained by making the value of m or M smaller than 1. The same applies to a combination of a high threshold transistor and a normal threshold transistor.

[0090]

As described in detail above, according to the present invention,
Since a reference voltage proportional to the potential difference between both ends of the first resistor is generated, a desired sufficiently low reference voltage can be obtained by adjusting the resistance value of the first resistor. Therefore, it can be easily adapted to future low power supply voltage products. Further, since no diode is required, the entire circuit can be simplified. In addition, conventional DRAM
Can be used, so that the manufacturing process can be prevented from being complicated.

[Brief description of the drawings]

FIG. 1 is a circuit diagram showing a constant voltage generation circuit according to a first example of the present invention.

FIG. 2 is a graph showing a relationship among a power supply voltage V _CC , a reference voltage V _REF , a power supply voltage V _CC, and a potential difference V _R1 across a resistor R1 in the first embodiment.

FIG. 3 shows I _ds −V _gs of N-channel MOS transistors MN1 and MN2 at various temperatures when m is 1.
It is a graph which shows a characteristic.

4 is a graph showing the I _ds -V _gs characteristics of N-channel MOS transistors MN1 and MN2 at various temperatures of the first embodiment.

FIG. 5 is a circuit diagram showing a constant voltage generation circuit according to a second example of the present invention.

FIG. 6 is a circuit diagram showing a constant voltage generation circuit according to a third embodiment of the present invention.

FIG. 7 shows I _ds − of N-channel MOS transistors MN1a and MN2a at various temperatures when m is 1.
It is a graph which shows _Vgs characteristic.

FIG. 8 is a circuit diagram showing a conventional reference voltage V _REF generation circuit.

FIG. 9 shows N-channel MOS transistors at various temperatures.
I of Star MN11 and MN12 _ds-V_gsGraph showing characteristics
FIG.

FIG. 10 shows a power supply voltage V in a conventional reference voltage generation circuit.
_CC and the reference voltage V _REF, is a graph showing the relationship between the potential difference across V _R11 of the power supply voltage V _{C C} and a resistor R11.

[Explanation of symbols]

MP1, MP2, MP3, MP4, MP5, MP6, M
P7, MP11, MP12, MP13; P channel MO
S transistor MN1, MN1a, MN2, MN2a, MN3, MN
4, MN5, MN11, MN12; N-channel MOS transistors R1, R2, R11, R12; resistors

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[Procedure amendment]

[Submission date] October 26, 1999 (1999.10.
26)

[Procedure amendment 1]

[Document name to be amended] Statement

[Correction target item name] Claims

[Correction method] Change

[Correction contents]

[Claims]

[Procedure amendment 2]

[Document name to be amended] Statement

[Correction target item name] 0022

[Correction method] Change

[Correction contents]

[0022]

A constant voltage generating circuit according to the present invention comprises a first and a second constant current source, and a first constant current source having a drain connected to the first constant current source and a source connected to the ground. 1
And a second transistor having a drain connected to the second constant current source and a gate connected to the gate of the first transistor, and a transistor connected between the source of the second transistor and ground. And a first resistor, wherein the absolute value of the threshold voltage of the first transistor is
Higher than the absolute value of the threshold voltage of the second transistor;
When the gate width of the first transistor is W1, the gate length is L1, and the gate width of the second transistor is W2 and the gate length is L2, a formula (W1 / L1) is obtained.
Is different from the value represented by the formula (W2 / L2), or the current flowing through the first transistor is represented by I
1. When a current flowing through the second transistor is I2, the current I1 is different from the current I2, and a constant voltage proportional to a potential difference between both ends of the first resistor is generated. .

[Procedure amendment 3]

[Document name to be amended] Statement

[Correction target item name] 0023

[Correction method] Change

[Correction contents]

In the present invention, the gate width and the gate length or the current of the first and second transistors are different, and the absolute value of the threshold voltage of the first transistor is the second transistor.
Since the absolute value of the threshold voltage of the transistor is higher than that of the transistor, a reference voltage having very little temperature dependency and power supply voltage dependency can be obtained. At this time, since the value of the reference voltage is proportional to the potential difference between both ends of the first resistor, a desired sufficiently small reference voltage can be obtained by adjusting the resistance value of the first resistor. . Further, since no diode is required, the circuit configuration is simplified.

[Procedure amendment 4]

[Document name to be amended] Statement

[Correction target item name] 0024

[Correction method] Deleted

[Procedure amendment 5]

[Document name to be amended] Statement

[Correction target item name] 0090

[Correction method] Change

[Correction contents]

[0090]

As described in detail above, according to the present invention,
Since a reference voltage proportional to the potential difference between both ends of the first resistor is generated, a desired sufficiently low reference voltage can be obtained by adjusting the resistance value of the first resistor. Therefore, it can be easily adapted to future low power supply voltage products. The diode is therefore not necessary, it is possible to simplify the overall circuitry. Further, since a transistor used in a conventional DRAM can be used, it is possible to prevent the manufacturing process from being complicated.

Claims (11)

[Claims] A first transistor having a drain connected to the first constant current source and a source connected to ground; and a drain connected to the second constant current source. And a second transistor having a gate connected to the gate of the first transistor, and a first resistor connected between the source of the second transistor and ground. Assuming that the gate width of the first transistor is W1, the gate length is L1, and the gate width of the second transistor is W2 and the gate length is L2, the value represented by the equation (W1 / L1) and the equation (W2 / L2) And generating a constant voltage proportional to a potential difference between both ends of the first resistor. 2. A first and second constant current source, a first transistor having a drain connected to the first constant current source and a source connected to ground, and a drain connected to the second constant current source. And a second transistor having a gate connected to the gate of the first transistor, and a first resistor connected between the source of the second transistor and ground. The current flowing through one transistor is I
1. When a current flowing through the second transistor is I2, the current I1 is different from the current I2, and a constant voltage proportional to a potential difference between both ends of the first resistor is generated. Constant voltage generation circuit. 3. The constant voltage generation circuit according to claim 1, wherein the absolute value of the threshold voltage of the first transistor is higher than that of the second transistor. 4. The temperature dependence of the threshold voltage of the first transistor is higher than that of the second transistor, and is expressed by the formula (((W1 / L1) / (W2 / L2)) × (I
4. The constant voltage generation circuit according to claim 1, wherein a value represented by (1 / I2)) is larger than 1. 5. The temperature dependence of the threshold voltage of the first transistor is lower than that of the second transistor, and is expressed by the formula (((W1 / L1) / (W2 / L2)) × (I
The constant voltage generation circuit according to any one of claims 1 to 3, wherein a value represented by (1 / I2)) is smaller than 1. 6. The method according to claim 1, wherein a current flowing from the first constant current source and a current flowing from the second constant current source are proportional to each other. The constant voltage generation circuit as described. 7. A third constant current source for supplying a current proportional to a current flowing from the first and second constant current sources, and
And a second resistor connected to a drain of the constant current source. 8. The first resistor and the second resistor,
The constant voltage generation circuit according to claim 7, wherein the constant voltage generation circuit is formed on the same conductive layer. 9. The constant voltage generating circuit according to claim 7, wherein each of the first to third constant current sources has a transistor for a constant current source whose gate is commonly connected. 10. At least one constant current source selected from the first to third constant current sources is a first holding unit that holds a drain-source voltage of the constant current source transistor at a predetermined voltage or less. The constant voltage generating circuit according to claim 9, further comprising a transistor for use. 11. The semiconductor device according to claim 1, further comprising a second holding transistor for holding a drain-source voltage of the first and second transistors at a predetermined voltage or less.
11. The constant voltage generation circuit according to any one of claims 10 to 10.