EP2643886B1 - Planar antenna having a widened bandwidth - Google Patents

Description

The present invention relates to a planar antenna with an enlarged bandwidth. It applies in particular to mobile communication terminals.

The invention applies, for example, to microwave planar antennas with an enlarged bandwidth.

Many devices, including mobile phones, use a planar micro-ribbon antenna for their flexible and easily integrable structure.

However, this antenna must meet certain criteria such as having a wide bandwidth, a high gain, a small footprint and it is low cost to integrate in these devices. These criteria often can not be met at the same time, especially for bandwidth, good performance (high gain) and reduced footprint. In particular, to have a good performance, the bandwidth of this antenna is generally low, of the order of 5%.

Several techniques based on the modification of the geometry of the antenna have been proposed to widen the bandwidth to the detriment of the size of the antenna. Other techniques rely on the use of lossy dielectric substrates, the insertion of slots on the radiating element, the use of the near context, and the use of materials having high impedance surfaces.

The document of the prior art US 7589676 which is considered to be the state of the art closest to the object of claim 1, discloses a slot-coupled antenna.

An example of such an antenna is given by the article "Stacked H-shaped microstrip patch antenna," published in 2004 in Antennas and Propagation, IEEE Transactions, pages 983-993, by J. Anguera et al.

In this article, there is described a patch antenna, comprising a first radiating element disposed above a ground plane and excited in its fundamental mode by a coaxial probe, and a second radiating element disposed above the first element and excited by the first radiating element by capacitive coupling so that the currents bloom in the first radiating element and excite to their all the second element. Metal studs allow the connection between the different layers separated from each other by a layer of air making office of dielectric to electrically isolate the conductive layers from each other.

In this article, the two radiating elements do not have the same size, the second radiating element is larger than the first radiating element. This results in a creation of two separate frequency bands.

The bandwidth of such an antenna is increased compared to a conventional structure but at the expense of the size of this antenna which is bulky. As a result, this type of antenna is very difficult to integrate because a relatively large antenna thickness is obtained for the purposes of integration into a communicating object.

One of the aims of the invention is to overcome all or part of the disadvantages of antennas of the state of the art by proposing an antenna which has both an enlarged bandwidth and a smaller footprint compared to antennas known in the art prior.

An object of the invention is to provide an antenna which has a good performance, ie an improved radiation efficiency.

Another object of the invention is to provide a thin-film antenna in planar technology also reducing its size so as to integrate it into an antenna network or any communication system.

Another object of the invention is to propose a dual-mode antenna, in other words two modes of polarization of the electromagnetic field propagating in the antenna, with two close resonance frequencies obtained thanks to a simple power supply / excitation device.

Another object of the invention is to propose an antenna with the two polarization modes orthogonal to each other, whose resulting orientation of the electromagnetic field changes as a function of frequency.

Another object of the invention is to propose an antenna having an input impedance compatible with a correct adaptation to the microwave devices.

Another object of the invention is to provide a simple antenna of realization and low cost favorable to a mass industrial production.

For this purpose, the invention relates to a planar antenna adapted to transmit or receive an electromagnetic wave, the antenna comprising at least a first conductive element disposed above a ground plane and separated from the latter, means for excitation of said at least first conductive element configured to excite two distinct orthogonal propagation modes (in particular two resonant modes), characterized in that said at least first conductive element is formed by a substrate comprising at least one thin layer of a material anisotropic with relative permeability greater than 10 for 2 GHz.

According to one embodiment of the antenna according to the invention, at least one slot is formed in the ground plane and allows said at least one conductive element to be electromagnetically coupled by at least one transmission line, said at least one slot being formed by a first opening extending in a direction forming a first angle of between 30 ° and 60 ° with the direction of the transmission line, and by a second opening extending in a direction forming a second angle between -30 ° and + 30 ° with the direction of the first opening.

An advantage of an antenna according to the invention lies in the fact that by virtue of the presence of an anisotropic material in a thin layer and / or at the arrangement of the openings with respect to an edge of the conducting or radiating element and their arrangement between them, it forces the electromagnetic field in the antenna to propagate in two orthogonal modes of propagation between them, separate and close, causing the antenna to have a single and single band wider in relation to the bandwidth of the antennas known, without complicating the structure and size of the antenna. This creates a dual mode antenna (or dual mode in English).

• chaque ouverture comprend un point proximal à un coin dudit au moins un premier élément conducteur situé à une distance maximale dudit coin sensiblement égale au tiers de la longueur de l'onde électromagnétique, avantageusement sensiblement égale au quart de cette longueur ;
• le point proximal est situé sensiblement sur la diagonale reliant ledit coin au coin opposé dudit au moins un premier élément conducteur ;
• ladite au moins une ligne de transmission croise chaque ouverture avec un angle compris entre 30° et 150° avec la direction dans laquelle s'étend l'ouverture ;
• ledit au moins un premier élément conducteur présente des dimensions différentes selon deux directions orthogonales (X, Y) ;
• ledit au moins premier élément conducteur est réalisé par un substrat comportant au moins une couche mince d'un matériau anisotrope à perméabilité relative supérieure à 10 pour 2 GHz ;
• le substrat peut comprendre en outre au moins une couche mince d'un matériau diélectrique à permittivité relative supérieure à 10 pour 2 GHz ;
• le substrat peut comprendre un empilement d'au moins une couche mince en matériau anisotrope et d'au moins une couche mince en matériau diélectrique alternées, l'épaisseur de la couche mince étant située entre λ/500 et λ/300 ;
• ledit au moins premier élément conducteur présente des dimensions égales selon deux directions orthogonales X, Y, avantageusement égales à la moitié de la longueur de l'onde électromagnétique ;
• l'antenne peut comprendre au moins un deuxième élément conducteur situé au dessus dudit au moins premier élément conducteur et séparé de ce dernier par une couche intermédiaire ;
• ledit au moins un premier élément conducteur et ledit au moins un deuxième élément conducteur ont les mêmes dimensions ;
• les ouvertures sont rapprochées pour ne former qu'une seule fente, et ladite au moins une ligne de transmission est disposée en regard de cette fente de manière à produire un couplage électromagnétique, à travers la première ouverture et à travers la deuxième ouverture, avec lesdits au moins un premier et un deuxième éléments conducteurs ;
• ladite seule fente forme un « L » et la ligne de transmission est disposée en regard du coin dudit « L » pour former, dans le plan desdits au moins éléments conducteurs, un angle compris entre 30° et 60° avec chacun des deux axes du « L », avantageusement un angle de 45°.

Embodiments of this planar antenna may include one or more of the following features:

• each opening comprises a point proximal to a corner of said at least one first conductive element located at a maximum distance from said wedge substantially equal to one-third of the length of the electromagnetic wave, advantageously substantially equal to a quarter of this length;
• the proximal point is located substantially on the diagonal connecting said corner to the opposite corner of said at least one conductive element;
• said at least one transmission line crosses each opening at an angle of between 30 ° and 150 ° with the direction in which the opening extends;
• said at least one conductive element has different dimensions in two orthogonal directions (X, Y);
• said at least first conductive element is made of a substrate comprising at least one thin layer of anisotropic material with relative permeability greater than 10 at 2 GHz;
• the substrate may further comprise at least one thin layer of a dielectric material with relative permittivity greater than 10 for 2 GHz;
• the substrate may comprise a stack of at least one thin layer of anisotropic material and at least one thin layer of alternating dielectric material, the thickness of the thin layer being between λ / 500 and λ / 300;
• said at least first conductive element has equal dimensions in two orthogonal directions X, Y, advantageously equal to half the length of the electromagnetic wave;
• the antenna may comprise at least one second conductive element situated above said at least first conductive element and separated from the latter by an intermediate layer;
• said at least one first conductive element and said at least one second conductive element have the same dimensions;
• the apertures are brought together to form a single slot, and said at least one transmission line is disposed opposite said slot so as to produce electromagnetic coupling, through the first aperture and through the second aperture, with said at least first and second conductive elements;
• said single slot forms an "L" and the transmission line is arranged facing the corner of said "L" to form, in the plane of said at least one conductive element, an angle of between 30 ° and 60 ° with each of the two axes of the "L", advantageously an angle of 45 °.

• l'utilisation des deux ouvertures à des positions situées au tiers, voire au quart de la longueur de l'onde électromagnétique émise ou reçue, ou l'utilisation de la fente en « L » dans l'alignement d'un coin d'un des éléments conducteurs ou rayonnants, permet d'exciter deux modes de propagation du champ électromagnétique de l'antenne ;
• l'utilisation, pour un des éléments conducteurs, d'un substrat composite magnéto-diélectrique anisotrope multi-alternances à perméabilité et permittivité relatives réglables, en particulier supérieures à 10 pour 2GHz, permet d'accroître la bande passante de l'antenne planaire tout en contribuant à sa miniaturisation ;
• l'utilisation d'une fente en « L » dans l'alignement d'un coin d'un des éléments conducteurs constitué un dispositif d'alimentation/excitation simple de réalisation et permet de n'avoir qu'un seul accès pour exciter les deux modes de propagation orthogonaux du champ électromagnétique dans l'antenne afin de maintenir un type de polarisation voulu ;
• isoler électriquement les éléments rayonnants ou conducteurs du plan de masse permet d'éviter de réaliser des plots verticaux reliant ces éléments au plan de masse ce qui simplifie la fabrication de l'antenne planaire et contribue également à la miniaturisation de l'antenne ;
• faire tourner la polarisation du champ électromagnétique en fonction de la fréquence d'un angle pouvant aller de 0° à 90°.

These embodiments also have the following advantages:

• the use of the two openings at positions one-third to one-quarter of the length of the electromagnetic wave emitted or received, or the use of the "L" slot in alignment with a corner of a conducting or radiating elements, makes it possible to excite two propagation modes of the electromagnetic field of the antenna;
• the use, for one of the conductive elements, of a multi-alternating anisotropic magneto-dielectric composite substrate with adjustable permeability and relative permittivity, in particular greater than 10 for 2 GHz, makes it possible to increase the bandwidth of the planar antenna while contributing to its miniaturization;
• the use of an "L" slot in the alignment of a corner of one of the conductive elements is a device for supply / excitation simple realization and allows to have only one access to excite the two orthogonal propagation modes of the electromagnetic field in the antenna to maintain a desired type of polarization;
• electrically isolating the radiating elements or conductors of the ground plane avoids making vertical studs connecting these elements to the ground plane which simplifies the manufacture of the planar antenna and also contributes to the miniaturization of the antenna;
• rotate the polarization of the electromagnetic field according to the frequency of an angle ranging from 0 ° to 90 °.

• la figure 1, une représentation en perspective d'un premier mode de réalisation d'une antenne selon l'invention ;
• les figures 2 et 3 sont une représentation en perspective et en coupe respectivement d'un deuxième mode de réalisation de l'antenne selon l'invention ;
• la figure 4, des courbes représentant l'évolution en fonction de la fréquence, la perméabilité complexe d'un matériau anisotrope utilisé pour former un des éléments conducteurs de l'antenne pour modifier les conditions de résonance selon une seule direction de l'antenne ;

• les figures 5a, 5b, une représentation schématique simplifiée d'exemples de modes d'alimentation d'une antenne selon l'invention ;
• les figures 6a et 6b, respectivement la partie réelle et la partie imaginaire de l'impédance d'entrée d'une antenne selon l'invention ;
• les figures 7a, 7b et 7c, des schémas simplifiés représentant trois types d'antenne différents, le premier type de la figure 7a étant connu de l'art antérieur ;
• la figure 8, des courbes représentant le coefficient de réflexion en fonction de la fréquence, pour les types d'antenne représentés en figures 7a, 7b, 7c ;
• la figure 9, une courbe représentant l'efficacité de rayonnement de l'antenne de la figure 7c en fonction de la fréquence ;
• les figures 10a et 10b, des schémas représentatifs de trois différents plans de coupe et de la distribution du champ électromagnétique se propageant dans l'antenne selon l'invention ;
• les figures 11 a et 11 b, l'évolution du rapport axial des composantes du champ électromagnétique en fonction de la fréquence ;
• les figures 12a et 12b, l'évolution de l'angle alpha entre un plan de coupe et une direction du champ électromagnétique en fonction de la fréquence
• les figures 13a à 13i, des exemples selon des schémas simplifiés de l'antenne de l'invention, selon la géométrie de l'antenne et la position de la fente (ou des ouvertures) par rapport à un bord de l'antenne.

Other characteristics will become apparent on reading the detailed description given by way of nonlimiting example, which follows, with reference to appended drawings which represent:

• the figure 1 , a perspective representation of a first embodiment of an antenna according to the invention;
• the Figures 2 and 3 are a representation in perspective and in section respectively of a second embodiment of the antenna according to the invention;
• the figure 4 , curves representing the evolution as a function of frequency, the complex permeability of an anisotropic material used to form one of the conductive elements of the antenna to modify the resonance conditions in a single direction of the antenna;

• the Figures 5a, 5b , a simplified schematic representation of examples of feeding modes of an antenna according to the invention;
• the Figures 6a and 6b respectively the real part and the imaginary part of the input impedance of an antenna according to the invention;
• the Figures 7a, 7b and 7c , simplified diagrams representing three different types of antenna, the first type of the figure 7a being known from the prior art;
• the figure 8 , curves representing the reflection coefficient as a function of frequency, for the types of antenna represented in Figures 7a, 7b, 7c ;
• the figure 9 , a curve representing the radiation efficiency of the antenna of the Figure 7c according to the frequency;
• the Figures 10a and 10b representative diagrams of three different section planes and the distribution of the electromagnetic field propagating in the antenna according to the invention;
• the figures 11 a and 11b, the evolution of the axial ratio of the components of the electromagnetic field as a function of frequency;
• the Figures 12a and 12b , the evolution of the alpha angle between a section plane and a direction of the electromagnetic field as a function of the frequency
• the Figures 13a to 13i , examples according to simplified diagrams of the antenna of the invention, according to the geometry of the antenna and the position of the slot (or openings) with respect to an edge of the antenna.

For convenience of representation, the figures are not to scale especially with regard to the thicknesses as well as the sizes of the openings.

In this description, the figures are oriented with respect to a reference XYZ comprising two orthogonal horizontal directions X and Y and a vertical direction Y. The terms "up" / "down", above "/" below "," on "/" under "are defined with respect to this direction Z.

In the following description, the features and functions well known to those skilled in the art are not described in detail.

In the figure 1 , a first embodiment of an antenna according to the present invention is shown in a perspective view.

The antenna 101 of the invention is a micro-ribbon planar antenna, able to emit and / or receive electromagnetic waves at a working frequency f _T corresponding to a wavelength λ _T. Typically, the frequency f _T is between 100 MHz and 100 GHz and preferably between 1 GHz and 10 GHz.

The planar antenna 101, preferably in micro-ribbon technology, emits essentially electromagnetic waves in the half-space above the XY plane. Here, the main transmit / receive direction is perpendicular to the XY plane and coincides with the Z direction.

Here, the antenna 101 comprises a stack, in the Z direction, of different layers extending essentially in a horizontal plane.

The stack comprises a first conductive or radiating element 111 disposed above a ground plane 115, or a substrate whose function is ground. In the particular case described here, the first conductive element is in the form of a horizontal plate, preferably substantially rectangular or substantially square, but may have other geometries as will be seen later.

In this embodiment, the first conductive element 111 has a horizontal front face exposed to electromagnetic radiation.

To electrically isolate the first conductive element 111 and the ground plane 115, the latter two are separated by a dielectric layer or a substrate 116 of a height h corresponding to the thickness of this layer, which is for example of the order of 500 to 700 μm.

In the example, the substrate 116 may be a dielectric thin film of the ROGERS type sold under the trademark ROGERS 4003 of relative permittivity equal to 3.55 and of a thickness equal to 0.8 mm. The ground plane 115 may be made of copper and may have a thickness of several micrometers, for example, from 9 μm to several mm.

A micro-ribbon transmission line is placed below the ground plane 115 to feed the first conductive or radiating element 111 through a slot 120 made in the ground plane 115.

Here, the transmission line may be a micro ribbon line printed on a ROGERS 4003 type substrate and with a characteristic impedance of 50 ohms. The dimensions of this line can be determined from the thickness and the permittivity of the substrate, for example, they can be 1.2 mm wide and 6 cm long.

A substrate layer, not shown, may be provided between the ground plane 115 and the transmission line 117 to maintain it below this plane and to electrically isolate it from the latter.

The ground plane 115 isolates the transmission line 117 from the radiating element 111 and limits the interference of the parasitic radiation on the radiation pattern of the antenna thereby providing polarization purity.

In a known manner, the transmission line, the electrical parameters and the dimensions of the various layers composing the antenna as well as the size of the slot are used to optimize the antenna.

According to the invention, the position of the slot 120 with respect to the conductive element and its shape have an effect on the performance of the antenna, in particular its bandwidth, as will be seen below.

According to the first embodiment of the invention, the first conductive or radiating element 111 is formed by an anisotropic magneto-dielectric composite substrate in a thin layer and with permeability and permittivity adjustable.

The material disclosed in the European patent application published under the number EP2200051 can, for example, be used in the context of the present invention to modify the resonance conditions of the conductive element 111.

More particularly, the first conductive element is formed by at least one layer of ferromagnetic material whose relative permeability is greater than 10 in the frequency band of interest, for example, for a frequency of 2 GHz, and whose thickness is strictly lower than the skin thickness of this ferromagnetic material. This thickness can be of the order of 25 to 80 nm.

A dielectric layer may be provided between this layer of ferromagnetic material and the ground plane 115 to electrically isolate this layer of the ground plane.

It is also possible that the composite substrate is made by a stack of thin dielectric layers, magnetic and conductive. This stack makes it possible to modify the resonance conditions of the conductive layer formed by the layer 111.

The material of the magnetic layers may be a ferromagnetic material used alone or coupled to an antiferromagnetic material.

For example, this composite material comprises a first stack of several ferromagnetic fine sub-layers superimposed on a thin insulating sub-layer itself superimposed on a second stack of several ferromagnetic fine sub-layers.

The stack of ferromagnetic fine sublayers may be composed of, for example, a first intermediate sublayer providing the interface between a first ferromagnetic sublayer and a dielectric underlayer, of a ferromagnetic sublayer, an antiferromagnetic sub-layer, a second ferromagnetic sub-layer, and a second intermediate sub-layer.

The first intermediate sublayer is for example made of ruthenium (Ru), tantalum (Ta) or platinum (Pt). Its thickness may be less than 10 nm.

The first ferromagnetic sublayer has a thickness less than the skin thickness of the ferromagnetic material and preferably less than half or one third of this skin thickness. Here, its thickness is less than 100 nm and preferably less than 50 or 25 nm. Such a choice of the thickness of the ferromagnetic sublayer limits the magnetic losses of the material.

Typically, this underlayer is made of an alloy of iron and / or cobalt and / or nickel. It may especially be a Cobalt FeCo iron alloy or a FeCoB alloy. Here it is a Fe ₆₅ Co ₃₅ alloy.

The antiferromagnetic sub-layer is for example made in a manganese alloy and in particular in a manganese and nickel alloy. For example, here it is a Nikel Magnesium Ni ₅₀ Mn ₅₀ alloy. The presence of the antiferromagnetic layer makes it possible to create an exchange coupling so that the material is self-polarized and does not therefore require the presence of an artificial external magnetic field.

Typically, the thickness of this sublayer is less than 100 nm and, for example, less than 50 nm.

The second ferromagnetic sublayer is for example identical to the first ferromagnetic sublayer. Similarly, the second intermediate sublayer is for example identical to the first sub-layer.

The insulating underlayer is made of a dielectric material having a relative permittivity greater than 10 and, preferably, greater than 100 in the frequency band of interest, for example at 2 or 3 GHz. This sublayer is typically made using a strontium (Sr) oxide and titanium (Ti). For example, it is titanium strontium (SrTiO ₃ ). The thickness of the dielectric underlayer is less than 10 μm or 1 μm. It is generally thicker than ferromagnetic and antiferromagnetic sublayers.

The second stack is for example identical to the first stack and will not be described in more detail.

According to a variant of this embodiment, the conductive element 111 and the dielectric layer separating this element from the ground plane can be replaced by an alternation of thin layers of anisotropic magnetic material with high permeability and thin layers of high-strength dielectric material. permittivity.

The typical thickness of the thin layers is advantageously between λ / 300 and λ / 100, where λ is the length of the wave emitted or received by the antenna, for example from a few tens to hundreds of nanometers.

The number of alternations can vary approximately from 1 to 10.

According to a second embodiment illustrated on the Figures 2 and 3 , the antenna 201 comprises a stack of two conductive elements 211 and 213 separated by an intermediate layer 212 and a dielectric layer 214 separating this stack from the ground plane 215. This conductive element contributes to radiate more effectively.

The conductive element 213 of the top of the stack is for example made of gold and has a horizontal front face exposed to electromagnetic radiation. Its thickness is for example 2 microns.

The intermediate layer 212 is made of silicon dioxide and acts as an electrical insulation between the two conductive elements. Its thickness is equal to 1 μm in the example, but the spacing between the first conductive element 211 and the second conductive element 213 may be greater, depending on the desired impedance matching level.

The dielectric layer 214 may comprise a substrate, for example glass.

The conductive element 211 is identical to the conductive element of the first embodiment. This conductive element may be made of conductive material of high conductivity or may be produced by a thin-film anisotropic magneto-dielectric composite substrate with adjustable permeability and permittivity, as will be seen below.

The stack of this second embodiment forms, in the example, a rectangular parallelepiped of length L equal to 35 mm, of identical width W, of height H equal to 500 μm, and disposed on the metal layer 215 forming the plane of mass surmounting a substrate layer 216 - in the example a ROGERS 4003 type substrate mentioned above with a thickness equal to 0.8 mm.

As will be seen below, with these dimensions the resonance frequency of the antenna for the fundamental mode TM ₁₀₀ is 2.1 GHz.

In the same way as in the first embodiment, a micro-ribbon transmission line 217 ( figure 3 ) is placed below the substrate layer 216 ( figure 3 ) for feeding the antenna through a slot 220 made in the ground plane 215.

An SMA connector may be used to power the antenna via the end of the transmission line 217.

The conductive or radiating elements are for example made of a conductive material whose conductivity is greater than 100 S / m and, preferably, greater than 1000 S / m or 1 MS / m. Here, the conductivity of the resonant elements 14 is greater than or equal to 5 MS / m.

To design an antenna with an enlarged bandwidth, the two conductive elements are metallic, and their dimensions in the X and Y directions are unequal. It is said that the antenna then has an asymmetry in its dimensions.

However, the dimensions of this antenna can remain identical (for a square antenna) and have an enlarged bandwidth by producing the conductive element 213 of a metallic material and the conductive element 211 into an anisotropic composite substrate.

The figure 4 illustrates, by curves, the complex permeability of the anisotropic magnetic composite material as a function of the frequency of the signal feeding the antenna. The first curve 401 represents the change as a function of the frequency of the permeability along a first axis in the plane of the antenna and the second curve 402 represents the change as a function of the frequency of the permeability of the material along an orthogonal axis the first axis of the curve 401, the two axes being in the plane of the conductive layer.

It appears that the anisotropic character of the material in thin layers results in the presence of different radioelectric properties along the two aforementioned axes, the relative permeability along the first axis being of the order of 200 at a frequency of 2 GHz, while is close to the unit along the second axis.

Therefore, the use of such a material to form one of the conductive layers of the antenna makes it possible to obtain two superposed conductive layers (layer 211 and layer 213, cf. figure 2 where the layer 211 is the conductive element 211 which is closest to the ground plane and the layer 213 is the conductive element 213 which receives the electromagnetic wave) square which have equal physical lengths (two layers each having dimensions according to the X and Y directions are equal) but have different electrical lengths, so as to widen the bandwidth. It should be noted that the conductive or radiating element 213 on the electromagnetic radiation side may have different dimensions than the conductive element 211.

In addition, it follows that the anisotropic composite material satisfies the needs of compactness and high integration of the antenna.

The Figures 5a and 5b schematically represent two simplified modes of supply of an antenna according to the invention, seen from below.

To facilitate the reading of these figures, only the conductive element 111 or 211 is shown.

According to Figures 5a and 5b , the antenna 500 comprises a conductive element 511 in the form of a patch having four edges of which only one of the edges is referenced in these figures.

On the figure 5a a first coupling power mode is shown.

A first opening 512a and a second opening 512b of fine rectangular shape are formed in the ground plane 551.

The first opening 512a extends in a direction forming an angle of between 30 ° and 60 ° with one of the edges 520 of the conductive member 511. Advantageously, said opening 512a forms an angle of 45 ° with this edge.

The second opening 512b extends in a direction forming an angle between -30 ° and + 30 ° with the direction of the first opening 512a.

Preferably, the two openings are each located at a maximum distance, equal to one third or even one quarter of the length of the electromagnetic wave, of a wedge 522 of the conductive element 511. They may both be close to each other. the same corner, or each close to a different corner.

The two openings 512a and 512b are located substantially on the diagonal connecting two opposite corners of the conductive element. They can be on the same diagonal and close to the same corner, or each close to an opposite side to the other. They can also be located on two different diagonals connecting two opposite opposite corners and close to the same edge 520 of the radiating or conductive element 511, or each disposed on these two diagonals close to two opposite edges of the conductive element 511.

The two openings can also cross and form a midpoint 512c near a corner 522 of the conductive element 511.

In this way, two modes of propagating an electromagnetic field to propagate in the antenna are forced.

The arrangement of these two openings is contrary to the arrangement of the openings according to the state of the art in which these openings in the ground plane are located towards the center of the conductive element or at a distance equal to half of the length of the electromagnetic wave emitted or received by the antenna, resulting in excitation of a single propagation mode or, if they exist, two propagation modes combined.

A transmission line 505 of micro-ribbon type is disposed obliquely under the ground plane 551 to feed the conductive element 511. This line crosses each opening at an angle between 30 ° and 150 ° with the direction in which extends the opening, the opening being chosen longer as the angle moves away from the value of 90 °. This length may be in the range of 1/6 to ½ of the width of the radiating element.

On the figure 5b a second preferred mode of supply is shown.

The two openings are close together and form an L-shaped slot 503 formed in the ground plane 551 and placed near a corner 522 of the patch 501.

The transmission line 505 is arranged obliquely under the patch, at an angle of approximately 45 ° with each of the branches 513a, 513b of the "L", so as to excite the antenna by coupling and cause the two orthogonal modes of propagation separated.

The transmission line 505 crosses and protrudes, of considerable length, the slot 503 at the angle of the "L", so as to ensure the impedance matching of the antenna. Typically, this exceeding of the length may be greater than λ / 20.

The transmission line 505 may intersect the slot 503 at an angle other than 45 °, but preferably in a range of 30 ° to 60 ° with one of the two branches 513a, 513b, so that each of the two modes be sufficiently powered.

Thus, if the transmission line is rotated about an axis orthogonal to the plane of the antenna and passing through a midpoint 514 between the outside corner of the "L" and the inside corner of the "L", one must at the same time adapt the length of each of the branches 513a, 513b to compensate for the imbalance generated by the angle different from 45 °. For example, if the angle between one of the branches 513a, 513b and the transmission line 505 decreases, the length of this branch should be increased so as to reinforce the mode of propagation due to this branch.

An advantage of this second power mode lies in the fact that only one excitation port is needed to bring the line of transmission 505 to excite the conductive element 511. As a result we have a simple power supply / excitation device to achieve.

Unlike the invention, the antenna of the state of the art needs, to excite two modes different from each other, two excitation ports, each of the ports allows a separate transmission line d bring the excitation to the conductive element. The known antenna may have only one transmission line, but in this case, two excitation accesses are necessary to have two modes, and a more bulky power supply circuit.

According to yet another embodiment of the antenna according to the invention, the supply is made by contact with a coaxial probe. The antenna may comprise a radiating element placed on the surface of a substrate surmounting a ground plane. The central core of a coaxial probe is preferably connected to a first axis of symmetry of the radiating element of the antenna (but not at its center), while the central core of a second coaxial probe is connected to a second axis of symmetry of the radiating element of the antenna (but not at its center) so as to excite two different orthogonal modes.

According to yet another mode of feeding an antenna according to the invention, the radiating element is directly powered by contact with microstrip lines.

According to yet another mode of feeding an antenna according to the invention, it is fed using a combination of different feed means, among which the use of probes, microstrip lines, or resonant slot.

The Figures 6a and 6b respectively the behavior as a function of the frequency of the real part and the imaginary part of the input impedance of an antenna according to the invention.

On the curve 601 showing the real part of the input impedance, a first resonance 611 at the frequency of 2.1 GHz representing the high resonance frequency of the antenna of the invention is observed and a second resonance 612 at a frequency of frequency of 2.04 GHz representing the low resonance frequency of this antenna.

These two frequencies of low and high resonances are obtained thanks to several parameters, for example, the dimensions of the conductive elements, the shape and position of the slot for exciting two fundamental propagation modes orthogonal to each other and distinct from the electromagnetic field propagating in the radiating elements.

The optimal operation of the antenna of the invention is obtained by the best compromise between all these parameters.

When the slot is rectangular and is located towards the middle of the radiating elements, only one mode is excited, or several different modes can exist but are confused. In other words, we do not control the excitation of these different modes.

The idea of the invention to design an antenna with modes of supply of the conductive elements the component, as described in relation to the Figures 5a and 5b allows you to control the propagation modes you want.

In addition, thanks to the design and composition of the conductive elements, the two propagation modes will generate two different resonant frequencies suitably positioned relative to each other to form a single operating frequency band, as will be seen below.

The Figures 7a, 7b and 7c represent, by diagrams, three different types of planar antenna, the Figures 7b and 7c showing simplified antenna diagrams according to the invention. W, H, L, Ms are the widths, lengths, heights of the conductive element and Ms one of the axes of propagation of the electromagnetic field

The first type of antenna, illustrated in figure 7a and known from the prior art, comprises a conductive element 701 of square shape and a rectangular slot 711 placed substantially towards the center of this element and made in the ground plane.

The slot has a length approximately equal to a quarter of the central wavelength of use of the antenna, and a width equal to about one tenth of this wavelength. The transmission line feeding the antenna cuts the slot 711, so as to excite the radiating elements of the antenna. The two orthogonal modes of propagation, if they exist, are then confused, so that the bandwidth is only equal to 1% (cf. figure 8 ).

For the second type of antenna according to the invention, illustrated in figure 7b the conductive element has a rectangular shape and the slot is an "L" slot 712 placed near a corner 722 of the radiating element 702.

The "L" slot 712 comprises a first branch 712a of the "L" parallel to the length of the radiating element and a second branch 712b of the "L" 712b perpendicular to the first branch 712a.

The corner 712c of the "L" is placed near a corner 722 of the radiating element, substantially on the diagonal connecting this corner 722 to the opposite corner 724 of the radiating element.

In addition, the first branch 712a is longer than the second branch 712b, in a ratio substantially equal to the ratio of length L / W between two adjacent sides of the radiating element. In other words, the longer the side of the antenna perpendicular to a branch of the radiating element, the longer the length of this branch is chosen large.

In this example, the antenna 702 does not include anisotropic material in one of its conductive layers; the asymmetrical dimensions of the radiating element, coupled to the unequal dimensions of the two branches of the "L" slot, make it possible to create two orthogonal propagation modes that are separate and close in frequency, as illustrated by FIG. figure 8 , and thus to widen the bandwidth to -6 dB of the antenna, the bandwidth at -6 dB of this antenna being approximately equal to 2.6%.

It should be noted that the point of the "L" slot which is proximal to the corner 722 of the antenna (in the example, the outer corner 712c of the "L") can be brought closer to the center of the radiating element 702. , but without departing from said corner of this element by a distance greater than one-third of the length of the electromagnetic wave, otherwise the two orthogonal modes will become closer in frequency until they become confused, and thus lose the beneficial effect of the frequency separation of the two modes.

Advantageously, the midpoint between the outer corner of the "L" and the inside corner of the "L", hereinafter referred to as the "middle point" of the slot, is located on the diagonal connecting two opposite corners of the radiating element and at a distance approximately equal to a quarter of the length of the electromagnetic wave.

The third type of antenna according to the invention, illustrated in Figure 7c , comprises a square shaped radiating element 703 comprising an "L" slot 713 placed near a corner of the conductive element 703. The side of this element 703 is approximately equal to half the length of the electromagnetic wave .

This conductive element 703 is made of a substrate of an anisotropic composite material, for example the material described in relation to the Figures 1 to 3 , allowing to modify, not the physical length of the radiating element, but the electric length of this element in a direction in the plane of this element.

By electrical length is meant the physical length divided by the square root of the product of the effective permeability with the effective permittivity of the material. $${\mathit{l}}_{\mathit{electric}}=\frac{{\mathit{l}}_{\mathit{physical}}}{\sqrt{{\mathit{\mu }}_{\mathit{effective}}{\mathit{\epsilon }}_{\mathit{effective}}}}$$

Permeability (or permittivity) is a magnitude that is such that its relationship to specific permeability (or permittivity) gives relative permeability (or permittivity).

In other words, instead of changing the physical length of the conductive element, as in figure 7b the effective permeability of the material included in one of the radiating elements is adjusted separately on each of the axes in the plane of the antenna.

Thanks to the use of the anisotropic properties of the material, each of the conductive elements of square shape and of the same dimensions leads to a different resonant frequency, the two frequencies being sufficiently close together for the bandwidth of the antenna to be widened.

Also, the dimensions of the branches 713a, 713b of the slot L, that is to say of its vertical component 713b and horizontal 713a, are chosen as a function of the permeability of the material in each of the directions corresponding to the branches of the L, and also according to the dimensions of the conductive elements, that is to say their width and length.

Similarly, the dimensions of each of the components 713a, 713b of the slot also depends on the position of the transmission line driving the excitation signal to the antenna, as explained above with respect to Figures 5a and 5b .

The bandwidth at -6 dB of this antenna is approximately equal to 4.3%.

It should be noted that the width of the bandwidth can be adjusted by adjusting the spacing between the two conductive layers 211, 213 (cf. Fig. 2 ) of the antenna (that is to say between the two radiating elements), the choice of the dimensions of the slot (s) and the choice of the permeability of the anisotropic material.

An advantage of the second and third type of antenna is that they each require only one access to excite the radiating elements, which facilitates the integration of the antenna in a circuit; indeed, a single transmission line, without additional circuitry is required.

Another advantage of these antennas is that the use of a single power supply to excite two orthogonal modes of propagation of the electromagnetic field makes it possible to maintain rectilinear polarization insofar as no phase shift is introduced between the two. propagation modes.

Another advantage of these antennas, illustrated later in figures 11 a and 11b, is that the polarization of the electromagnetic field propagating in the antenna changes as a function of the frequency of the signal.

An advantage of the third type of antenna is that the reduction of the electrical length of one of the two conductive layers, thanks to the permeability of the material, contributes to the miniaturization of the antenna because it is no longer necessary to to increase a dimension (cf. figure 7b ) to change the electrical length of a radiating element.

In addition, only a small thickness of insulation is required between the two conductive layers to remove the eddy currents, which allows to obtain an antenna of very small thickness, so a small footprint.

As a corollary, the broadening of the bandwidth of the antenna can advantageously be used to reduce the physical length of the antenna when a narrow band is sufficient for the intended application.

The figure 8 represents, via different curves, the reflection coefficient as a function of frequency, for the types of antenna represented in Figures 7a, 7b, 7c .

A first curve 801 represents the evolution as a function of the modulus frequency of the reflection coefficient, denoted S ₁₁ , of the first type of antenna represented in FIG. figure 7a . A single negative peak 811 appears because the two modes of propagation are merged; the propagation conditions being identical on both axes of the antenna.

A second curve 802 represents the evolution as a function of the frequency of the modulus of the reflection coefficient of the second type of antenna represented in FIG. figure 7b . This second curve 802 shows that two negative peaks 821, 822 appear.

The appearance of the first peak 821, separately from the second peak 822, is due to the elongation of one of the dimensions of the antenna. Each of these peaks 821, 822 corresponds to a propagation mode of the electromagnetic wave; two orthogonal propagation modes are therefore separated in frequency, because of the different physical dimensions of the antenna of the figure 7b .

With the appearance of these two orthogonal modes separated, the bandwidth at -6 dB is much wider than for the first antenna of the figure 7a .

It is necessary that the parameters of the antenna such as, for example, the dimensions of the slot, the dimensions of the antenna, the spacing between the two conductive layers, are chosen so that the two modes are not too far apart. frequency, otherwise the bandwidth is split into two disjoint parts corresponding to the two peaks 821, 822.

A third curve 803 represents the evolution as a function of the frequency of the modulus of the reflection coefficient of the third type of antenna represented in FIG. Figure 7c . As on the second curve 802, it is found on this third curve 803 that two negative peaks 831, 832 appear.

The appearance of the first peak 831, separately from the second peak 832, is due to the use of an anisotropic magnetic material modifying the resonance conditions in a direction of the antenna.

Two orthogonal propagation modes are therefore separated in frequency, thanks to the use of this anisotropic material. With the appearance of these two orthogonal modes separated, the bandwidth at -6 dB for this third antenna is even wider than for the second antenna 702 of the figure 7b .

However, in this particular case, there is a decrease in the value of the peaks 822, 832 with respect to the value of the peak 811 due to the position of the excitation (slot, transmission line) with respect to the conductive elements.

The two curves 802 and 803 have a plateau approximately around a frequency close to 2 GHz and which is at -6 dB. This plateau can be lowered to values below -6 dB, for example at -10 dB (corresponding to the value of the bandwidth for certain communication standards), by playing on the parameters such as the composition and the dimensions of the conductive elements. or radiating, the provisions of the slot and the transmission line between them and their respective geometry, and the disposition of the slot relative to a corner of one of the radiating elements.

To evaluate the performance of the antenna according to the invention, the figure 9 shows a curve 901 representing the radiation efficiency of the antenna of the Figure 7c depending on the frequency of the excitation signal of this antenna. It appears that the antenna of the figure 9 reveals a strong disparity as a function of frequency. The conductivity of the anisotropic material plays an important role in the performance of the antenna, since depending on the quality of the conductive element produced by this material, a different efficiency is obtained.

It is noted that the efficiency is very good at the high resonance frequency which corresponds to the mode not solicited by the anisotropic material. It is however less important when one goes towards the low resonant frequency. This is due to the ohmic losses of the material due to the eddy currents created in the conductive layer by the variation over time of the electromagnetic field.

As announced above, one of the advantages of the invention, more particularly the antenna according to the second embodiment provided with a single excitation access for the transmission line, lies in the fact that the polarization of the electromagnetic field is propagating in the antenna according to the invention changes according to the frequency and varies at an angle ranging from 0 ° to 90 °.

In the state of the art, two excitation accesses lead to two distinct polarizations of the electromagnetic field. It is realized that, thanks to the invention, namely having an excitation port and two orthogonal polarization modes, a rotating polarization is obtained.

To understand this phenomenon, we will describe the Figures 10 to 12 .

The Figures 10a and 10b represent simplified diagrams representative of three different cutting planes of the antenna of the invention and the distribution of the electromagnetic field propagating in this antenna.

The planes P1, P2 and P3 are defined as references to highlight the polarization variations as a function of frequency. They are such that the plane P1 coincides with the plane of the direction X, the plane P3 coinciding with the plane of the direction Y, the plane P2 being situated between the two.

More particularly, when the electromagnetic field is defined in known polar or cylindrical coordinates, the planes P1 and P2 define between them an angle equal to the angle φ and the plane P3 and the plane of the direction Z define an angle equal to angle θ. The plane P1 is such that φ = 0 °, for the plane P2 φ = 45 ° and for the plane P3, φ = 90 °.

The electromagnetic field, in particular the component E of the field has two components, one E _φ along the horizontal plane including the angle φ and _θ E is the vertical plane comprising the θ angle.

The polarization mode of the electromagnetic field chosen in this example is the rectilinear polarization. Other polarizations may be envisaged, such as elliptical polarization or circular polarization, for example.

It should be noted that in the plane P1, we find the low resonance frequency of the antenna corresponding to the propagation mode of the electromagnetic field propagating in the antenna provided with the anisotropic material. In the plane P3, we find the high resonant frequency of the antenna corresponding to the mode of propagation of the field propagating in the antenna without the influence of the anisotropic material (which intervenes only on a single direction). In the plane P2, the two modes of propagation of the field coexist.

In known manner, an axial ratio is defined which is, for an elliptical polarization, the ratio between the major axis of the ellipse on the minor axis of this ellipse. If one approximates the elliptic polarization by a rectilinear polarization, this ratio is worth either 0 or infinity all depends on the axis on which one is placed.

The figures 11 a to 12b show the evolution as a function of the frequency of the axial ratio of the electromagnetic field propagating in the antenna according to the invention.

It can be seen that the axial ratio for the plane P1 is low for the high resonance frequency of the antenna and then increases as one approaches the low resonance frequency of the antenna.

Conversely, the ratio for plane P3 decreases as the frequency decreases from the high resonance frequency to the low resonant frequency.

A common point exists between the two axial ratios of the planes P1 and P3 and corresponds to a point for which this ratio is zero. This point is located between the two frequencies, where the two components E _θ and E _φ are equal. This common point corresponds to an angle φ = 45 °.

On the figure 11b it can be seen that for f ₀ , the low resonance frequency E _θ is greater than E _φ , so that the influence of the anisotropic nature of the material on the antenna of the invention is noted. For f ₁ E _θ is equal to E _φ , and for f ₂ , corresponding to the high resonance frequency, for which the anisotropic nature of the material does not occur, E _{θ is} smaller than E _φ .

On the Figures 12a and 12b , the evolution of the angle alpha as a function of the frequency is represented. This angle alpha is defined by the angle between the plane P1 and the direction of the field E, in other words according to a first axis 1211 orthogonal to the direction of propagation of the field and the axis 1221 of the electromagnetic field of the signal propagated in the antenna ( figure 12b ).

The figure 12a illustrates the evolution of the polarization of this field as a function of frequency, by showing the evolution of the angle alpha as a function of frequency. It can be seen that at the low resonance frequency f ₀ equal to 2.04 GHz, the angle alpha is approximately equal to 20 °, then the angle alpha increases to approximately 45 ° at f ₁ = 2.07 GHz and almost 90 ° at the high resonance frequency f ₂ = 2.1 GHz.

The Figures 13a to 13i represent variants of the embodiments of the antenna according to the invention. These variants have been partly described in relation to Figures 5a, 5b and 7b and 7c .

Here, either there are two separate separate openings and two transmission lines arranged in two directions orthogonal to each other to excite the two orthogonal propagation modes of the electromagnetic field propagating in the antenna, or there is a single slot with a single line of transmission exciting both modes.

Note that the two openings can come together to make a single "T" slot, as shown in the illustration. figure 13a , or an "L" slot, as shown on the figure 13d , but whose branches 712a, 713a and 712b and 713b are symmetrical with respect to the branches of the slot illustrated on the Figures 7b and 7c .

It can also be noted that it is possible to obtain a triangle-shaped slot as illustrated on the figure 13e whose adjacent sides are in alignment with a corner of the conductive element.

It is also possible to use a radiating element of circular shape for which two openings are necessary to have the excitation of the two orthogonal propagation modes, as illustrated on the figure 13h . Note, however, that the configurations of Figures 13b, 13c, 13f and 13g may apply for the circular radiating element.

A geometry of the radiating element in ellipse as illustrated on the figure 13i allows to have only one slot for the excitation of the orthogonal modes. Indeed, here the ellipse has two distinct dimensions (a major axis and a small axis), so it is possible to have only one slot instead of two openings. This slot can have any geometry, provided to respect the position of the slot according to the invention. In the example of the figure 13i this slot has the shape of an arc.

Many other embodiments are possible.

Many shapes are possible for each radiating element. For example, it may be a square pellet, orthogonal, diamond-shaped or a dipole. Generally, this shape has an axis of symmetry with respect to an axis orthogonal to the plane in which the essential of this radiating element extends.

In a simplified embodiment, the second stack and the dielectric underlayer of the radiating element 111, 211 are omitted. In an even more simplified embodiment, the conductive or radiating element consists of a single thin sublayer of ferromagnetic material whose thickness is less than the skin thickness of this ferromagnetic material.

Alternatively, other materials may be used as the dielectric. For example, it may be an oxide of barium (Ba) and titanium (Ti), in particular barium titanium BaTiO ₃ , hafnium oxide (Hf), in particular HfO ₂ , or tantalum (Ta), especially Ta ₂ O ₅ (ferroelectric). Perovskites, however, are preferred, such as BaTiO ₃ or SrTiO ₃ for example, which have a higher relative permittivity (of the order of 100 versus 10 for barium or hafnium oxides at 2 or 3 GHz).

Other materials are also possible for the antiferromagnetic layer such as a PtMn or IrMn alloy and more generally any manganese-based alloy or iron oxides or cobalt or nickel.

For the ferromagnetic layer, the alloys CoFeB, FeN and CoFeN will be preferred, but other materials are possible, especially all the alloys combining two or three of the elements chosen from iron, cobalt and nickel. These alloys may optionally be doped, for example boron or nitrogen. They may also be associated with other elements such as Al, Si, Ta, Hf, Zr.

The radiating conductor 213 may be a single wire.

Moreover, at least two antennas according to the invention can be grouped together in an antenna array for any type of communication system to increase the efficiency of the radiation as well as the gain of the antenna.

Claims (13)

1. A planar antenna (201) designed to transmit or receive an electromagnetic wave, said antenna comprising at least one first conducting element (211, 511) disposed above a ground plane (215) and separated therefrom, means (505, 565, 575, 585, 589) for exciting said at least one first conducting element, which means are configured to excite two distinct orthogonal propagation modes, characterised in that said at least one first conducting element (211, 511) is formed by a substrate comprising at least one thin layer of an anisotropic material with relative permeability of more than 10 for 2 GHz.
2. The planar antenna according to claim 1, wherein said substrate comprises at least one thin layer of a dielectric material with relative permittivity of more than 10 for 2 GHz.
3. The planar antenna according to claim 1 or 2, wherein said substrate comprises a stack of at least one thin layer of anisotropic material and at least one thin layer of dielectric material that are alternating, with the thickness of said thin layers being between λ/300 to λ/100.
4. The planar antenna according to any one of claims 1 to 3, wherein said at least one first conducting element (211) has equal dimensions along two orthogonal directions (X, Y), advantageously equal to half of the length of the guided electromagnetic wave.
5. The planar antenna according to any one of claims 1 to 3, wherein said at least one first conducting element (211, 511, 702) has different dimensions along two orthogonal directions (X, Y).
6. The planar antenna according to any one of the preceding claims, characterised in that it comprises at least one second conducting element (213) located above said at least one first conducting element (211, 511, 702, 703) and separated from said first conducting element by an intermediate layer (212).
7. The planar antenna according to claim 6, wherein said at least one first conducting element (211, 511, 702, 703) and said at least one second conducting element (213) have the same dimensions.
8. The planar antenna according to any one of the preceding claims, wherein at least one slot (120, 220) is formed in said ground plane (115, 215, 501) and is configured so that at least one first conducting element (111, 211) is fed by electromagnetic coupling by at least one transmission line (117, 217, 505), characterised in that said at least one slot (120, 220) is formed by a first opening (512a, 513a, 712a, 713a) extending in a direction forming an angle of between 30° and 60° with said transmission line (505), and by a second opening (512b, 513b, 712b, 713b) extending along a direction forming a second angle of between -30° and +30° with the direction of said first opening.
9. The planar antenna according to claim 8, wherein each opening (512a, 513a, 712a, 713a; 512b, 513b, 712b, 713b) comprises a proximal point (512c, 712c) at a corner (522, 722) of said at least one first conducting element (111, 211, 511, 702, 703) substantially located at a maximum distance from said corner that is equal to a third of the length of the electromagnetic wave, advantageously equal to a quarter of this length of the electromagnetic wave.
10. The planar antenna according to claim 9, wherein said proximal point (512c, 712c) is substantially located on the diagonal linking said corner to the opposite corner (524, 724) of said at least one first conducting element (111, 211, 511, 702, 703).
11. The planar antenna according to any one of claims 8 to 10, wherein said at least one transmission line (117, 217, 505) crosses each opening with an angle of between 30° and 150° with the direction in which the opening extends.
12. The planar antenna according to any one of claims 8 to 11, wherein said openings (513a, 712a, 713a; 513b, 712b, 713b) are brought together so as to form a single slot (512, 712, 713) in the alignment of a corner (522, 722) of said at least one first conducting element, and said at least one transmission line (505) is disposed facing said slot so as to produce an electromagnetic coupling, through said first opening (513a, 712a, 713a) and through said second opening (513b, 712b, 713b), with said at least one first and one second conducting elements (211, 511, 702, 703; 213).
13. The planar antenna according to claim 12, wherein said single slot forms an "L"-shape (512, 712, 713) and said transmission line (505) is disposed facing the corner of said "L"-shape so as to form, in the plane of said at least one conducting element (211, 511, 702, 703; 213), an angle of between 30° and 60° with each of the two axes of the "L"-shape, advantageously forming an angle of 45° with these axes.

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