EP1794613A1

EP1794613A1 - Radar system comprising a heterodyne mixer for the improved detection of short-range signals

Info

Publication number: EP1794613A1
Application number: EP05771892A
Authority: EP
Inventors: Thomas Walter; Juergen Hildebrandt; Joachim Hauk; Martin Reiche
Original assignee: Robert Bosch GmbH
Current assignee: Robert Bosch GmbH
Priority date: 2004-09-25
Filing date: 2005-08-03
Publication date: 2007-06-13
Also published as: US20080117099A1; WO2006032569A1; DE102004046632A1

Abstract

The invention relates to an antenna radar system comprising a short-range antenna path (310 - 365), and a long-range antenna path (210 305) separate from the short-range antenna path (310 - 365). According to the invention, a push-pull mixer (345 - 360) is especially arranged in the short-range antenna path. A heterodyne frequency translation or mix of the signals to be processed can be carried out by means of the push-pull mixer (345 - 360) in the short-range antenna path.

Description

RADAR SYSTEM WITH HETERODYNAMIC MIXER FOR IMPROVED DETECTION OF INTERMEDIATE SIGNALS

State of the art

The invention relates to an antenna radar system which can preferably be used in the automotive industry and to a method for its operation according to the preambles of the respective independent claims.

In the field of automotive technology so far almost exclusively

Long Range Radar (LRR) radar systems are used for remote detection of detection targets. However, there is also an increasing demand for the use of Nahradarsystemen (Short Range Radar = SRR) with close range detection, for example, to carry out distance measurements in vehicle convoys (automatic starting in traffic jam rides or the like.) Or for use as a parking aid.

The detection field for short-range applications generally has a much larger aperture angle compared to long-range applications. However, due to a smaller so-called EIRP value in short-range applications, these also have a shorter range. The said EIRP (Equivalent Isotropy Radiated Power) value represents a purely arithmetic quantity and indicates with which transmission power one would have to supply an antenna uniformly radiating in all spatial directions (isotropically) in order to achieve the same power flux density in the far field as with a focusing directional antenna in your Main transmission direction. For these reasons, it is nearly impossible to provide a common antenna aperture for the LRR and SRR functions.

In the known in the art, suitable for the automotive sector Antennenradarsystemen the received signal by means of unbalanced

Homodynes down frequency offset ("down-mixed") This causes noise in the amplitude-modulated output signal, thus significantly limiting the sensitivity of the radar system to near-range objects.

In addition, antenna radar systems already used outside the automotive industry and optimized for near range detection / detection currently only achieve a minimum measuring distance in the range of 0.5 m. In the above-mentioned driving situations (traffic jams, etc.), however, the smallest possible minimum measuring distance in the range of a few decimeters is desired.

The invention is therefore based on the object to further develop an antenna radar system of the type described above to the effect that said short-range weakness of the known systems is eliminated. However, this development should be based as far as possible on existing antenna radar systems in order to keep the development and production costs as low as possible.

Advantages of the invention

The invention proposes, in an antenna radar system concerned in one

Nahbereichsantennenpfad provide a push-pull mixer, which uses a same or at least very similar intermediate frequency as there known as provided phase-locked loop (PLL).

The inventively proposed Antennenradarsystem can be by means of

Operate push-pull mixer in the near range by means of heterodyne mixing, wherein the frequency shelves of potential detection targets are far outside the phase noise of a local oscillator (LO) arranged in the Nahbereichsantennaspfad come and the amplitude modulation noise (AM noise) is suppressed on the LO path.

The push-pull mixer thus suppresses the most present on the LO path amplitude modulation noise, which together with the carrier frequency of

Transmission signal is automatically mixed in arranged around the intermediate frequency sidebands. As is known, in the case of amplitude-modulated signals, the carrier frequency itself does not vary in amplitude. Rather, the modulation occurs in the form of signal components with frequencies slightly above and below the carrier frequency, which signal components are commonly referred to as "sidebands".

As a result, short-range measurements with a resolution of a few decimeters are made possible by means of the invention.

In a preferred embodiment, the LO of the push-pull mixer is fed with the fourth (4th) harmonic of a reference oscillator; however, instead of using the fourth harmonic, two frequency doublers may also be provided, which has the added advantage that maximum LO power can be maintained for the balanced mixer.

In the field of automotive technology, monostatic antennas are used in radar systems affected here, in which the signals transmitted and emitted (so-called "RX / TX feeds") use a common antenna lens.The polarization axes of these two signals usually have one in the radar systems mentioned Angle of

45 °, to ensure that received from an oncoming, equipped with a same radar vehicle outgoing signals are received with respect to the own received signal cross-polarized. Due to this measure, disturbing interference between the signals of the two vehicles is effectively suppressed. The antenna radar system according to the invention can be designed for this purpose so that the apertures of the distance and Nahbereichsfunktion are operated cross-polarized, wherein by means of switchable transmitter preamplifier in the transmission path of the remote and Nahbereichsradarfunktion a temporal multiplexing remote / Nahmode is realized. Due to the known antenna characteristic of radar antennas, ie the - A -

given main and side lobes in the radiation and Einstrahlcharakteristik, it would namely without the measures mentioned to a crosstalk (coupling) between these two functions. Aufrund the cross polarization for the Nahbereichs¬ and the Fernbereichsfunktion a very effective decoupling between these two functions is achieved, so that these functions readily in a single

Antenna radar system can be integrated.

By means of the inventively proposed heterodyne frequency conversion (mixture), an existing predominantly long-range antenna radar system (LRR) can be expanded by a high-resolution close-range detection, for example a

To provide combination system for both the near and the far range.

The antenna radar concept according to the invention can be used with the advantages mentioned in addition to monostatic antennas in bistatic antennas, which are known to have separate transmission and reception paths. Already due to this path separation crosstalk of the transmission signal is minimized in the receiver.

drawing

The invention will be described in more detail below, with reference to the attached drawing, by means of exemplary embodiments from which further features and advantages of the invention emerge.

In detail in the drawing:

1 is a schematic diagram of a receiver circuit with heterodyne detection according to the prior art;

Fig. 2a, legs an overview of the principle occurring in down and up mixers by means of a mixer shown in Figure 1; Fig. 3 is an electronic circuit diagram of a preferred embodiment of the erfϊndungsgemäßen Antennenradarsystems;

Fig. 4a, btypische signal waveforms in the time domain of a single-mixer (a) and a

Push-pull mixer (b) in comparison; and

5 shows a transmission power mask of a combination LRR-SRR sensor according to the invention.

Description of exemplary embodiments

As in the known high-frequency (RF) transmission / reception technology, for example of communications engineering, the antenna radar systems included here have a phase-locked loop (phase-locked loop) enabling the reception of very short wavelengths, with the associated relatively high spatial resolution. PLL ') 70, which is modulated by a digital divider N and which has an integrated voltage controlled oscillator (VCO), which is used to generate a carrier signal. As can be seen from FIG. 1, the VCO additionally acts as a so-called local oscillator for the mixer 20, for example for the receiver, which down-converts or converts the high-frequency received signal f_E to a lower intermediate frequency f_ZF. This principle of frequency conversion or frequency reduction has been used for many decades in radio receivers.

As is known, a distinction is made between direct detection and heterodyne detection in the signal-receiving detectors mentioned above. In direct detection, an incoming received signal is further processed directly, whereas in heterodyne detection an additional signal f_LO fed by a local oscillator (LO) 30 is superimposed on the received signal f_E. These two frequencies are mixed to obtain the said intermediate frequency (f_ZF). The intermediate frequency f_ZF is in a frequency range that can be easily amplified and the

Use of frequency selection circuits with the desired bandwidth. Such a heterodyne receiver is shown schematically in FIG. Normally, the intermediate frequency is fixed and only the oscillator 30 and the input circuit 10 are matched. The heart of the circuit is the mixer 20, in which the received signal f_E is superimposed with the oscillator frequency f_LO. If one were to introduce a frequency-independent resistor directly at the output of the mixer stage 20, the four following different frequencies would result:

a. the input frequency f_E b. the oscillator frequency f_LO c. the sum frequency f_E + f_LO and d. the difference frequency f_LO - f_E (= intermediate frequency f_ZF).

In general, only interested in the intermediate frequency f_ZF and therefore one tuned to f_ZF Bandfϊlter 40 in the output line of the mixer 20. From there it goes to further amplification and selection to a downstream intermediate frequency amplifier 50 and turn this downstream, for the eventual demodulation of amplitude modulated input signal detector 60.

Accordingly, by means of the heterodyne receiver shown schematically in FIG. 1, the input signal f_E is converted or mixed before the demodulation by means of the mixer 20 to the fixed intermediate frequency f_ZF. Accordingly, on the (not shown here) transmitter side, the modulation is often performed not on the transmission frequency, but also on a smaller intermediate frequency and the resulting

Signal raised to the desired transmission frequency. The necessary beat frequency is also supplied by a VCO covering, for example, the frequency range of 300 to 450 MHz.

The above-described frequency conversion is carried out either by means of "upmixers" or by means of "downmixers", depending on whether the desired output signal should be above or below the input signal. Such a mixer provides grds. a three-port with the inputs for the input frequency f_E and the local oscillator frequency f_LO and with an output for the intermediate frequency f_ZF, wherein the mixing a non- represents a linear process in which at least two of said variables are multiplied together. An ideal mixer behaves between the gates, E 'and, ZF' as a matched, lossy two-port, which simply makes a frequency shift in addition. At the input 'LO', the local oscillator signal with the frequency f_LO is supplied, which determines the difference between f_E and f_ZF and is generally much stronger than the other two signals. For the relationship between the three frequencies mentioned, F ZF = +/- (f_E - f_LO).

In a down-converter, the input signal at the frequency f_E is higher-frequency than the desired output signal f_ZF. Depending on whether f_E is greater or less than f_LO, the positive or negative sign applies in the above equation. The relationship between these three frequencies is shown in FIGS. 2a and 2b. Fig. 2a shows the frequencies occurring at a down-converter, whereas Fig. 2b comprises the frequencies resulting from an up-converter. The downward arrows correspond to input signals, whereas the upward arrows

Arrows represent output signals.

In addition to the desired output frequency fl, the so-called "image frequency" f_SP, which has a frequency of f_SP = f_LO-f_ZF, also occurs in particular (see FIG. 2b).

The meaning of the image frequency (in FIG. 2c) is that an external signal having the image frequency, at a given local oscillator frequency, is converted (mixed) into the same intermediate frequency f_ZF as the desired input signal of the frequency f_E. Therefore, the image frequency is usually by means of a suitable

Input filter ausgefϊltert. The disturbing "image frequency" f_SP represents a mirror-image (at a distance of the intermediate frequency) from the oscillator frequency and usually undesirable second reception option. For the image frequency f_SP is therefore the context f_SP = f_E - (2 * f_ZF).

The antenna radar system according to the invention shown in Fig. 3 comprises at the same time a far-range (LRR) function comprised by reference numerals 210-305 and a short-range (SRR) function comprised by reference numerals 310-365 and reference numerals 230 and 237. The LRR function 210-305 and the SRR Function 310 - 365 are synchronously in the present exemplary embodiment, ie not operated in time multiplex operation by means of a changeover switch, multiplexer or the like. It should be noted, however, that the present invention is basically also applicable to such time-division multiplex systems.

The said synchronous operation is only possible with each other without interference between the two functions, since cross-polarization takes place between the SRR function and the LRR function in this exemplary embodiment, which results in sufficient signal-technical isolation between the two functions. In cross-polarization, the polarized signals of the SRR function and the LRR

Function operated in a conventional manner polarized perpendicular to each other, which is prevented in many situations that the signals can overlap constructively or destructively at all.

The feeds indicated only schematically in Fig. 3, i. the Tx / Rx feed 290-305 the

LRR function as well as the Tx feed fed via a first patch antenna 237 and the Rx feed of the SRR function supplied via a second patch antenna 365, which provide the actual transmission / reception function of the monostatic antenna, are each connected by a 3, which is not shown in FIG. 3. The Tx feed supplied via the first patch antenna 237 enters into a "patch array"

Bias bias 235 powered preamplifier 230.

With regard to the patch arrays, it should be noted that their technical details are not important in the present context. Such a patch array for a radio-frequency antenna is, for example, in the simultaneously filed patent application

(Applicant's file R. 307998), to which reference is made in its entirety in the present context. Said feeds 290 - 305, 237 and 365 are spatially separated from each other for the reasons mentioned.

The circuit shown in Fig. 3 will now be described in more detail. First, the near-end antenna path (SRR path) 310-365 will be described. An input signal 200 supplied by a first phase-locked loop (not shown) (PLL) initially serves to operate a voltage-controlled oscillator (VCO) 205. The oscillation signal generated by the VCO (in this case transmitting VCO) 205 is generated by means of a power divider 210, 215 fed to the short-range transmission antenna 237. This input signal 200 is fed by means of a preferably capacitive coupling element 310 to a mixer 320 whose input signal in turn comes from a source 340. For this, the fourth harmonic of a stable reference oscillator 340, which in the present embodiment oscillates at a frequency of 4 * 18.65 GHz = 74.6 GHz, and the signal decoupled from the VCO, which in this case has a frequency of 76.5

GHz +/- 125 MHz, mixed at a serially arranged in the SRR path 310 - 365 diode 320. Said fourth harmonic is generated in the present embodiment from the signal supplied by the reference oscillator 340 by means of two series-connected frequency doublers 330, 335.

Consequently, the intermediate frequency resulting from this down-conversion in this example is 76.5 GHz - 74.6 GHz = 1900 MHz. The already mentioned image frequency, which, as mentioned above, has a particularly critical effect on interfering signals when they are mixed to the same intermediate frequency, is 72.7 GHz in the present example.

The frequency generated at 330, 335 and 340 is applied to a balanced mixer 345-360. The exact mode of operation of push-pull mixer 345-360 will be described in more detail below with reference to FIGS. 4a and 4b.

Because of the heterodyne downconversion of the intermediate frequency signal and the antenna signal supplied via the SRR feed 365, which are also described below, the frequency dependencies of potential detection targets fall far outside the phase noise of the LO 330, 335 and 340. The phase noise is mixed, for example, by reflection at the RX feed 365 in a DC-near frequency range.

By contrast, the AM noise of the LO 330, 335 and 340 is mixed directly into the DC-near frequency range by rectification in the mixer. The AM suppression done in push-pull mixer by the cancellation due to the different polarity of the two diodes.

In the technical implementation of the invention, the basic modulation form of known systems can be maintained, thus largely relying on existing electronics

(VCO, PLL, reference StaLO, etc.) can be accessed.

It should also be noted that in order to achieve the aforementioned characteristics in alternative pulsed radar (rather than the present continuous wave radar), in contrast to the approach described above, an implementation of fast switches, their drivers, and high precision variable delay electronics would be required. However, the components mentioned are very expensive.

The far-field antenna (LRR) path 210 - 305 has four unaligned single-diode mixers 270 - 285 for downconverting the signal provided by the VCO 205 here. The mixing diodes 270-285 each lie separately in the path of each Tx / Rx feed 290-305. The mixing diodes 270-285 functionally correspond to switches which are opened and closed in time with the oscillator 205. The Tx signals arrive via a further four patch antennas 290, likewise designed as patch arrays.

305 to a focusing unit (e.g., lens) and radiated therefrom. The reflected components reach the patch antennas 290-305 via the focusing unit and are mixed into the baseband by means of the mixing diodes 270-285. The low-frequency IF signal resulting from the down-conversion by means of the mixing diodes 270-285 is then applied via the patch antennas 290-305 and the

Mixer diodes 270-285, in turn, the TP 202 - 265, the entire received power converging TP structure supplied - in turn, a second, supplied with a bias voltage 225 preamplifier 220.

Figures 4a and 4b illustrate the operation of a single-diode mixer (Figure 4a) and a balanced mixer (Figure 4b) in direct comparison. Respective corresponding components are provided for convenience with corresponding, above-painted reference numerals. As can be seen from Fig. 4a, an AC voltage applied to the input US by means of a transformer coil 400 (or 400 ') first, depending on the application up or down transformed. In the upper leg a (mixer) diode 410 is arranged, whereas on the lower leg both a local oscillator (LO) 420 and a LO downstream resistor 430 are arranged. As a result of the above-described superposition of the input signal US and of the oscillator signal U LO, a voltage U IF oscillating with the intermediate frequency ZF is dropped across a load resistor RL 440 arranged at the output. The oscillator signal U LO periodically controls the diode 410 non-linearly. The input signal US sees in the diode 410 a linear, temporally periodically variable network, ie the LO 420

"Pumps" the diode 410.

As can be seen in Figure 4b, unlike the single-ended mixer, the balanced mixer has two symmetrically connected (i.e., balanced) diodes 410 ', 415 which are driven in the same direction by an LO 420'. The LO 420 'and a resistance 430' associated therewith are arranged on an additional line branch arranged symmetrically (in the middle) with respect to the upper and lower line branches. The resulting isolation between LO 420 'and IF suppresses the LO noise at the IF gate.

In the right half of Fig. 4 are shown typical resulting in the two mixers output voltage waveforms. The uppermost diagram shows the voltage curves of the predetermined signals U S 470 and U LO 460. In the left half of FIG. 4, the upper diagram shows the conventional circuit diagram of a single-diode mixer and the lower diagram shows the circuit diagram of a conventional push-pull mixer.

FIG. 5 shows a transmission power mask (EIRP over frequency) for the far range with frequencies of 76-77 GHz as well as the near range with frequencies of 79-81 GHz. With known antenna radar systems objects can be measured in a maximum distance of about 100 - 200 m. In this area, the highest sensitivity of the system must be guaranteed. For a ramp duration of, for example, .DELTA.t = 5 ms and a frequency deviation of .DELTA.f = 250 MHz would be at a static detection target (ie without Doppler shift) at a distance of 100 m a Frequency deviation in the homodyne intermediate frequency band of fz _F = 2 * 100 m * Δf / Δt / c = 33 kHz.

Claims

claims

Antenna radar system comprising a short - range antenna path (230, 237, 310 - 365) and a far - field antenna path (210 - 305) separated from the short - range antenna path (230, 237, 310 - 365), characterized in that a balanced mixer (345 - 360 ) is arranged.

2. Antenna radar system according to claim 1, characterized in that the push-pull mixer (345 - 360) is connected to a local oscillator (340), wherein between the Gegentakmischer (345 - 360) and the local oscillator (340) two series-connected frequency doubler (330, 335) are arranged.

3. Antennenradarsystem according to claim 1, characterized in that the push-pull mixer (345 - 360) is supplied with the provided by a reference oscillator fourth harmonic of a reference frequency of preferably 18.65 GHz.

4. Antennenradarsystem according to claim 3, characterized in that the reference frequency for the mixing of the intermediate frequency in the Nahbereichsantennenpfad (230, 237, 310 - 365) is generated by means of a mixer.

5. Antennenradarsystem according to claim 4, characterized in that the

Mixer supplies a second phase-locked loop.

6. Antennenradarsystem according to claim 1, characterized by

Polarizing means by which the apertures of the near-field antenna path (310- 365) and the far-field antenna path (210-305) can be cross-polarized.

7. Antenna radar system according to one of the preceding claims, characterized in that in the transmission path of the Nahbereichsantennenpfads (310 - 365) and the

Remote area antenna paths (210-305) are arranged, by means of which a temporal multiplex between the operation of the short-range antenna path (310 - 365) and the operation of the remote area antenna path (210 - 305) is realized.

8. Antennenradarsy system according to one of the preceding claims, characterized by a line network, by means of which the Nahbereichsantennenpfad (310 - 365) and the Fernbereichsantennenpfad (210 - 305) are operated synchronously.

9. Antenna radar system according to one of the preceding claims, characterized by a changeover switch, preferably a multiplexer, by means of which the Nahbereichsantennenpfad (310 - 365) and the Fernbereichsantennenpfad (210 - 305) are operated alternately.

10. A method of operating an antenna radar system with a

A short range antenna path (310-365) and a far end antenna path (210-305) separated from the short range antenna path (310-365) according to any one of the preceding claims, characterized in that the short range antenna path (310-365) is operated by heterodyne mixing.

11. The method according to claim 10, characterized in that the heterodyne

Mixture of the Nahantenantennenpfads (310 - 365) by means of a push-pull mixer (345 - 360) takes place.

12. The method according to claim 10 or 11, characterized in that the

Apertures of the Nahantenantennenpfads (310 - 365) and the Fernbereichsantennenpfads (210 - 305) are operated cross-polarized.

13. The method according to claim 10, characterized in that the transmission path of the Nahbereichsantennenpfads (310 - 365) and the Fernbereichsantennenpfads (210 - 305) are operated with a time multiplex.