DE3502398C2 - Method and device for measuring noise and interference signals at a radar receiver - Google Patents

Description

The invention is based on a method and a device for measuring noise and interference signals of a coherence Pulse radar receiver according to the preamble of the claim 1 or claim 4.

The basics of Doppler radar systems for the visual representation of moving targets, for example aircraft, are described in detail in the relevant literature. Can be referred z. B. on the textbook by MH SKOLNIK "Radar Handbook", 1970, McGraw Hill. The Doppler radar systems for the visual display of moving targets must work satisfactorily even in the event of interference from unwanted radar signals. Undesirable radar signals include, in particular, the signals reflected on the floor and on buildings, echo signals from ground vehicles, signals reflected on precipitation, the so-called "angel" signals and the like. These unwanted radar signals correspond to objects that move at low or moderate speed, ie at a speed of less than 30 ms ^-1 . Radar systems of this type must also remain operational in such a hostile environment where a possible attacker uses passive countermeasures, for example dowels, or electronic countermeasures by emitting continuous or time-chopped electromagnetic interference signals. Radar systems may also experience electromagnetic interference, such as interference caused by other devices operating nearby.

In the processing units for the output signals of the Radar receiver is the radial radar range in distance fields divided, whose unit width essentially the range resolution for that from the antenna radiated signals corresponds. The interference ratio in one given direction of the antenna can be as a ratio of Number of disturbed distance fields to the number of considered Distance fields can be defined. This interference ratio strives in the case of continuous interference signals high level, i.e. H. in the case of a medium thermal Noise level exceeding noise level, towards one.

The device for detecting movable echoes on the output side of the receiver. It is then necessary to determine the mean of the Noise signal level of the thermal generated by the receiver Knowing noise exactly because of the size of this parameter the sensitivity limit of the detection device for moving Sets goals. Exact knowledge of the mean the noise signal level also allows control of System false alarm rate. So the problem arises that these noise signals of thermal origin are present of disturbances that have to be measured on the one hand by unwanted radar signals and on the other hand by possible Interference signals are caused.  

A pulse Doppler radar device is included in EP 0 037 075 Az a means of fixed-sign suppression and post-integration containing evaluation circuit described in which for Error monitoring and correction a constant Threshold relating to false alarm is used, which depends on the average noise level. To generate a circuit adapted to the noise are those of a Post-integration circuit supplied noise signals one Series connection supplied that a changeover switch, an averager, a multiplier and a memory contains. The noise signals affect a first input of the switch. The output of the memory is one second input of the switch returned. The store is followed by a comparator through which the memory content with the noise signals supplied to the series connection is compared. The switch on the input side is switched over a gate controlled by the comparator. After education an initial mean or an initial Threshold by multiplying the mean with an appropriate factor will be obtained new ones Average and threshold values are calculated. When calculating this the incoming noise signals are further values compared with the previously calculated threshold. Becomes this threshold value through a respective incoming noise signal is exceeded, the averager is over the Switch instead of the relevant noise signal in the output-side memory containing previous threshold fed. In the same way, further funds and threshold values determined. Through this iterative middle and thresholding is intended to influence target echoes and pulsed interference signals are reduced.

The invention has for its object the method and to further develop the device of the type mentioned at the beginning, that apart from that guaranteed in the absence of interference signals accurate measurement of the level of the thermal noise signals it is also ensured in a simple manner that in If there are any interference signals, their presence  signaled perfectly and the respective fault situation reliable can be specified.

With the method according to the invention, this object is achieved by
that the determined sequence n of M noise samples A _{m is provided} at the output of a central element of the Doppler filter bank,
that the count value L of a counter is increased by one in each case if a noise sample value A _{m is} greater than the threshold value K1 · μ _n-1 ,
that the counter value L representing the exceedances at m = M-1 is compared with a predetermined reference value L₀ which depends on an admissible interference ratio,
that the mean value μ _n for the noise samples A _{m of} the current sequence n is set to the calculated value μ _c and output when the count value L is less than the reference value L₀, and
that when the count value L is larger than the reference value L₀, the calculated value μ _{c is} replaced by a predetermined value μ₀, this predetermined value μ₀ is output and, in addition, a signaling data quantity is output which indicates that the count value L is larger than the reference value L₀ and thus a large number of samples A _{m is} due to an interference signal.

On the basis of this design, characteristic features of the interference signals which provide information about the respective interference situation are always determined using the simplest means. By replacing the calculated mean value μ _c in the event that the threshold value exceeds the reference value L₀ by the predetermined value μ₀ and output this complementary predetermined value, the complementary value also immediately indicates that a large number of sample values A _m is due to an interference signal.

The one according to the invention which is suitable for carrying out this method Device is characterized in that it has a comprises first and a second measuring channel, which together are coupled and have a common input, the connected to a central element of the Doppler filter bank is that the first measurement channel is the means for calculation of the mean of the noise signals contains that the second Measuring channel includes means for measuring an interference ratio, and that means are provided for the calculated average to be replaced by a predetermined value if that Interference ratio has a value that is greater than one predetermined reference value.

The first measuring channel thus essentially contains means for Calculate the mean value of the noise signal level during the second measuring channel means for measuring the interference ratio contains. In this work according to the above-mentioned procedure The Doppler filter is preferably the measuring device centered on a frequency that is substantially equal to that Is half the repetition rate of the radar system.

Advantageous developments of the invention are Subject of the subclaims.

Embodiments of the invention result from the following description and from the figures of the drawing to which reference is made. In the drawing shows

. Figure 1 shows the plane detection area of a coherent pulse radar system;

Fig. 2 is a matrix of distance fields of this radar system;

Fig. 3 is a matrix of Doppler-distance / speed fields at the output of a doppler filter bank;

Fig. 4 is a block diagram of a radar receiver containing a Doppler filter bank;

Figure 5 shows a typical amplitude-frequency response curve of the central element of the Doppler filter bank.

Fig. 6 is a block diagram of an embodiment of a device for measuring the noise and interference signals according to the invention;

Fig. 7 is a timing diagram of the sequence control signals for the measuring device;

Fig. 8 is an illustration to illustrate the operation of the measuring device;

Fig. 9 is an alternative embodiment of the measuring device;

Fig. 10 shows a further embodiment of the measuring device; and

Fig. 11 is a flowchart which illustrates the different applied in the present process operations.

The planar detection zone of a radar system shown in Fig. 1 has a maximum detection distance which is set to a value R _max such that R _{max is} equal to Tr · c / 2 or less than this value, where Tr is the repetition period of those radiated from the antenna Pulse signals and c is the propagation speed of the electromagnetic waves. This distance R _max is divided into M distance fields or windows of the elementary duration R, which is equal to δc / 2, where δ is the duration corresponding to the range resolution of the emitted signals. In a coherence radar system, the interval of coherent processing (CIP as an abbreviation for coherent interval processing) contains a number P of successively transmitted pulses with the same repetition period and the same wavelength. The number n of CIT in the flat detection zone is given by the following relationship:

where Fr = 1 / Tr
Ta the rotation period of the antenna
and M is the number of range fields, which in turn is equal to Rmax / ΔR.

The angular movement R of the antenna during an interval coherent processing (CIT) is through the following relationship given:

and this angular movement R is generally smaller than the nominal width R _{o of} the antenna bundle. Depending on the radar system under consideration, it can emit the same pulses continuously or emit bursts of pulses of a number P of pulses, the repetition frequency of these pulses changing from one burst to the next, or the repetition frequency is maintained and the frequency Fo of the carrier oscillation from one burst to the next changed to at least partially avoid the occurrence of undetected speeds.

Figure 2 shows a matrix of range fields for an interval of coherent processing (CIT) to P pulses. Along the axis τ, which corresponds to the radar time axis, M columns of range windows are arranged, which are denoted by R to (M-1), and along the t time axis, P rows of range windows are arranged, which are R to (P- 1) are designated. An elementary distance window is denoted by the term a _{m · p} , where a is a complex quantity that is determined by its signal components I and Q. If the radar system works with coded bursts, i.e. if the product of the repetition frequency and the wavelength of the carrier oscillation changes from one burst to the next, there is a certain time gap between two successive bursts n and (n + 1).

Fig. 3 shows the matrix of the speed window for a salvo of the ordinal number n. The distance windows are only shown in this figure that the repetition frequency Fr centered on the half of the value. The central Doppler filter FDc corresponds to these distance windows, in which the value of c is equal to N / 2 or approximately equal to this value if the Doppler filter bank comprises N elements. The bandwidth of the Doppler filter is limited by the number P of pulses that are taken into account in a CIP. By appropriate selection of the parameters of the radar system, in particular repetition frequency Fr, carrier frequency Fo of the emitted pulses and number P of pulses of a CIP, the output signal of the filter FDc can be freed from any undesired radar signal, and only the echo signals from moving targets with the Doppler frequency shift Fr / 2 and odd multiples of the same can pass through the filter so that the magnitude of these echo signals from moving targets, when this Doppler filter is viewed over M distance windows, is negligible to the noise level of the receiver and, moreover, is reduced by the measuring method described below becomes.

Fig. 4 is a block diagram of an embodiment of the means for generating a sequence of M noise samples centered on half the value of the repetition frequency of the radar system. The output signal s (t) of the intermediate frequency amplifier of the radar receiver is applied to the input of a coherence detector 1 . The output of this coherence detector is connected to an analog / digital encoder 2 which supplies the signal components I and Q of each of the signals am, p which are present in a corresponding distance field. In the course of the successive bursts of pulses, the coded signals supplied by the analog / digital encoder are alternately stored in two memories 3 a and 3 b, which can be addressed for writing and reading, via an electronic switch SW ₁ . The memories 3 a and 3 b, which can be designed as RAM memory (random access memory or working memory), each have a capacity of 2MP words to K bits, where K is the number of bits that are derived from the analog / digital - Encoder 2 are supplied. The outputs of these memories 3 a and 3 b are connected to the inputs of an electronic switch SW2, which operates in the opposite phase to the input switch SW1.

The output of the electronic switch SW2 is connected to the inputs of a Doppler filter bank 4, 5 , which contains a number N of adjacent elements, only a central element of which is shown in FIG. 4. These filters are complex filters and their outputs are connected to an operator 5 which enables the calculation of the amount for a respective sample Am of the noise component that has passed through the Doppler filter.

Figure 5 is a typical normalized amplitude / frequency response curve of a Doppler filter centered at half the repetition frequency Fr of the radar system in the event that the number P of pulses in a salvo is limited to 8 or 10 pulses. It should be noted that the echo signals from slow targets whose Doppler frequency shift is less than 0.35 Fr are completely suppressed. In the example shown, for a value P in the order of 10 pulses / CIP, the echo signals of objects whose radial speed V _{r is given} by the following relationship:

not passed to the noise signal measuring device, on the output side of the central Doppler Filters lies.

Fig. 6 is a block diagram of an embodiment of the device for measuring the receiver noise. This measuring device comprises a first measuring channel 10 and a second measuring channel 11 . These two measuring channels 10, 11 have a common input which is connected to the output of the operator 5 of the Doppler filter bank 4, 5 , as shown in FIG. 4. This operator 5 supplies the sequences of M coded noise samples Am, which are digitally coded at K levels. The first measuring channel 10 is used to calculate the mean value μ _{c of} the M noise samples A _m , while the second channel 11 compares the noise samples A _m with a threshold value K ₁ · μ _n-1 , as will be described later.

The first measuring channel 10 comprises the following elements connected in series:

• - a first switch 12 with a first input (1) to which the input noise samples Am _{m are} applied, a second input (2) and a control input (c);
• - an operator who does the mean calculation performed and formed by a weighted with the factor M digital integrator 13 , which is provided with a clock input (c · p) and a zero reset input (RZ);
• - A memory register 14 with a loading input (LD) for the calculated mean value μ _{c of} the integrator 13 forming the result; and
• - A second switch 22 with a first input (1), which is connected to the output of the memory register 14 , a second input (2), to which a voltage signal of a predetermined value μ₀ is applied, and a control input (C); the output of this second switch 22 provides as output data size n associated with mean μ, _n of the average value μ follows _n-1, which for the preceding sequence n-1 of M noise samples A _m has been set, the current to the current sequence The mean value μ _{n is} set to μ _c or μ₀, depending on the result that is supplied by the second measuring channel 11 .

The second measuring channel 11 essentially comprises a level comparator 15 , the reference input of which receives a threshold value K ₁ · μ _n-1 . This level comparator 15 supplies a control signal at its output when a respective input noise sample value A _m exceeds the threshold value; this control signal is applied on the one hand to the control input of the first changeover switch 12 , which is contained in the first measuring channel 10 , and on the other hand to a counter circuit which contains the following elements connected in series:

• a counter 16 with a counting capacity of approximately M bits or less;
• a decoder 17 for decoding the count value L of the counter 16 to determine whether the count value L has exceeded a predetermined reference variable Lgröße in the course of a sequence of M noise samples A _m ;
• a memory register 18 in the form of a flip-flop whose task is to store the result supplied by decoder 17 at the end of a sequence of M samples A _m ; this memory register 18 supplies at its output (Q) an output data quantity E, which can assume two states and is fed to the control input (c) of the second switch 15 , which is contained in the first measuring channel 10 . The state of the output data variable E allows the presence of interference signals to be displayed if a count value L representative of the exceedances has been determined for the M input values A _m that is greater than the reference value L₀.

The sequence control of the measuring device is ensured by the output signals of a clock signal generator 20 , the operation of which takes place synchronously with the clock signal generator of the transmitter of the radar system.

Fig. 7 is a timing chart of the clock signals supplied from the clock signal generator 20. The sequence or sequence with the ordinal number n of the M noise samples A₀ to A _M-1 is shown in relation to the clock signals CK1, CK2 and CK3. The clock signal CK1 enables the integrator 13 to successively sum up a current noise sample Am to the accumulated sum of previous m samples. The clock signal CK2 controls the loading of the result of the integrator 13 into the memory register 14 and the setting of the memory register 18 formed by a flip-flop. The clock signal CK3 controls the zero reset of the integrator 13 and the counter 16 before the arrival of the M noise samples of the subsequent sequence n + 1.

Fig. 8 illustrates the operation of the measuring device shown in Fig. 6. The M noise samples A₀ to A _{M + 1} are quantified quantities, the mean value of which is μ _c . If a sample value A _m exceeds the threshold value K1 · μ _n-1 , this sample value A _{m is} replaced by the value μ _n-1 , which was obtained during the preceding sequence n-1. The predetermined value μ₀ considerably exceeds the threshold value K1 · μ _n-1 , which is generated by the device. In this Fig. 8, the count value L of the counter 16 is shown in the event that this count value L is smaller than the reference value L₀. In the event that the count value L exceeds the reference value L₀, the threshold value K1 · μ _{n-1 is} replaced by the threshold value K1 · μ₀, whereby - if the level of the interference and interference signals decreases - the mean value of the thermal receiver noise is quickly found again can be.

According to a preferred embodiment, a linear / logarithmic encoder 21 is arranged between the output of the central element of the Doppler filter 4, 5 and the common input of the measuring device, as shown in FIG. 9. It follows from this that the multiplier operator 19 shown in FIG. 6 has to be replaced by an adding operator and the size K1 has to be adapted accordingly.

On the other hand, it can be useful to distinguish between the three interference situations that are encountered in a radar system: absence of interference sources, continuous interference sources or chopped interference. The corresponding operation is easily realized by the circuit shown in Fig. 10, or also by an equivalent circuit. The input of the decoder 17 a is connected to the corresponding outputs of the counter 16 detecting the overshoot, and this decoder 17 a supplies an output data size of 2 bits, which is stored in the memory register 18 a. The output data quantities E are defined as follows: (NB) for the absence of interference sources; (BD) in the event of the presence of time-chopped disturbances; (BC) in the event of continuous disturbances. The output data size E = (NB) and E = (BD) or E = (BC) can then be combined with one another in order to obtain the three fault situations specified above.

In the described embodiment of a device for measuring the noise signals or interference signals of a Radar receiver is the device through a specialized Processor formed; but there are also others Embodiments of such a processor possible, in particular an embodiment with a programmable Processor.

Fig. 11 is a flowchart showing the sequential operations of a radar receiver are performed in the method for measuring the noise signals and interference signals in order to calculate the value of μ _c, which represents the average of the noise signals or noise of the receiver, and the output data size E to determine which indicates the fault situation. At the end of a sequence n-1 of M samples A _m one has the preceding value μ _n-1 , which represents the mean value μ _c , which was calculated during this sequence n-1, or a fixed predetermined value μ₀, which is always is greater than the calculated value μ _c . Furthermore, at the beginning of each sequence of M samples A _m, the results of the averaging operations for the noise signal levels and the counting of the exceedances of the interference signal levels were reset to zero. During the subsequent sequence with the atomic number n, a new value μ _{c is} calculated by cumulating the values of the input signals and replacing this value with the value μ _n-1 each time the respective noise sample Am is greater than the threshold value K1 Μ _n-1 . At the same time, the number of times the threshold value K1 .mu.n _{-1 is} exceeded is evaluated in order to then classify the disturbance situations that were encountered in the course of the sequence under consideration. As shown in the flowchart, the count L representing the number of overshoots can be classified according to the following three malfunction situations encountered:

• - not disturbed L ≦ ωτ L ₀ ≦ ωτ L ₁ = NB
• - chopped disturbance L ₀ ≦ ωτ L ≦ ωτ L ₁ = BD
• - continuous disturbance L ₁ ≦ ωτ L = BC

The precision achieved when measuring the Average of the thermal generated by the radar system Noise is high because in the course of each of the consecutive measurements a large number of noise samples is taken into account, those Noise samples, which are likely to or interference signal by which most likely mean of the noise to be replaced. Furthermore, the disturbances resulting from unwanted Radar signals already result before the Doppler filter eliminated, and the echoes of moving targets which have crossed this Doppler filter, affect only negligibly on the measurement result.  

The invention is applicable to radar systems using a constant repetition frequency or operated with coded pulse sequences, with coherence pulses or related impulses as well as impulses work for a short time, or with impulses of large duration Duration encoded for later compression in the receiver are.

A particular application of the invention is radar systems, where an optimal detection sensitivity for moving Goals are striven for.

Claims (6)

1. Method for measuring noise and interference signals of a coherence pulse radar receiver with an associated Doppler filter bank, in which
a sequence n of M noise samples A _{m is} determined,
comparing the noise samples A _m with a threshold value K1 · μ _n-1, n-1 is proportional to a noise samples for the On a preceding sequence set mean μ _n-1,
the respective noise sample value A _{m is} maintained if it is smaller than the threshold value K1 .mu.n _-1 ,
the respective noise sample A _m by n-1 set for the noise samples of the preceding sequence of mean value μ _n-1 is replaced as the threshold value K1 when it is larger · μ _n-1,
the noise sample A _m in question is added to the sum obtained since the beginning of the sequence n: the subsequent sample value A _{m + 1 is} entered if the ordinal number m is less than M, and a value if m = M-1 is calculated,
characterized by

• that the determined sequence n of M noise samples A _{m is provided} at the output of a central element of the Doppler filter bank,
• that the count value L of a counter is increased by one in each case if a noise sample value A _{m is} greater than the threshold value K1 .mu.n _-1 ,
• - that the counter value L representing the exceedances at m = M-1 is compared with a predetermined reference value L₀ which depends on a permissible interference ratio,
• - That the mean value μ _n for the noise samples A _{m of} the current sequence n is set to the calculated value μ _c and output when the count value L is less than the reference value L₀, and
• - that if the count value L is greater than the reference value L₀, the calculated value μ _{c is} replaced by a predetermined value μ₀, this predetermined value μ₀ is output and, in addition, a signaling data quantity is output which indicates that the count value L is greater than the reference value L₀ and thus a large number of samples A _{m is} due to an interference signal.

2. The method according to claim 1, characterized in that the amplitudes of the noise samples provided at the output of a central element of the Doppler filter bank A _m are encoded in logarithmic form. 3. The method according to claim 1 or 2, characterized in that the count value L representing the number of overshoots is classified with respect to several predetermined reference values L₀ to L _n . 4. Device for measuring noise and interference signals of a coherence pulse radar receiver, to which a Doppler filter bank is assigned, with means for calculating the mean value of the noise signals, characterized in that it has a first and a second measuring channel ( 10, 11 ) has which are coupled to one another and have a common input which is connected to a central element of the Doppler filter bank ( 4, 5 ) that the first measuring channel ( 10 ) has the means ( 13 ) for calculating the mean value (μ _c ) of the Noise signals contains that the second measuring channel ( 11 ) comprises means ( 16 ) for measuring an interference ratio, and that means ( 22 ) are provided to replace the calculated mean (μ _c ) by a predetermined value (μ₀) when the interference ratio has a value that is greater than a predetermined reference value (L₀). 5. The device according to claim 4, characterized in that it additionally has a linear / logarithmic encoder ( 21 ) between the central element of the Doppler filter bank ( 4, 5 ) and the common input of the two measuring channels ( 10, 11 ) is arranged. 6. The device according to claim 4, characterized in that the first measuring channel ( 10 ) contains the following elements connected in series: a first switch ( 12 ), an integrator ( 13 ) for weighted integration, a memory register ( 14 ) and a second Changeover switch ( 22 ); that the second measuring channel ( 11 ) contains the following elements connected in series: a level comparator ( 15 ), a digital counter ( 16 ), a decoder ( 17 ) and a memory register ( 18 ); that the outputs of the level comparator (15) and of the memory register (18) to the control inputs of the first (12) or the second switch (22) are connected so that the applied at an input from the associated storage register (14), second switch (22 ) has a further input to which a voltage signal with a predetermined value (μ₀) is applied, that the first switch ( 12 ) is acted upon at an input by the central element of the Doppler filter bank ( 4, 5 ) and that the output of the second Switch ( 22 ) is connected on the one hand to a further input of the first switch ( 12 ) and on the other hand via a multiplier operator ( 19 ) to the reference input of the level comparator ( 15 ).