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CN111146801A - Zero-sequence current suppression method for common direct-current bus double-inverter photovoltaic power generation system - Google Patents

Zero-sequence current suppression method for common direct-current bus double-inverter photovoltaic power generation system Download PDF

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CN111146801A
CN111146801A CN201911240421.0A CN201911240421A CN111146801A CN 111146801 A CN111146801 A CN 111146801A CN 201911240421 A CN201911240421 A CN 201911240421A CN 111146801 A CN111146801 A CN 111146801A
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CN111146801B (en
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张兴
王宝基
洪剑峰
赵文广
曹仁贤
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Hefei University of Technology
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of AC power input into DC power output; Conversion of DC power input into AC power output
    • H02M7/42—Conversion of DC power input into AC power output without possibility of reversal
    • H02M7/44—Conversion of DC power input into AC power output without possibility of reversal by static converters
    • H02M7/48—Conversion of DC power input into AC power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/483—Converters with outputs that each can have more than two voltages levels
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Abstract

本发明公开了一种共直流母线双逆变器光伏发电系统的零序电流抑制方法。本发明针对共直流母线双逆变器光伏发电系统中的零序电流，提出采用比例谐振调节器并协同基于零序注入的120度解耦载波调制策略来抑制该零序电流。通过采用120度解耦载波调制策略可以抑制零序电流的高频分量，通过对零序电流进行基于比例谐振调节器的闭环控制，并以控制环的输出调节调制波的零序注入分量，可以抑制零序电流中由逆变器死区等非线性因素造成的低频零序电流。相比于现有基于空间矢量调制策略的零序电流抑制方法，本发明无需扇区判断、开关时间计算和查表等过程，极大地减小了运算量，更易于工程实现。

Figure 201911240421

The invention discloses a zero-sequence current suppression method for a common DC busbar double inverter photovoltaic power generation system. Aiming at the zero-sequence current in the dual-inverter photovoltaic power generation system of the common DC bus, the present invention proposes to suppress the zero-sequence current by adopting a proportional resonance regulator and cooperating with a 120-degree decoupling carrier modulation strategy based on zero-sequence injection. By adopting a 120-degree decoupling carrier modulation strategy, the high-frequency component of the zero-sequence current can be suppressed. By performing a closed-loop control based on a proportional resonant regulator on the zero-sequence current, and adjusting the zero-sequence injection component of the modulated wave with the output of the control loop, it is possible to Suppress the low-frequency zero-sequence current caused by nonlinear factors such as inverter dead zone in the zero-sequence current. Compared with the existing zero-sequence current suppression method based on the space vector modulation strategy, the present invention does not require processes such as sector judgment, switching time calculation and table lookup, which greatly reduces the amount of computation and is easier to implement in engineering.

Figure 201911240421

Description

Zero-sequence current suppression method for common direct-current bus double-inverter photovoltaic power generation system
Technical Field
The invention belongs to the field of electrical engineering and relates to a double-inverter control technology based on an open winding structure, in particular to a zero-sequence current suppression method of a common direct-current bus double-inverter photovoltaic power generation system.
Background
The double-two-level inverter topology based on the open-winding transformer structure can be equivalent to a three-level inverter, but compared with the traditional midpoint clamp type three-level inverter, the inverter has the advantages of high direct-current voltage utilization rate, strong redundancy, no need of a clamping diode, no need of midpoint balance control and the like, and has a certain application prospect in the field of photovoltaic power generation. However, when the double-inverter topology shares the dc bus, because both the ac side and the dc side have direct electrical connection, a zero sequence loop exists in the system, and zero sequence current is generated under the excitation of zero sequence voltage in the system, which causes the current quality to be poor and the system loss to be increased.
In order to suppress the zero-sequence current, experts and scholars at home and abroad propose methods such as an article entitled "a space-vector modulation scheme for a dual zero-level inverter fed open-end winding indication motor drive for the excitation of zero-sequence currents", Somasekhar V T, Gopakumar K, shivatumar E g, EPE Journal, 2002,12(2):1-19. ("a zero-sequence current suppression spatial vector modulation strategy applied to dual inverter driving of a winding induction motor", "european power electronics and drive Journal, volume 12, volume 2, 2002, 1-19) proposes that only a vector with zero-sequence voltage is selected to synthesize a reference voltage, which can simultaneously control high and low-frequency zero-sequence currents generated by modulation, but when considering non-linear voltage reduction factors such as non-linear dead zone, low-sequence tube voltage reduction, and the like, the low-frequency zero-sequence current of the system is still large;
an article entitled "Effect of Zero-Vector Placement in a Dual-Inverter Fed Open-EndWindingInducation-Motor Drive With a Decoupplied Space-Vector PWM Stratage", Veermraju T.Somasekhar, Srirama Srinivas, and Kommu Kranti Kumar, IEEETransactions on Industrial Electronics 2008, 55(6):2497 2505. ("the Effect of Zero Vector position when Open-winding Induction Motor Dual Inverter Drive employs Decoupled spatial voltage Vector modulation", "IEEE Industrial Electronics society, 2008, Vol.55 vol.6, pp.2050) proposes that the Zero Vector position is adjusted so that the average of the Zero sequence voltage in one sampling period is Zero, the Zero sequence current generated by the modulation can be suppressed, but the low frequency non-linear voltage reduction dead zone is not taken into account, and the like factors of the low frequency voltage reduction, the low-frequency zero-sequence current of the system is still large;
the article entitled "Dual-Space Vector Control of Open-End Winding Permanent magnet Motor Drive by Dual Inverter", An, quintao, Jin Liu, Zhuang Peng, and Lizhi Sun, "IEEE Transactions on Power Electronics, 2016, 31(12):8329-8342. (" Dual Space Vector Control of a Dual Inverter-driven Open Winding Permanent magnet synchronous Motor "," IEEE Power Electronics journal, volume 31 volume 12, 8329-8342 page) proposes a closed-loop Control of low-frequency zero-sequence currents by means of PI regulators and Space Vector modulation with zero Vector position reset, which can suppress low-frequency zero-sequence currents caused by non-linear factors such as dead zones, tube voltage drop, etc., but the high-frequency zero-sequence current is mainly suppressed by the inductance in the high-frequency system, and the photovoltaic system adopts a large-frequency filtering scheme, in addition, the scheme needs complex sector judgment, switching time calculation, zero vector distribution factor calculation, table lookup and the like, and the calculation amount is large;
the problem is that the zero sequence current suppression technology of the common direct current bus open winding permanent magnet synchronous motor based on the proportional resonance control, the article of the constant force, year honing, Zhouyije, the report of the electrotechnical Commission, No. 31, No. 22, 35-44 pages in 2016 proposes that the zero sequence current is suppressed by adopting a proportional resonance regulator and 180-degree decoupling sine pulse width modulation, the calculation amount of the scheme is small, but the high-frequency zero sequence current is still suppressed through the inductance in the system, and the method is not suitable for a photovoltaic grid-connected system with smaller zero sequence inductance;
an article entitled "Analysis and suppression of zero sequence circulating current inverting PMSM drives with common DC bus", Zhan, Hanlin, Zi-qiang Zhu, and miijana Odavic, "IEEE Transactions on industrial Applications, 2017, 53(4), 3609-;
in summary, the prior art mainly has the following disadvantages:
1. the existing method for inhibiting zero-sequence current only by adopting a modulation strategy cannot inhibit low-frequency zero-sequence current caused by non-linear factors such as dead zones, tube voltage drop and the like;
2. the existing method for resetting the space vector modulation strategy based on the proportional-integral regulator and the zero vector position can inhibit low-frequency zero-sequence current, but does not consider the inhibition of high-frequency zero-sequence current, and the modulation strategy has larger operand;
3. the existing method based on the proportional resonant regulator and the 180-degree decoupling carrier modulation strategy can inhibit low-frequency zero-sequence current, the modulation strategy computation amount is small, but the inhibition of high-frequency zero-sequence current in the system is not considered;
4. the existing 120-degree decoupling modulation strategy method based on the self-adaptive PR regulator and the adjustable zero vector action time can simultaneously inhibit the high-frequency and low-frequency zero-sequence currents of the system, but the adopted modulation strategy has larger calculation amount;
disclosure of Invention
The invention aims to solve the problem of zero sequence current of a common direct current bus double-inverter photovoltaic power generation system, and provides a zero sequence current suppression method based on a proportional resonant regulator and zero sequence injection 120-degree decoupling carrier modulation strategy.
In order to realize the purpose of the invention, the adopted technical scheme is as follows:
a zero sequence current suppression method for a common DC bus double-inverter photovoltaic power generation system comprises a photovoltaic array PV and a DC capacitor CdcThe three-phase power supply system comprises a first three-phase two-level voltage source inverter INV1, a second three-phase two-level voltage source inverter INV2, a three-phase filter inductor L, a three-phase filter capacitor C and a three-phase open winding transformer T; the photovoltaic array PV and the direct current capacitor CdcParallel connection; the first three-phase two-level voltage source inverter INV1 is connected in parallel with the DC side of the second three-phase two-level voltage source inverter INV2 and is connected with the DC capacitor CdcAre connected together; the primary three-phase winding of the three-phase open winding transformer T is in an open state, and the A-phase winding has two terminals which are respectively marked as a terminal A1And terminal A2The winding of phase B has two terminals, respectively denoted as terminal B1And terminal B2The C-phase winding has two terminals, respectively denoted as terminal C1And terminal C2Setting terminal A1Terminal B1And terminal C1The three terminals are arranged on the same side of the primary winding of the three-phase open winding transformer T and are used as input terminals of the primary winding of the three-phase open winding transformer T, and the terminal A2Terminal B2And terminal C2The three terminals are used as output terminals of the primary winding of the three-phase open winding transformer T on the other side of the primary winding of the three-phase open winding transformer T; the three-phase filter inductor L is provided with 6 terminals, two terminals of each phase are respectively marked as a terminal A3And terminal A4And the two terminals of phase B are respectively denoted as terminal B3And terminal B4And the two terminals of the C phase are respectively marked as terminals C3And terminal C4Setting terminal A3Terminal B3And terminal C3At the same side of the three-phase filter inductor L and using the three terminals as the input terminal of the three-phase filter inductor L, the terminal A4Terminal B4And terminal C4The three terminals are used as output terminals of the three-phase filter inductor L on the other side of the three-phase filter inductor L; the three-phase filter capacitor C has 6 terminals, two terminals per phase, and two terminals AIs marked as terminal A5And terminal A6And the two terminals of phase B are respectively denoted as terminal B5And terminal B6And the two terminals of the C phase are respectively marked as terminals C5And terminal C6Setting terminal A5Terminal B5And terminal C5At the same side of the three-phase filter capacitor C and using the three terminals as the input terminals of the three-phase filter capacitor C, the terminal A6Terminal B6And terminal C6The three terminals are used as output terminals of the three-phase filter capacitor C on the other side of the three-phase filter capacitor C; terminal A of the three-phase filter inductor L3Terminal B3And terminal C3Is connected to the AC output side of the first three-phase two-level voltage source inverter INV1, terminal A4Terminal B4And terminal C4Respectively connected with terminals A of primary windings of three-phase open winding transformer T1Terminal B1And terminal C1And terminal A of three-phase filter capacitor C5Terminal B5And terminal C5Connecting; terminal A of primary winding of T-shaped three-phase open winding transformer2Terminal B2Terminal C2Respectively connected with terminals A of three-phase filter capacitor C6Terminal B6And terminal C6After being connected, the three-phase two-level voltage source inverter is connected to the alternating current output side of the second three-phase two-level voltage source inverter INV 2; the secondary side of the three-phase open winding transformer T is connected into a power grid E in a star connection or a triangular connection mode;
the zero sequence current suppression method comprises the following steps:
step 1, collecting direct-current side voltage v of a double-inverter photovoltaic power generation systemdcD.c. side current idcCollecting and recording the voltage of the three-phase filter capacitor C as the voltage v of the three-phase filter capacitorca、vcb、vccCollecting the current of the input end of the three-phase filter inductor L and recording the current as the bridge arm side inductor current ia、ib、ic；
Step 2, according to the voltage v on the direct current side obtained in the step 1dcAnd a direct side current idcObtaining the direct current side voltage of the double-inverter photovoltaic power generation system after maximum power point trackingInstruction vdc_ref(ii) a Then obtaining an active current instruction i of the double-inverter photovoltaic power generation system through a direct-current side voltage closed-loop control equationd_ref；
The direct current side voltage closed-loop control equation is as follows:
Figure BDA0002306067800000041
in the formula, Kp_vdcIs the proportionality coefficient, K, of a voltage-loop PI regulator on the DC sidei_vdcThe integral coefficient of the direct-current side voltage loop PI regulator is shown, and s is a Laplace operator;
step 3, obtaining the three-phase filter capacitor voltage v according to the step 1ca、vcb、vccObtaining the phase angle theta of the three-phase filter capacitor voltage and the dq component v of the three-phase filter capacitor voltage through a phase-locked loopcd、vcq；
The calculation equation of the voltage phase angle theta of the three-phase filter capacitor is as follows:
Figure BDA0002306067800000042
Figure BDA0002306067800000043
wherein θ ' is the phase angle v ' of the three-phase filter capacitor voltage obtained in the previous control period 'cqThe q component, omega, obtained by performing synchronous rotation coordinate transformation on the phase angle theta' of the three-phase filter capacitor voltage calculated according to the previous control period0Rated angular frequency, K, of three-phase filter capacitor voltagep_PLLIs the proportionality coefficient, K, of a phase-locked loop PI regulatori_PLLThe integral coefficient of the phase-locked loop PI regulator is obtained;
dq component v of the three-phase filter capacitor voltagecd、vcqThe calculation equation of (a) is:
Figure BDA0002306067800000051
step 4, according to the bridge arm side inductive current i obtained in the step 1a、ib、icAnd step 3, obtaining the dq component i of the bridge arm side inductive current through a single synchronous rotation coordinate transformation equation according to the phase angle theta of the three-phase filter capacitor voltage obtained in the stepd、iqAnd zero sequence current i0；
The transformation equation of the single synchronous rotation coordinate is as follows:
Figure BDA0002306067800000052
step 5, firstly setting a reactive current instruction iq_refThen according to the active current command i obtained in the step 2d_refAnd 3, obtaining dq component v of the three-phase filter capacitor voltage in the step 3cd、vcqAnd dq component i of the bridge arm side inductive current obtained in step 4d、iqAnd obtaining dq component v of the master control signal of the double-inverter photovoltaic power generation system through a current closed-loop control equationd、vq；
The current closed-loop control equation is as follows:
Figure BDA0002306067800000053
in the formula, Kp_iIs the proportionality coefficient, K, of a current loop PI regulatori_iThe integral coefficient of the current loop PI regulator is shown, omega is the fundamental angular frequency, and L is the filter inductance value;
step 6, according to the dq component v of the total control signal of the double-inverter photovoltaic power generation system obtained in the step 5d、vqMultiply by 2/v, respectivelydcObtaining per unit dq component m of the master control signal of the double-inverter photovoltaic power generation systemd、mqObtaining a per-unit dq component m of the control signal of the first three-phase two-level voltage source inverter INV1 through a control signal decoupling equationd1、mq1And a per-unit dq component m of a control signal of the second three-phase two-level voltage source inverter INV2d2、mq2；
Per unit dq component m of total control signal of the double-inverter photovoltaic power generation systemd、mqThe calculation formula of (2) is as follows:
Figure BDA0002306067800000061
the control signal decoupling equation is as follows:
Figure BDA0002306067800000062
Figure BDA0002306067800000063
step 7, obtaining a per-unit dq component m of the control signal of the first three-phase two-level voltage source inverter INV1 according to step 6d1、mq1And a per-unit dq component m of a control signal of the second three-phase two-level voltage source inverter INV2d2、mq2And step 3, obtaining a per-unit three-phase control component m of the control signal of the first three-phase two-level voltage source inverter INV1 through a single synchronous rotation coordinate inverse transformation equation according to the phase angle theta of the three-phase filter capacitor voltage obtained in the step 3a1、mb1、mc1And a unified three-phase control component m of a control signal of the second three-phase two-level voltage source inverter INV2a2、mb2、mc2；
The single synchronous rotation coordinate inverse transformation equation is as follows:
Figure BDA0002306067800000064
Figure BDA0002306067800000071
step 8, firstly setting a reference instruction i of the zero sequence current0_refThen according to the zero sequence current i obtained in the step 40Zero sequence current closed loop control equationObtaining a zero sequence adjusting signal delta k;
the zero-sequence current closed-loop control equation is as follows:
Figure BDA0002306067800000072
in the formula, Kp_0Is the proportionality coefficient, K, of a zero-sequence current loop proportional resonant regulatorr_0Is the resonance coefficient, omega, of a zero-sequence current loop proportional resonant regulatorcIs the bandwidth, omega, of a proportional resonant regulatorrIs the resonant frequency of the proportional resonant regulator;
step 9, obtaining a per-unit three-phase control component m according to the control signal of the first three-phase two-level voltage source inverter INV1 obtained in step 7a1、mb1、mc1Taking the maximum value and the minimum value of the three values and recording the maximum value and the minimum value as mmax1And mmin1(ii) a A unit three-phase control component m according to the control signal of the second three-phase two-level voltage source inverter INV2 obtained in step 7a2、mb2、mc2Taking the maximum value and the minimum value of the three values and recording the maximum value and the minimum value as mmax2And mmin2(ii) a Then, according to the zero sequence adjusting signal Δ k obtained in step 8, a zero sequence injection component calculation equation is performed to obtain a zero sequence injection component m of the first three-phase two-level voltage source inverter INV1z1And a zero-sequence injection component m of a second three-phase two-level voltage source inverter INV2z2；
The zero-sequence injection component calculation equation is respectively as follows:
mz1＝2Δk-(0.5+Δk)mmax1-(0.5-Δk)mmin1
mz2＝-2Δk-(0.5-Δk)mmax2-(0.5+Δk)mmin2
step 10, injecting a zero sequence component m of the first three-phase two-level voltage source inverter INV1 obtained in step 9z1Respectively compared with the unit three-phase control component m of the control signal of the first three-phase two-level voltage source inverter INV1 obtained in step 7a1、mb1、mc1Add to obtain the first threeModulated wave signal of two-phase voltage source inverter INV1
Figure BDA0002306067800000073
Figure BDA0002306067800000081
Injecting the zero sequence of the second three-phase two-level voltage source inverter INV2 obtained in the step 9 into the component mz2Respectively compared with the unit three-phase control component m of the control signal of the second three-phase two-level voltage source inverter INV2 obtained in step 7a2、mb2、mc2Adding to obtain modulated wave signal of second three-phase two-level voltage source inverter INV2
Figure BDA0002306067800000082
Figure BDA0002306067800000083
Then PWM control signals PWM1 and PWM2 for driving switching tubes of the first three-phase two-level voltage source inverter INV1 and the second three-phase two-level voltage source inverter INV2 are generated through comparison with the triangular carrier waves respectively;
the modulated wave signal
Figure BDA0002306067800000084
And modulating the wave signal
Figure BDA0002306067800000085
The calculation equations of (a) are:
Figure BDA0002306067800000086
Figure BDA0002306067800000087
compared with the prior art, the invention has the beneficial effects that:
1. the method can inhibit high-frequency and low-frequency zero-sequence currents in the system at the same time, and is particularly suitable for the application field with smaller zero-sequence inductance in the system, such as the photovoltaic grid-connected field;
2. compared with the space vector modulation strategy in the prior art, the adopted modulation strategy only needs to calculate the zero sequence injection component, and adopts carrier modulation, so that the processes of sector judgment, switching time calculation, zero vector allocation time calculation, table look-up and the like are not needed, the operation amount is greatly reduced, and the realization is easier.
Drawings
Fig. 1 is a main circuit block diagram of a common dc bus double-inverter photovoltaic power generation system in an embodiment of the invention.
Fig. 2 is a control block diagram of the zero sequence current suppression method in the embodiment of the present invention.
Fig. 3 is a block diagram of a zero sequence injection component calculation link of two inverters in the zero sequence current suppression method according to the embodiment of the present invention.
Fig. 4 is a simulation waveform of the front and rear zero-sequence currents of the zero-sequence current suppression method provided by the present invention.
Fig. 5 is a simulation waveform of front and rear bridge arm currents cut into the zero sequence current suppression method provided by the present invention.
Fig. 6 is a fourier analysis diagram of the front bridge arm current cut into the zero sequence current suppression method provided by the present invention.
Fig. 7 is a fourier analysis diagram of the rear bridge arm current cut into the zero sequence current suppression method provided by the present invention.
Detailed Description
The invention is described in further detail below with reference to the figures and examples.
Fig. 1 is a main circuit block diagram of a common dc bus double-inverter photovoltaic power generation system in an embodiment of the invention. As shown in fig. 1, the common dc bus dual-inverter photovoltaic power generation system according to the present invention includes a photovoltaic array PV and a dc capacitor CdcThe three-phase power supply system comprises a first three-phase two-level voltage source inverter INV1, a second three-phase two-level voltage source inverter INV2, a three-phase filter inductor L, a three-phase filter capacitor C and a three-phase open winding transformer T.
The photovoltaic array PV and the direct current capacitor CdcAnd (4) connecting in parallel.
The above-mentionedThe first three-phase two-level voltage source inverter INV1 is connected in parallel with the DC side of the second three-phase two-level voltage source inverter INV2 and is connected with the DC capacitor CdcAnd are joined together.
The primary three-phase winding of the three-phase open winding transformer T is in an open state, and the A-phase winding has two terminals which are respectively marked as a terminal A1And terminal A2The winding of phase B has two terminals, respectively denoted as terminal B1And terminal B2The C-phase winding has two terminals, respectively denoted as terminal C1And terminal C2Setting terminal A1Terminal B1And terminal C1The three terminals are arranged on the same side of the primary winding of the three-phase open winding transformer T and are used as input terminals of the primary winding of the three-phase open winding transformer T, and the terminal A2Terminal B2And terminal C2And the three terminals are arranged on the other side of the primary winding of the three-phase open winding transformer T and are used as output terminals of the primary winding of the three-phase open winding transformer T.
The three-phase filter inductor L is provided with 6 terminals, two terminals of each phase are respectively marked as a terminal A3And terminal A4And the two terminals of phase B are respectively denoted as terminal B3And terminal B4And the two terminals of the C phase are respectively marked as terminals C3And terminal C4Setting terminal A3Terminal B3And terminal C3At the same side of the three-phase filter inductor L and using the three terminals as the input terminal of the three-phase filter inductor L, the terminal A4Terminal B4And terminal C4And the three terminals are arranged on the other side of the three-phase filter inductor L and are used as output terminals of the three-phase filter inductor L.
The three-phase filter capacitor C has 6 terminals, two terminals of each phase, and two terminals of the A phase are respectively marked as a terminal A5And terminal A6And the two terminals of phase B are respectively denoted as terminal B5And terminal B6And the two terminals of the C phase are respectively marked as terminals C5And terminal C6Setting terminal A5Terminal B5And terminal C5The three terminals are arranged on the same side of the three-phase filter capacitor C and are used as input terminals and terminals of the three-phase filter capacitor CA6Terminal B6And terminal C6And the three terminals are arranged on the other side of the three-phase filter capacitor C and are used as output terminals of the three-phase filter capacitor C.
Terminal A of the three-phase filter inductor L3Terminal B3And terminal C3Is connected to the AC output side of the first three-phase two-level voltage source inverter INV1, terminal A4Terminal B4And terminal C4Are respectively connected with primary winding terminals A of a three-phase open winding transformer T1Terminal B1And terminal C1And a three-phase filter capacitor C terminal A5Terminal B5And terminal C5Connecting; terminal A of primary winding of T-shaped three-phase open winding transformer2Terminal B2Terminal C2Respectively connected with terminals A of three-phase filter capacitor C6Terminal B6And terminal C6After being connected, the three-phase two-level voltage source inverter is connected to the alternating current output side of the second three-phase two-level voltage source inverter INV 2; and the secondary side of the three-phase open winding transformer T is connected into a power grid E in a star connection or a triangular connection mode.
In the embodiment of the invention, the double inverters adopt a double-two-level inverter topology, the total rated power of the system is 30kW, the rated power of each inverter is 15kW, the switching frequency is 5kHz, the dead time of a switching tube is 3 mus, and the direct-current capacitor CdcThe inductance L of the three-phase filter is 0.5mF, the L of the three-phase filter inductance is 3.6mH, the C of the three-phase filter capacitance is 10 muF, the phase voltage transformation ratio of the open winding transformer T is 364V/380V, the short-circuit impedance is 3%, and the effective value of the grid line voltage is 380V/50 Hz.
Referring to fig. 2 and fig. 3, the zero sequence current suppression method of the present invention includes the following steps:
step 1, collecting direct-current side voltage v of a double-inverter photovoltaic power generation systemdcD.c. side current idcCollecting and recording the voltage of the three-phase filter capacitor C as the voltage v of the three-phase filter capacitorca、vcb、vccCollecting the current of the input end of the three-phase filter inductor L and recording the current as the bridge arm side inductor current ia、ib、ic。
Step 2, according to the product obtained in step 1Voltage v at the DC sidedcAnd a direct side current idcObtaining a direct-current side voltage instruction v of the double-inverter photovoltaic power generation system after Maximum Power Point Tracking (MPPT)dc_ref(ii) a Then obtaining an active current instruction i of the double-inverter photovoltaic power generation system through a direct-current side voltage closed-loop control equationd_ref；
The direct current side voltage closed-loop control equation is as follows:
Figure BDA0002306067800000101
in the formula, Kp_vdcIs the proportionality coefficient, K, of a voltage-loop PI regulator on the DC sidei_vdcAnd s is a Laplace operator. In this embodiment, vdc_ref＝620V，Kp_vdc＝2，Ki_vdc＝50。
Step 3, obtaining the three-phase filter capacitor voltage v according to the step 1ca、vcb、vccObtaining the phase angle theta of the three-phase filter capacitor voltage and the dq component v of the three-phase filter capacitor voltage through a phase-locked loop (PLL)cd、vcq。
The calculation equation of the voltage phase angle theta of the three-phase filter capacitor is as follows:
Figure BDA0002306067800000111
Figure BDA0002306067800000112
wherein θ ' is the phase angle v ' of the three-phase filter capacitor voltage obtained in the previous control period 'cqThe q component, omega, obtained by performing synchronous rotation coordinate transformation on the phase angle theta' of the three-phase filter capacitor voltage calculated according to the previous control period0Rated angular frequency, K, of three-phase filter capacitor voltagep_PLLIs the proportionality coefficient, K, of a phase-locked loop PI regulatori_PLLIs the integral coefficient of the phase-locked loop PI regulator. In the present embodiment, ω0＝100πrad/s，Kp_PLL＝0.1，Ki_PLL＝0.005。
Dq component v of the three-phase filter capacitor voltagecd、vcqThe calculation equation of (a) is:
Figure BDA0002306067800000113
step 4, according to the bridge arm side inductive current i obtained in the step 1a、ib、icAnd step 3, obtaining the dq component i of the bridge arm side inductive current through a single synchronous rotation coordinate transformation equation according to the phase angle theta of the three-phase filter capacitor voltage obtained in the stepd、iqAnd zero sequence current i0。
The transformation equation of the single synchronous rotation coordinate is as follows:
Figure BDA0002306067800000114
step 5, firstly setting a reactive current instruction iq_refThen according to the active current command i obtained in the step 2d_refAnd 3, obtaining dq component v of the three-phase filter capacitor voltage in the step 3cd、vcqAnd dq component i of the bridge arm side inductive current obtained in step 4d、iqAnd obtaining dq component v of the master control signal of the double-inverter photovoltaic power generation system through a current closed-loop control equationd、vq。
The current closed-loop control equation is as follows:
Figure BDA0002306067800000121
in the formula, Kp_iIs the proportionality coefficient, K, of a current loop PI regulatori_iThe integral coefficient of the current loop PI regulator is shown, omega is the angular frequency of the fundamental wave, and L is the inductance value of the filter. In this embodiment, ω ═ 100 π rad/s, Kp_i＝5，Ki_i＝200。
Step 6, according to the total control of the double-inverter photovoltaic power generation system obtained in the step 5Dq component v of the signald、vqMultiply by 2/v, respectivelydcObtaining per unit dq component m of the master control signal of the double-inverter photovoltaic power generation systemd、mqObtaining a per-unit dq component m of the control signal of the first three-phase two-level voltage source inverter INV1 through a control signal decoupling equationd1、mq1And a per-unit dq component m of a control signal of the second three-phase two-level voltage source inverter INV2d2、mq2。
Per unit dq component m of total control signal of the double-inverter photovoltaic power generation systemd、mqThe calculation formula of (2) is as follows:
Figure BDA0002306067800000122
the control signal decoupling equation is as follows:
Figure BDA0002306067800000123
Figure BDA0002306067800000124
step 7, obtaining a per-unit dq component m of the control signal of the first three-phase two-level voltage source inverter INV1 according to step 6d1、mq1And a per-unit dq component m of a control signal of the second three-phase two-level voltage source inverter INV2d2、mq2And step 3, obtaining a per-unit three-phase control component m of the control signal of the first three-phase two-level voltage source inverter INV1 through a single synchronous rotation coordinate inverse transformation equation according to the phase angle theta of the three-phase filter capacitor voltage obtained in the step 3a1、mb1、mc1And a unified three-phase control component m of a control signal of the second three-phase two-level voltage source inverter INV2a2、mb2、mc2。
The single synchronous rotation coordinate inverse transformation equation is as follows:
Figure BDA0002306067800000131
Figure BDA0002306067800000132
step 8, firstly setting a reference instruction i of the zero sequence current0_refThen according to the zero sequence current i obtained in the step 40And obtaining a zero sequence adjusting signal delta k through a zero sequence current closed-loop control equation.
The zero-sequence current closed-loop control equation is as follows:
Figure BDA0002306067800000133
in the formula, Kp_0Proportional coefficient, K, of a zero sequence current loop Proportional Resonance (PR) regulatorr_0Is the resonance coefficient, omega, of a zero sequence current loop PR regulatorcFor the bandwidth of the PR regulator, ωrIs the resonant frequency of the PR modulator. In this embodiment, i0_ref＝0，Kp_0＝0.1，Kr_0＝10，ωc＝3rad/s，ωr＝300πrad/s。
Step 9, obtaining a per-unit three-phase control component m according to the control signal of the first three-phase two-level voltage source inverter INV1 obtained in step 7a1、mb1、mc1Taking the maximum value and the minimum value of the three values and recording the maximum value and the minimum value as mmax1And mmin1(ii) a A unit three-phase control component m according to the control signal of the second three-phase two-level voltage source inverter INV2 obtained in step 7a2、mb2、mc2Taking the maximum value and the minimum value of the three values and recording the maximum value and the minimum value as mmax2And mmin2(ii) a Then, according to the zero sequence adjusting signal Δ k obtained in step 8, a zero sequence injection component calculation equation is performed to obtain a zero sequence injection component m of the first three-phase two-level voltage source inverter INV1z1And a zero-sequence injection component m of a second three-phase two-level voltage source inverter INV2z2。
The zero-sequence injection component calculation equation is respectively as follows:
mz1＝2Δk-(0.5+Δk)mmax1-(0.5-Δk)mmin1
mz2＝-2Δk-(0.5-Δk)mmax2-(0.5+Δk)mmin2
step 10, injecting a zero sequence component m of the first three-phase two-level voltage source inverter INV1 obtained in step 9z1Respectively compared with the unit three-phase control component m of the control signal of the first three-phase two-level voltage source inverter INV1 obtained in step 7a1、mb1、mc1Adding to obtain modulated wave signal of first three-phase two-level voltage source inverter INV1
Figure BDA0002306067800000141
Figure BDA0002306067800000142
Injecting the zero sequence of the second three-phase two-level voltage source inverter INV2 obtained in the step 9 into the component mz2Respectively compared with the unit three-phase control component m of the control signal of the second three-phase two-level voltage source inverter INV2 obtained in step 7a2、mb2、mc2Adding to obtain modulated wave signal of second three-phase two-level voltage source inverter INV2
Figure BDA0002306067800000143
Figure BDA0002306067800000144
Then PWM control signals PWM1 and PWM2 for driving switching tubes of the first three-phase two-level voltage source inverter INV1 and the second three-phase two-level voltage source inverter INV2 are generated through comparison with the triangular carrier waves respectively.
The modulated wave signal
Figure BDA0002306067800000145
And modulating the wave signal
Figure BDA0002306067800000146
The calculation equations of (a) are:
Figure BDA0002306067800000147
Figure BDA0002306067800000148
fig. 4 is a simulation waveform of zero sequence current before and after the zero sequence current suppression method according to specific parameters of the embodiment of the present invention is performed, and it can be found that the high frequency component of the zero sequence current is suppressed but the low frequency component is larger only by decoupling modulation of 120 degrees before the zero sequence current suppression method according to the present invention is performed, and both the high frequency component and the low frequency component of the zero sequence current are suppressed after the zero sequence current suppression method according to the present invention is performed. Fig. 5 is a simulation waveform of front and rear bridge arm currents of the zero-sequence current suppression method according to the embodiment of the present invention, fig. 6 is a fourier analysis diagram of the front bridge arm current of the zero-sequence current suppression method according to the present invention, fig. 7 is a fourier analysis diagram of the rear bridge arm current of the zero-sequence current suppression method according to the present invention, and it can be found from fig. 5, fig. 6 and fig. 7 that the first 3 harmonics of the zero-sequence current suppression method according to the present invention account for 4.75% of the bridge arm current, and the 3 harmonics of the zero-sequence current suppression method according to the present invention only account for 0.17% of the bridge arm current, which greatly improves the electric energy quality of the bridge arm current. The simulation results show the correctness and the effectiveness of the zero sequence current suppression method of the common direct current bus double-inverter photovoltaic power generation system.

Claims (1)

1.一种共直流母线双逆变器光伏发电系统的零序电流抑制方法，其中本方法所涉及的共直流母线双逆变器光伏发电系统包括光伏阵列PV、直流电容Cdc、第一三相两电平电压源型逆变器INV1、第二三相两电平电压源型逆变器INV2、三相滤波电感L、三相滤波电容C和一台三相开绕组变压器T；所述光伏阵列PV与直流电容Cdc并联；所述第一三相两电平电压源型逆变器INV1与第二三相两电平电压源型逆变器INV2直流侧并联，且与直流电容Cdc并接在一起；所述三相开绕组变压器T的原边三相绕组呈打开状态，A相绕组有两个端子，分别记为端子A1和端子A2，B相绕组有两个端子，分别记为端子B1和端子B2，C相绕组有两个端子，分别记为端子C1和端子C2，设定端子A1、端子B1和端子C1在三相开绕组变压器T原边绕组的同一侧并以该三个端子作为三相开绕组变压器T原边绕组的输入端子，端子A2、端子B2和端子C2在三相开绕组变压器T原边绕组的另一侧并以该三个端子作为三相开绕组变压器T原边绕组的输出端子；所述三相滤波电感L有6个端子，每相两个端子，A相两个端子分别记为端子A3和端子A4，B相两个端子分别记为端子B3和端子B4，C相两个端子分别记为端子C3和端子C4，设定端子A3、端子B3和端子C3在三相滤波电感L的同一侧并以该三个端子作为三相滤波电感L的输入端子，端子A4、端子B4和端子C4在三相滤波电感L的另一侧并以该三个端子作为三相滤波电感L的输出端子；所述三相滤波电容C有6个端子，每相两个端子，A相两个端子分别记为端子A5和端子A6，B相两个端子分别记为端子B5和端子B6，C相两个端子分别记为端子C5和端子C6，设定端子A5、端子B5和端子C5在三相滤波电容C的同一侧并以该三个端子作为三相滤波电容C的输入端子，端子A6、端子B6和端子C6在三相滤波电容C的另一侧并以该三个端子作为三相滤波电容C的输出端子；所述三相滤波电感L的端子A3、端子B3和端子C3连接至第一三相两电平电压源型逆变器INV1的交流输出侧，端子A4、端子B4和端子C4分别与三相开绕组变压器T原边绕组的端子A1、端子B1和端子C1以及三相滤波电容C的端子A5、端子B5和端子C5相连接；所述三相开绕组变压器T原边绕组的端子A2、端子B2、端子C2分别与三相滤波电容C的端子A6、端子B6和端子C6相连接后，再连接至第二三相两电平电压源型逆变器INV2的交流输出侧；所述三相开绕组变压器T的副边通过星形连接或三角形连接并入电网E；1. A zero-sequence current suppression method for a common DC bus double inverter photovoltaic power generation system, wherein the common DC bus double inverter photovoltaic power generation system involved in the method comprises a photovoltaic array PV, a DC capacitor C dc , a first three A phase two-level voltage source inverter INV1, a second three-phase two-level voltage source inverter INV2, a three-phase filter inductor L, a three-phase filter capacitor C and a three-phase open-winding transformer T; the The photovoltaic array PV is connected in parallel with the DC capacitor C dc ; the first three-phase two-level voltage source inverter INV1 is connected in parallel with the DC side of the second three-phase two-level voltage source inverter INV2, and is connected with the DC capacitor C dc are connected together in parallel; the primary three-phase winding of the three-phase open-winding transformer T is in an open state, the A-phase winding has two terminals, which are respectively denoted as terminal A 1 and terminal A 2 , and the B-phase winding has two terminals. , respectively denoted as terminal B 1 and terminal B 2 , the C-phase winding has two terminals, denoted as terminal C 1 and terminal C 2 respectively, set terminal A 1 , terminal B 1 and terminal C 1 in the three-phase open winding transformer The same side of the T primary winding and the three terminals are used as the input terminals of the three - phase open-winding transformer T primary winding. One side and the three terminals are used as the output terminals of the primary winding of the three-phase open-winding transformer T; the three-phase filter inductor L has 6 terminals, two terminals for each phase, and the two terminals of the A phase are respectively recorded as terminals A 3 and terminal A 4 , the two terminals of phase B are respectively recorded as terminal B 3 and terminal B 4 , the two terminals of phase C are respectively recorded as terminal C 3 and terminal C 4 , set terminal A 3 , terminal B 3 and terminal C 3 are on the same side of the three-phase filter inductor L and use the three terminals as the input terminals of the three-phase filter inductor L, and the terminal A 4 , the terminal B 4 and the terminal C 4 are on the other side of the three-phase filter inductor L and use the three terminals as the input terminals of the three-phase filter inductor L. The three terminals are used as the output terminals of the three-phase filter inductor L; the three-phase filter capacitor C has 6 terminals, two terminals for each phase, the two terminals of the A phase are respectively recorded as terminal A 5 and terminal A 6 , and the two terminals of the B phase are respectively recorded as terminal A 5 and terminal A 6 . The two terminals are respectively recorded as terminal B 5 and terminal B 6 , and the two terminals of phase C are respectively recorded as terminal C 5 and terminal C 6 . The three terminals are used as the input terminals of the three-phase filter capacitor C, and the terminals A 6 , B 6 and C 6 are on the other side of the three-phase filter capacitor C and the three terminals are used as the three-phase filter capacitor C. The output terminal of the three-phase filter inductor L; the terminal A 3 , the terminal B 3 and the terminal C 3 of the three-phase filter inductor L are connected to the AC output side of the first three-phase two-level voltage source inverter INV1, and the terminal A 4 and the terminal B 4 and terminal C 4 are respectively connected with terminal A 1 , terminal B 1 and terminal C 1 of the primary winding of the three-phase open-winding transformer T and terminal A 5 , terminal B 5 and terminal C 5 of the three-phase filter capacitor C; The three-phase open-winding transformer T After the terminal A 2 , the terminal B 2 and the terminal C 2 of the primary winding are respectively connected with the terminal A 6 , the terminal B 6 and the terminal C 6 of the three-phase filter capacitor C, they are then connected to the second three-phase two-level voltage source. The AC output side of the inverter INV2; the secondary side of the three-phase open-winding transformer T is connected to the grid E through a star connection or a delta connection; 其特征在于，所述零序电流抑制方法包括如下步骤：It is characterized in that, the zero-sequence current suppression method includes the following steps: 步骤1，采集双逆变器光伏发电系统的直流侧电压vdc、直流侧电流idc，采集三相滤波电容C的电压并记为三相滤波电容电压vca、vcb、vcc，采集三相滤波电感L输入端的电流并记为桥臂侧电感电流ia、ib、ic；Step 1, collect the DC side voltage v dc and the DC side current i dc of the dual-inverter photovoltaic power generation system, collect the voltage of the three-phase filter capacitor C and record it as the three-phase filter capacitor voltage v ca , v cb , v cc , collect The current at the input end of the three-phase filter inductor L is also recorded as the bridge arm side inductor currents i a , ib , and ic ; 步骤2，根据步骤1中得到的直流侧电压vdc和直流侧电流idc，经最大功率点跟踪后得到双逆变器光伏发电系统的直流侧电压指令vdc_ref；然后经直流侧电压闭环控制方程得到双逆变器光伏发电系统的有功电流指令id_ref；Step 2: According to the DC side voltage v dc and the DC side current i dc obtained in step 1, after the maximum power point tracking, the DC side voltage command v dc_ref of the dual-inverter photovoltaic power generation system is obtained; then the DC side voltage is closed-loop controlled. The equation obtains the active current command id_ref of the dual-inverter photovoltaic power generation system; 所述直流侧电压闭环控制方程为：The DC side voltage closed-loop control equation is:
Figure FDA0002306067790000021
Figure FDA0002306067790000021
 式中，Kp_vdc为直流侧电压环PI调节器的比例系数，Ki_vdc为直流侧电压环PI调节器的积分系数，s为拉普拉斯算子；In the formula, K p_vdc is the proportional coefficient of the DC side voltage loop PI regulator, K i_vdc is the integral coefficient of the DC side voltage loop PI regulator, and s is the Laplace operator; 步骤3，根据步骤1中得到的三相滤波电容电压vca、vcb、vcc，经锁相环得到三相滤波电容电压的相角θ和三相滤波电容电压的dq分量vcd、vcq；Step 3: According to the three-phase filter capacitor voltages v ca , v cb , and v cc obtained in step 1, the phase angle θ of the three-phase filter capacitor voltage and the dq components v cd , v of the three-phase filter capacitor voltage are obtained through the phase-locked loop. cq ; 所述三相滤波电容电压相角θ的计算方程为：The calculation equation of the three-phase filter capacitor voltage phase angle θ is:
Figure FDA0002306067790000022
Figure FDA0002306067790000022
Figure FDA0002306067790000023
Figure FDA0002306067790000023
 式中，θ′为上一控制周期得到的三相滤波电容电压的相角，v′cq为三相滤波电容电压根据上一控制周期计算得到的三相滤波电容电压的相角θ′做同步旋转坐标变换得到的q分量，ω0为三相滤波电容电压的额定角频率，Kp_PLL为锁相环PI调节器的比例系数，Ki_PLL为锁相环PI调节器的积分系数；In the formula, θ' is the phase angle of the three-phase filter capacitor voltage obtained in the previous control cycle, v' cq is the phase angle θ' of the three-phase filter capacitor voltage calculated according to the previous control cycle. The q component obtained by the rotational coordinate transformation, ω 0 is the rated angular frequency of the three-phase filter capacitor voltage, K p_PLL is the proportional coefficient of the phase-locked loop PI regulator, and K i_PLL is the integral coefficient of the phase-locked loop PI regulator; 所述三相滤波电容电压的dq分量vcd、vcq的计算方程为：The calculation equations of the dq components v cd and v cq of the three-phase filter capacitor voltage are:
Figure FDA0002306067790000031
Figure FDA0002306067790000031
 步骤4，根据步骤1中得到的桥臂侧电感电流ia、ib、ic和步骤3中得到的三相滤波电容电压的相角θ，经单同步旋转坐标变换方程得到桥臂侧电感电流的dq分量id、iq和零序电流i0；Step 4: According to the bridge arm side inductance currents i a , ib , ic obtained in step 1 and the phase angle θ of the three-phase filter capacitor voltage obtained in step 3 , the bridge arm side inductance is obtained through the single synchronous rotation coordinate transformation equation. the dq components of the current id , i q and the zero sequence current i 0 ; 所述单同步旋转坐标变换方程为：The single synchronous rotation coordinate transformation equation is:
Figure FDA0002306067790000032
Figure FDA0002306067790000032
 步骤5，先设定无功电流指令iq_ref，然后根据步骤2中得到的有功电流指令id_ref、步骤3中得到的三相滤波电容电压的dq分量vcd、vcq和步骤4中得到的桥臂侧电感电流的dq分量id、iq，通过电流闭环控制方程得到双逆变器光伏发电系统的总控制信号的dq分量vd、vq；Step 5, first set the reactive current command i q_ref , and then according to the active current command i d_ref obtained in step 2, the dq components v cd , v cq of the three-phase filter capacitor voltage obtained in step 3 and the obtained in step 4 The dq components id and i q of the inductor current on the bridge arm side are obtained through the current closed-loop control equation to obtain the dq components v d and v q of the total control signal of the dual-inverter photovoltaic power generation system; 所述电流闭环控制方程为:The current closed-loop control equation is:
Figure FDA0002306067790000033
Figure FDA0002306067790000033
 式中，Kp_i为电流环PI调节器的比例系数，Ki_i为电流环PI调节器的积分系数，ω为基波角频率，L为滤波电感值；In the formula, K p_i is the proportional coefficient of the current loop PI regulator, K i_i is the integral coefficient of the current loop PI regulator, ω is the fundamental angular frequency, and L is the filter inductance value; 步骤6，根据步骤5中得到的双逆变器光伏发电系统的总控制信号的dq分量vd、vq，分别乘以2/vdc后得到双逆变器光伏发电系统的总控制信号的标幺化dq分量md、mq，再经控制信号解耦方程得到第一三相两电平电压源型逆变器INV1的控制信号的标幺化dq分量md1、mq1和第二三相两电平电压源型逆变器INV2的控制信号的标幺化dq分量md2、mq2；Step 6: According to the dq components v d and v q of the total control signal of the dual-inverter photovoltaic power generation system obtained in step 5, multiply by 2/v dc to obtain the total control signal of the dual-inverter photovoltaic power generation system. The per-unit dq components m d , m q are then obtained through the control signal decoupling equation to obtain the per-unit dq components m d1 , m q1 and the second control signal of the first three-phase two-level voltage source inverter INV1 Per-unit dq components m d2 and m q2 of the control signal of the three-phase two-level voltage source inverter INV2; 所述双逆变器光伏发电系统的总控制信号的标幺化dq分量md、mq的计算公式为：The formulas for calculating the per-unit dq components m d and m q of the total control signal of the dual-inverter photovoltaic power generation system are:
Figure FDA0002306067790000041
Figure FDA0002306067790000041
 所述控制信号解耦方程为：The control signal decoupling equation is:
Figure FDA0002306067790000042
Figure FDA0002306067790000042
Figure FDA0002306067790000043
Figure FDA0002306067790000043
 步骤7，根据步骤6中得到的第一三相两电平电压源型逆变器INV1的控制信号的标幺化dq分量md1、mq1、第二三相两电平电压源型逆变器INV2的控制信号的标幺化dq分量md2、mq2和步骤3中得到的三相滤波电容电压的相角θ，经单同步旋转坐标反变换方程得到第一三相两电平电压源型逆变器INV1的控制信号的标幺化三相控制分量ma1、mb1、mc1和第二三相两电平电压源型逆变器INV2的控制信号的标幺化三相控制分量ma2、mb2、mc2；Step 7, according to the per-unit dq components m d1 , m q1 of the control signal of the first three-phase two-level voltage source inverter INV1 obtained in step 6, and the second three-phase two-level voltage source inverter The per-unit dq components m d2 and m q2 of the control signal of the device INV2 and the phase angle θ of the three-phase filter capacitor voltage obtained in step 3 are obtained through the single synchronous rotation coordinate inverse transformation equation to obtain the first three-phase two-level voltage source The per-unitized three-phase control components of the control signals of the inverter INV1 and the per- unitized three-phase control components of the control signals of the second three-phase two-level voltage source inverter INV2 m a2 , m b2 , m c2 ; 所述单同步旋转坐标反变换方程为：The inverse transformation equation of the single synchronous rotation coordinate is:
Figure FDA0002306067790000044
Figure FDA0002306067790000044
Figure FDA0002306067790000045
Figure FDA0002306067790000045
 步骤8，先设定零序电流的参考指令i0_ref，然后根据步骤4中得到的零序电流i0，经零序电流闭环控制方程得到零序调节信号Δk；Step 8, first set the zero-sequence current reference command i 0_ref , and then obtain the zero-sequence adjustment signal Δk through the zero-sequence current closed-loop control equation according to the zero-sequence current i 0 obtained in step 4; 所述零序电流闭环控制方程为：The zero-sequence current closed-loop control equation is:
Figure FDA0002306067790000051
Figure FDA0002306067790000051
 式中，Kp_0为零序电流环比例谐振调节器的的比例系数，Kr_0为零序电流环比例谐振调节器的的谐振系数，ωc为比例谐振调节器的带宽，ωr为比例谐振调节器的谐振频率；In the formula, K p_0 is the proportional coefficient of the zero-sequence current loop proportional resonant regulator, K r_0 is the resonant coefficient of the zero-sequence current loop proportional resonant regulator, ω c is the bandwidth of the proportional resonant regulator, ω r is the proportional resonance the resonant frequency of the regulator; 步骤9，根据步骤7中得到的第一三相两电平电压源型逆变器INV1的控制信号的标幺化三相控制分量ma1、mb1、mc1，取三者中的最大值和最小值并分别记为mmax1和mmin1；根据步骤7中得到的第二三相两电平电压源型逆变器INV2的控制信号的标幺化三相控制分量ma2、mb2、mc2，取三者中的最大值和最小值并分别记为mmax2和mmin2；然后再根据步骤8中得到的零序调节信号Δk，经零序注入分量计算方程后分别得到第一三相两电平电压源型逆变器INV1的零序注入分量mz1和第二三相两电平电压源型逆变器INV2的零序注入分量mz2；Step 9: According to the per-unitized three-phase control components ma1 , m b1 , and m c1 of the control signal of the first three-phase two-level voltage source inverter INV1 obtained in step 7, take the maximum value of the three and the minimum value and denoted as m max1 and m min1 respectively; according to the per-unitized three-phase control components m a2 , m b2 , m c2 , take the maximum value and the minimum value among the three and record them as m max2 and m min2 respectively; then according to the zero-sequence adjustment signal Δk obtained in step 8, after the zero-sequence injection component calculation equation, the first three the zero-sequence injection component m z1 of the phase two-level voltage source inverter INV1 and the zero-sequence injection component m z2 of the second three-phase two-level voltage source inverter INV2; 所述零序注入分量计算方程分别为：The calculation equations of the zero-sequence injection components are: mz1＝2Δk-(0.5+Δk)mmax1-(0.5-Δk)mmin1 m z1 =2Δk-(0.5+Δk)m max1 -(0.5-Δk)m min1 mz2＝-2Δk-(0.5-Δk)mmax2-(0.5+Δk)mmin2 m z2 =-2Δk-(0.5-Δk)m max2 -(0.5+Δk)m min2 步骤10，将步骤9中得到的第一三相两电平电压源型逆变器INV1的零序注入分量mz1分别与步骤7中得到的第一三相两电平电压源型逆变器INV1的控制信号的标幺化三相控制分量ma1、mb1、mc1相加得到第一三相两电平电压源型逆变器INV1的调制波信号
Figure FDA0002306067790000052
将步骤9中得到的第二三相两电平电压源型逆变器INV2的零序注入分量mz2分别与步骤7中得到的第二三相两电平电压源型逆变器INV2的控制信号的标幺化三相控制分量ma2、mb2、mc2相加得到第二三相两电平电压源型逆变器INV2的调制波信号
Figure FDA0002306067790000053
然后再分别经过与三角载波比较生成驱动第一三相两电平电压源型逆变器INV1和第二三相两电平电压源型逆变器INV2开关管的PWM控制信号PWM1和PWM2；Step 10: The zero-sequence injection component m z1 of the first three-phase two-level voltage source inverter INV1 obtained in step 9 is respectively combined with the first three-phase two-level voltage source inverter obtained in step 7. The per-unitized three-phase control components m a1 , m b1 , and m c1 of the control signal of INV1 are added to obtain the modulated wave signal of the first three-phase two-level voltage source inverter INV1
Figure FDA0002306067790000052
Control the zero-sequence injection component m z2 of the second three-phase two-level voltage source inverter INV2 obtained in step 9 and the second three-phase two-level voltage source inverter INV2 obtained in step 7 respectively. The per-unitized three-phase control components m a2 , m b2 , and m c2 are added to obtain the modulated wave signal of the second three-phase two-level voltage source inverter INV2
Figure FDA0002306067790000053
Then, the PWM control signals PWM1 and PWM2 for driving the switches of the first three-phase two-level voltage source inverter INV1 and the second three-phase two-level voltage source inverter INV2 are respectively generated by comparing with the triangular carrier; 所述调制波信号
Figure FDA0002306067790000054
和调制波信号
Figure FDA0002306067790000055
的计算方程分别为：The modulated wave signal
Figure FDA0002306067790000054
and modulated wave signal
Figure FDA0002306067790000055
The calculation equations are:
Figure FDA0002306067790000056
Figure FDA0002306067790000056
Figure FDA0002306067790000061
Figure FDA0002306067790000061
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