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CN111146801A - Zero-sequence current suppression method for common direct-current bus double-inverter photovoltaic power generation system - Google Patents

Zero-sequence current suppression method for common direct-current bus double-inverter photovoltaic power generation system Download PDF

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CN111146801A
CN111146801A CN201911240421.0A CN201911240421A CN111146801A CN 111146801 A CN111146801 A CN 111146801A CN 201911240421 A CN201911240421 A CN 201911240421A CN 111146801 A CN111146801 A CN 111146801A
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CN111146801B (en
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张兴
王宝基
洪剑峰
赵文广
曹仁贤
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Hefei University of Technology
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
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    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/483—Converters with outputs that each can have more than two voltages levels
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Abstract

The invention discloses a zero sequence current suppression method of a common direct current bus double-inverter photovoltaic power generation system. Aiming at the zero-sequence current in the common direct-current bus double-inverter photovoltaic power generation system, the invention provides a method for restraining the zero-sequence current by adopting a proportional resonance regulator and cooperating with a 120-degree decoupling carrier modulation strategy based on zero-sequence injection. The high-frequency component of the zero-sequence current can be inhibited by adopting a 120-degree decoupling carrier modulation strategy, the low-frequency zero-sequence current caused by non-linear factors such as an inverter dead zone and the like in the zero-sequence current can be inhibited by carrying out closed-loop control based on a proportional resonant regulator on the zero-sequence current and regulating the zero-sequence injection component of a modulation wave by the output of a control loop. Compared with the existing zero sequence current suppression method based on the space vector modulation strategy, the method provided by the invention does not need the processes of sector judgment, switching time calculation, table look-up and the like, greatly reduces the operation amount, and is easier to realize in engineering.

Description

Zero-sequence current suppression method for common direct-current bus double-inverter photovoltaic power generation system
Technical Field
The invention belongs to the field of electrical engineering and relates to a double-inverter control technology based on an open winding structure, in particular to a zero-sequence current suppression method of a common direct-current bus double-inverter photovoltaic power generation system.
Background
The double-two-level inverter topology based on the open-winding transformer structure can be equivalent to a three-level inverter, but compared with the traditional midpoint clamp type three-level inverter, the inverter has the advantages of high direct-current voltage utilization rate, strong redundancy, no need of a clamping diode, no need of midpoint balance control and the like, and has a certain application prospect in the field of photovoltaic power generation. However, when the double-inverter topology shares the dc bus, because both the ac side and the dc side have direct electrical connection, a zero sequence loop exists in the system, and zero sequence current is generated under the excitation of zero sequence voltage in the system, which causes the current quality to be poor and the system loss to be increased.
In order to suppress the zero-sequence current, experts and scholars at home and abroad propose methods such as an article entitled "a space-vector modulation scheme for a dual zero-level inverter fed open-end winding indication motor drive for the excitation of zero-sequence currents", Somasekhar V T, Gopakumar K, shivatumar E g, EPE Journal, 2002,12(2):1-19. ("a zero-sequence current suppression spatial vector modulation strategy applied to dual inverter driving of a winding induction motor", "european power electronics and drive Journal, volume 12, volume 2, 2002, 1-19) proposes that only a vector with zero-sequence voltage is selected to synthesize a reference voltage, which can simultaneously control high and low-frequency zero-sequence currents generated by modulation, but when considering non-linear voltage reduction factors such as non-linear dead zone, low-sequence tube voltage reduction, and the like, the low-frequency zero-sequence current of the system is still large;
an article entitled "Effect of Zero-Vector Placement in a Dual-Inverter Fed Open-EndWindingInducation-Motor Drive With a Decoupplied Space-Vector PWM Stratage", Veermraju T.Somasekhar, Srirama Srinivas, and Kommu Kranti Kumar, IEEETransactions on Industrial Electronics 2008, 55(6):2497 2505. ("the Effect of Zero Vector position when Open-winding Induction Motor Dual Inverter Drive employs Decoupled spatial voltage Vector modulation", "IEEE Industrial Electronics society, 2008, Vol.55 vol.6, pp.2050) proposes that the Zero Vector position is adjusted so that the average of the Zero sequence voltage in one sampling period is Zero, the Zero sequence current generated by the modulation can be suppressed, but the low frequency non-linear voltage reduction dead zone is not taken into account, and the like factors of the low frequency voltage reduction, the low-frequency zero-sequence current of the system is still large;
the article entitled "Dual-Space Vector Control of Open-End Winding Permanent magnet Motor Drive by Dual Inverter", An, quintao, Jin Liu, Zhuang Peng, and Lizhi Sun, "IEEE Transactions on Power Electronics, 2016, 31(12):8329-8342. (" Dual Space Vector Control of a Dual Inverter-driven Open Winding Permanent magnet synchronous Motor "," IEEE Power Electronics journal, volume 31 volume 12, 8329-8342 page) proposes a closed-loop Control of low-frequency zero-sequence currents by means of PI regulators and Space Vector modulation with zero Vector position reset, which can suppress low-frequency zero-sequence currents caused by non-linear factors such as dead zones, tube voltage drop, etc., but the high-frequency zero-sequence current is mainly suppressed by the inductance in the high-frequency system, and the photovoltaic system adopts a large-frequency filtering scheme, in addition, the scheme needs complex sector judgment, switching time calculation, zero vector distribution factor calculation, table lookup and the like, and the calculation amount is large;
the problem is that the zero sequence current suppression technology of the common direct current bus open winding permanent magnet synchronous motor based on the proportional resonance control, the article of the constant force, year honing, Zhouyije, the report of the electrotechnical Commission, No. 31, No. 22, 35-44 pages in 2016 proposes that the zero sequence current is suppressed by adopting a proportional resonance regulator and 180-degree decoupling sine pulse width modulation, the calculation amount of the scheme is small, but the high-frequency zero sequence current is still suppressed through the inductance in the system, and the method is not suitable for a photovoltaic grid-connected system with smaller zero sequence inductance;
an article entitled "Analysis and suppression of zero sequence circulating current inverting PMSM drives with common DC bus", Zhan, Hanlin, Zi-qiang Zhu, and miijana Odavic, "IEEE Transactions on industrial Applications, 2017, 53(4), 3609-;
in summary, the prior art mainly has the following disadvantages:
1. the existing method for inhibiting zero-sequence current only by adopting a modulation strategy cannot inhibit low-frequency zero-sequence current caused by non-linear factors such as dead zones, tube voltage drop and the like;
2. the existing method for resetting the space vector modulation strategy based on the proportional-integral regulator and the zero vector position can inhibit low-frequency zero-sequence current, but does not consider the inhibition of high-frequency zero-sequence current, and the modulation strategy has larger operand;
3. the existing method based on the proportional resonant regulator and the 180-degree decoupling carrier modulation strategy can inhibit low-frequency zero-sequence current, the modulation strategy computation amount is small, but the inhibition of high-frequency zero-sequence current in the system is not considered;
4. the existing 120-degree decoupling modulation strategy method based on the self-adaptive PR regulator and the adjustable zero vector action time can simultaneously inhibit the high-frequency and low-frequency zero-sequence currents of the system, but the adopted modulation strategy has larger calculation amount;
disclosure of Invention
The invention aims to solve the problem of zero sequence current of a common direct current bus double-inverter photovoltaic power generation system, and provides a zero sequence current suppression method based on a proportional resonant regulator and zero sequence injection 120-degree decoupling carrier modulation strategy.
In order to realize the purpose of the invention, the adopted technical scheme is as follows:
a zero sequence current suppression method for a common DC bus double-inverter photovoltaic power generation system comprises a photovoltaic array PV and a DC capacitor CdcThe three-phase power supply system comprises a first three-phase two-level voltage source inverter INV1, a second three-phase two-level voltage source inverter INV2, a three-phase filter inductor L, a three-phase filter capacitor C and a three-phase open winding transformer T; the photovoltaic array PV and the direct current capacitor CdcParallel connection; the first three-phase two-level voltage source inverter INV1 is connected in parallel with the DC side of the second three-phase two-level voltage source inverter INV2 and is connected with the DC capacitor CdcAre connected together; the primary three-phase winding of the three-phase open winding transformer T is in an open state, and the A-phase winding has two terminals which are respectively marked as a terminal A1And terminal A2The winding of phase B has two terminals, respectively denoted as terminal B1And terminal B2The C-phase winding has two terminals, respectively denoted as terminal C1And terminal C2Setting terminal A1Terminal B1And terminal C1The three terminals are arranged on the same side of the primary winding of the three-phase open winding transformer T and are used as input terminals of the primary winding of the three-phase open winding transformer T, and the terminal A2Terminal B2And terminal C2The three terminals are used as output terminals of the primary winding of the three-phase open winding transformer T on the other side of the primary winding of the three-phase open winding transformer T; the three-phase filter inductor L is provided with 6 terminals, two terminals of each phase are respectively marked as a terminal A3And terminal A4And the two terminals of phase B are respectively denoted as terminal B3And terminal B4And the two terminals of the C phase are respectively marked as terminals C3And terminal C4Setting terminal A3Terminal B3And terminal C3At the same side of the three-phase filter inductor L and using the three terminals as the input terminal of the three-phase filter inductor L, the terminal A4Terminal B4And terminal C4The three terminals are used as output terminals of the three-phase filter inductor L on the other side of the three-phase filter inductor L; the three-phase filter capacitor C has 6 terminals, two terminals per phase, and two terminals AIs marked as terminal A5And terminal A6And the two terminals of phase B are respectively denoted as terminal B5And terminal B6And the two terminals of the C phase are respectively marked as terminals C5And terminal C6Setting terminal A5Terminal B5And terminal C5At the same side of the three-phase filter capacitor C and using the three terminals as the input terminals of the three-phase filter capacitor C, the terminal A6Terminal B6And terminal C6The three terminals are used as output terminals of the three-phase filter capacitor C on the other side of the three-phase filter capacitor C; terminal A of the three-phase filter inductor L3Terminal B3And terminal C3Is connected to the AC output side of the first three-phase two-level voltage source inverter INV1, terminal A4Terminal B4And terminal C4Respectively connected with terminals A of primary windings of three-phase open winding transformer T1Terminal B1And terminal C1And terminal A of three-phase filter capacitor C5Terminal B5And terminal C5Connecting; terminal A of primary winding of T-shaped three-phase open winding transformer2Terminal B2Terminal C2Respectively connected with terminals A of three-phase filter capacitor C6Terminal B6And terminal C6After being connected, the three-phase two-level voltage source inverter is connected to the alternating current output side of the second three-phase two-level voltage source inverter INV 2; the secondary side of the three-phase open winding transformer T is connected into a power grid E in a star connection or a triangular connection mode;
the zero sequence current suppression method comprises the following steps:
step 1, collecting direct-current side voltage v of a double-inverter photovoltaic power generation systemdcD.c. side current idcCollecting and recording the voltage of the three-phase filter capacitor C as the voltage v of the three-phase filter capacitorca、vcb、vccCollecting the current of the input end of the three-phase filter inductor L and recording the current as the bridge arm side inductor current ia、ib、ic；
Step 2, according to the voltage v on the direct current side obtained in the step 1dcAnd a direct side current idcObtaining the direct current side voltage of the double-inverter photovoltaic power generation system after maximum power point trackingInstruction vdc_ref(ii) a Then obtaining an active current instruction i of the double-inverter photovoltaic power generation system through a direct-current side voltage closed-loop control equationd_ref；
The direct current side voltage closed-loop control equation is as follows:
Figure BDA0002306067800000041
in the formula, Kp_vdcIs the proportionality coefficient, K, of a voltage-loop PI regulator on the DC sidei_vdcThe integral coefficient of the direct-current side voltage loop PI regulator is shown, and s is a Laplace operator;
step 3, obtaining the three-phase filter capacitor voltage v according to the step 1ca、vcb、vccObtaining the phase angle theta of the three-phase filter capacitor voltage and the dq component v of the three-phase filter capacitor voltage through a phase-locked loopcd、vcq；
The calculation equation of the voltage phase angle theta of the three-phase filter capacitor is as follows:
Figure BDA0002306067800000042
Figure BDA0002306067800000043
wherein θ ' is the phase angle v ' of the three-phase filter capacitor voltage obtained in the previous control period 'cqThe q component, omega, obtained by performing synchronous rotation coordinate transformation on the phase angle theta' of the three-phase filter capacitor voltage calculated according to the previous control period0Rated angular frequency, K, of three-phase filter capacitor voltagep_PLLIs the proportionality coefficient, K, of a phase-locked loop PI regulatori_PLLThe integral coefficient of the phase-locked loop PI regulator is obtained;
dq component v of the three-phase filter capacitor voltagecd、vcqThe calculation equation of (a) is:
Figure BDA0002306067800000051
step 4, according to the bridge arm side inductive current i obtained in the step 1a、ib、icAnd step 3, obtaining the dq component i of the bridge arm side inductive current through a single synchronous rotation coordinate transformation equation according to the phase angle theta of the three-phase filter capacitor voltage obtained in the stepd、iqAnd zero sequence current i0；
The transformation equation of the single synchronous rotation coordinate is as follows:
Figure BDA0002306067800000052
step 5, firstly setting a reactive current instruction iq_refThen according to the active current command i obtained in the step 2d_refAnd 3, obtaining dq component v of the three-phase filter capacitor voltage in the step 3cd、vcqAnd dq component i of the bridge arm side inductive current obtained in step 4d、iqAnd obtaining dq component v of the master control signal of the double-inverter photovoltaic power generation system through a current closed-loop control equationd、vq；
The current closed-loop control equation is as follows:
Figure BDA0002306067800000053
in the formula, Kp_iIs the proportionality coefficient, K, of a current loop PI regulatori_iThe integral coefficient of the current loop PI regulator is shown, omega is the fundamental angular frequency, and L is the filter inductance value;
step 6, according to the dq component v of the total control signal of the double-inverter photovoltaic power generation system obtained in the step 5d、vqMultiply by 2/v, respectivelydcObtaining per unit dq component m of the master control signal of the double-inverter photovoltaic power generation systemd、mqObtaining a per-unit dq component m of the control signal of the first three-phase two-level voltage source inverter INV1 through a control signal decoupling equationd1、mq1And a per-unit dq component m of a control signal of the second three-phase two-level voltage source inverter INV2d2、mq2；
Per unit dq component m of total control signal of the double-inverter photovoltaic power generation systemd、mqThe calculation formula of (2) is as follows:
Figure BDA0002306067800000061
the control signal decoupling equation is as follows:
Figure BDA0002306067800000062
Figure BDA0002306067800000063
step 7, obtaining a per-unit dq component m of the control signal of the first three-phase two-level voltage source inverter INV1 according to step 6d1、mq1And a per-unit dq component m of a control signal of the second three-phase two-level voltage source inverter INV2d2、mq2And step 3, obtaining a per-unit three-phase control component m of the control signal of the first three-phase two-level voltage source inverter INV1 through a single synchronous rotation coordinate inverse transformation equation according to the phase angle theta of the three-phase filter capacitor voltage obtained in the step 3a1、mb1、mc1And a unified three-phase control component m of a control signal of the second three-phase two-level voltage source inverter INV2a2、mb2、mc2；
The single synchronous rotation coordinate inverse transformation equation is as follows:
Figure BDA0002306067800000064
Figure BDA0002306067800000071
step 8, firstly setting a reference instruction i of the zero sequence current0_refThen according to the zero sequence current i obtained in the step 40Zero sequence current closed loop control equationObtaining a zero sequence adjusting signal delta k;
the zero-sequence current closed-loop control equation is as follows:
Figure BDA0002306067800000072
in the formula, Kp_0Is the proportionality coefficient, K, of a zero-sequence current loop proportional resonant regulatorr_0Is the resonance coefficient, omega, of a zero-sequence current loop proportional resonant regulatorcIs the bandwidth, omega, of a proportional resonant regulatorrIs the resonant frequency of the proportional resonant regulator;
step 9, obtaining a per-unit three-phase control component m according to the control signal of the first three-phase two-level voltage source inverter INV1 obtained in step 7a1、mb1、mc1Taking the maximum value and the minimum value of the three values and recording the maximum value and the minimum value as mmax1And mmin1(ii) a A unit three-phase control component m according to the control signal of the second three-phase two-level voltage source inverter INV2 obtained in step 7a2、mb2、mc2Taking the maximum value and the minimum value of the three values and recording the maximum value and the minimum value as mmax2And mmin2(ii) a Then, according to the zero sequence adjusting signal Δ k obtained in step 8, a zero sequence injection component calculation equation is performed to obtain a zero sequence injection component m of the first three-phase two-level voltage source inverter INV1z1And a zero-sequence injection component m of a second three-phase two-level voltage source inverter INV2z2；
The zero-sequence injection component calculation equation is respectively as follows:
mz1＝2Δk-(0.5+Δk)mmax1-(0.5-Δk)mmin1
mz2＝-2Δk-(0.5-Δk)mmax2-(0.5+Δk)mmin2
step 10, injecting a zero sequence component m of the first three-phase two-level voltage source inverter INV1 obtained in step 9z1Respectively compared with the unit three-phase control component m of the control signal of the first three-phase two-level voltage source inverter INV1 obtained in step 7a1、mb1、mc1Add to obtain the first threeModulated wave signal of two-phase voltage source inverter INV1
Figure BDA0002306067800000073
Figure BDA0002306067800000081
Injecting the zero sequence of the second three-phase two-level voltage source inverter INV2 obtained in the step 9 into the component mz2Respectively compared with the unit three-phase control component m of the control signal of the second three-phase two-level voltage source inverter INV2 obtained in step 7a2、mb2、mc2Adding to obtain modulated wave signal of second three-phase two-level voltage source inverter INV2
Figure BDA0002306067800000082
Figure BDA0002306067800000083
Then PWM control signals PWM1 and PWM2 for driving switching tubes of the first three-phase two-level voltage source inverter INV1 and the second three-phase two-level voltage source inverter INV2 are generated through comparison with the triangular carrier waves respectively;
the modulated wave signal
Figure BDA0002306067800000084
And modulating the wave signal
Figure BDA0002306067800000085
The calculation equations of (a) are:
Figure BDA0002306067800000086
Figure BDA0002306067800000087
compared with the prior art, the invention has the beneficial effects that:
1. the method can inhibit high-frequency and low-frequency zero-sequence currents in the system at the same time, and is particularly suitable for the application field with smaller zero-sequence inductance in the system, such as the photovoltaic grid-connected field;
2. compared with the space vector modulation strategy in the prior art, the adopted modulation strategy only needs to calculate the zero sequence injection component, and adopts carrier modulation, so that the processes of sector judgment, switching time calculation, zero vector allocation time calculation, table look-up and the like are not needed, the operation amount is greatly reduced, and the realization is easier.
Drawings
Fig. 1 is a main circuit block diagram of a common dc bus double-inverter photovoltaic power generation system in an embodiment of the invention.
Fig. 2 is a control block diagram of the zero sequence current suppression method in the embodiment of the present invention.
Fig. 3 is a block diagram of a zero sequence injection component calculation link of two inverters in the zero sequence current suppression method according to the embodiment of the present invention.
Fig. 4 is a simulation waveform of the front and rear zero-sequence currents of the zero-sequence current suppression method provided by the present invention.
Fig. 5 is a simulation waveform of front and rear bridge arm currents cut into the zero sequence current suppression method provided by the present invention.
Fig. 6 is a fourier analysis diagram of the front bridge arm current cut into the zero sequence current suppression method provided by the present invention.
Fig. 7 is a fourier analysis diagram of the rear bridge arm current cut into the zero sequence current suppression method provided by the present invention.
Detailed Description
The invention is described in further detail below with reference to the figures and examples.
Fig. 1 is a main circuit block diagram of a common dc bus double-inverter photovoltaic power generation system in an embodiment of the invention. As shown in fig. 1, the common dc bus dual-inverter photovoltaic power generation system according to the present invention includes a photovoltaic array PV and a dc capacitor CdcThe three-phase power supply system comprises a first three-phase two-level voltage source inverter INV1, a second three-phase two-level voltage source inverter INV2, a three-phase filter inductor L, a three-phase filter capacitor C and a three-phase open winding transformer T.
The photovoltaic array PV and the direct current capacitor CdcAnd (4) connecting in parallel.
The above-mentionedThe first three-phase two-level voltage source inverter INV1 is connected in parallel with the DC side of the second three-phase two-level voltage source inverter INV2 and is connected with the DC capacitor CdcAnd are joined together.
The primary three-phase winding of the three-phase open winding transformer T is in an open state, and the A-phase winding has two terminals which are respectively marked as a terminal A1And terminal A2The winding of phase B has two terminals, respectively denoted as terminal B1And terminal B2The C-phase winding has two terminals, respectively denoted as terminal C1And terminal C2Setting terminal A1Terminal B1And terminal C1The three terminals are arranged on the same side of the primary winding of the three-phase open winding transformer T and are used as input terminals of the primary winding of the three-phase open winding transformer T, and the terminal A2Terminal B2And terminal C2And the three terminals are arranged on the other side of the primary winding of the three-phase open winding transformer T and are used as output terminals of the primary winding of the three-phase open winding transformer T.
The three-phase filter inductor L is provided with 6 terminals, two terminals of each phase are respectively marked as a terminal A3And terminal A4And the two terminals of phase B are respectively denoted as terminal B3And terminal B4And the two terminals of the C phase are respectively marked as terminals C3And terminal C4Setting terminal A3Terminal B3And terminal C3At the same side of the three-phase filter inductor L and using the three terminals as the input terminal of the three-phase filter inductor L, the terminal A4Terminal B4And terminal C4And the three terminals are arranged on the other side of the three-phase filter inductor L and are used as output terminals of the three-phase filter inductor L.
The three-phase filter capacitor C has 6 terminals, two terminals of each phase, and two terminals of the A phase are respectively marked as a terminal A5And terminal A6And the two terminals of phase B are respectively denoted as terminal B5And terminal B6And the two terminals of the C phase are respectively marked as terminals C5And terminal C6Setting terminal A5Terminal B5And terminal C5The three terminals are arranged on the same side of the three-phase filter capacitor C and are used as input terminals and terminals of the three-phase filter capacitor CA6Terminal B6And terminal C6And the three terminals are arranged on the other side of the three-phase filter capacitor C and are used as output terminals of the three-phase filter capacitor C.
Terminal A of the three-phase filter inductor L3Terminal B3And terminal C3Is connected to the AC output side of the first three-phase two-level voltage source inverter INV1, terminal A4Terminal B4And terminal C4Are respectively connected with primary winding terminals A of a three-phase open winding transformer T1Terminal B1And terminal C1And a three-phase filter capacitor C terminal A5Terminal B5And terminal C5Connecting; terminal A of primary winding of T-shaped three-phase open winding transformer2Terminal B2Terminal C2Respectively connected with terminals A of three-phase filter capacitor C6Terminal B6And terminal C6After being connected, the three-phase two-level voltage source inverter is connected to the alternating current output side of the second three-phase two-level voltage source inverter INV 2; and the secondary side of the three-phase open winding transformer T is connected into a power grid E in a star connection or a triangular connection mode.
In the embodiment of the invention, the double inverters adopt a double-two-level inverter topology, the total rated power of the system is 30kW, the rated power of each inverter is 15kW, the switching frequency is 5kHz, the dead time of a switching tube is 3 mus, and the direct-current capacitor CdcThe inductance L of the three-phase filter is 0.5mF, the L of the three-phase filter inductance is 3.6mH, the C of the three-phase filter capacitance is 10 muF, the phase voltage transformation ratio of the open winding transformer T is 364V/380V, the short-circuit impedance is 3%, and the effective value of the grid line voltage is 380V/50 Hz.
Referring to fig. 2 and fig. 3, the zero sequence current suppression method of the present invention includes the following steps:
step 1, collecting direct-current side voltage v of a double-inverter photovoltaic power generation systemdcD.c. side current idcCollecting and recording the voltage of the three-phase filter capacitor C as the voltage v of the three-phase filter capacitorca、vcb、vccCollecting the current of the input end of the three-phase filter inductor L and recording the current as the bridge arm side inductor current ia、ib、ic。
Step 2, according to the product obtained in step 1Voltage v at the DC sidedcAnd a direct side current idcObtaining a direct-current side voltage instruction v of the double-inverter photovoltaic power generation system after Maximum Power Point Tracking (MPPT)dc_ref(ii) a Then obtaining an active current instruction i of the double-inverter photovoltaic power generation system through a direct-current side voltage closed-loop control equationd_ref；
The direct current side voltage closed-loop control equation is as follows:
Figure BDA0002306067800000101
in the formula, Kp_vdcIs the proportionality coefficient, K, of a voltage-loop PI regulator on the DC sidei_vdcAnd s is a Laplace operator. In this embodiment, vdc_ref＝620V，Kp_vdc＝2，Ki_vdc＝50。
Step 3, obtaining the three-phase filter capacitor voltage v according to the step 1ca、vcb、vccObtaining the phase angle theta of the three-phase filter capacitor voltage and the dq component v of the three-phase filter capacitor voltage through a phase-locked loop (PLL)cd、vcq。
The calculation equation of the voltage phase angle theta of the three-phase filter capacitor is as follows:
Figure BDA0002306067800000111
Figure BDA0002306067800000112
wherein θ ' is the phase angle v ' of the three-phase filter capacitor voltage obtained in the previous control period 'cqThe q component, omega, obtained by performing synchronous rotation coordinate transformation on the phase angle theta' of the three-phase filter capacitor voltage calculated according to the previous control period0Rated angular frequency, K, of three-phase filter capacitor voltagep_PLLIs the proportionality coefficient, K, of a phase-locked loop PI regulatori_PLLIs the integral coefficient of the phase-locked loop PI regulator. In the present embodiment, ω0＝100πrad/s，Kp_PLL＝0.1，Ki_PLL＝0.005。
Dq component v of the three-phase filter capacitor voltagecd、vcqThe calculation equation of (a) is:
Figure BDA0002306067800000113
step 4, according to the bridge arm side inductive current i obtained in the step 1a、ib、icAnd step 3, obtaining the dq component i of the bridge arm side inductive current through a single synchronous rotation coordinate transformation equation according to the phase angle theta of the three-phase filter capacitor voltage obtained in the stepd、iqAnd zero sequence current i0。
The transformation equation of the single synchronous rotation coordinate is as follows:
Figure BDA0002306067800000114
step 5, firstly setting a reactive current instruction iq_refThen according to the active current command i obtained in the step 2d_refAnd 3, obtaining dq component v of the three-phase filter capacitor voltage in the step 3cd、vcqAnd dq component i of the bridge arm side inductive current obtained in step 4d、iqAnd obtaining dq component v of the master control signal of the double-inverter photovoltaic power generation system through a current closed-loop control equationd、vq。
The current closed-loop control equation is as follows:
Figure BDA0002306067800000121
in the formula, Kp_iIs the proportionality coefficient, K, of a current loop PI regulatori_iThe integral coefficient of the current loop PI regulator is shown, omega is the angular frequency of the fundamental wave, and L is the inductance value of the filter. In this embodiment, ω ═ 100 π rad/s, Kp_i＝5，Ki_i＝200。
Step 6, according to the total control of the double-inverter photovoltaic power generation system obtained in the step 5Dq component v of the signald、vqMultiply by 2/v, respectivelydcObtaining per unit dq component m of the master control signal of the double-inverter photovoltaic power generation systemd、mqObtaining a per-unit dq component m of the control signal of the first three-phase two-level voltage source inverter INV1 through a control signal decoupling equationd1、mq1And a per-unit dq component m of a control signal of the second three-phase two-level voltage source inverter INV2d2、mq2。
Per unit dq component m of total control signal of the double-inverter photovoltaic power generation systemd、mqThe calculation formula of (2) is as follows:
Figure BDA0002306067800000122
the control signal decoupling equation is as follows:
Figure BDA0002306067800000123
Figure BDA0002306067800000124
step 7, obtaining a per-unit dq component m of the control signal of the first three-phase two-level voltage source inverter INV1 according to step 6d1、mq1And a per-unit dq component m of a control signal of the second three-phase two-level voltage source inverter INV2d2、mq2And step 3, obtaining a per-unit three-phase control component m of the control signal of the first three-phase two-level voltage source inverter INV1 through a single synchronous rotation coordinate inverse transformation equation according to the phase angle theta of the three-phase filter capacitor voltage obtained in the step 3a1、mb1、mc1And a unified three-phase control component m of a control signal of the second three-phase two-level voltage source inverter INV2a2、mb2、mc2。
The single synchronous rotation coordinate inverse transformation equation is as follows:
Figure BDA0002306067800000131
Figure BDA0002306067800000132
step 8, firstly setting a reference instruction i of the zero sequence current0_refThen according to the zero sequence current i obtained in the step 40And obtaining a zero sequence adjusting signal delta k through a zero sequence current closed-loop control equation.
The zero-sequence current closed-loop control equation is as follows:
Figure BDA0002306067800000133
in the formula, Kp_0Proportional coefficient, K, of a zero sequence current loop Proportional Resonance (PR) regulatorr_0Is the resonance coefficient, omega, of a zero sequence current loop PR regulatorcFor the bandwidth of the PR regulator, ωrIs the resonant frequency of the PR modulator. In this embodiment, i0_ref＝0，Kp_0＝0.1，Kr_0＝10，ωc＝3rad/s，ωr＝300πrad/s。
Step 9, obtaining a per-unit three-phase control component m according to the control signal of the first three-phase two-level voltage source inverter INV1 obtained in step 7a1、mb1、mc1Taking the maximum value and the minimum value of the three values and recording the maximum value and the minimum value as mmax1And mmin1(ii) a A unit three-phase control component m according to the control signal of the second three-phase two-level voltage source inverter INV2 obtained in step 7a2、mb2、mc2Taking the maximum value and the minimum value of the three values and recording the maximum value and the minimum value as mmax2And mmin2(ii) a Then, according to the zero sequence adjusting signal Δ k obtained in step 8, a zero sequence injection component calculation equation is performed to obtain a zero sequence injection component m of the first three-phase two-level voltage source inverter INV1z1And a zero-sequence injection component m of a second three-phase two-level voltage source inverter INV2z2。
The zero-sequence injection component calculation equation is respectively as follows:
mz1＝2Δk-(0.5+Δk)mmax1-(0.5-Δk)mmin1
mz2＝-2Δk-(0.5-Δk)mmax2-(0.5+Δk)mmin2
step 10, injecting a zero sequence component m of the first three-phase two-level voltage source inverter INV1 obtained in step 9z1Respectively compared with the unit three-phase control component m of the control signal of the first three-phase two-level voltage source inverter INV1 obtained in step 7a1、mb1、mc1Adding to obtain modulated wave signal of first three-phase two-level voltage source inverter INV1
Figure BDA0002306067800000141
Figure BDA0002306067800000142
Injecting the zero sequence of the second three-phase two-level voltage source inverter INV2 obtained in the step 9 into the component mz2Respectively compared with the unit three-phase control component m of the control signal of the second three-phase two-level voltage source inverter INV2 obtained in step 7a2、mb2、mc2Adding to obtain modulated wave signal of second three-phase two-level voltage source inverter INV2
Figure BDA0002306067800000143
Figure BDA0002306067800000144
Then PWM control signals PWM1 and PWM2 for driving switching tubes of the first three-phase two-level voltage source inverter INV1 and the second three-phase two-level voltage source inverter INV2 are generated through comparison with the triangular carrier waves respectively.
The modulated wave signal
Figure BDA0002306067800000145
And modulating the wave signal
Figure BDA0002306067800000146
The calculation equations of (a) are:
Figure BDA0002306067800000147
Figure BDA0002306067800000148
fig. 4 is a simulation waveform of zero sequence current before and after the zero sequence current suppression method according to specific parameters of the embodiment of the present invention is performed, and it can be found that the high frequency component of the zero sequence current is suppressed but the low frequency component is larger only by decoupling modulation of 120 degrees before the zero sequence current suppression method according to the present invention is performed, and both the high frequency component and the low frequency component of the zero sequence current are suppressed after the zero sequence current suppression method according to the present invention is performed. Fig. 5 is a simulation waveform of front and rear bridge arm currents of the zero-sequence current suppression method according to the embodiment of the present invention, fig. 6 is a fourier analysis diagram of the front bridge arm current of the zero-sequence current suppression method according to the present invention, fig. 7 is a fourier analysis diagram of the rear bridge arm current of the zero-sequence current suppression method according to the present invention, and it can be found from fig. 5, fig. 6 and fig. 7 that the first 3 harmonics of the zero-sequence current suppression method according to the present invention account for 4.75% of the bridge arm current, and the 3 harmonics of the zero-sequence current suppression method according to the present invention only account for 0.17% of the bridge arm current, which greatly improves the electric energy quality of the bridge arm current. The simulation results show the correctness and the effectiveness of the zero sequence current suppression method of the common direct current bus double-inverter photovoltaic power generation system.

Claims (1)

1. A zero sequence current suppression method for a common DC bus double-inverter photovoltaic power generation system comprises a photovoltaic array PV and a DC capacitor CdcThe three-phase power supply system comprises a first three-phase two-level voltage source inverter INV1, a second three-phase two-level voltage source inverter INV2, a three-phase filter inductor L, a three-phase filter capacitor C and a three-phase open winding transformer T; the photovoltaic array PV and the direct current capacitor CdcParallel connection; the first three-phase two-level voltage source inverter INV1 and the secondThe DC side of the three-phase two-level voltage source inverter INV2 is connected in parallel and is connected with the DC capacitor CdcAre connected together; the primary three-phase winding of the three-phase open winding transformer T is in an open state, and the A-phase winding has two terminals which are respectively marked as a terminal A1And terminal A2The winding of phase B has two terminals, respectively denoted as terminal B1And terminal B2The C-phase winding has two terminals, respectively denoted as terminal C1And terminal C2Setting terminal A1Terminal B1And terminal C1The three terminals are arranged on the same side of the primary winding of the three-phase open winding transformer T and are used as input terminals of the primary winding of the three-phase open winding transformer T, and the terminal A2Terminal B2And terminal C2The three terminals are used as output terminals of the primary winding of the three-phase open winding transformer T on the other side of the primary winding of the three-phase open winding transformer T; the three-phase filter inductor L is provided with 6 terminals, two terminals of each phase are respectively marked as a terminal A3And terminal A4And the two terminals of phase B are respectively denoted as terminal B3And terminal B4And the two terminals of the C phase are respectively marked as terminals C3And terminal C4Setting terminal A3Terminal B3And terminal C3At the same side of the three-phase filter inductor L and using the three terminals as the input terminal of the three-phase filter inductor L, the terminal A4Terminal B4And terminal C4The three terminals are used as output terminals of the three-phase filter inductor L on the other side of the three-phase filter inductor L; the three-phase filter capacitor C has 6 terminals, two terminals of each phase, and two terminals of the A phase are respectively marked as a terminal A5And terminal A6And the two terminals of phase B are respectively denoted as terminal B5And terminal B6And the two terminals of the C phase are respectively marked as terminals C5And terminal C6Setting terminal A5Terminal B5And terminal C5At the same side of the three-phase filter capacitor C and using the three terminals as the input terminals of the three-phase filter capacitor C, the terminal A6Terminal B6And terminal C6The three terminals are used as output terminals of the three-phase filter capacitor C on the other side of the three-phase filter capacitor C; the above-mentionedTerminal A of three-phase filter inductor L3Terminal B3And terminal C3Is connected to the AC output side of the first three-phase two-level voltage source inverter INV1, terminal A4Terminal B4And terminal C4Respectively connected with terminals A of primary windings of three-phase open winding transformer T1Terminal B1And terminal C1And terminal A of three-phase filter capacitor C5Terminal B5And terminal C5Connecting; terminal A of primary winding of T-shaped three-phase open winding transformer2Terminal B2Terminal C2Respectively connected with terminals A of three-phase filter capacitor C6Terminal B6And terminal C6After being connected, the three-phase two-level voltage source inverter is connected to the alternating current output side of the second three-phase two-level voltage source inverter INV 2; the secondary side of the three-phase open winding transformer T is connected into a power grid E in a star connection or a triangular connection mode;
the zero sequence current suppression method is characterized by comprising the following steps of:
step 1, collecting direct-current side voltage v of a double-inverter photovoltaic power generation systemdcD.c. side current idcCollecting and recording the voltage of the three-phase filter capacitor C as the voltage v of the three-phase filter capacitorca、vcb、vccCollecting the current of the input end of the three-phase filter inductor L and recording the current as the bridge arm side inductor current ia、ib、ic；
Step 2, according to the voltage v on the direct current side obtained in the step 1dcAnd a direct side current idcObtaining a direct-current side voltage instruction v of the double-inverter photovoltaic power generation system after maximum power point trackingdc_ref(ii) a Then obtaining an active current instruction i of the double-inverter photovoltaic power generation system through a direct-current side voltage closed-loop control equationd_ref；
The direct current side voltage closed-loop control equation is as follows:
Figure FDA0002306067790000021
in the formula, Kp_vdcIs the proportionality coefficient, K, of a voltage-loop PI regulator on the DC sidei_vdcThe integral coefficient of the direct-current side voltage loop PI regulator is shown, and s is a Laplace operator;
step 3, obtaining the three-phase filter capacitor voltage v according to the step 1ca、vcb、vccObtaining the phase angle theta of the three-phase filter capacitor voltage and the dq component v of the three-phase filter capacitor voltage through a phase-locked loopcd、vcq；
The calculation equation of the voltage phase angle theta of the three-phase filter capacitor is as follows:
Figure FDA0002306067790000022
Figure FDA0002306067790000023
wherein θ ' is the phase angle v ' of the three-phase filter capacitor voltage obtained in the previous control period 'cqThe q component, omega, obtained by performing synchronous rotation coordinate transformation on the phase angle theta' of the three-phase filter capacitor voltage calculated according to the previous control period0Rated angular frequency, K, of three-phase filter capacitor voltagep_PLLIs the proportionality coefficient, K, of a phase-locked loop PI regulatori_PLLThe integral coefficient of the phase-locked loop PI regulator is obtained;
dq component v of the three-phase filter capacitor voltagecd、vcqThe calculation equation of (a) is:
Figure FDA0002306067790000031
step 4, according to the bridge arm side inductive current i obtained in the step 1a、ib、icAnd step 3, obtaining the dq component i of the bridge arm side inductive current through a single synchronous rotation coordinate transformation equation according to the phase angle theta of the three-phase filter capacitor voltage obtained in the stepd、iqAnd zero sequence current i0；
The transformation equation of the single synchronous rotation coordinate is as follows:
Figure FDA0002306067790000032
step 5, firstly setting a reactive current instruction iq_refThen according to the active current command i obtained in the step 2d_refAnd 3, obtaining dq component v of the three-phase filter capacitor voltage in the step 3cd、vcqAnd dq component i of the bridge arm side inductive current obtained in step 4d、iqAnd obtaining dq component v of the master control signal of the double-inverter photovoltaic power generation system through a current closed-loop control equationd、vq；
The current closed-loop control equation is as follows:
Figure FDA0002306067790000033
in the formula, Kp_iIs the proportionality coefficient, K, of a current loop PI regulatori_iThe integral coefficient of the current loop PI regulator is shown, omega is the fundamental angular frequency, and L is the filter inductance value;
step 6, according to the dq component v of the total control signal of the double-inverter photovoltaic power generation system obtained in the step 5d、vqMultiply by 2/v, respectivelydcObtaining per unit dq component m of the master control signal of the double-inverter photovoltaic power generation systemd、mqObtaining a per-unit dq component m of the control signal of the first three-phase two-level voltage source inverter INV1 through a control signal decoupling equationd1、mq1And a per-unit dq component m of a control signal of the second three-phase two-level voltage source inverter INV2d2、mq2；
Per unit dq component m of total control signal of the double-inverter photovoltaic power generation systemd、mqThe calculation formula of (2) is as follows:
Figure FDA0002306067790000041
the control signal decoupling equation is as follows:
Figure FDA0002306067790000042
Figure FDA0002306067790000043
step 7, obtaining a per-unit dq component m of the control signal of the first three-phase two-level voltage source inverter INV1 according to step 6d1、mq1And a per-unit dq component m of a control signal of the second three-phase two-level voltage source inverter INV2d2、mq2And step 3, obtaining a per-unit three-phase control component m of the control signal of the first three-phase two-level voltage source inverter INV1 through a single synchronous rotation coordinate inverse transformation equation according to the phase angle theta of the three-phase filter capacitor voltage obtained in the step 3a1、mb1、mc1And a unified three-phase control component m of a control signal of the second three-phase two-level voltage source inverter INV2a2、mb2、mc2；
The single synchronous rotation coordinate inverse transformation equation is as follows:
Figure FDA0002306067790000044
Figure FDA0002306067790000045
step 8, firstly setting a reference instruction i of the zero sequence current0_refThen according to the zero sequence current i obtained in the step 40Obtaining a zero-sequence adjusting signal delta k through a zero-sequence current closed-loop control equation;
the zero-sequence current closed-loop control equation is as follows:
Figure FDA0002306067790000051
in the formula, Kp_0Is the proportionality coefficient, K, of a zero-sequence current loop proportional resonant regulatorr_0Is zero sequence currentResonance coefficient, omega, of a ring proportional resonant regulatorcIs the bandwidth, omega, of a proportional resonant regulatorrIs the resonant frequency of the proportional resonant regulator;
step 9, obtaining a per-unit three-phase control component m according to the control signal of the first three-phase two-level voltage source inverter INV1 obtained in step 7a1、mb1、mc1Taking the maximum value and the minimum value of the three values and recording the maximum value and the minimum value as mmax1And mmin1(ii) a A unit three-phase control component m according to the control signal of the second three-phase two-level voltage source inverter INV2 obtained in step 7a2、mb2、mc2Taking the maximum value and the minimum value of the three values and recording the maximum value and the minimum value as mmax2And mmin2(ii) a Then, according to the zero sequence adjusting signal Δ k obtained in step 8, a zero sequence injection component calculation equation is performed to obtain a zero sequence injection component m of the first three-phase two-level voltage source inverter INV1z1And a zero-sequence injection component m of a second three-phase two-level voltage source inverter INV2z2；
The zero-sequence injection component calculation equation is respectively as follows:
mz1＝2Δk-(0.5+Δk)mmax1-(0.5-Δk)mmin1
mz2＝-2Δk-(0.5-Δk)mmax2-(0.5+Δk)mmin2
step 10, injecting a zero sequence component m of the first three-phase two-level voltage source inverter INV1 obtained in step 9z1Respectively compared with the unit three-phase control component m of the control signal of the first three-phase two-level voltage source inverter INV1 obtained in step 7a1、mb1、mc1Adding to obtain modulated wave signal of first three-phase two-level voltage source inverter INV1
Figure FDA0002306067790000052
Injecting the zero sequence of the second three-phase two-level voltage source inverter INV2 obtained in the step 9 into the component mz2Respectively compared with the unit three-phase control component m of the control signal of the second three-phase two-level voltage source inverter INV2 obtained in step 7a2、mb2、mc2Adding to obtain modulated wave signal of second three-phase two-level voltage source inverter INV2
Figure FDA0002306067790000053
Then PWM control signals PWM1 and PWM2 for driving switching tubes of the first three-phase two-level voltage source inverter INV1 and the second three-phase two-level voltage source inverter INV2 are generated through comparison with the triangular carrier waves respectively;
the modulated wave signal
Figure FDA0002306067790000054
And modulating the wave signal
Figure FDA0002306067790000055
The calculation equations of (a) are:
Figure FDA0002306067790000056
Figure FDA0002306067790000061
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