CN117941243A - Power conversion device, motor drive device, and refrigeration cycle application device - Google Patents

Abstract

The power conversion device (2) is provided with a converter (10) for rectifying a power supply voltage applied from an AC power supply (1), a capacitor (20) connected to the output end of the converter (10), an inverter (30) connected to both ends of the capacitor (20), and a control device (100) for controlling the operation of the inverter (30). A control device (100) performs control to reduce the ripple of a capacitor output current outputted from a capacitor (20) to an inverter (30) during driving of a load.

Description

Power conversion device, motor drive device, and refrigeration cycle application device

Technical Field

The present invention relates to a power conversion device, a motor driving device, and a refrigeration cycle application apparatus that supply ac power to a motor that drives a load.

Background

The power conversion device includes a converter that rectifies a power supply voltage applied from an ac power supply, a capacitor connected to an output terminal of the converter, and an inverter that converts a dc voltage output from the capacitor into an ac voltage and applies the ac voltage to the motor.

The following patent document 1 discloses the following technology: torque pulsation, which is a pulsation component of load torque, is appropriately compensated according to a state of a motor driving the compressor, thereby suppressing an increase in power consumption.

Prior art literature

Patent literature

Patent document 1: japanese patent laid-open publication 2016-178814

Disclosure of Invention

Problems to be solved by the invention

In an air conditioner, which is one of application products of refrigeration cycle application equipment, in order to suppress a failure due to a harmonic component contained in a power supply current, restrictions concerning the harmonic of the power supply current are prescribed. For example, in japan, a limit value is defined for a higher harmonic of a power supply current by Japanese Industrial Standards (JIS).

However, in the technique described in patent document 1, harmonics of the power supply current are not considered. Therefore, when the technique of patent document 1 is used to generate a compensation component of torque ripple of the motor at a frequency that is not synchronized with the power supply frequency, there is a problem that the power supply current becomes unbalanced between positive and negative polarities thereof, and a harmonic component of the power supply current increases.

The present invention has been made in view of the above circumstances, and an object of the present invention is to provide a power conversion device capable of suppressing an increase in harmonic components of a power supply current.

Means for solving the problems

In order to solve the above-described problems and achieve the object, the power conversion device of the present invention is a power conversion device for supplying ac power to a motor that drives a load. The power conversion device has a converter that rectifies a power supply voltage applied from an ac power supply, and a capacitor connected to an output terminal of the converter. The power conversion device further includes an inverter connected to both ends of the capacitor, and a control device for controlling the operation of the inverter. The control device performs control to reduce ripple of the capacitor output current output from the capacitor to the inverter at the time of driving the load.

Effects of the invention

According to the power conversion device of the present invention, an effect is exhibited that an increase in harmonic components of a power supply current can be suppressed.

Drawings

Fig. 1 is a diagram showing a configuration example of a power conversion device according to embodiment 1.

Fig. 2 is a diagram showing a configuration example of an inverter included in the power conversion device according to embodiment 1.

Fig. 3 is a block diagram showing a configuration example of a control device included in the power conversion device according to embodiment 1.

Fig. 4 is a diagram 1 for explaining the problem of the present application.

Fig. 5 is a diagram of fig. 2 for explaining the problem of the present application.

Fig. 6 is a block diagram showing a configuration example of a voltage command value calculation unit included in the power conversion device according to embodiment 1.

Fig. 7 is a block diagram showing a configuration example of the delta-axis current command value generation unit included in the voltage command value calculation unit according to embodiment 1.

Fig. 8 is a block diagram showing a1 st configuration example of a capacitor output current control unit included in the voltage command value calculation unit according to embodiment 1.

Fig. 9 is a block diagram showing a2 nd configuration example of the capacitor output current control unit included in the voltage command value calculation unit according to embodiment 1.

Fig. 10 is a diagram for explaining the effect of pulsation reduction control in embodiment 1.

Fig. 11 is a diagram showing an example of a hardware configuration of a control device included in the power conversion device according to embodiment 1.

Fig. 12 is a diagram showing a configuration example of the refrigeration cycle application apparatus according to embodiment 2.

Detailed Description

Hereinafter, a power conversion device, a motor driving device, and a refrigeration cycle application apparatus according to an embodiment of the present invention will be described in detail with reference to the accompanying drawings.

Embodiment 1

Fig. 1 is a diagram showing a configuration example of a power conversion device 2 according to embodiment 1. Fig. 2 is a diagram showing a configuration example of an inverter 30 included in the power conversion device 2 according to embodiment 1. The power conversion device 2 is connected to the ac power source 1 and the compressor 8. The compressor 8 is an example of a load having a characteristic in which the load torque periodically fluctuates when driven. The compressor 8 has a motor 7. An example of the motor 7 is a three-phase permanent magnet synchronous motor. The power conversion device 2 converts a power supply voltage applied from the ac power supply 1 into an ac voltage having a desired amplitude and phase and applies the ac voltage to the motor 7. The power conversion device 2 includes a reactor 4, a converter 10, a capacitor 20, an inverter 30, a voltage detection unit 82, a current detection unit 84, and a control device 100. The motor driving device 50 is constituted by the electric power conversion device 2 and the motor 7 included in the compressor 8.

The converter 10 has 4 diodes D1, D2, D3, D4. The 4 diodes D1 to D4 are bridged to constitute a rectifying circuit. The converter 10 rectifies a power supply voltage applied from the ac power supply 1 by a rectifier circuit composed of 4 diodes D1 to D4. In the converter 10, one end of the input side is connected to the ac power supply 1 via the reactor 4, and the other end of the input side is connected to the ac power supply 1. In addition, in the converter 10, the output side is connected to the capacitor 20.

The converter 10 may have a rectifying function and a boosting function for boosting the rectifying voltage. The converter having the voltage boosting function may be configured to have 1 or more transistor elements in addition to or instead of the diode, or 1 or more switching elements in which the transistor elements and the diode are connected in anti-parallel. The arrangement and connection of the transistor elements and the switching elements in the converter having the step-up function are well known, and the description thereof is omitted.

The capacitor 20 is connected to the output of the converter 10 via dc buses 22a, 22 b. The dc bus 22a is a positive dc bus, and the dc bus 22b is a negative dc bus. The capacitor 20 smoothes the rectified voltage applied from the converter 10. As the capacitor 20, an electrolytic capacitor, a film capacitor, and the like are exemplified.

The inverter 30 is connected to both ends of the capacitor 20 via dc buses 22a and 22 b. The inverter 30 converts the dc voltage smoothed by the capacitor 20 into an ac voltage for the compressor 8, and applies the ac voltage to the motor 7 of the compressor 8. The voltage applied to the motor 7 is a 3-phase alternating voltage of variable frequency and voltage value.

As shown in fig. 2, the inverter 30 has an inverter main circuit 310 and a driving circuit 350. The inverter main circuit 310 includes switching elements 311 to 316. Rectifying elements 321 to 326 for current return are connected in anti-parallel to the switching elements 311 to 316, respectively.

In the inverter main circuit 310, IGBTs (Insulated Gate Bipolar Transistor: insulated gate bipolar transistors), MOSFETs (Metal-Oxide-Semiconductor Field-Effect Transistor: metal Oxide semiconductor field effect transistors), and the like are assumed as the switching elements 311 to 316, but any element may be used as long as it can perform switching. In addition, when the switching elements 311 to 316 are MOSFETs, since the MOSFETs have parasitic diodes in structure, the same effect can be obtained even if the rectifying elements 321 to 326 for current return are not connected in anti-parallel.

As a material for forming the switching elements 311 to 316, not only silicon (Si) but also silicon carbide (SiC), gallium nitride (GaN), diamond, or the like, which is a wide band gap semiconductor, may be used. By forming the switching elements 311 to 316 using a wide band gap semiconductor, loss can be further reduced.

The driving circuit 350 generates driving signals Sr1 to Sr6 based on PWM (Pulse Width Modulation: pulse width modulation) signals Sm1 to Sm6 output from the control device 100. The driving circuit 350 controls the on/off of the switching elements 311 to 316 by the driving signals Sr1 to Sr6. Thus, the inverter 30 can apply a 3-phase ac voltage with a variable frequency and a variable voltage to the motor 7 via the output lines 331 to 333.

The PWM signals Sm1 to Sm6 are signals having a signal level of a logic circuit, for example, a magnitude of 0V to 5V. The PWM signals Sm1 to Sm6 are signals for setting the ground potential of the control device 100 to the reference potential. On the other hand, the drive signals Sr1 to Sr6 are signals having voltage levels required for controlling the switching elements 311 to 316, for example, a magnitude of-15V to +15v. The drive signals Sr1 to Sr6 are signals in which the potential of the emitter terminal, which is the negative terminal of the switching element, corresponding to each other is set to the reference potential.

The voltage detection unit 82 detects the voltage across the capacitor 20, thereby detecting the bus voltage Vdc. The bus voltage Vdc is the voltage between the dc buses 22a, 22 b. The voltage detection unit 82 includes, for example, a voltage dividing circuit that divides a voltage by resistors connected in series. The voltage detection unit 82 converts the detected bus voltage Vdc into a voltage suitable for processing in the control device 100, for example, a voltage of 5V or less, using a voltage dividing circuit, and outputs the converted voltage to the control device 100 as a voltage detection signal, which is an analog signal. The voltage detection signal outputted from the voltage detection unit 82 to the control device 100 is converted from an analog signal to a digital signal by an AD (Analog to Digital: analog to digital) conversion unit (not shown) in the control device 100, and is used for internal processing in the control device 100.

The current detection unit 84 has a shunt resistor inserted into the dc bus 22 b. The current detection unit 84 detects the capacitor output current idc using a shunt resistor. The capacitor output current idc is an input current to the inverter 30, that is, a current input from the capacitor 20 to the inverter 30. The current detection unit 84 outputs the detected capacitor output current idc as a current detection signal, which is an analog signal, to the control device 100. The current detection signal output from the current detection unit 84 to the control device 100 is converted from an analog signal to a digital signal by an AD conversion unit, not shown, in the control device 100, and is used for internal processing in the control device 100.

The control device 100 generates the PWM signals Sm1 to Sm6 described above, and controls the operation of the inverter 30. Specifically, the control device 100 changes the angular frequency ωe and the voltage value of the output voltage of the inverter 30 according to the PWM signals Sm1 to Sm 6.

The angular frequency ωe of the output voltage of the inverter 30 defines the rotational angular velocity at the electrical angle of the motor 7. In the present specification, the rotational angular velocity is also denoted by the same reference symbol ωe. The rotational angular velocity ωm at the mechanical angle of the motor 7 is equal to the value obtained by dividing the rotational angular velocity ωe at the electrical angle of the motor 7 by the pole pair number P. Therefore, there is a relationship between the rotational angular velocity ωm at the mechanical angle of the motor 7 and the angular frequency ωe of the output voltage of the inverter 30, which is expressed by the following expression (1). In the present specification, the rotational angular velocity may be simply referred to as "rotational velocity", and the angular frequency may be simply referred to as "frequency".

ωm＝ωe/P…(1)

Next, the configuration of the control device 100 will be described. Fig. 3 is a block diagram showing a configuration example of the control device 100 included in the power conversion device 2 according to embodiment 1. The control device 100 includes an operation control unit 102 and an inverter control unit 110.

The operation control unit 102 receives command information Qe from the outside and generates a frequency command value ωe based on the command information Qe. As shown in the following expression (2), the frequency command value ωe can be obtained by multiplying the rotation speed command value ωm, which is a command value of the rotation speed of the motor 7, by the pole pair number P.

ωe*＝ωm*×P…(2)

When controlling an air conditioner as a refrigeration cycle application device, the control device 100 controls the operation of each unit of the air conditioner based on the instruction information Qe. The command information Qe is, for example, information indicating a temperature detected by a temperature sensor not shown, a set temperature indicated by a remote control as an operation unit not shown, selection information of an operation mode, instruction information of operation start and operation end, and the like. The operation modes include heating, cooling, and dehumidification. The operation control unit 102 may be located outside the control device 100. That is, the control device 100 may be configured to obtain the frequency command value ωe from the outside.

The inverter control unit 110 includes a current restoration unit 111, a 3-phase 2-phase conversion unit 112, a γ -axis current command value generation unit 113, a voltage command value calculation unit 115, an electric phase calculation unit 116, a 2-phase 3-phase conversion unit 117, and a PWM signal generation unit 118.

The current restoring unit 111 restores the phase currents iu, iv, iw flowing through the motor 7 based on the capacitor output current idc detected by the current detecting unit 84. The current restoring unit 111 can restore the phase currents iu, iv, iw by sampling the detected value of the capacitor output current idc detected by the current detecting unit 84 at a timing determined based on the PWM signals Sm1 to Sm6 generated by the PWM signal generating unit 118. Further, current detectors may be provided on the output lines 331 to 333, and the phase currents iu, iv, iw may be directly detected and input to the 3-phase 2-phase conversion unit 112. In this configuration, the current restoring unit 111 is not required.

The 3-phase 2-phase conversion unit 112 converts the phase currents iu, iv, iw restored by the current restoration unit 111 into current values of γ -axis current iγ as an excitation current and γ - δ -axis current iδ as a torque current, using an electric phase θe generated by an electric phase calculation unit 116 described later.

The γ -axis current command value generation unit 113 generates a γ -axis current command value iγ, which is an excitation current command value, from the δ -axis current iδ. More specifically, the γ -axis current command value generation unit 113 obtains a current phase angle at which the output torque of the motor 7 is equal to or greater than a set value or a maximum value from the δ -axis current iδ, and calculates a γ -axis current command value iγ from the obtained current phase angle. Instead of the output torque of the motor 7, a motor current flowing through the motor 7 may be used. In this case, the γ -axis current command value iγ is calculated from a current phase angle at which the motor current flowing through the motor 7 is equal to or less than a set value or a minimum value.

Fig. 3 shows a configuration in which the γ -axis current command value iγ is obtained from the δ -axis current iδ, but the configuration is not limited to this configuration. Instead of the δ -axis current iδ, a γ -axis current command value iγ may be obtained from the γ -axis current iγ. The γ -axis current command value generation unit 113 may determine the γ -axis current command value iγ by controlling the flux weakening.

The voltage command value calculation unit 115 generates a γ -axis voltage command value vγ and a δ -axis voltage command value vδ based on the frequency command value ωe obtained from the operation control unit 102, the γ -axis current iγ and the δ -axis current iδ obtained from the 3-phase 2-phase conversion unit 112, and the γ -axis current command value iγ obtained from the γ -axis current command value generation unit 113. Further, the voltage command value calculation unit 115 estimates the frequency estimation value ωest from the γ -axis voltage command value vγ, the δ -axis voltage command value vδ, the γ -axis current iγ, and the δ -axis current iδ.

The electric phase calculation unit 116 integrates the frequency estimation value ωest obtained from the voltage command value calculation unit 115, thereby calculating the electric phase θe.

The 2-phase 3-phase conversion unit 117 converts the voltage command values of the two-phase coordinate system, which are the gamma-axis voltage command values vγ and the delta-axis voltage command values vδ, obtained from the voltage command value calculation unit 115, into the 3-phase voltage command values Vu, vv, vw, which are the output voltage command values of the three-phase coordinate system, using the electrical phase θe obtained from the electrical phase calculation unit 116.

The PWM signal generation unit 118 compares the 3-phase voltage command values Vu, vv, vw obtained from the 2-phase 3-phase conversion unit 117 with the bus voltage Vdc detected by the voltage detection unit 82, thereby generating PWM signals Sm1 to Sm6. The PWM signal generation unit 118 can stop the motor 7 by not outputting the PWM signals Sm1 to Sm6.

Next, the reason why the problem of the present application occurs will be described. Fig. 4 and 5 are fig. 1 and 2, respectively, for explaining the object of the present application. The problems of the present application are briefly described in the section [ problems to be solved by the present application ], but are described in further detail herein.

First, when the load is a load having torque pulsation, such as a single-rotation compressor, a scroll compressor, or a double-rotation compressor, control to compensate for the torque pulsation is performed as described in the section of [ background art ]. This control is also referred to as "vibration suppression control". In general vibration suppression control, a torque current compensation value is generated so that the output torque of the motor 7 follows the torque pulsation, and the inverter 30 is controlled. However, if this control is simply performed, as described in the section of [ the problem to be solved by the invention ], the capacitor output current idc becomes unbalanced between the positive and negative power supply currents, and there is a problem that the harmonic component of the power supply current increases.

In fig. 4 and 5, waveforms of the power supply voltage Vin, the power supply current Iin, and the capacitor output current idc are shown in order from the upper part. The horizontal axis of fig. 4 and 5 represents time.

In the middle part of fig. 4, a state is shown in which the peak value of the waveform on the positive side and the peak value of the waveform on the negative side in the power supply current Iin are different, that is, the peak value is unbalanced between the positive and negative of the polarity of the power supply current Iin. When such unbalance occurs, as shown below, the capacitor output current idc pulsates. As a result, the power supply current Iin contains a large amount of harmonic components.

The present inventors have found that the larger the load torque and the smaller the inertia of the load, the larger the ripple of the capacitor output current idc, and that the larger the load torque, the larger the ripple of the capacitor output current idc is exhibited in the vibration suppression control. Further, the present inventors found that the ripple of the capacitor output current idc of the single rotary compressor is large compared to the double rotary compressor and the scroll compressor.

Further, an ideal state in which the capacitor output current idc is constant is shown in the lower part of fig. 5. In this ideal state, as shown in the middle part of fig. 5, the peak value of the waveform on the positive side and the peak value of the waveform on the negative side in the power supply current Iin are equal, and no imbalance between the positive and negative sides in the power supply current Iin is generated. Therefore, the harmonic component that may be included in the power supply current Iin is very small compared with the case of fig. 4.

As described above, the higher harmonic component that may be contained in the power supply current Iin is related to the ripple of the capacitor output current idc. Therefore, the voltage command value calculation unit 115 included in the control device 100 according to embodiment 1 performs control to reduce the ripple of the capacitor output current idc when the load is driven. The specific structure is shown in fig. 6 and 7.

Fig. 6 is a block diagram showing a configuration example of the voltage command value calculation unit 115 included in the control device 100 according to embodiment 1. As shown in fig. 6, the voltage command value calculation unit 115 includes a frequency estimation unit 501, subtraction units 502, 509, 510, a delta-axis current command value generation unit 503, a capacitor output current control unit 504, a gamma-axis current control unit 511, and a delta-axis current control unit 512. Fig. 7 is a block diagram showing a configuration example of the delta current command value generation unit 503 included in the voltage command value calculation unit 115 according to embodiment 1. Fig. 7 also illustrates a subtracting section 502 located in front of the delta-axis current command value generating section 503.

The frequency estimation unit 501 estimates the frequency of the voltage applied to the motor 7 from the γ -axis current iγ, the δ -axis current iδ, the γ -axis voltage command value vγ, and the δ -axis voltage command value vδ, and outputs the estimated frequency as a frequency estimation value ωest.

The subtracting unit 502 calculates a difference (ωe×ωest) between the frequency estimation value ωest estimated by the frequency estimating unit 501 and the frequency command value ωe.

The capacitor output current control unit 504 generates a delta-axis current compensation value iδ—lcc from the capacitor output current idc obtained from the current detection unit 84. The delta-axis current compensation value idelta_lcc is a component for reducing the control amount of the ripple component of the capacitor output current idc. The details of the delta-axis current compensation value idelta_lcc are described later. In the present specification, the capacitor output current control unit may be referred to simply as a "current control unit", and the δ -axis current compensation value may be referred to simply as a "current compensation value".

The delta-axis current command value generation unit 503 generates a delta-axis current command value iδ as a torque current command value in the rotation coordinate system. More specifically, the delta-axis current command value generation unit 503 performs proportional-integral operation, i.e., PI (Proportional Integral: proportional-integral) control on the difference (ωe×ωest) calculated by the subtraction unit 502, and obtains a delta-axis current command value iδ that brings the difference (ωe×ωest) to zero. The delta-axis current command value generating unit 503 generates and outputs a delta-axis current command value iδ obtained by correcting or compensating the delta-axis current command value iδ, based on the delta-axis current command value iδ and the delta-axis current compensation value iδ—lcc obtained from the capacitor output current control unit 504.

Fig. 7 shows an example of the structure of the delta current command value generation unit 503. As shown in fig. 7, the delta current command value generation unit 503 includes a speed control unit 610 and a compensation unit 620. The speed control unit 610 is a control unit that generates a current command value based on the frequency deviation. The speed control unit 610 includes a proportional control unit 611, an integral control unit 612, and an addition unit 613, and the compensation unit 620 includes an addition unit 621.

In the speed control unit 610, the proportion control unit 611 performs proportion control on the difference (ωe×ωest) between the frequency command value ωe and the frequency estimation value ωest obtained from the subtracting unit 502, and outputs a proportion term iδ—p. The integration control unit 612 performs integration control on the difference (ωe×ωest) between the frequency command value ωe and the frequency estimation value ωest obtained from the subtracting unit 502, and outputs an integral term iδ—i. The adder 613 adds the proportional term iδ_p obtained from the proportional controller 611 to the integral term iδ_i obtained from the integral controller 612, and generates a δ -axis current command value iδ. The adder 621 adds the delta-axis current command value iδ generated by the speed controller 610 and the delta-axis current compensation value iδ—lcc obtained from the capacitor output current controller 504 to generate a delta-axis current command value iδ. In the case of distinguishing between the delta-axis current command value iδ output from the speed control unit 610 and the delta-axis current command value iδ output from the delta-axis current command value generation unit 503, the delta-axis current command value iδ is referred to as a "1st delta-axis current command value", and the delta-axis current command value iδ is referred to as a "2nd delta-axis current command value".

As described above, the δ -axis current command value generation unit 503 performs control for matching the frequency estimation value ωest with the frequency command value ωe and suppressing the ripple of the capacitor output current idc.

Returning to fig. 6, the subtracting unit 509 performs a difference (iγ×iγ) between the γ -axis current iγ and the γ -axis current command value iγ. The subtracting unit 510 calculates a difference (iδ—iδ) between the δ -axis current iδ and the δ -axis current command value iδ.

The gamma-axis current control unit 511 performs a proportional-integral operation on the difference (iγ×iγ) calculated by the subtraction unit 509, and generates a gamma-axis voltage command value vγ that approximates the difference (iγ×iγ) to zero. The γ -axis current control unit 511 generates such a γ -axis voltage command value vγ, and thereby performs control such that the γ -axis current iγ matches the γ -axis current command value iγ.

The delta-axis current control unit 512 performs a proportional-integral operation on the difference (iδ—iδ) calculated by the subtraction unit 510, and generates a delta-axis voltage command value vδ that approximates the difference (iδ—iδ) to zero. The delta-axis current control unit 512 generates such a delta-axis voltage command value vδ, and thereby performs control such that the delta-axis current iδ matches the delta-axis current command value iδ. As described above, the delta-axis current command value iδ input to the delta-axis current control unit 512 includes the delta-axis current compensation value iδ—lcc obtained from the capacitor output current control unit 504. Therefore, the delta-axis current control unit 512 controls the inverter 30 based on the delta-axis voltage command value vδ generated based on the delta-axis current compensation value iδ—lcc, whereby the ripple of the capacitor output current idc can be suppressed.

Next, a structure of the capacitor output current control unit 504 will be described. Fig. 8 is a block diagram showing the 1 st configuration example of the capacitor output current control unit 504 included in the voltage command value calculation unit 115 according to embodiment 1. The capacitor output current control unit 504 includes an arithmetic unit 550, a cosine arithmetic unit 551, a sine arithmetic unit 552, multiplication units 553, 554, 561, 562, low-pass filters 555, 556, subtraction units 557, 558, frequency control units 559, 560, and an addition unit 563.

The calculation unit 550 calculates a phase angle θx related to the frequency ωx of the pulse component of interest in the capacitor output current idc. The ripple component of interest is a ripple component to be reduced among a plurality of ripple components included in the capacitor output current idc. The phase angle θx is determined by the frequency ωx of the pulsation component to be reduced, the clock frequency of the processor performing the processing, and the like.

The cosine calculating unit 551 calculates a cosine value cos θx from the phase angle θx. The sine calculation unit 552 calculates a sine value sin θx from the phase angle θx.

The multiplication unit 553 multiplies the capacitor output current idc by the cosine value cos θx, and calculates the cosine component idc·cos θx of the capacitor output current idc. The multiplication unit 554 multiplies the capacitor output current idc by a sine value sin θx to calculate a sine component idc·sin θx of the capacitor output current idc. The cosine component idc·cos θx and the sine component idc·sin θx calculated by the multiplying units 553 and 554 include a harmonic component, which is a pulse component having a frequency higher than ωx, in addition to a pulse component having a frequency ωx.

The low-pass filters 555, 556 are first-order lag filters whose transfer function is represented by 1/(1+s·tf). s is the Laplacian and Tf is the time constant. The time constant Tf is determined to remove the pulsation component of a frequency higher than the frequency ωx. In addition, "removing" includes a case where a part of the pulsation component is attenuated, i.e., reduced. The time constant Tf may be set by the operation control unit 102 according to the speed command value, and the operation control unit 102 may notify the low-pass filters 555 and 556 of the time constant Tf, or may be held by the low-pass filters 555 and 556. The low-pass filters 555 and 556 are examples of the first-order lag filter, and may be a moving average filter or the like, as long as the ripple component on the high-frequency side can be removed, and the type of filter is not limited.

The low-pass filter 555 performs low-pass filtering on the cosine component idc·cos θx, removes a component having a frequency higher than the frequency ωx, and outputs a low-frequency component idc_c. The low frequency component idc_c is a cosine component of the frequency ωx out of the ripple component of the capacitor output current idc.

The low-pass filter 556 performs low-pass filtering on the sinusoidal component idc·sin θx, removes a component having a frequency higher than the frequency ωx, and outputs a low-frequency component idc_s. The low frequency component idc_s is a sinusoidal component of frequency ωx among the ripple component of the capacitor output current idc.

The subtracting unit 557 calculates a difference (idc_c-0) between the low frequency component idc_c output from the low pass filter 555 and zero. The subtracting unit 558 calculates a difference (idc_s-0) between the low frequency component idc_s output from the low pass filter 556 and zero.

The frequency control unit 559 integrates the difference (idc_c-0) calculated by the subtraction unit 557, and calculates a cosine component iδ_trq_c of the current command value for bringing the difference (idc_c-0) to zero. By generating the cosine component iδ_trq_c in this way, the frequency control unit 559 performs control for matching the low frequency component idc_c with zero. The integration operation is an example, and the proportional-integral operation may be performed instead of the integration operation.

The frequency control unit 560 integrates the difference (idc_s-0) calculated by the subtraction unit 558, and calculates a sinusoidal component iδ_trq_s of the current command value that brings the difference (idc_s-0) to zero. By generating the sinusoidal component iδ_trq_s in this way, the frequency control unit 560 performs control for matching the low frequency component idc_s with zero. The integration operation is an example, and the proportional-integral operation may be performed instead of the integration operation.

The multiplying unit 561 multiplies the cosine component iδ_trq_c outputted from the frequency control unit 559 by the cosine value cos θx to generate iδ_trq_c·cos θx. The multiplication unit 562 multiplies the sinusoidal component iδ_trq_s outputted from the frequency control unit 560 by a sinusoidal value sin θx to generate iδ_trq_s·sin θx.

The adder 563 obtains the sum of iδ_trq_c·cos θx output from the multiplier 561 and iδ_trq_s·sin θx output from the multiplier 562. The capacitor output current control unit 504 outputs the value obtained by the adder 563 as a delta-axis current compensation value iδ—lcc.

As described above, the control device 100 according to embodiment 1 performs control to reduce the ripple of the capacitor output current idc output from the capacitor 20 to the inverter 30 during the driving of the compressor 8. By this control, the power supply current Iin can be prevented from becoming unbalanced between positive and negative polarities. This can suppress an increase in the harmonic component that may be contained in the power supply current Iin.

Next, a method of reducing a ripple component, particularly, a dominant frequency, from among a plurality of ripple components included in the capacitor output current idc will be described.

First, motor power, which is electric power applied from the inverter 30 to the motor 7, is denoted by Pm. The motor power Pm can be expressed by the following expression (3).

The meaning of the symbol represented by the above formula (3) is as follows.

Vγ: gamma axis voltage in motor 7

Vδ: delta-axis voltage in motor 7

Iγ: gamma-axis current flowing through motor 7

Iδ: delta-axis current flowing through motor 7

Ra: phase resistance in motor 7

Ωe: frequency (electric angle) of output voltage of inverter 30

Lγ: gamma axis inductance in motor 7

Lδ: delta-axis inductance in motor 7

Induced voltage constant in motor 7

In the above equation (3), the delta-axis current contributes more to the motor torque than the gamma-axis current. Therefore, the following expression (4) is obtained when the term of the γ -axis current iγ, that is, the term of the reluctance torque is ignored, considering that the influence of the γ -axis current iγ is small.

When the electric power supplied from the capacitor 20 to the inverter 30 is represented by Pdc, pm=pdc can be considered. Therefore, according to the above expression (4), the capacitor output current idc can be expressed by the following expression (5).

Here, consider a motor load in which 1 load torque pulsation occurs in 1 cycle of the mechanical angle of the motor 7. Examples of such motor loads are single-rotation compressors, scroll compressors. When such a motor load is driven, the most dominant frequency component among the ripple components included in the capacitor output current idc is the mechanical 1f component. The mechanical 1f component is a component 1 times the mechanical angular frequency of the motor 7, that is, the mechanical angular frequency.

In addition, although not dominant as in the machine 1f component, the pulsation component of the power supply frequency, which is a frequency depending on the mechanical angular frequency of the motor 7 and the power supply voltage Vin, also becomes large. The frequency of the pulsation component can be expressed by the following expression (6).

Power frequency x m-mechanical angular frequency x n |

(M is an integer of 0 or more, n is a positive integer)

…(6)

Here, the dominant component of the bus voltage Vdc is a component 2 times the power supply frequency. This component is denoted as "power supply 2f". Therefore, when the component 1f is excluded, the following components become components with large pulsation. In addition, the symbol "|a|" represents the absolute value of the numerical value a.

Machinery 2f (m=0, n=2)

Power supply 2 f-machine 1f (m=2, n=1)

Power supply 2 f-machine 2f (m=2, n=2)

In addition, in the capacitor output current idc, components contributing to the influence of higher harmonics of the power supply current Iin are components having large absolute values and components having low frequencies.

When the rotation speed becomes high, the machine 2f component becomes large, and when the rotation speed becomes low, the i power source 2 f-machine 1f component becomes small. As described above, the lower the frequency is, the larger the influence of the higher harmonic contributing to the power supply current Iin is, and therefore, when there is a limit in the number of pulsation components to be reduced, the frequency components to be reduced are changed according to the rotation speed. For example, when the number of pulsation components to be reduced is 2, the control device 100 reduces the following pulsation components.

The rotation speed is lower than the threshold speed: component 1f and component 2f

The rotation speed is higher than the threshold speed: machine 1f component, |power supply 2 f-machine 1 f|component

Fig. 8 shows a configuration example of the capacitor output current control unit 504 in the case where the number of ripple components to be reduced is 1, but the same configuration can be made in the case where the number of ripple components to be reduced is 2 or more. Fig. 9 is a block diagram showing a2 nd configuration example of the capacitor output current control unit 504 included in the voltage command value calculation unit 115 according to embodiment 1. Fig. 9 shows a configuration example in which the number of pulsation components to be reduced is 3, as an example.

In fig. 9, the same or equivalent components as those in fig. 8 are denoted by the same reference numerals, and the components are denoted by simplified symbols or characters, and some of the components are omitted from illustration. In fig. 9, "m1f" represents "machine 1f", "m2f" represents "machine 2f", and "|in2f—m1f|" represents "|power supply 2 f-machine 1f|".

In the upper control system of fig. 9, a1 st δ -axis current compensation value iδm1f is calculated from the phase angle θm1f of the pulsation component of the machine 1 f. In the control system in the middle of fig. 9, the 2δ -axis current compensation value iδm2f is calculated from the phase angle θm2f of the pulsation component of the machine 2 f. In the lower control system of fig. 9, a 3 rd δ -axis current compensation value iδy (=iδ _|in2f-m1f|) is calculated from the phase angle θy (=θ _|in2f-m1f|) of the pulsating component of the |power source 2 f-machine 1 f|. These 1 st δ -axis current compensation value iδm1f, 2 nd δ -axis current compensation value iδm2f, and 3 rd δ -axis current compensation value iδy are added by the adder 564 and output as δ -axis current compensation value iδ—lcc.

Fig. 9 shows a case where the number of pulsation components to be reduced is 3, but is not limited to 3. Even when the number of pulsation components to be reduced is 2 or 4 or more, the control system for generating the δ -axis current compensation value iδ—lcc can be configured in parallel and added at the final stage, as in fig. 9.

Fig. 10 is a diagram for explaining the effect of pulsation reduction control in embodiment 1. The waveforms of the power supply current and the capacitor output current in the case where the capacitor output current control unit 504 is not provided in the voltage command value calculation unit 115 in fig. 6 are shown in the left part of fig. 10. In addition, waveforms of the power supply current and the capacitor output current in the case where the capacitor output current control unit 504 is shown in the right part of fig. 10, that is, in the case where the voltage command value calculation unit 115 of fig. 6 is used.

In the case where the capacitor output current control unit 504 is not provided, as shown in the left part of fig. 10, the ripple of the capacitor output current increases. This shows peak variation of the power supply current and increases the harmonic component contained in the power supply current. In contrast, when the capacitor output current control unit 504 is provided, the ripple of the capacitor output current is reduced as shown in the right part of fig. 10. This shows that the peak value of the power supply current is substantially constant and that the harmonic component contained in the power supply current is reduced.

As described above, the power conversion device according to embodiment 1 performs control to reduce ripple of the capacitor output current output from the capacitor to the inverter at the time of driving the load. By this control, it is possible to avoid the power supply current from becoming unbalanced between positive and negative of its polarity. This can suppress an increase in harmonic components that may be contained in the power supply current. Further, since an imbalance state between positive and negative in the power supply current is suppressed, it is easy to adapt to the power supply harmonic standard. Thus, it is not necessary to change or modify the circuit constant of the converter and the switching method of the converter, and therefore, a motor drive device with low cost and high reliability can be obtained. In addition, since the power factor of the power supply is also increased by reducing the power supply harmonics, it is not necessary to flow an unnecessary current. This can increase the efficiency of the inverter side, and a motor drive device with high efficiency can be obtained.

In the above control, when there is a limit to the number of pulsation components to be reduced, the control device preferably performs control to reduce the pulsation component that depends at least on the mechanical angular frequency of the motor, among the pulsation components included in the capacitor output current. Further, among the pulsation components depending on the mechanical angular frequency, the control device more preferably reduces the 1 st pulsation component particularly generated due to the mechanical angular frequency. The 1 st pulsation component is dominant in the case where the rotational speed of the motor is low, medium and high. Therefore, if the 1 st ripple component can be reduced, it is possible to greatly contribute to reduction of the higher harmonics of the power supply current. An example of a1 st pulsation component is the machine 1f component. Further, as the pulsation component to be reduced, a 2 nd pulsation component generated by a frequency 2 times the mechanical angular frequency may be added. An example of a 2 nd pulse component is the mechanical 2f component.

In the above control, when there is a limit to the number of pulsation components to be reduced, the control device preferably performs control to reduce at least 1 of the pulsation components depending on both the power supply frequency and the mechanical angular frequency, in addition to the 1 st pulsation component. One of the pulsation components depending on both the power supply frequency and the mechanical angular frequency is a3 rd pulsation component generated due to the absolute value of the frequency of the difference between the frequency of 2 times the power supply frequency and the mechanical angular frequency. The other of the pulse components depending on both the power supply frequency and the mechanical angular frequency is a4 th pulse component generated by the absolute value of the frequency of the difference between the frequency of 2 times the power supply frequency and the frequency of 2 times the mechanical angular frequency. An example of the 3 rd pulse component is the |power supply 2 f-machine 1f| component, and an example of the 4 th pulse component is the |power supply 2 f-machine 2f| component.

In the control described above, the control device preferably changes the pulsation component to be reduced according to the mechanical angular frequency. The magnitude relation between the 2 nd pulse component and the 4 th pulse component is reversed according to the mechanical angular frequency. Therefore, if the pulsation component to be reduced is changed according to the mechanical angular frequency, efficient control with high reduction effect can be performed even in the case where the scale of the control system is small.

In the control described above, the control device preferably performs control to preferentially reduce the pulse components having lower frequencies for the 2 nd pulse component, the 3 rd pulse component, and the 4 th pulse component. As described above, the component contributing to the large influence of the higher harmonic of the power supply current is a pulsation component having a low frequency. Therefore, if the control is performed to preferentially reduce the pulsation component having a lower frequency, the efficient control with a high reduction effect can be performed even when the scale of the control system is small.

Next, a hardware configuration of the control device 100 included in the power conversion device 2 will be described. Fig. 11 is a diagram showing an example of a hardware configuration of a control device 100 included in the power conversion device 2 according to embodiment 1. The control device 100 is implemented by a processor 201 and a memory 202.

The Processor 201 is a CPU (Central Processing Unit: central processing unit, also called a central processing unit, a processing unit, an arithmetic unit, a microprocessor, a microcomputer, a Processor, a DSP (DIGITAL SIGNAL Processor: digital signal Processor)) or a system LSI (LARGE SCALE Integration: large scale Integration). The Memory 202 can be a nonvolatile or volatile semiconductor Memory such as RAM (Random Access Memory: random access Memory), ROM (Read Only Memory), flash Memory, EPROM (Erasable Programmable Read Only Memory: erasable programmable Read Only Memory), EEPROM (registered trademark) (ELECTRICALLY ERASABLE PROGRAMMABLE READ-Only Memory). The memory 202 is not limited to these, and may be a magnetic disk, an optical disk, a high-density disk, a mini disk, or a DVD (DIGITAL VERSATILE DISC: digital versatile disk).

Embodiment 2

Fig. 12 is a diagram showing a configuration example of a refrigeration cycle application apparatus 900 according to embodiment 2. The refrigeration cycle application apparatus 900 of embodiment 2 has the power conversion device 2 described in embodiment 1. The refrigeration cycle application apparatus 900 according to embodiment 2 can be applied to a product having a refrigeration cycle, such as an air conditioner, a refrigerator, a freezer, and a heat pump water heater. In fig. 12, the same reference numerals as those in embodiment 1 are given to the components having the same functions as those in embodiment 1.

The refrigeration cycle apparatus 900 is provided with a compressor 901 incorporating the motor 7 according to embodiment 1, a four-way valve 902, an indoor heat exchanger 906, an expansion valve 908, and an outdoor heat exchanger 910 via a refrigerant pipe 912.

A compression mechanism 904 that compresses a refrigerant and a motor 7 that operates the compression mechanism 904 are provided inside the compressor 901.

The refrigeration cycle application apparatus 900 can perform a heating operation or a cooling operation by switching operation of the four-way valve 902. The compression mechanism 904 is driven by a variable speed controlled motor 7.

In the heating operation, as shown by solid arrows, the refrigerant is pressurized by the compression mechanism 904, sent out, and returned to the compression mechanism 904 through the four-way valve 902, the indoor heat exchanger 906, the expansion valve 908, the outdoor heat exchanger 910, and the four-way valve 902.

In the cooling operation, as indicated by the broken-line arrows, the refrigerant is pressurized by the compression mechanism 904, sent out, and returned to the compression mechanism 904 through the four-way valve 902, the outdoor heat exchanger 910, the expansion valve 908, the indoor heat exchanger 906, and the four-way valve 902.

During the heating operation, the indoor heat exchanger 906 functions as a condenser to release heat, and the outdoor heat exchanger 910 functions as an evaporator to absorb heat. During cooling operation, the outdoor heat exchanger 910 functions as a condenser to release heat, and the indoor heat exchanger 906 functions as an evaporator to absorb heat. The expansion valve 908 decompresses and expands the refrigerant.

The configuration described in the above embodiment shows an example, and the configuration can be combined with other known techniques, and a part of the configuration can be omitted or changed without departing from the gist.

Description of the reference numerals

1: An alternating current power supply; 2: a power conversion device; 4: a reactor; 7: a motor; 8: a compressor; 10: a converter; 20: a capacitor; 22a, 22b: a direct current bus; 30: an inverter; 50: a motor driving device; 82: a voltage detection unit; 84: a current detection unit; 100: a control device; 102: an operation control unit; 110: an inverter control unit; 111: a current restoration unit; 112: a 3-phase 2-phase conversion unit; 113: a gamma-axis current command value generation unit; 115: a voltage command value calculation unit; 116: an electric phase operation unit; 117: a 2-phase 3-phase conversion unit; 118: a PWM signal generation unit; 201: a processor; 202: a memory; 310: an inverter main circuit; 311 to 316: a switching element; 321-326: a rectifying element; 331-333: an output line; 350: a driving circuit; 501: a frequency estimation unit; 502. 509, 510, 557, 558: a subtracting section; 503: a delta-axis current command value generation unit; 504: a capacitor output current control unit; 511: a gamma-axis current control unit; 512: a delta-axis current control unit; 550: an arithmetic unit; 551: a cosine operation unit; 552: a sine calculation unit; 553. 554, 561, 562: a multiplication unit; 555. 556, are formed: a low pass filter; 559. 560: a frequency control unit; 563. 564, 613, 621: an addition unit; 610: a speed control unit; 611: a proportion control part; 612: an integral control unit; 620: a compensation unit; 900: refrigeration cycle application equipment; 901: a compressor; 902: a four-way valve; 904: a compression mechanism; 906: an indoor heat exchanger; 908: an expansion valve; 910: an outdoor heat exchanger; 912: refrigerant piping; d1, D2, D3, D4: a diode.

Claims (9)

1. A power conversion device that supplies ac power to a motor that drives a load, wherein the power conversion device has:

A converter that rectifies a power supply voltage applied from an ac power supply;

a capacitor connected to an output terminal of the converter;

an inverter connected to both ends of the capacitor; and

A control device for controlling the operation of the inverter,

The control device performs control to reduce pulsation of a capacitor output current output from the capacitor to the inverter at the time of driving the load.

2. The power conversion device according to claim 1, wherein,

The control device performs control to reduce a pulsation component that depends on a mechanical angular frequency of the motor, among pulsation components included in the capacitor output current.

3. The power conversion device according to claim 2, wherein,

The control device reduces a pulsation component that depends on both a power supply frequency, which is a frequency of the power supply voltage, and the mechanical angular frequency.

4. The power conversion device according to claim 3, wherein,

The 1 st pulsation component generated by the mechanical angular frequency and the 2 nd pulsation component generated by the frequency 2 times of the mechanical angular frequency are contained in the pulsation component depending on the mechanical angular frequency,

The 3 rd pulse component generated by the absolute value of the frequency of the difference between the frequency of the power frequency and the mechanical angular frequency and the 4 th pulse component generated by the absolute value of the frequency of the difference between the frequency of the power frequency and the frequency of the 2 times of the mechanical angular frequency are included in the pulse components depending on both the power frequency and the mechanical angular frequency,

The control device reduces the 1 st pulse component and reduces at least 1 pulse component of the 2 nd pulse component, the 3 rd pulse component, and the 4 th pulse component.

5. The power conversion device according to claim 4, wherein,

The control device changes the pulsation component to be reduced according to the mechanical angular frequency.

6. The power conversion device according to claim 4 or 5, wherein,

The control device controls the pulse components having a low frequency to be preferentially reduced for the 2 nd pulse component, the 3 rd pulse component, and the 4 th pulse component.

7. The power conversion apparatus according to any one of claims 2 to 6, wherein,

The control device comprises:

a speed control unit that generates a current command value based on the frequency deviation; and

A current control unit that generates a current compensation value for reducing the ripple component,

The current compensation value is superimposed on a current command value output from the speed control unit.

8. A motor drive apparatus having the power conversion apparatus according to any one of claims 1 to 7.

9. A refrigeration cycle application apparatus having the power conversion device according to any one of claims 1 to 7.