CN115214394B - Electric vehicle dynamic wireless charging system and lateral offset power fluctuation suppression method - Google Patents
Electric vehicle dynamic wireless charging system and lateral offset power fluctuation suppression method Download PDFInfo
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- B60L53/00—Methods of charging batteries, specially adapted for electric vehicles; Charging stations or on-board charging equipment therefor; Exchange of energy storage elements in electric vehicles
- B60L53/10—Methods of charging batteries, specially adapted for electric vehicles; Charging stations or on-board charging equipment therefor; Exchange of energy storage elements in electric vehicles characterised by the energy transfer between the charging station and the vehicle
- B60L53/12—Inductive energy transfer
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- B—PERFORMING OPERATIONS; TRANSPORTING
- B60—VEHICLES IN GENERAL
- B60L—PROPULSION OF ELECTRICALLY-PROPELLED VEHICLES; SUPPLYING ELECTRIC POWER FOR AUXILIARY EQUIPMENT OF ELECTRICALLY-PROPELLED VEHICLES; ELECTRODYNAMIC BRAKE SYSTEMS FOR VEHICLES IN GENERAL; MAGNETIC SUSPENSION OR LEVITATION FOR VEHICLES; MONITORING OPERATING VARIABLES OF ELECTRICALLY-PROPELLED VEHICLES; ELECTRIC SAFETY DEVICES FOR ELECTRICALLY-PROPELLED VEHICLES
- B60L53/00—Methods of charging batteries, specially adapted for electric vehicles; Charging stations or on-board charging equipment therefor; Exchange of energy storage elements in electric vehicles
- B60L53/10—Methods of charging batteries, specially adapted for electric vehicles; Charging stations or on-board charging equipment therefor; Exchange of energy storage elements in electric vehicles characterised by the energy transfer between the charging station and the vehicle
- B60L53/12—Inductive energy transfer
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- B—PERFORMING OPERATIONS; TRANSPORTING
- B60—VEHICLES IN GENERAL
- B60L—PROPULSION OF ELECTRICALLY-PROPELLED VEHICLES; SUPPLYING ELECTRIC POWER FOR AUXILIARY EQUIPMENT OF ELECTRICALLY-PROPELLED VEHICLES; ELECTRODYNAMIC BRAKE SYSTEMS FOR VEHICLES IN GENERAL; MAGNETIC SUSPENSION OR LEVITATION FOR VEHICLES; MONITORING OPERATING VARIABLES OF ELECTRICALLY-PROPELLED VEHICLES; ELECTRIC SAFETY DEVICES FOR ELECTRICALLY-PROPELLED VEHICLES
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- B60L53/60—Monitoring or controlling charging stations
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
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Abstract
本发明提供一种电动汽车动态无线充电系统及横向偏移功率波动抑制方法,系统包括电能发射端和电能接收端,电能发射端采用多段能量发射导轨逐一并排设置,每一段能量发射导轨包括沿第一方向绕制的能量发射线圈和与能量发射线圈串联且沿第一方向相反方向绕制的原边补偿线圈,电能接收端包括能量接收线圈和可变集成补偿电感,可变集成补偿电感包括作为副边补偿网络的交流电感线圈和用于改变交流电感线圈等效自感的两个直流控制绕组,两个直流控制绕组按照相反的绕向分别设置在所述交流电感线圈的内外两侧。其效果是:在实现导轨切换时功率波动抑制的基础上进一步提高横向抗偏移性,使用较小的直流电流实现了横向偏移时系统输出功率跌落的补偿。
The invention provides a dynamic wireless charging system for an electric vehicle and a method for suppressing lateral offset power fluctuations. The system includes an electric energy transmitting end and an electric energy receiving end. An energy transmitting coil wound in one direction and a primary side compensation coil wound in series with the energy transmitting coil and opposite to the first direction, the power receiving end includes an energy receiving coil and a variable integrated compensation inductance, and the variable integrated compensation inductance includes as The AC inductance coil of the secondary side compensation network and two DC control windings for changing the equivalent self-inductance of the AC inductance coil, the two DC control windings are respectively arranged on the inner and outer sides of the AC inductance coil according to opposite winding directions. The effect is: on the basis of suppressing power fluctuations during guide rail switching, the lateral offset resistance is further improved, and the compensation for system output power drops during lateral offsets is realized by using a small DC current.
Description
技术领域Technical Field
本发明涉及无线电能传输技术,具体涉及一种电动汽车动态无线充电系统及横向偏移功率波动抑制方法。The present invention relates to wireless power transmission technology, and in particular to a dynamic wireless charging system for electric vehicles and a method for suppressing lateral offset power fluctuations.
背景技术Background Art
无线电能传输(wireless power transfer,WPT)通过磁场、电场、激光及微波等媒介,实现电能的非电气接触传输。该技术可以有效地解决传统有线取电方式引起的设备灵活性受限和安全隐患的问题。目前,在电动汽车、消费电子产品、家用电器等应用领域,国内外的专家学者对该技术已展开了研究并获得了较多的理论成果。Wireless power transfer (WPT) achieves non-electrical contact transmission of electrical energy through media such as magnetic fields, electric fields, lasers and microwaves. This technology can effectively solve the problems of limited equipment flexibility and safety hazards caused by traditional wired power supply methods. At present, domestic and foreign experts and scholars have conducted research on this technology in application fields such as electric vehicles, consumer electronics, and household appliances, and have obtained many theoretical results.
针对电动汽车而言,为了实现其在行驶过程中的持续无线供电,目前已有人提出了电动汽车动态无线充电系统,通过采用多段短导轨形式的无线能量发生线圈实现无线能量动态传输。同时,针对动态充电过程中的功率波动问题,本研究团队曾提出了一种通过磁集成抑制功率波动的电动汽车动态无线充电系统,具体见中国发明专利:202111524549.7,但是在使用过程中发现,电动汽车在充电道路上行驶过程中,不可避免地会出现横向偏移,且往往会持续在某个横向偏移的状态下沿道路持续移动,测试发现,虽然现有技术针对导轨的切换所引起的功率跌落实现了功率补偿,但是在横向偏移下输出功率仍然有所下降,如图1所示,横向偏移y=5cm和y=10cm情况下,虽然在接收线圈位置移动过程中,等效互感基本维持稳定,但是在不同的偏移情况下,其等效互感仍会发生变化,且偏移越远,等效互感越低,因此,还需要继续研究一种可以实现横向偏移功率波动抑制的系统及方法。For electric vehicles, in order to achieve continuous wireless power supply during driving, some people have proposed a dynamic wireless charging system for electric vehicles, which realizes dynamic wireless energy transmission by adopting wireless energy generating coils in the form of multiple short rails. At the same time, in response to the problem of power fluctuations during dynamic charging, this research team has proposed a dynamic wireless charging system for electric vehicles that suppresses power fluctuations through magnetic integration. For details, see Chinese invention patent: 202111524549.7. However, it is found during use that electric vehicles will inevitably have lateral displacements while driving on the charging road, and they often continue to move along the road in a certain lateral displacement state. Tests have found that although the prior art has achieved power compensation for the power drop caused by the switching of the rails, the output power still decreases under lateral displacement. As shown in Figure 1, under the lateral displacement of y=5cm and y=10cm, although the equivalent mutual inductance basically remains stable during the movement of the receiving coil position, its equivalent mutual inductance will still change under different displacement conditions, and the farther the displacement, the lower the equivalent mutual inductance. Therefore, it is necessary to continue to study a system and method that can suppress lateral displacement power fluctuations.
发明内容Summary of the invention
为了解决横向偏移状态下导轨切换引起的功率跌落问题,本发明的首要目的在于提出一种电动汽车动态无线充电系统,使其能够在现有基础上,实现电动汽车横向偏移所引起的功率波动抑制。In order to solve the power drop problem caused by rail switching in a lateral offset state, the primary purpose of the present invention is to propose a dynamic wireless charging system for electric vehicles, so that it can suppress power fluctuations caused by lateral offset of electric vehicles on the existing basis.
为了实现上述目的,本发明所采用的具体技术方案如下:In order to achieve the above object, the specific technical solutions adopted by the present invention are as follows:
一种电动汽车动态无线充电系统,包括电能发射端和电能接收端,其关键在于,所述电能发射端采用多段能量发射导轨逐一并排设置,每一段能量发射导轨包括沿第一方向绕制的能量发射线圈和与所述能量发射线圈串联且沿第一方向相反方向绕制的原边补偿线圈,所述电能接收端包括能量接收线圈和可变集成补偿电感,所述可变集成补偿电感包括作为副边补偿网络的交流电感线圈和用于改变所述交流电感线圈等效自感的两个直流控制绕组,所述两个直流控制绕组按照相反的绕向分别设置在所述交流电感线圈的内外两侧。A dynamic wireless charging system for electric vehicles comprises an electric energy transmitting end and an electric energy receiving end, wherein the electric energy transmitting end adopts a plurality of sections of energy transmitting rails arranged side by side one by one, each section of the energy transmitting rail comprises an energy transmitting coil wound along a first direction and a primary compensation coil connected in series with the energy transmitting coil and wound in a direction opposite to the first direction, the electric energy receiving end comprises an energy receiving coil and a variable integrated compensation inductor, the variable integrated compensation inductor comprises an AC inductor coil as a secondary compensation network and two DC control windings for changing the equivalent self-inductance of the AC inductor coil, and the two DC control windings are respectively arranged on the inner and outer sides of the AC inductor coil in opposite winding directions.
可选地,在两个直流控制绕组上加载有直流电源,其中内侧直流控制绕组与外侧直流控制绕组的流经的电流所产生的磁场方向相反。Optionally, a DC power source is loaded on the two DC control windings, wherein the magnetic fields generated by the currents flowing through the inner DC control winding and the outer DC control winding are in opposite directions.
可选地,所述内侧直流控制绕组与所述外侧直流控制绕组的匝数相同。Optionally, the inner DC control winding and the outer DC control winding have the same number of turns.
可选地,相邻两段能量发射导轨之间按第一预设值设置有水平间距。Optionally, a horizontal spacing is set between two adjacent sections of the energy emission rail according to a first preset value.
可选地,所述电能发射端设置有原边LCC谐振补偿网络,在所述能量发射线圈上还设置有原边串联谐振电容。Optionally, the electric energy transmitting end is provided with a primary LCC resonant compensation network, and a primary series resonant capacitor is also provided on the energy transmitting coil.
可选地,所述能量发射线圈按矩形线圈水平绕制,所述原边补偿线圈按同一平面绕制在所述能量发射线圈的内部,所述能量接收线圈按矩形线圈水平绕制,其外轮廓的宽度与所述能量发射线圈宽度相同,所述可变集成补偿电感中的交流电感线圈和两个直流控制绕组按同一平面绕制在所述能量接收线圈的内部。Optionally, the energy transmitting coil is horizontally wound as a rectangular coil, the primary compensation coil is wound inside the energy transmitting coil in the same plane, the energy receiving coil is horizontally wound as a rectangular coil, the width of its outer contour is the same as the width of the energy transmitting coil, and the AC inductance coil and two DC control windings in the variable integrated compensation inductor are wound inside the energy receiving coil in the same plane.
可选地,在所述能量发射线圈和所述原边补偿线圈的底面设置有原边磁芯板,所述能量接收线圈和所述可变集成补偿电感的顶面设置有副边磁芯板和屏蔽板。Optionally, a primary magnetic core plate is provided on the bottom surface of the energy transmitting coil and the primary compensation coil, and a secondary magnetic core plate and a shielding plate are provided on the top surface of the energy receiving coil and the variable integrated compensation inductor.
基于上述系统,本发明还提出一种电动汽车动态无线充电系统的横向偏移功率波动抑制方法,其关键在于,包括以下步骤:Based on the above system, the present invention also proposes a method for suppressing lateral offset power fluctuation of a dynamic wireless charging system for an electric vehicle, the key of which is that it includes the following steps:
S1:检测电动汽车横向偏移量;S1: Detect the lateral deviation of the electric vehicle;
S2:根据横向偏移量确定第一互感系数的跌落情况,所述第一互感系数为相邻两段能量发射线圈与能量接收线圈之间互感系数;S2: determining a drop in a first mutual inductance coefficient according to the lateral offset, where the first mutual inductance coefficient is a mutual inductance coefficient between two adjacent energy transmitting coils and an energy receiving coil;
S3:通过调整两个直流控制绕组的驱动电流来改变第二互感系数,使得系统输出功率维持预定范围内,所述第二互感系数为所述能量接收线圈与所述可变集成补偿电感之间的互感系数。S3: Changing the second mutual inductance by adjusting the driving current of the two DC control windings so that the system output power is maintained within a predetermined range, wherein the second mutual inductance is the mutual inductance between the energy receiving coil and the variable integrated compensation inductor.
可选地,根据系统参数配置,通过仿真预先建立横向偏移量与所述第一互感系数的映射关系,同时通过仿真预先建立所述两个直流控制绕组的驱动电流与所述第二互感系数的映射关系。Optionally, according to system parameter configuration, a mapping relationship between the lateral offset and the first mutual inductance is pre-established through simulation, and a mapping relationship between the driving currents of the two DC control windings and the second mutual inductance is pre-established through simulation.
可选地,步骤S3中按照
Re来确定第一互感系数与第二互感系数的变化关系,其中Pout为输出功率,MABs为第一互感系数,Msf3为第二互感系数,ω为系统工作角频率,Lf为原边LCC谐振补偿网络中的补偿电感值,Lse为副边谐振网络的等效电感值,Re为等效负载,逆变器交流输出电压的相量表达式,当电动汽车横向偏移导致第一互感系数MABs发生变化时,通过调整两个直流控制绕组的驱动电流来改变第二互感系数Msf3,使得所述输出功率Pout维持预定范围内。Optionally, in step S3, Re is used to determine the changing relationship between the first mutual inductance coefficient and the second mutual inductance coefficient, where P out is the output power, M ABs is the first mutual inductance coefficient, Msf3 is the second mutual inductance coefficient, ω is the system operating angular frequency, L f is the compensation inductance value in the primary LCC resonant compensation network, L se is the equivalent inductance value of the secondary resonant network, Re is the equivalent load, The phasor expression of the inverter AC output voltage, when the electric vehicle lateral displacement causes the first mutual inductance M ABs to change, the second mutual inductance M sf3 is changed by adjusting the driving current of the two DC control windings so that the output power P out is maintained within a predetermined range.本发明的效果是:The effects of the present invention are:
本发明提出的一种电动汽车动态无线充电系统及横向偏移功率波动抑制方法,基于可变集成电感线圈的应用,在实现导轨切换时功率波动抑制的基础上进一步提高横向抗偏移性,使用较小的直流电流实现了横向偏移时系统输出功率跌落的补偿。The present invention proposes a dynamic wireless charging system for electric vehicles and a method for suppressing lateral displacement power fluctuations. Based on the application of a variable integrated inductor coil, the lateral displacement resistance is further improved on the basis of suppressing power fluctuations during rail switching, and a smaller DC current is used to compensate for the system output power drop during lateral displacement.
附图说明BRIEF DESCRIPTION OF THE DRAWINGS
为了更清楚地说明本发明具体实施方式或现有技术中的技术方案,下面将对具体实施方式或现有技术描述中所需要使用的附图作简单地介绍。In order to more clearly illustrate the specific implementation of the present invention or the technical solution in the prior art, the drawings required for use in the specific implementation or the description of the prior art are briefly introduced below.
图1为现有技术中不同横向偏移下等效互感的变化曲线;FIG1 is a variation curve of equivalent mutual inductance under different lateral offsets in the prior art;
图2为具体实施例中提供的电动汽车动态无线充电系统结构示意图;FIG2 is a schematic diagram of the structure of a dynamic wireless charging system for an electric vehicle provided in a specific embodiment;
图3为具体实施例中提供的电动汽车动态无线充电系统电路原理图;FIG3 is a circuit diagram of a dynamic wireless charging system for an electric vehicle provided in a specific embodiment;
图4为次级侧补偿网络的等效去耦电路;FIG4 is an equivalent decoupling circuit of the secondary side compensation network;
图5为考虑同侧电感耦合的等效电路;FIG5 is an equivalent circuit considering the inductive coupling on the same side;
图6为可变集成电感线圈的结构示意图;FIG6 is a schematic diagram of the structure of a variable integrated inductor coil;
图7为可变集成电感线圈的绕组和电流方向示意图;FIG7 is a schematic diagram of the winding and current direction of the variable integrated inductor coil;
图8为磁性材料的典型磁化曲线;Figure 8 is a typical magnetization curve of a magnetic material;
图9为具有极化直流偏置磁场磁化的磁滞回线;FIG9 is a hysteresis loop of magnetization with a polarized DC bias magnetic field;
图10为具体实施例中可变电感磁集成耦合机构的设计流程图;FIG10 is a design flow chart of a variable inductance magnetic integrated coupling mechanism in a specific embodiment;
图11为磁集成耦合机构的结构主视图和尺寸;FIG11 is a front view of the structure and dimensions of the magnetic integrated coupling mechanism;
图12为磁集成耦合机构的结构俯视图和尺寸;FIG12 is a top view of the structure and dimensions of the magnetic integrated coupling mechanism;
图13为耦合系数k与间隙距离g的关系图;FIG13 is a graph showing the relationship between the coupling coefficient k and the gap distance g;
图14为匝数ndc1和匝数ndc2对次级侧线圈和控制线圈之间的互感差影响;FIG14 shows the effect of the number of turns n dc1 and the number of turns n dc2 on the mutual inductance difference between the secondary coil and the control coil;
图15为集成控制线圈中的复合互感设计区域;FIG15 is a composite mutual inductance design area in an integrated control coil;
图16为在不同匝数组合下集成补偿电感线圈的自感及其与接收线圈的互感;FIG16 shows the self-inductance of the integrated compensation inductor and its mutual inductance with the receiving coil under different combinations of turns;
图17在内外侧匝数相同情况下集成补偿电感线圈的自感及其与接收线圈的互感;FIG17 shows the self-inductance of the integrated compensation inductor and its mutual inductance with the receiving coil when the number of inner and outer turns is the same;
图18为集成控制线圈中不同直流电流对磁通密度模的影响;Figure 18 shows the effect of different DC currents in the integrated control coil on the magnetic flux density modulus;
图19为不同接收线圈位置下系统的逆变器输出波形;FIG19 is the inverter output waveform of the system at different receiving coil positions;
图20为横向偏移10cm下沿发射线圈移动时仿真的输出功率和效率。FIG20 shows the simulated output power and efficiency when moving along the transmitting coil with a lateral offset of 10 cm.
附图标记:1-能量发射线圈,11-原边补偿线圈,2-能量接收线圈,21-交流电感线圈,22-直流控制绕组。Figure numerals: 1 - energy transmitting coil, 11 - primary compensation coil, 2 - energy receiving coil, 21 - AC inductor coil, 22 - DC control winding.
具体实施方式DETAILED DESCRIPTION
下面将结合附图对本发明技术方案的实施例进行详细的描述。以下实施例仅用于更加清楚地说明本发明的技术方案,因此只作为示例,而不能以此来限制本发明的保护范围。The following embodiments of the technical solution of the present invention are described in detail in conjunction with the accompanying drawings. The following embodiments are only used to more clearly illustrate the technical solution of the present invention, and are therefore only used as examples, and cannot be used to limit the protection scope of the present invention.
需要注意的是,除非另有说明,本申请使用的技术术语或者科学术语应当为本发明所属领域技术人员所理解的通常意义。It should be noted that, unless otherwise specified, the technical terms or scientific terms used in this application should have the common meanings understood by those skilled in the art to which the present invention belongs.
如图2所示,本实施例提供了一种电动汽车动态无线充电系统,包括电能发射端和电能接收端,所述电能发射端采用多段能量发射导轨逐一并排设置,每一段能量发射导轨包括沿第一方向绕制的能量发射线圈1和与所述能量发射线圈1串联且沿第一方向相反方向绕制的原边补偿线圈11,所述电能接收端包括能量接收线圈2和可变集成补偿电感,所述可变集成补偿电感包括作为副边补偿网络的交流电感线圈21和用于改变所述交流电感线圈21等效自感的两个直流控制绕组22,所述两个直流控制绕组22按照相反的绕向分别设置在所述交流电感线圈21的内外两侧。As shown in FIG2 , this embodiment provides a dynamic wireless charging system for an electric vehicle, including a power transmitting end and a power receiving end, wherein the power transmitting end is arranged side by side with multiple sections of energy transmitting rails, each section of the energy transmitting rails includes an
具体实施时,相邻两段能量发射导轨之间按第一预设值设置有水平间距,能量发射线圈1按矩形线圈水平绕制,所述原边补偿线圈11按同一平面绕制在所述能量发射线圈1的内部,所述能量接收线圈2按矩形线圈水平绕制,其外轮廓的宽度与所述能量发射线圈1宽度相同,所述可变集成补偿电感中的交流电感线圈21和两个直流控制绕组22按同一平面绕制在所述能量接收线圈2的内部。通常在能量发射线圈1和所述原边补偿线圈11的底面设置有原边磁芯板,所述能量接收线圈2和所述可变集成补偿电感的顶面设置有副边磁芯板和屏蔽板。In a specific implementation, a horizontal spacing is set between two adjacent sections of the energy transmitting rail according to a first preset value, the
在本实施例中,所述电能发射端设置有原边LCC谐振补偿网络,在所述能量发射线圈上还设置有原边串联谐振电容。In this embodiment, the power transmitting end is provided with a primary LCC resonant compensation network, and a primary series resonant capacitor is also provided on the energy transmitting coil.
结合上文描述,系统电路结构如图3所示,图中Uin是逆变器交流输出电压,RL是负载电阻,线圈LA1和LA2反向串联连接作为第一个发射器,对应为第一段能量发射导轨中的能量发射线圈1和原边补偿线圈11;线圈LB1和LB2反向串联连接作为第二个发射器,对应为第二段能量发射导轨中的能量发射线圈1和原边补偿线圈11;线圈Ls和Lf3集成作为接收器,对应为能量接收线圈2和可变集成补偿电感,各个集成线圈和主耦合线圈共用同一个铁氧体和屏蔽板。L1和L2是补偿电感,C1、C2、Cf1、Cf2、Cp1、Cp2、Cs、Cf3是补偿电容,Co是滤波电容。从图3可以看出,系统中有15个重要的耦合互感,MA1B2、MA1B1、MA2B2、MA2B1、MA1A2、MB1B2、MA1f3、MA2f3、MB1f3、MB2f3、MA1s、MA2s、MB1s、MB2s和Msf3。其中,MA1B2、MA1B1、MA2B2、MA2B1、MA1A2、MB1B2为两组初级侧线圈之间的互感。MA1f3、MA2f3、MB1f3、MB2f3、MA1s、MA2s、MB1s、MB2s为次级侧线圈与初级侧线圈之间的互感。Msf3为能量接收线圈2与可变集成补偿电感中交流电感线圈21之间的互感。IA和IB是发射线圈的电流,Is是接收线圈的电流。I1和I2分别为两个补偿网络的逆变器输出电流,I3是负载输出电流,ω是系统工作频率。Combined with the above description, the system circuit structure is shown in Figure 3, in which U in is the inverter AC output voltage, RL is the load resistance, coils LA1 and LA2 are connected in reverse series as the first transmitter, corresponding to the
为了便于理解和描述,系统中涉及的互感定义为:For ease of understanding and description, the mutual inductance involved in the system is defined as:
MA1A2和MB1B2是电路中直接串联的内部互感,通常也是固定的。因此,它们可以直接等效于电路中的元件如下:M A1A2 and M B1B2 are the internal mutual inductances directly connected in series in the circuit, and are usually fixed. Therefore, they can be directly equivalent to the components in the circuit as follows:
在电能接收端,能量接收线圈与集成电感线圈之间的内部互感通过等效T型网络实现去耦,如图4所示,Msf3形成的T型网络会影响次级侧的电路阻抗,这也提供了提高输出功率的途径。At the power receiving end, the internal mutual inductance between the energy receiving coil and the integrated inductor coil is decoupled through an equivalent T-type network, as shown in FIG4 . The T-type network formed by Msf3 affects the circuit impedance on the secondary side, which also provides a way to increase the output power.
Lf1和Lf2表示串联支路的等效电感,分别表示为: Lf1 and Lf2 represent the equivalent inductance of the series branch, which are respectively expressed as:
谐振网络配置为:The resonant network configuration is:
此外,考虑同侧线圈之间耦合的简化电路如图5所示。In addition, a simplified circuit considering the coupling between coils on the same side is shown in FIG5 .
得到KVL方程:The KVL equation is obtained:
通过求解方程,计算得到发射线圈电流和输出电流的值为:By solving the equation, the values of the transmitting coil current and the output current are calculated as:
根据式(6),能量发射线圈电流和输出电流的表达式都不包含负载Re,因此保持了初级侧和次级侧电路的恒定电流输出特性。According to equation (6), the expressions of the energy transmitting coil current and the output current do not include the load Re , thus maintaining the constant current output characteristics of the primary and secondary side circuits.
考虑到系统参数的对称性,相应的参数设置为:Considering the symmetry of the system parameters, the corresponding parameters are set as:
Lf1=Lf2=Lf (7)L f1 = L f2 = L f (7)
为了使表达式更简洁,交叉互感表示为:To make the expression more concise, the cross inductance is expressed as:
输出功率表示为:The output power is expressed as:
按照本团队前期提交中国发明专利:202111524549.7所提及的系统设计及优化方案,可以实现电动汽车行驶过程中具备稳定的MABs,在无横向偏移下的系统导轨切换时的输出功率波动得到有效的抑制。According to the system design and optimization scheme mentioned in the Chinese invention patent 202111524549.7 submitted by our team earlier, stable MABS can be achieved during the driving of electric vehicles, and the output power fluctuation during the system rail switching without lateral deviation can be effectively suppressed.
结合图1分析结果可以看出,电动汽车在横向偏移状态下移动过程中,MABs在较低值下保持相对稳定。根据式(9),可以通过调整Msf3来提高功率传输能力,从而抑制横向偏移下的整体输出功率跌落。Combined with the analysis results in Figure 1, it can be seen that when the electric vehicle moves in the lateral offset state, M ABs remains relatively stable at a low value. According to formula (9), the power transmission capacity can be improved by adjusting M sf3 , thereby suppressing the overall output power drop under lateral offset.
但是,基于对具有磁集成耦合机构的LCC-LCC无线传能系统的分析,Msf3的调整必然会影响谐振状态。根据式(5)式计算得到的逆变器总输出电流表示为:However, based on the analysis of the LCC-LCC wireless energy transfer system with magnetic integrated coupling mechanism, the adjustment of Msf3 will inevitably affect the resonant state. The total output current of the inverter calculated according to formula (5) is expressed as:
其中,Lses=Lse+Msf3,Lsed=Lse-Lf3系统的总输入阻抗表示为:Among them, Lses = Lse + Msf3 , Lsed = Lse - Lf3. The total input impedance of the system is expressed as:
逆变器输出电压和电流之间的相位角表示为γ,可以用于反映系统的谐振状态。γ的正值和负值分别代表感性偏移和容性偏移。此外,其绝对值表示系统偏离谐振点的程度。γ的正切值表示为:The phase angle between the inverter output voltage and current is denoted as γ, which can be used to reflect the resonant state of the system. The positive and negative values of γ represent inductive and capacitive offsets, respectively. In addition, its absolute value indicates the degree to which the system deviates from the resonant point. The tangent value of γ is expressed as:
Msf3和Lf3是由集成电感线圈引入的参数,这两个参数对输出功率和ZVS条件的影响需要进一步深入分析。Msf3的变化量定义为ΔMsf3,Lf3的变化量定义为ΔLf3,式(9)改写为:M sf3 and L f3 are parameters introduced by the integrated inductor coil. The impact of these two parameters on the output power and ZVS conditions needs further in-depth analysis. The change of M sf3 is defined as ΔM sf3 , and the change of L f3 is defined as ΔL f3 . Equation (9) can be rewritten as:
根据式(13),输出功率受ΔMsf3影响,它可用作控制变量。同时,输出功率不受ΔLf3影响,这与次级侧的恒定输出电流特性一致。According to equation (13), the output power is affected by ΔM sf3 , which can be used as a control variable. At the same time, the output power is not affected by ΔL f3 , which is consistent with the constant output current characteristic of the secondary side.
式(12)改写为:Formula (12) can be rewritten as:
其中,
in,在式(14)中,T1是原始ZVS配置部分,T2是由ΔMsf3和ΔLf3产生的附加部分,它们共同构成了系统输入阻抗角的正切值,决定了谐振状态和ZVS条件。为了在电动汽车在道路上行驶期间获得相对稳定且优化的ZVS工作状态,T2期望相对于配置的T1而言足够小以至于可以忽略。因此,通过仔细合适的设计,可以提高横向偏移下的输出功率,并使系统能够工作在合理的ZVS条件。In equation (14), T1 is the original ZVS configuration part, and T2 is the additional part generated by ΔM sf3 and ΔL f3 , which together constitute the tangent value of the system input impedance angle, determining the resonant state and ZVS condition. In order to obtain a relatively stable and optimized ZVS working state during the driving of electric vehicles on the road, T2 is expected to be small enough to be ignored relative to the configured T1 . Therefore, through careful and appropriate design, the output power under lateral offset can be improved and the system can operate under reasonable ZVS conditions.
针对可变集成补偿电感而言,具体实施时,如图6、图7所示,集成电感绕组Nac缠绕在中间位置,两个直流控制绕组Ndc1和Ndc2分别绕制在内侧和外侧,在两个直流控制绕组上加载有直流电源,其中内侧直流控制绕组与外侧直流控制绕组的流经的电流所产生的磁场方向相反。For the variable integrated compensation inductor, in a specific implementation, as shown in FIG6 and FIG7 , the integrated inductor winding Nac is wound in the middle position, and the two DC control windings Ndc1 and Ndc2 are respectively wound on the inside and the outside, and a DC power supply is loaded on the two DC control windings, wherein the magnetic field generated by the current flowing through the inner DC control winding and the outer DC control winding is in opposite directions.
由于交流绕组产生的磁通量流经直流绕组,因此在两个直流绕组中会产生不期望的电压。由于非线性B-H行为,电感器和控制绕组之间的理想解耦很困难。但是直流绕组中感应的交流电压可以通过以下措施得到缓解。Since the magnetic flux generated by the AC winding flows through the DC winding, an undesirable voltage is generated in the two DC windings. Due to the nonlinear B-H behavior, ideal decoupling between the inductor and the control winding is difficult. However, the AC voltage induced in the DC winding can be mitigated by the following measures.
(1)两个直流绕组分别紧挨着交流绕组的内外边缘,使其与交流绕组具有紧密的耦合特性。因此,对直流电的控制将更加有效。(1) The two DC windings are close to the inner and outer edges of the AC winding, making them closely coupled with the AC winding. Therefore, the control of DC power will be more effective.
(2)两个直流绕组反向串联。考虑到电流的方向,在中间绕组中,感应电压在相反方向上,这使得直流线圈上的电压彼此抵消。(2) The two DC windings are connected in series in opposite directions. Considering the direction of the current, the induced voltage in the middle winding is in the opposite direction, which makes the voltages on the DC coils cancel each other.
磁芯可以增大线圈的电感,磁芯的磁导率影响电感值。对于集成电感线圈,磁场密度标记为B,磁场强度标记为H。当交流电通过线圈时,磁芯的磁化过程可以表示为H和B的关系曲线,如图8所示。该过程包括不饱和区、过渡区和饱和区。在初始时刻,H和B的值都为零。在不饱和区域,H和B具有很强的正相关和非线性关系。在过渡区域,B的增加相对缓慢。在饱和区域下,B趋于稳定。H和B之间关系曲线的斜率反映了磁化磁芯的速度,对应于磁芯材料的等效相对磁导率μ。The magnetic core can increase the inductance of the coil, and the magnetic permeability of the magnetic core affects the inductance value. For the integrated inductor coil, the magnetic field density is marked as B and the magnetic field strength is marked as H. When the alternating current passes through the coil, the magnetization process of the magnetic core can be expressed as a relationship curve between H and B, as shown in Figure 8. This process includes the unsaturated region, the transition region, and the saturated region. At the initial moment, the values of H and B are both zero. In the unsaturated region, H and B have a strong positive correlation and nonlinear relationship. In the transition region, the increase of B is relatively slow. In the saturation region, B tends to be stable. The slope of the relationship curve between H and B reflects the speed of the magnetized magnetic core, which corresponds to the equivalent relative permeability μ of the magnetic core material.
电感的一般定义是总磁链与通过绕组的电流之比,表示为:The general definition of inductance is the ratio of the total flux linkage to the current through the winding, expressed as:
在表达式中,Ψ(t)是总磁链。Φ是磁通量,与磁芯材料的相对磁导率正相关。N是导线的匝数,i是绕组中的电流。结合图9和式(15),可变集成电感的自感与磁芯材料的等效相对磁导率正相关。In the expression, Ψ(t) is the total flux linkage. Φ is the magnetic flux, which is positively correlated with the relative permeability of the core material. N is the number of turns of the wire, and i is the current in the winding. Combining Figure 9 and equation (15), the self-inductance of the variable integrated inductor is positively correlated with the equivalent relative permeability of the core material.
在无线电能传输系统运行期间,磁芯的相对磁导率通常是恒定的。特别是直流电流产生的极化偏置会影响磁芯的等效相对磁导率。极化直流偏置磁场磁化的磁滞回线如图9所示。当磁芯被极化直流偏置磁场磁化时,交流磁化产生一个小的柳叶刀形磁滞回线,随着交流H场的减小,磁滞回线减小为一条直线。该环路的面积通常较小,其平均斜率可以反映增量相对磁导率Δμ:During the operation of the wireless power transmission system, the relative permeability of the core is usually constant. In particular, the polarization bias generated by the DC current will affect the equivalent relative permeability of the core. The hysteresis loop of the polarized DC bias magnetic field magnetization is shown in Figure 9. When the core is magnetized by the polarized DC bias magnetic field, the AC magnetization produces a small lancet-shaped hysteresis loop, which decreases to a straight line as the AC H field decreases. The area of this loop is usually small, and its average slope can reflect the incremental relative permeability Δμ:
结合电感的定义和直流电对磁芯磁化过程的影响,因此,利用电感磁芯的磁导率来调整电感是可行的。Combining the definition of inductance and the influence of direct current on the magnetization process of the magnetic core, it is feasible to adjust the inductance by using the magnetic permeability of the inductor core.
结合上文描述,本实施例还给出了磁集成耦合机构的设计流程,在实施过程中,为了减小外侧控制线圈的加入对接收线圈自感的影响,并放大直流磁场的影响,外侧控制线圈与能量接收线圈之间的距离应足够小。另外,为了使控制线圈上的总感应电压尽可能小,内侧控制线圈的尺寸应尽可能接近外部控制线圈,因此内侧控制线圈和接收线圈之间的距离也应足够小。综合以上考虑,所提出的磁耦合机构的设计流程如图10所示。Combined with the above description, this embodiment also provides a design process of the magnetic integrated coupling mechanism. During the implementation process, in order to reduce the influence of the addition of the outer control coil on the self-inductance of the receiving coil and amplify the influence of the DC magnetic field, the distance between the outer control coil and the energy receiving coil should be small enough. In addition, in order to make the total induced voltage on the control coil as small as possible, the size of the inner control coil should be as close as possible to the outer control coil, so the distance between the inner control coil and the receiving coil should also be small enough. Taking the above into consideration, the design process of the proposed magnetic coupling mechanism is shown in Figure 10.
首先,根据应用场景需求和相关标准设计能量接收线圈,综合考虑耦合系数k和实际应用情况设计能量发射线圈。First, the energy receiving coil is designed according to the application scenario requirements and relevant standards, and the energy transmitting coil is designed by comprehensively considering the coupling coefficient k and the actual application situation.
其次,在确定主耦合线圈的情况下,设计原边集成的原边补偿线圈,以在得到不同的接收器位置稳定的MABs。Secondly, with the main coupling coil determined, the primary compensation coil of the primary integration is designed to obtain stable MABs at different receiver positions.
第三,在能量发射线圈与原边补偿线圈的基础上,设计了副边补偿电感线圈以保证不同接收器位置的MABf≤εMABs。Third, based on the energy transmitting coil and the primary compensation coil, a secondary compensation inductor coil is designed to ensure that M ABf ≤ εM ABs at different receiver positions.
第四,合理设置控制线圈内外匝数的关系,使总感应电压足够小,并通过遍历法获得了内外控制线圈的一系列匝数选取区域。Fourth, the relationship between the number of inner and outer turns of the control coil is reasonably set to make the total induced voltage small enough, and a series of turn selection areas of the inner and outer control coils are obtained through the traversal method.
第五,根据横向偏移时互感的下降和输出功率的表达式,确定了互感Msf3的变化范围,并据此设计了控制线圈的匝数。Fifth, according to the expression of the decrease of mutual inductance and output power during lateral displacement, the variation range of mutual inductance Msf3 is determined, and the number of turns of the control coil is designed accordingly.
第六,根据Lf3和Msf3变化之间的关系,根据式(14)评估ZVS条件。Sixth, based on the relationship between the changes in Lf3 and Msf3 , the ZVS condition is evaluated according to equation (14).
最后,在对主耦合线圈和集成线圈进行仿真设计的基础上,在Maxwell建立磁耦合机构模型以获得系统参数。Finally, based on the simulation design of the main coupling coil and the integrated coil, a magnetic coupling mechanism model is established in Maxwell to obtain the system parameters.
使用Maxwell建模的整个系统结构如图11和图12所示,表1中定义了描述耦合机构尺寸的参数。从而建立一个4.5kW功率等级的典型磁耦合机构以验证设计方法。The entire system structure modeled using Maxwell is shown in Figures 11 and 12, and the parameters describing the dimensions of the coupling mechanism are defined in Table 1. Thus, a typical magnetic coupling mechanism with a power level of 4.5kW is established to verify the design method.
表1可变电感磁集成耦合机构的参数定义Table 1 Parameter definition of variable inductance magnetic integrated coupling mechanism
针对初始磁耦合机构的设计而言,根据SAE J2954 WPT3Z3 VA标准设计接收器的结构,能量接收线圈的尺寸设计为“400mm*400mm*5mm”,匝数nw3设计为10。铁氧体板和铝屏蔽板分别用于磁场增强和电磁屏蔽。次级侧铁氧体板的尺寸为“400mm*400mm*10mm”,铝屏蔽的尺寸为“420mm*420mm*5mm”,传输距离为d=150mm。For the design of the initial magnetic coupling mechanism, the receiver structure is designed according to the SAE J2954 WPT3Z3 VA standard. The size of the energy receiving coil is designed to be "400mm*400mm*5mm", and the number of turns nw3 is designed to be 10. The ferrite plate and aluminum shielding plate are used for magnetic field enhancement and electromagnetic shielding, respectively. The size of the secondary side ferrite plate is "400mm*400mm*10mm", the size of the aluminum shielding is "420mm*420mm*5mm", and the transmission distance is d=150mm.
通常耦合系数和成本是设计主磁耦合机构的两个主要指标,在本系统中,耦合系数k与功率传递性能密切相关。对于一定长度的道路,长度较短的传输线圈需要更多的谐振补偿网络、磁芯和成本较高的Litz线。以发射线圈为例,充分考虑了电动汽车的长度、成本、漏磁和耦合系数,设计了初始磁耦合机构。在这种情况下,发射线圈的尺寸设计为“900mm*450mm*5mm”,匝数nw1设计为6。Usually the coupling coefficient and cost are the two main indicators for designing the main magnetic coupling mechanism. In this system, the coupling coefficient k is closely related to the power transfer performance. For a certain length of road, a shorter transmission coil requires more resonant compensation networks, magnetic cores, and more expensive Litz wires. Taking the transmitting coil as an example, the initial magnetic coupling mechanism is designed by fully considering the length, cost, leakage flux, and coupling coefficient of the electric vehicle. In this case, the size of the transmitting coil is designed to be "900mm*450mm*5mm", and the number of turns nw1 is designed to be 6.
在本系统中,两个相邻发射线圈之间的交叉耦合将导致激活线圈受到附近未激活线圈的影响。此外,过多的交叉耦合会使串联谐振补偿电容器在实际应用中难以配置。间隙距离越大,互感下降越大,输出功率波动越大。为了避免过度的交叉耦合,应优化两个相邻发射线圈之间的间隙距离g。耦合系数k和间隙距离g之间的关系如图13所示。结果表明,在间隙距离达到30mm后,k值可以忽略不计,并缓慢减小,因此间隙距离g设置为30mm。In this system, the cross-coupling between two adjacent transmitting coils will cause the activated coil to be affected by the nearby unactivated coil. In addition, too much cross-coupling will make the series resonant compensation capacitor difficult to configure in practical applications. The larger the gap distance, the greater the mutual inductance drop and the greater the output power fluctuation. In order to avoid excessive cross-coupling, the gap distance g between two adjacent transmitting coils should be optimized. The relationship between the coupling coefficient k and the gap distance g is shown in Figure 13. The results show that after the gap distance reaches 30mm, the k value can be ignored and slowly decreases, so the gap distance g is set to 30mm.
集成反向线圈的尺寸设计为“650mm*190mm*5mm”,匝数nw2设计为3。结果表明,在整个移动范围内,输出功率波动保持在±4%以内。因此,在没有横向偏移的情况下,可以有效地抑制发射线圈过渡期间的功率波动。此外,还进一步设计了次级侧集成补偿电感线圈以优化ZVS条件,集成补偿电感线圈的尺寸设计为“170mm*170mm*5mm”,匝数nw4设计为6。The size of the integrated reverse coil is designed to be "650mm*190mm*5mm", and the number of turns nw2 is designed to be 3. The results show that the output power fluctuation is kept within ±4% over the entire moving range. Therefore, the power fluctuation during the transition of the transmitting coil can be effectively suppressed without lateral offset. In addition, the secondary-side integrated compensation inductor coil is further designed to optimize the ZVS condition. The size of the integrated compensation inductor coil is designed to be "170mm*170mm*5mm", and the number of turns nw4 is designed to be 6.
针对集成控制线圈而言,集成控制线圈的加入不可避免地会带来与耦合机构中其他单极线圈的耦合,根据理论分析应削弱这些耦合。由于其与初级侧线圈之间的互感足够小,因此在设计中仅考虑其与次级侧线圈之间的互感。As for the integrated control coil, the addition of the integrated control coil will inevitably bring about coupling with other monopolar coils in the coupling mechanism. According to theoretical analysis, these couplings should be weakened. Since the mutual inductance between it and the primary side coil is small enough, only the mutual inductance between it and the secondary side coil is considered in the design.
根据纽曼公式,发射线圈和接收线圈之间的互感可以表示为:According to Newman's formula, the mutual inductance between the transmitting coil and the receiving coil can be expressed as:
其中Na、Nb、
分别为线圈匝数,分别为发射线圈和接收线圈的微元。rab是和之间的距离。μ0是真空的磁导率。Among them, Na , Nb , are the number of coil turns, and are the infinitesimals of the transmitting coil and the receiving coil, respectively. and The distance between them. μ 0 is the magnetic permeability of vacuum.根据式(17),对于固定位置的集成控制线圈,匝数是影响互感的决定性因素。因此,通过遍历内侧和外侧控制线圈的匝数,得到不同匝数组合配置下集成控制线圈和次级侧线圈之间的互感。集成补偿电感线圈和集成控制线圈之间的互感分别为Md1f3和Md2f3。接收线圈和集成控制线圈之间的互感分别为Md1和Md2。集成补偿电感线圈和集成控制线圈的等效互感差值表示为ΔMdf3。接收线圈和集成控制线圈的等效互感差值表示为ΔMds。According to formula (17), for the integrated control coil at a fixed position, the number of turns is the decisive factor affecting the mutual inductance. Therefore, by traversing the number of turns of the inner and outer control coils, the mutual inductance between the integrated control coil and the secondary coil under different combinations of turns is obtained. The mutual inductance between the integrated compensation inductor coil and the integrated control coil is M d1f3 and M d2f3 , respectively. The mutual inductance between the receiving coil and the integrated control coil is M d1 and M d2 , respectively. The difference in equivalent mutual inductance between the integrated compensation inductor coil and the integrated control coil is expressed as ΔM df3 . The difference in equivalent mutual inductance between the receiving coil and the integrated control coil is expressed as ΔM ds .
结合图14可以看出,图14(a)和图14(b)中绘制了受匝数ndc1和匝数ndc2影响的次级侧线圈和控制线圈之间的互感差。由于Is和I3之间存在90度的相位差,接收线圈和集成电感线圈产生的总等效感应电压可视为二者的最大值。在4.5kW功率等级的样机设计中,为了确保较小的感应电压,复合互感差值最大值(ΔMdf3,ΔMds)可以设置在6μH以下。为了确定内外侧集成控制线圈匝数的选择范围,通过叠加获得集成控制线圈中的复合互感,并绘制在图15中。在图15中,小于6μH的区域被标记为匝数的设计范围。As can be seen from Figure 14, the mutual inductance difference between the secondary side coil and the control coil affected by the number of turns n dc1 and the number of turns n dc2 is plotted in Figure 14 (a) and Figure 14 (b). Due to the 90-degree phase difference between I s and I 3 , the total equivalent induced voltage generated by the receiving coil and the integrated inductor coil can be regarded as the maximum value of the two. In the prototype design of the 4.5kW power level, in order to ensure a smaller induced voltage, the maximum value of the composite mutual inductance difference (ΔM df3 , ΔM ds ) can be set below 6μH. In order to determine the selection range of the number of turns of the inner and outer integrated control coils, the composite mutual inductance in the integrated control coil is obtained by superposition and plotted in Figure 15. In Figure 15, the area less than 6μH is marked as the design range of the number of turns.
理论上,可变集成电感可以在设计范围内的任意匝数进行调整,只是其中所需的直流电流范围不同。为了在较小的控制电流下实现对输出功率的控制,通过仿真进一步优化匝数配置。根据图15,在设计范围内,当外部匝数nd1=3时,可选择的内部匝数nd2的范围相对较大。因此,首先分析nd1=3时内外匝的系统特性,在不同匝数组合下集成补偿电感线圈的自感及其与接收线圈之间的互感绘制在图16中。在图中,上方曲线代表Lf3的变化,下方曲线代表Msf3的变化,结果显示了两个特征。首先,在设计范围内的匝数配置下,Lf3和Msf3有效地随直流电流的增加而变化。第二,相对于nd1,nd2越小,相应的Lf3和Msf3受直流电流的影响越明显。考虑到设计目的,随着直流电流的增加,预计Lf3变化较小,Msf3变化较大。因此,为了在自感和互感之间实现折衷,在设计范围内将nd1和nd2设置为相等。Theoretically, the variable integrated inductor can be adjusted to any number of turns within the design range, except that the required DC current range is different. In order to achieve control of the output power under a smaller control current, the number of turns configuration is further optimized through simulation. According to Figure 15, within the design range, when the external number of turns n d1 = 3, the range of the selectable internal number of turns n d2 is relatively large. Therefore, the system characteristics of the internal and external turns when n d1 = 3 are first analyzed, and the self-inductance of the integrated compensation inductor coil and its mutual inductance with the receiving coil under different combinations of turns are plotted in Figure 16. In the figure, the upper curve represents the change of L f3 , and the lower curve represents the change of Msf3 . The results show two characteristics. First, under the number of turns configuration within the design range, L f3 and Msf3 effectively change with the increase of DC current. Second, relative to n d1 , the smaller the n d2 , the more obvious the influence of DC current on the corresponding L f3 and Msf3 . Considering the design purpose, it is expected that L f3 will change less and Msf3 will change more with the increase of DC current. Therefore, to achieve a compromise between self-inductance and mutual inductance, nd1 and nd2 are set equal within the design range.
在内外侧匝数相同情况下集成补偿电感线圈的自感及其与接收线圈的互感如图17所示。结果表明,不同的匝数配置对应不同的Msf3调整范围。在本例4.5kW功率级的原型设计中,根据图16中横向偏移y=10cm时的互感跌落,选取nd1=5和nd2=5。The self-inductance of the integrated compensation inductor and its mutual inductance with the receiving coil when the number of turns inside and outside are the same are shown in Figure 17. The results show that different turn configurations correspond to different adjustment ranges of Msf3 . In the prototype design of the 4.5kW power level in this example, nd1 = 5 and nd2 = 5 are selected according to the mutual inductance drop when the lateral offset y = 10cm in Figure 16.
为了进一步验证直流电流产生的极化偏置对磁芯等效相对磁导率的影响,对带有不同大小直流电流的集成控制线圈的磁通密度模分布进行有限元仿真分析。基于前文设计的系统参数,仿真得到的磁通密度模如图18所示。In order to further verify the influence of polarization bias generated by DC current on the equivalent relative permeability of the magnetic core, a finite element simulation analysis is performed on the magnetic flux density norm distribution of the integrated control coil with DC currents of different magnitudes. Based on the system parameters designed in the previous article, the magnetic flux density norm obtained by simulation is shown in Figure 18.
根据图18中的仿真结果,集成控制线圈中的直流电流可以有效地改变集成电感线圈的磁通密度模,改变了磁芯材料的相对磁导率,从而改变了集成补偿电感线圈的自感值。因此,电动汽车动态无线充电系统中横向偏移状态下导轨切换引起的输出功率波动可以得到有效抑制。According to the simulation results in Figure 18, the DC current in the integrated control coil can effectively change the magnetic flux density modulus of the integrated inductor coil, change the relative magnetic permeability of the magnetic core material, and thus change the self-inductance value of the integrated compensation inductor coil. Therefore, the output power fluctuation caused by the rail switching in the lateral offset state in the dynamic wireless charging system of electric vehicles can be effectively suppressed.
为了进一步验证理论分析的正确性,下面构建了一个4.5kW功率级EVDWC系统原型。通过仿真和实验验证该方法的可行性。In order to further verify the correctness of the theoretical analysis, a 4.5kW power level EVDWC system prototype is built. The feasibility of the method is verified through simulation and experiments.
表2耦合机构仿真参数和计算的谐振参数Table 2 Coupling mechanism simulation parameters and calculated resonance parameters
根据表2中的仿真参数建立系统电路仿真模型,逆变器输出电压和负载输出电压设置为400V。横向偏移设置为y=10cm,集成控制线圈中通有10A的直流电流。在不同接收线圈位置的逆变器输出波形如图19所示,其中vinv是逆变器输出电压,iinvA和iinvB是对应于两个变送器的逆变器输出电流,iinv是iinvA和iinvB的叠加总电流。逆变器的输出电压相位略微超前于总电流,在限制谐振电路中循环的无功功率从而降低功率损耗的同时,还保证了正常参数配置下的ZVS条件。The system circuit simulation model is established according to the simulation parameters in Table 2, and the inverter output voltage and load output voltage are set to 400V. The lateral offset is set to y = 10cm, and a DC current of 10A is passed through the integrated control coil. The inverter output waveforms at different receiving coil positions are shown in Figure 19, where v inv is the inverter output voltage, i invA and i invB are the inverter output currents corresponding to the two transmitters, and i inv is the superimposed total current of i invA and i invB . The output voltage phase of the inverter is slightly ahead of the total current, which not only limits the reactive power circulating in the resonant circuit and thus reduces power loss, but also ensures the ZVS condition under normal parameter configuration.
在10cm的横向偏移情况下,不同接收线圈位置的输出功率和效率的仿真结果如图20所示。结果表明,在不同的接收位置处效率均保持在较高水平。且在接收线圈移动过程中,整体的输出功率得到提高,横向偏移下的功率波动得到抑制。相对于平均功率,输出功率波动被抑制在±5%以内。The simulation results of the output power and efficiency at different receiving coil positions with a lateral offset of 10 cm are shown in Figure 20. The results show that the efficiency remains at a high level at different receiving positions. In addition, during the movement of the receiving coil, the overall output power is improved, and the power fluctuation under lateral offset is suppressed. Relative to the average power, the output power fluctuation is suppressed within ±5%.
综上可以看出,基于上述理论分析和仿真验证,本实施例中还可以提出一种电动汽车动态无线充电系统的横向偏移功率波动抑制方法,包括以下步骤:In summary, based on the above theoretical analysis and simulation verification, a method for suppressing lateral offset power fluctuation of a dynamic wireless charging system for an electric vehicle can be proposed in this embodiment, including the following steps:
S1:检测电动汽车横向偏移量;S1: Detect the lateral deviation of the electric vehicle;
S2:根据横向偏移量确定第一互感系数的跌落情况,所述第一互感系数为相邻两段能量发射线圈与能量接收线圈之间互感系数;具体实施时,根据系统参数配置,通过仿真预先建立横向偏移量与所述第一互感系数的映射关系S2: Determine the drop of the first mutual inductance coefficient according to the lateral offset, where the first mutual inductance coefficient is the mutual inductance coefficient between two adjacent energy transmitting coils and energy receiving coils; in specific implementation, according to the system parameter configuration, a mapping relationship between the lateral offset and the first mutual inductance coefficient is pre-established through simulation
S3:通过调整两个直流控制绕组的驱动电流来改变第二互感系数,使得系统输出功率维持预定范围内,所述第二互感系数为所述能量接收线圈与所述可变集成补偿电感之间的互感系数。具体实施时,通过仿真预先建立所述两个直流控制绕组的驱动电流与所述第二互感系数的映射关系。S3: Changing the second mutual inductance coefficient by adjusting the driving current of the two DC control windings so that the system output power is maintained within a predetermined range, wherein the second mutual inductance coefficient is the mutual inductance coefficient between the energy receiving coil and the variable integrated compensation inductor. In specific implementation, a mapping relationship between the driving current of the two DC control windings and the second mutual inductance coefficient is pre-established through simulation.
在步骤S3中即可按照
来确定第一互感系数与第二互感系数的变化关系,当电动汽车横向偏移导致第一互感系数MABs发生变化时,通过调整两个直流控制绕组的驱动电流来改变第二互感系数Msf3,使得所述输出功率Pout维持预定范围内。In step S3, To determine the changing relationship between the first mutual inductance and the second mutual inductance, when the electric vehicle lateral offset causes the first mutual inductance M ABs to change, the second mutual inductance M sf3 is changed by adjusting the driving current of the two DC control windings so that the output power P out is maintained within a predetermined range.通过采用上述方法,可以使用较小的直流电流实现系统输出功率跌落的补偿。By adopting the above method, a smaller DC current can be used to compensate for the system output power drop.
最后需要说明的是,以上各实施例仅用以说明本发明的技术方案,而非对其限制;尽管参照前述各实施例对本发明进行了详细的说明,本领域的普通技术人员应当理解:其依然可以对前述各实施例所记载的技术方案进行修改,或者对其中部分或者全部技术特征进行等同替换;而这些修改或者替换,并不使相应技术方案的本质脱离本发明各实施例技术方案的范围,这样的变换均应涵盖在本发明的权利要求和说明书的范围当中。Finally, it should be noted that the above embodiments are only used to illustrate the technical solutions of the present invention, rather than to limit it. Although the present invention has been described in detail with reference to the above embodiments, those skilled in the art should understand that the technical solutions described in the above embodiments can still be modified, or some or all of the technical features can be replaced by equivalents. These modifications or replacements do not make the essence of the corresponding technical solutions deviate from the scope of the technical solutions of the embodiments of the present invention, and such changes should be included in the scope of the claims and description of the present invention.
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