NO337399B1 - Digital bearing module and detection method - Google Patents

Description

This invention relates to a sonar system and a method for detecting with the sonar system. In particular, the invention relates to a sonar system and method of detection for use as a hobby fish finder.

A sonar system typically consists of a transmitter, a transducer, a receiver and a display. The transmitter of the sonar system sends an electrical pulse. The transducer converts the electrical pulse into a sound wave and sends the sound wave through the water. When the sound wave hits an object, the sound bounces back and returns to the sonar system. The sonar return wave hits the transducer of the sonar system and the transducer converts the sonar return wave into an electrical signal. The receiver filters, amplifies, and detects the sonar return signal. The receiver converts the sonar return signal into a digital signal after its detection and sends the amplified signal to the display.

US6418080B2 describes an underwater detection apparatus consisting of transducer, transmitter, broadband receiver circuit, preamplifier, mixer, filter, main amplifier, A/D converter and display.

US6445646B2 describes a signal processing method and apparatus, as well as sonar systems.

US4000650A describes a device in which ultrasonic pulses are sent by a transducer from the top of a tank or container and reflected by the material of which it is composed and received by the same

the transducer.

EP261731B1 describes a method and device for measuring a bottom depth below a water surface, by echoing sound from a vessel moving on the surface. Using a transmitter/receiver unit on the vessel equipped with a measuring probe, ultrasonic pulses are sent to the seabed and pulses reflected from here are received, and the time difference between sending and receiving the same pulse is measured and stored.

The receiver section of a conventional sonar system comprises an analogue filter and an analogue detector. Such conventional sonar receiver sections can be built using a Philips SA604 "High Performance Low Power FM IF System". The Philips SA604 is a monolithic low power FM intermediate frequency (IF) system incorporating two limiting IF amplifiers, a quadrature detector, attenuation, a logarithmic received signal strength indicator, and a voltage regulator. The Philips S A604 is typically used in radios for use as an intermediate amplifier. The Philips SA604 includes a Received Signal Strength Indicator (RSSI) to output a logarithmic representation of the signal strength on the intermediate frequency (IF) input.

A disadvantage of conventional sonar systems is that they are manually tuned during production and allow a later adjustment of the center frequency. As a result, tuning may not be optimized for use with a specific transducer.

A further disadvantage of conventional sonar systems is that the bandwidth is fixed. A fixed bandwidth makes it impossible for the sonar system to extract an optimal time/frequency response in both shallow and deep water, and thus affects the resolution.

Conventional sonar systems have a dynamic range of approximately 120 dB. An analog to digital (A/D) converter with a minimum resolution of 20 bits is required to digitally sample sonar return waves within this dynamic range. The cost of a commercially available A/D converter with 20 bit resolution is too high for practical use in a hobby fish finder.

It is an object of the invention to provide a sonar system which performs a digital filtering and a digital detection of a sonar carrier wave.

A further object of the invention is to provide a sonar system with high sensitivity and a wide dynamic range.

A further object of the invention is to provide a sonar system with a programmable frequency.

Another object of the invention is to provide a sonar system with an automatic tuning that can be optimized for use with a specific transducer.

Another object of the invention is to provide a sonar system with a programmable bandwidth.

Another purpose of the invention is to provide a sonar system with an optimal time/frequency response in both shallow and deep water.

Another object of the invention is to provide a sonar receiver section with commercially available components which have low costs and are intended for use in a hobby fishing finder.

The problems and disadvantages of the conventional sonar systems described above are overcome, and other advantages are provided, with the present invention which is characterized by the features set forth in the accompanying independent patent claims 1 or 14.

Features of embodiments of the present invention according to device patent claim 1 or method patent claim 14 are indicated in the non-independent device patent claims 2-13 and 15-27 respectively in the non-independent method patent claim 28.

The invention includes a sonar system with a high sensitivity, a wide dynamic range, a programmable frequency, and a programmable bandwidth. The invention also includes a method for performing a digital filtering and a digital detection of a sonar return wave.

The invention relates to a digital bearing module. The digital sounding module includes a sonar carrier wave producing device. A bandpass filter, a preamplifier to provide a high sensitivity and cover a wide dynamic range, and an analog to digital converter to provide a digital implementation of a superheterodyne detector and produce an intermediate frequency. The digital gauge module also includes a programmable logic device for controlling a gain of the preamplifier and for digitally filtering the intermediate frequency and a microprocessor.

The invention also relates to a method for performing a digital filtering and a digital detection of a sonar return wave. This method of detecting a sonar carrier wave includes receiving a sonar carrier wave, limiting the sonar carrier wave with a bandpass filter, and amplifying the filtered sonar carrier wave with a preamplifier. Then, the method includes feeding the amplified sonar carrier wave to an analog to digital converter to produce an intermediate frequency of the sonar carrier wave. So by controlling the preamplifier and analog to digital converter, a programmable logic device is performed. This step is followed by digital filtering of the sonar carrier and digital detection of the sonar carrier.

The invention will now be described with reference to the drawings where Figure 1 illustrates a sonar system in accordance with the invention. Figure 2 illustrates a functional diagram of a digital bearing module in accordance with the invention.

Figure 3 illustrates a harmonic sampling of the sonar return signal.

Figure 4 illustrates an algorithm for approximating the Euclidean distance between vectors that are 90 degrees out of phase. Figure 6 illustrates the logarithmic function implemented as a piecewise linear lo<gg>2() function. Figure 7 illustrates a transducer mounted on the hull of a vessel with echo reflections from three targets at different depths. Figures 8A to 8D illustrate the modification of the echo return signal by the signal processing of the present invention.

The invention includes a sonar system with a high sensitivity, a wide dynamic range, a programmable frequency, and a programmable bandwidth. The invention also includes a method for performing a digital filtering and a digital detection of a sonar return signal.

Figure 1 illustrates a sonar system 100 in accordance with the invention. The sonar system 100 includes a sonar sound wave 101, a sonar return wave 102, a transducer 103, a sonar return signal 104, a digital bearing module (DSM) 150, and a display 124. The DSM 150 includes a bandpass filter 106, a filtered return signal 108, a preamplifier 110, an amplified feedback signal 112, an analog to digital (A/D) converter 115, an intermediate frequency (If) feedback signal 116, a programmable logic device (PLD) 118, a digitized feedback signal 120, a microprocessor 122, a compressed signal 123, and a preamplifier gain curve 126 , illustrated in Figure 8C.

As illustrated in figure 7, the transducer 103 of the sonar system 100 mounted on the vessel 99 sends a sonar sound wave 101 into the water. When the sonar wave 101 hits an object, such as for example objects A, B and C, the sonar sound wave 101 is reflected and returns to the transducer 103 of the sonar system 100. The sonar return waves 102 hit the transducer 103 of the sonar system 100.

The transducer 103 can be either a through-hull, an in-hull, or a window-mounted transducer. The transducer 103 emits a sonar acoustic wave 101 at either 200 kHz or 50 kHz in the exemplary embodiments, any sonar wave may operate in accordance with the teachings of the present invention. The transducer 103 receives the sonar return wave 102, as a series of reflections of the sonar wave 101 from objects in the water and from the bottom of the body of water if this is within the range of the sonar detection capabilities. By measuring the return time of the echo, the propagated distance can be determined. The received echo is transformed into an electrical sonar return signal 104. The transducer 103 then sends the sonar return signal 104 to the DSM 150. The general characteristics of a typical sonar return signal 104 with three echoes are illustrated in extremely simplified form in Figure 8A.

The PLD controls the systems and adjusts such parameters as expected sharp sound depth depending on return echo characteristics or user-selected sonar parameters. In order to transmit the initial unmodified signal with the least clipping, the first bandpass filter 106 is set with the widest bandpass. The programmable bandpass filter that adjusts according to the depth of the application is in the PLD. A narrow bandpass filter is chosen by PLD to give good performance in deep water and a wider bandpass is chosen for shallow water. The transducer 103 has sharp sounds sent out every two seconds to twenty-six times per second, less often in deeper water (once every 20 seconds) to allow echo return time and more often (up to twenty-six times per second) in shallow water. PLD adapts a narrow bandpass with less frequent pulses and a wider bandpass with a more frequent pulse rate.

The signal 104, Figure 8A, has a carrier frequency of 50 to 200 kHz depending on the frequency of the transducer 103. Areas of increased magnitude in the waveform (A, B and C) indicate echoes of the signal reflected by objects in the water. Three echoes are indicated for simplified illustrative purposes. A return sonar can have more or fewer echoes. The position along the wave is indicative of the depth of the object. Echo A is a reflection from an object that is closer to the surface than object B, which in turn is closer than object C. The size m and the duration / of the area of increased size are indicative of the size of the object causing the reflected echo. However, due to the characteristics of acoustic sonar echo reflection, the echo reflections of objects of similar sizes can vary significantly in size depending on the distance to the object.

The DSM 150 is the receiver of the sonar system 100. A preferred embodiment of the DSM 150 is a digital implementation of a superheterodyne detector with a programmable frequency and a programmable bandwidth. The DSM 150 provides a programmable frequency range of 10 kHz to 200 kHz. DSM 150 receives the sonar return signal 104 from the transducer 103 and the sonar return signal 104 is fed to the bandpass filter 106 where it is bandpass limited.

The bandpass filter 106 is a typical broadband bandpass filter. The bandpass filter 106 is centered around 50 kHz or 200 kHz (depending on the frequency of the signal from transducer 1 as determined by PLD 118, based on water conditions such as depth) in a preferred embodiment. The choice of frequency is controlled by the PLD 118, based on various factors, as described below. This can easily be modified to filter any frequency between 10 kHz and 200 kHz. The bandpass filter 106 supplies a filtered signal, 8B to the preamplifier 110. The bandpass filter 106 sends the filtered return signal 108 to the preamplifier 110 for modification of the preamplifier in accordance with a variable gain.

The preamplifier 110 is a two-stage differential variable gain amplifier using general purpose operational amplifiers. The gain of the amplifier is set by the gain curve 126. The preamplifier 110 provides the DSM 150 with highly sensitive and wide dynamic range. The high sensitivity and wide dynamic range of the preamplifier 110 allows the use of a low-cost A/D converter with a lower resolution while maintaining the necessary dynamic range of the DSM 150. The preamplifier 110 is digitally controlled by the PLD 118. The preamplifier 110 amplifies the filtered return signal 108 based on preamplifier gain curve 126 which is sent by PLD 118.

Gain curve 126, Figure 8C, begins at a minimum gain level and increases the gain provided to amplifier 110 over time. In this way, echoes from greater depths receive significantly greater amplification than echoes from shallower depths. The shape of the gain curve, an important feature of the present invention, is believed to reduce the significant difference in magnitude between signals from shallow and deep echoes. Since sonar is significantly attenuated in water, the deeper the echo reflection source, the greater the attenuation. By providing a gain that increases with time, the difference in echo size can be partially compensated for. The size difference illustrated in Figure 8A has been partially compensated as illustrated in Figure 8D. The preamplifier 110 outputs the compensated return signal 112, Figure 8D, to the A/D converter 114.

The preamplifier gain curve 126 increases with depth and allows continuous reprogramming of a programmable filter, which provides optimal performance at any depth/pulse width. The preamplifier gain curve 126 provides a sensitivity timing control function that reduces surface grating and background noise by reducing gain in shallow water.

The A/D converter 114 is an analog to digital converter. A preferred embodiment of the A/D converter 114 is an LTC<®>1861 from Linear Technology Corporation. The LTC<®>1861 is a 12 bit A/D converter that offers a software-selectable two-channel multiplexer. The amplified carrier wave is transformed into digital by an A/D converter. The rate at which the carrier is converted to digital acts as a harmonic subsampling and the carrier is transformed into an intermediate frequency when digitized. The A/D converter 114 converts the amplified return signal 112 to an intermediate frequency sonar return signal 116. The A/D converter 114 outputs and sends the intermediate frequency sonar return signal 116 to the PLD 118.

PLD 118 is a programmable logic device and works as a coprocessor for main processor 122. A preferred embodiment of PLD 118 is a XILINX Spartan Ile XC2S300E. The PLD 118 uses a simplified implementation of a non-coherent window in-phase/quadrance (IQ) detector to digitally filter the intermediate frequency sonar return signal 116. Procedures for digitally filtering the intermediate frequency sonar return signal 116 are illustrated in Figures 2 through 5 and described in more detail below. The PLD 118 outputs the preamplifier gain curve 126 to control the preamplifier 110 and outputs a digitized signal 120 to the microprocessor 122 .

The microprocessor 122 is illustrated in Figure 6 and described in further detail below.

The display 124 receives the compressed signal 123 from the microprocessor 122 and outputs a display that may include the water depth, temperature and/or speed as measured by the transducer 103, and a dual or split frequency fishfinder display as detected by the DSM 150.

Figure 2 illustrates a functional diagram of a DSM 150 in accordance with the invention. The functional diagram of the DSM 150 includes the sonar return signal 104, the bandpass filter 106, the filtered return signal 108, the preamplifier 110, the amplified signal 112, an A/D converter 114, the intermediate frequency sonar return signal 116, the PLD 118, and the digitized return signal 120. The PLD 118 includes a window filter 206, a window return signal 208, a power measurement function 210, a power signal 212, a signal compression function 214, the preamplifier gain curve 126.

The digital signal in an example embodiment is filtered using a single point Fourier transform and a programmable Kaiser window. The size signal is transformed into a logarithmic representation.

The sonar return signal 104 is bandpass limited by the bandpass filter 106. The filtered return signal 108 is amplified by the preamplifier 110 based on the preamplifier gain curve 126 as sent from the PLD 118. The amplified return signal 112 is fed into the A/D converter 114, where it is mixed with a cosine (|> signal sent from PLD 118 and transformed into the intermediate frequency sonar return signal 116. PLD 118 sets the collection rate to ensure that the sonar return signal is above the Nyquist frequency. The effect of this subsampling is a frequency shift that is mathematically described as "mixing with a cosine ((> signal). PLD 118 detects the IF sonar return signal 116 using a simplified implementation of a non-coherent window in-phase / quadrature (I/Q) detector. The PLD 118 performs a Fourier transform of the intermediate frequency sonar return signal 116. The Fourier transform transforms the intermediate frequency sonar return signal 116 into a complex return signal that includes a true value which is an in-phase e detected signal and an imaginary value which is a quadrature detected signal. PLD 118 then multiplies the complex return signal with window filter 206. Window filter 206 sets the bandwidth and frequency rolloff. A preferred embodiment of the window filter 206 is a Kaiser window.

The power measurement function 210 measures the sonar return signal 104 by taking the absolute value of the window return signal 208 at a set frequency point.

Signal compression function 214 compresses the dynamic range of DSM 150 by taking the log2til power signal 212 and outputs the digitized return signal 120.

The center frequency of the DSM 150 can be changed by adjusting the value of the window filter 206.

The time/frequency response of the DSM 150 can be changed by adjusting the window function w[n] of the window filter 206. Figures 3A and 3B illustrate harmonic sampling of the sonar return signal. The DSM 150 samples the sonar wave using harmonic sampling. The DSM 150 "subsamples" sonar carriers containing frequencies above the Nyquist sampling rate (one half of the sampling rate, fs) as shown in Figure 3A. Such undersampling of the sonar return signal is possible since the sonar carrier wave is a baseband signal. Sonar carriers containing frequencies below the Nyquist rate are directly sampled, as shown in Figure 3B. The A/D converter 114 samples the amplified return signal 112 and converts the amplified return signal 112 into an intermediate frequency sonar return signal 116. Figure 4 illustrates an algorithm for approximating a phase shift. It is not necessary to calculate both the in-phase and quadrature components of the intermediate frequency sonar return et 116. An approximation can instead be made by approximating each component as a 90 degree phase shift of the other. Such an approximation reduces the complexity of the calculations by half with only a few resulting minor errors. PLD 118 approximates this phase shift by taking a Fourier transform 400 of the w[«]2(;n<?)/*r signal from the window filter 206 (shown in Figure 2), where sign(x) is equal to 1 if x > 0 , 0 if x = 0 and -1 if x < 0. The square of an in-phase detected signal 402 is taken by a quadrature function 404. A delay element 415 produces a pseudo-quadrature signal 416 to the in-phase detected in the signal 402. The square of the the pseudo quadrature signal 416 is then taken by a square function 417. The Euclidean distance 412 is calculated by taking the square root 410 of the sum 406 of the square of the in-phase signal 405 and the square of the quadrature signal 418. The digitized sonar return signal 120 is then compressed and output by the log2 function 414.

Figure 5 illustrates an algorithm for approximating the Euclidean distance between 90 degrees out of the phase vectors. This approximation also provides a significant reduction in complexity without any significant sampling loss to the sonar return signal.

The filtered digital signal is compared with the previous result, and an approximate magnitude is calculated where magnitude = max(y[n-l],y[n-l]) + 0.5<*>min(y[n],y[n-l] ). This magnitude signal is transformed into a logarithmic representation using an approximate log2(x) calculation. The approximated log2 is a piecewise linear approximation of a log2 function where the log value is linearly approximated between all integer log2 values.

The PLD 118 approximates the Euclidean distance 506 by first taking a Fourier transform 400 of the ^u>[«]2(;n<?)/<*r> signal from the window filter 206 (shown in Figure 2),

where the sign (x) is equal to 1 if x > 0, 0 if x = 0, and -1 if x < 0. The length of the longest in-phase detected signal 502 is then added to half the length of the shortest in-phase detected signal 503 to produce the approximation of the Euclidean distance 506. The digitized sonar return signal 120 is then compressed and output by the log2 function 414.

Figure 6 illustrates the logarithmic function implemented as a piecewise linear log2() function. A left shift of the magnitude signal from the digital sonar module is performed until a "1" appears in the most significant bit. The signal is then sent through a multiplexer and a register.

Claims (28)

1. Digital sounding module, characterized in that it comprises: a sonar transducer for transmitting acoustic sonar waves with a selected frequency and pulse rate and for receiving echoes from the transmitted acoustic sonar waves reflected from objects and for converting the received acoustic sonar waves into a first electrical signal, a bandpass filter for providing a first wide bandpass filtering on the first signal to produce a second signal, an amplifier for controlled amplification of the second signal to produce a third signal, an analog to digital converter for receiving the third signal and providing a digital sampling this signal, a programmable logic device for providing a variable gain control signal to the amplifier for normalizing the amplitude of reflected echoes within the second signal by increasing gain in proportion to the distance of said echo reflection source from the sonar transducer, and a microprocessor. 2. Digital gauge module according to claim 1, characterized in that the bandpass filter is a broadband filter. 3. Digital gauge module according to claim 2, characterized in that the bandpass filter is centered around 50 kHz or 20 kHz. 4. Digital gauge module according to claim 2, characterized in that the bandpass filter is modified to filter any frequency between 10 kHz and 200 kHz. 5. Digital sounding module according to claim 1, characterized in that the amplifier is programmable and provides an optimal performance at any depth/pulse width. 6. Digital bearing module according to claim 1, characterized in that the variable gain control signal normalizes the amplitude of a sonar pulse by increasing the gain over time from the initial reception of a pulse echo to the end of reception of a pulse echo. 7. Digital gauge module according to claim 6, characterized in that a gain to the amplifiers is set via a digitally programmable feedback resistor. 8. Digital sounding module according to claim 7, characterized in that the amplifier provides a sensitivity time control function which reduces surface grating and background noise by reducing the amplification in shallow water. 9. Digital gauge module according to claim 1, characterized in that the amplifier is digitally controlled by a gain curve sent from the programmable logic device. 10. Digital sounding module according to claim 9, characterized in that the gain curve increases with depth and allows a continuous reprogramming of a programmable filter. 11. Digital gauge module according to claim 1, characterized in that the analog to digital converter comprises: a conversion rate to convert the third signal into a digitized return signal which acts as a harmonic subsampling, in which the third signal is converted to an intermediate frequency. 12. Digital gauge module according to claim 1, characterized in that the programmable logic device works as a compressor for the main processor. 13. Digital gauge module according to claim 12, characterized in that the programmable logic device performs a simplified implementation of a non-coherent window in-phase/quadrance detector to digitally filter the intermediate frequency return signal. 14. Method of detecting a sonar carrier wave, the method comprising: receiving a sonar carrier wave, limiting the sonar carrier wave with a bandpass filter, amplifying the filtered sonar carrier wave with an amplifier, feeding the amplified sonar carrier wave to an analog to digital converter to produce an intermediate frequency of the sonar carrier wave , controlling the amplifier and analog to digital converter with a programmable logic device, digitally filtering the sonar carrier wave, and digitally detecting the sonar carrier wave. 15. Digital bearing module according to claim 1, in which the third signal is an amplified carrier wave. 16. Digital bearing module according to claim 1, wherein the programmable logic device also provides a harmonic subsampling rate, coscj), to the analog to digital converter such that the reflected transmitted sonar wave is above the Nyquist frequency. 17. Digital bearing module according to claim 1, in which the microprocessor provides an output for displaying the reflected echoes. 18. Digital bearing module according to claim 1, in which the programmable logic device provides an approximated log2 function, in which the log value is linearly approximated between all integer log2 values. 19. Digital gauge module according to claim 18, wherein the programmable logic device further comprises a window filter to transmit a window return signal to a power measurement function, which power measurement function transmits a power signal to a signal compression function, which signal compression function transmits a signal to the microprocessor. 20. Digital gauge module according to claim 1, in which the analog to digital converter is a 12 bit analog to digital converter. 21. Digital sounding module according to claim 11, in which the harmonic subsampling is performed by subsampling the third signal containing frequencies above the Nyquist sampling rate and directly sampling the third signals containing frequencies below the Nyquist rate. 22. Digital sounding module according to claim 11, wherein the programmable logic device implements a non-coherent windowed in-phase/quadrature, I/Q, detector and performs Fourier transformation of the third signal at intermediate frequency, which Fourier transformation transforms the third signal at intermediate frequency into a complex return signal including a real value which is an in-phase detected signal and an imaginary value which is a quadrature detected signal, which programmable logic device then multiplies the complex return signal with a window filter, which window filter sets the bandwidth and frequency, which programmable logic facility provision a cosc|> signal to the analog-to-digital converter, which programmable logic device sets the harmonic sub-sampling rate of the analog-to-digital converter such that the reflected transmitted sonar wave is above the Nyquist frequency, and a piecewise linear approximation of a log2 function, in which the log value is linearly approximated between all integer log2 values. 23. Digital sounding module according to claim 22, wherein the programmable logic device executes an algorithm for approximating a phase shift, wherein this approximation is performed by approximating either the in-phase component or the quadrature component of the third intermediate frequency signal as a 90 degree shift of the second, and the approximation cuts the complexity of the calculations in half. 24. Digital bearing module according to claim 23, in which the programmable logic device approximates the phase shift by taking a Fourier transform of the -signal from the window filter, where sign(x) is equal to 1 if x>0, 0 if x=0 and -1 if x<0. 25 Digital bearing module according to claim 23, wherein the programmable logic device approximates a Euclidean distance by taking a Fourier transform of the '^ dw\ ntyJne)' k signal from the window filter, where sign(x) is equal to 1 if x>0, 0 if x =0 and -1 if x<0, and then add the length of the longest in-phase detected signal to half the length of the shortest in-phase detected signal to produce the approximation of the second Euclidean distance. 26. Digital bearing module according to claim 22, wherein the window filter is a programmable Kaiser window filter. 27. The digital gauge module of claim 26, wherein the programmable logic device provides a programmable Kaiser window filter to transmit a window return signal to a power measurement function, which power measurement function transmits a power signal to a signal compression function, which signal compression function transmits a signal to a microprocessor. 28. Method according to claim 14, further comprehensive wherein the bandpass filter is an adjustable bandpass filter, wherein the amplifier is a two-stage differential variable gain amplifier wherein the gain of the amplifier is set by a gain curve received from the programmable logic device, wherein the feeding comprises feeding the amplified sonar carrier wave into the analog to digital converter which analog to digital converter converts the amplified sonar carrier wave at a harmonic subsampling rate, coscj), supplied from the programmable logic device to produce a digital intermediate frequency of the sonar carrier wave, wherein the programmable logic device controls the amplifier by performing the detection and filtering on the intermediate frequency sonar carrier with a non-coherent windowed in-phase/quadrance detector, compressing the digitized sonar carrier wave with a log2 function, and providing output from the programmable logic device to a microprocessor and a display.