An IC Operational Transconductance Amplifier (OTA) With Power Capability
An IC Operational Transconductance Amplifier (OTA) With Power Capability
An IC Operational Transconductance Amplifier (OTA) With Power Capability
[ /Title (AN60 77) /Subject (An IC Operational Transc onductance Amplifier (OTA) With Power Capability) /Autho r () /Keywords (Intersil Corporation, power switch, power amplifier, programmable power switch) /Creator ()
V+
Z OUTPUT
Q1 Q2
NON-INVERTING INPUT
5 IABC
V-
What Is an OTA?
The OTA, operational transconductance amplifier, concept is as basic as the transistor; once understood, it will broaden the designer's horizons to new boundaries and make realizable designs that were previously unobtainable. Figure 1 shows an equivalent diagram of the OTA. The differential input circuit is the same as that found on many modern operational amplifiers. The remainder of the OTA is composed of current mirrors as shown in Figure 2. The geometry of these mirrors is such that the current gain is unity. Thus, by highly degenerating the current mirrors, the output current is precisely defined by the differential input amplifier. Figure 3 shows the output current transfer characteristic of the amplifier. The shape of this characteristic remains constant and is independent of supply voltage. Only the maximum current is modified by the bias current.
7 2 RIN V+ OTA
FIGURE 3. THE OUTPUT CURRENT TRANSFER CHARACTERISTIC OF THE OTA IS THE SAME AS THAT OF AN IDEALIZED DIFFERENTIAL AMPLIFIER
ein
gm ein
+
The major controlling factor in the OTA is the input amplifier bias current IABC; as explained in Figure 1, the total output current and gm are controlled by this current. In addition, the input bias current, input resistance, total supply current, and output resistance are all proportional to this amplifier bias current. These factors provide the key to the performance of this most flexible device, an idealized differential amplifier, i.e., a circuit in which differential input to single ended output conversion can be realized. With this knowledge of the basics of the OTA, it is possible to explore some of the applications of the device.
DC Gain Control
The methods of providing DC gain control functions are numerous. Each has its advantage: simplicity, low cost, high level control, low distortion. Many manufacturers who have nothing better to offer propose the use of a four quadrant
the improvement in linearity of the transfer characteristic. Reduced input impedance does result from this shunt connection. Similar techniques could be used on the OTA output, but then the output signal would be reduced and the correction circuitry further removed from the source of non linearity. It must be emphasized that the input circuitry is differential.
7 6 5 4 3 2 1 0 0.1 10A 500A IABC CA3080A S/N RATIO DIODE CURRENT = 0mA
THD (PERCENT)
100 80 60 40 20 0 1.0 10 100 1.0V INPUT VOLTAGE (mV) S/N RATIO (dB) S/N RATIO (dB) S/N RATIO (dB)
THD
OTA CA3080A 3 +
FIGURE 5A.
7 DIODE CURRENT = 0.5mA 6 5 4 3 10A 2 1 0 THD 40 20 0 500A 100 80 60
51
THD (PERCENT)
As long as the differential input signal to the OTA remains under 50mV peak-to-peak, the deviation from a linear transfer will remain under 5 percent. Of course, the total harmonic distortion will be considerably less than this value. Signal excursions beyond this point only result in an undesired compressed output. The reason for this compression can be seen in the transfer characteristic of the differential amplifier in Figure 3. Also shown in Figure 3 is a curve depicting the departure from a linear line of this transfer characteristic. The actual performance of the circuit shown in Figure 4 is plotted in Figure 5. Both signal to noise ratio and total harmonic distortion are shown as a function of signal input. Figures 5B and 5C show how the signal handling capability of the circuit is extended through the connection of diodes on the input as shown in Figure 6 [2]. Figure 7 shows total system gain as a function of amplifier bias current for several values of diode current. Figure 8 shows an oscilloscope reproduction of the CA3080 transfer characteristic as applied to the circuit of Figure 4. The oscilloscope reproduction of Figure 9 was obtained with the circuit shown in Figure 6. Note 2
10
100
1V
10V
FIGURE 5B.
7 DIODE CURRENT = 1mA 6 5 4 3 2 1 0 IABC CA3080A S/N RATIO 10A THD 500A 100 80 60 40 20 0 10V
FIGURE 5C. FIGURE 5. PERFORMANCE CURVES FOR THE CIRCUIT OF FIGURES 4 AND 6
2K
Transistors from CA3046 array. AGC System with extended input range. FIGURE 6. A CIRCUIT SHOWING HOW THE SIGNAL HANDLING CAPABILITY OF THE CIRCUIT OF FIGURE 4 CAN BE EXTENDED THROUGH THE CONNECTION OF DIODES ON THE INPUT
0.01
100
200
500
FIGURE 7. TOTAL SYSTEM GAIN vs AMPLIFIER BIAS CURRENT FOR SEVERAL VALUES OF DIODE CURRENT
Horizontal: 0.5V/Div. Vertical: 50A/Div., IABC = 100A, Diode Current = 1mA FIGURE 9. CA3080 TRANSFER CHARACTERISTIC FOR THE CIRCUIT OF FIGURE 6
The CA3094
The CA3094 offers a unique combination of characteristics that suit it ideally for use as a programmable gain block for audio power amplifiers. It is a transconductance amplifier in which gain and open-loop bandwidth can be controlled between wide limits. The device has a large reserve of output-current capability, and breakdown and power dissipation ratings sufficiently high to allow it to drive a complementary pair of transistors. For example, a 12W power amplifier stage (8 load) can be driven with peak currents of 35mA (assuming a minimum output transistor beta of 50) and supply voltages of 18V. In this application, the CA3094A is operated substantially below its supply voltage rating of 44V max. and its dissipation rating of 1.6W max. Also in this application, a high value of open-loop gain suggests the possibility of precise adjustment of frequency response characteristics by adjustment of impedances in the feedback networks.
+ -
R4 R1
R5
DIFFERENTIAL INPUT
R2 R3 + R6
OUTPUT R 1 = R3 R4 R5 = R6 R7
R7
+V
6 4 V-
OUTPUT RL 10K
Cost Advantages
In addition to the savings resulting from reduced parts count and circuit size, the use of the CA3094 leads to further savings in the power supply system. Typical values of power supply rejection and common-mode rejection are 90dB and 100dB, respectively. An amplifier with 40dB of gain and 90dB of power supply rejection would require 316mV of power supply ripple to produce 1mV of hum at the output. Thus, no further filtering is required other than that given by the energy storage capacitor at the output of the rectifier system.
A = gm RL at 500A, IABC: gm 10mS. A = 10mS x 10K = 100. FIGURE 11. A DIFFERENTIAL TO SINGLE ENDED CONVERSION CIRCUIT WITHOUT PRECISION RESISTORS
EO
FIGURE 12. BLOCK DIAGRAM OF A SYSTEM USING A LOSSER TYPE TONE CONTROL CIRCUIT
ESIG = 40mV ESIG = 40mV EO = 4V ATOTAL = 60dB
AMPLIFIER WITH FEEDBACK TONE CONTROLS EN = 5 x 10-6 EN = 4.03 x 10-3 = 990 AT MAX VOL
EO EN
4 4.03 x 10-3
FIGURE 13. A SYSTEM IN WHICH TONE CONTROLS ARE IMPLICIT IN THE FEEDBACK CIRCUIT OF THE POWER AMPLIFIER
A bias arrangement that can be accomplished at lower cost than those already described replaces the Vbe multiplier with a 1N5391 diode in series with an 8.2 resistor. This arrangement does not provide the degree of bias stability of the Vbe multiplier, but is adequate for many applications.
Tone-Controls
The tone controls, the essential elements of the feedback system, are located in two sets of parallel paths. The bass network includes R3, R4, R5, C4, and C5. C6 blocks the DC from the feedback network so that the DC gain from input to the feedback takeoff point is unity. The residual DC output voltage at the speaker terminals is then
R 11 + R 12 R 1 -------------------------- I ABC R 12
Output Biasing
Instead of the usual two-diode arrangement for establishing idling currents in Q1 and Q2, a Vbe Multiplier, transistor Q3, is used. This method of biasing establishes the voltage between the base of Q1 and the base of Q2 at a constant multiple of the base to emitter voltage of a single transistor while maintaining a low variational impedance between its collector and emitter (see Appendix A). If transistor Q3 is mounted in intimate thermal contact with the output units, the operating temperature of the heat sink forces the Vbe of Q3 up and down inversely with heat-sink temperature. The voltage bias between the bases of Q1 and Q2 varies inversely with heat sink temperature and tends to keep the idling current in Q1 and Q2 constant.
The treble network consists of R7, R8, R9, R10, C7, C8, C9, and C10. Resistors R7 and R9 limit the maximum available cut and boost, respectively. The boost limit is useful in curtailing heating due to finite turn-off time in the output units. The limit is also desirable when there are tape recorders nearby. The cut limit aids the stability of the amplifier by cutting the loop gain at higher frequencies where phase shifts become significant. In cases in which absolute stability under all load conditions is required, it may be necessary to insert a small inductor in the output lead to isolate the circuit from capacitive loads. A 3H inductor (1A) in parallel with a 22 resistor is adequate. The derivation of circuit constants is shown in Appendix B. Curves of control action versus electrical rotation are also given.
C10
T1 4700F D1 D2
0.12F R9 68
120V AC
C1 ES R1 2
5F +
D3 Q1 2N6292
7 1
4700 F
D4
CA3094
R11 330
R12 47
R2 1.8M
5 R6 680K
CUT (CCW)
JUMPER
FIGURE 14. A COMPLETE POWER AMPLIFIER USING THE CA3094 AND THREE ADDITIONAL TRANSISTORS
+36V VCC R5 1.2M 0.01F 820 5F + 1.8K 0.001F R1 220 1W +
220K 5% 68
0.12F
TREBLE 15K
5.6K
47
220K 5%
25F 1/2VCC - +
Figure 16 is a plot of the measured response of the complete amplifier at the extremes of tone control rotation. A comparison of Figure 16 with the computed curves of Figure B4 (Appendix B) shows good agreement. The total harmonic distortion of the amplifier with an unregulated power supply is shown in Figure 17; IM distortion is plotted in Figure 18. Hum and noise are typically 700V at the output, or 83dB down.
60 55 50 45 EO /ES (dB) 40 35 30 25 20 15 10 10 2 BASS CUT 3 68 2 100 3 68 2 1000 3 68 2 10K 3 68 100K TREBLE CUT FLAT BASS BOOST TREBLE BOOST
FREQUENCY (Hz)
FIGURE 16. THE MEASURED RESPONSE OF THE AMPLIFIER AT EXTREMES OF TONE CONTROL ROTATION
1.0 TOTAL HARMONIC DISTORTION (%) 0.9 0.8 0.7 0.6 0.5 0.4 0.3 0.2 0.1 0 2 4 6 8 10 1kHz 26kHz 16kHz 4kHz 12
2kHz
FIGURE 17. TOTAL HARMONIC DISTORTION OF THE AMPLIFIER WITH AN UNREGULATED POWER SUPPLY
4kHz
16kHz
26kHz
1kHz
The derivative of Equation A1 with respect to I yields the incremental impedance of the Vbe multiplier:
dE 1 R1 K3 R2 BR1 ---------- = Z = ------------ + 1 + ------------------------- ------------------------dI +1 ( + 1 )R 2 R 2 I e + K 3
50F
(EQ. A3)
EOUT
(EQ. A5)
Ie = IS e
0.002F 680pF 1.5M 3.9K + 5F 1F 3 ES 56K 5 56K 2 CA3080 + 4
7 6 EOUT
Using the values shown in Figure 14, plus data on the 2N6292 (a typical transistor that could be used in the circuit), the dynamic impedance of the circuit at a total current of 40mA is found to be 4.6. In the actual design of the Vbe multiplier, the value of IR2 must be greater than Vbe or the transistor will never become forward biased.
50F
(EQ. A1)
E1 R1 I Vbe
The asymptotic high-frequency gain is obtained by letting S increase without limit in each expression:
R1 + R2 Bass Boost: A HIGH = -------------------R2
R2
Ie
R1 + R2 Bass Cut: A HIGH = -------------------R2 C 3 + C 4 Treble Boost: A HIGH = 1 + C 1 -------------------- C3 C4 C1 C4 C 2 + -------------------C1 + C4 Treble Cut: A HIGH = ---------------------------------C1 + C2
The value of Vbe is itself dependent on the emitter current of the transistor, which is, in turn, dependent on the input current I since:
V be I e = I --------R2 (EQ. A2)
Bass Circuit: A LOW ( Boost ) = 10A MID A MID A LOW ( Cut ) = ------------10 Treble Circuit: A HIGH ( Boost ) = 10A MID A MID A HIGH ( Cut ) = ------------10
C1 R2
C2 R1
+
ES
R3
R1 EO
R2
R3
+ ES
EO
( R2 R3 + R1 R3 ) R 1 + R 2 + R 3 1 + SC ----------------------------------------1 ( R + R + R )A = ---------------------------------- 1 2 3 R2 --------------------------------------------------------------1 + SR 3 C 1 R1 + R2 + R3 A LOW FREQUENCY = ---------------------------------R2 FIGURE B1 (A). BASS BOOST
C3
( R2 R3 + R1 R3 ) 1 + SC 2 ----------------------------------------( R1 + R2 + R3 ) R 1 + R 2 + R 3 A = ---------------------------------- -------------------------------------------------------------- R2 + R3 R2 R3 1 + SC2 -------------------- R 2 + R 3 R1 + R2 + R3 A LOW FREQUENCY = ---------------------------------R2 + R3 FIGURE B1 (B). BASS CUT
C2 REFF
C1
C4 REFF EO C1
C4
+ ES
+ ES
EO
( C1 C4 + C3 C4 + C1 C3 ) C 1 + C 4 1 + SR EFF --------------------------------------------------------------A = -------------------- ( C1 + C4 ) C 4 --------------------------------------------------------------------------------------------1 + SR EFF C 3 C1 + C4 A LOW FREQUENCY = -------------------C4 FIGURE B1 (C). TREBLE BOOST
( C1 C4 + C2 C4 + C1 C2 ) C 1 + C 4 1 + SR EFF --------------------------------------------------------------A = -------------------- ( C1 + C4 ) C 4 --------------------------------------------------------------------------------------------1 + SR EFF ( C 1 + C 2 ) C1 + C4 A LOW FREQUENCY = -------------------C4 FIGURE B1 (D). TREBLE CUT
FIGURE B1. FOUR OPERATIONAL AMPLIFIER CIRCUIT CONFIGURATIONS AND THE GAIN EXPRESSIONS FOR EACH
C2 C1 +
C2 R3 ( R1 + R2 ) and C 1 R = ----------------------------------------3 R1 + R2 + R3
C1
CW R2
R1
CCW
C4
R1 R2 = -------------------- ( C 1 + C 2 ) R1 + R2
1 2 1 4 2 4 1 2 and R 2 C 3 = -------------------- ---------------------------------------------------------------
R R (C C + C C + C C ) ( C1 + C4 ) R 1 + R 2
C3 C2 +
since C 1 100C 2 , C2 = C 3 and C 1 = 10C 4 , R1 = 9R 2 To make the controls work in the circuit of Figure 14, breaks were set at 1000Hz: for the base control 0.1C1 R 3 = -------------------------and for the treble control R 1 C 3 = -------------------------1 2 1000 1 2 1000
FIGURE B2 (B). TREBLE CONTROL FIGURE B2. CUT AND BOOST BASS AND TREBLE CONTROLS THAT HAVE THE CHARACTERISTICS OF THE CIRCUITS IN FIGURE B1
ES 0.02 F
EO
Figure B3 is a plot of the response with bass and treble tone controls combined at various settings of both controls. The values shown are the practical ones used in the actual design. Figure B4 shows the information of Figure B3 replotted as a function of electrical rotation. The ideal taper for each control would be the complement of the 100Hz plot for the bass control and the 10kHz response for the treble control. The mechanical center should occur at the crossover point in each case.
35 Q = 0.99 30 25
Q =0.914 20 Q = 0.85 15 10 5 0 Q=0 10 100 Q = 0.75 Q = 0.5 Q = 0.2 P = 0.92 P = 0.8 P = 0.5 P = 0.3 P = 0.1 P = 0.05 1000 FREQUENCY (Hz) 10K
P=0 100K
FIGURE B3. A PLOT OF THE RESPONSE OF THE CIRCUIT OF FIGURE 14 WITH BASS AND TREBLE TONE CONTROLS COMBINED AT VARIOUS SETTINGS OF BOTH CONTROLS
10
FIGURE B4 (A).
FIGURE B4 (B). FIGURE B4. THE INFORMATION OF FIGURE B3 PLOTTED AS A FUNCTION OF ELECTRICAL ROTATION
References
For Intersil documents available on the internet, see web site http://www.intersil.com/ [1] AN6668 Application Note, Applications of the CA3080 and CA3080A High Performance Operational Transconductance Amplifiers, H. A. Wittlinger, Intersil Corporation. [2] A New Wide-Band Amplifier Technique, B. Gilbert, IEEE Journal of Solid State Circuits, Vol. SC-3, No. 4, December, 1968. [3] Trackability, James A. Kogar, Audio, December, 1966.
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