346 IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS, VOL. 34, NO.

2, MARCH/APRIL 1998

Sensorless Vector Controller for a

Synchronous Reluctance Motor
Milutin G. Jovanović, Robert E. Betz, Member, IEEE, and Don Platt, Member, IEEE

Abstract— A new high-performance sensorless speed vector The others carried out detailed studies of different estimation
controller that implements the maximum torque per ampere algorithms and, while significant from a theoretical viewpoint,
control strategy for the inverter-driven synchronous reluctance the results are preliminary and with no experimental support
machine is presented in this paper. It is based on a parameter-
dependent technique for on-line estimation of rotor position [4], [5], [7].
and angular velocity at the control rate. The current ripple The basic principle used by most techniques [1], [3]–[6] is
principle is used to estimate position. The estimates are fed to the variation of stator inductances with rotor position, which
a conventional closed-loop observer to predict the new position allows position estimation down to zero speed. This variation
and angular velocity. The very high accuracy of the sensorless
is enhanced with larger saliency ratios and can be detected
control algorithm at both low and high speeds is confirmed by
experimental results. in the switching ripples on the current waveforms [4]–[6]
or by the magnetic coupling coefficients between windings
Index Terms— Angular velocity observer, sensorless control,
[3]. These two approaches are combined in [1] for both low
synchronous reluctance machines.
and high speeds, respectively. A Kalman filter is then used to
obtain the optimal position and velocity estimates. An effective
I. INTRODUCTION flux-oriented speed controller based on the torque vector
control principle [7] is described in [2]. Unlike the previous
T HE high-performance rotor-oriented vector control of
the cageless inverter-fed synchronous reluctance machine
(Syncrel) requires an accurate knowledge of rotor position
techniques, it does not require a rotor position information at
all, but only needs to know the flux position. Reference [8]
to convert the measurable stator quantities into their rotating outlines the versatile sensorless scheme potentially applicable
frame equivalents. The position information is traditionally to all salient ac machines.
provided by measurements using costly transducers, such as The main contribution of the work presented in [2], [6], and
optical encoders or magnetic resolvers. In order to make the [8] is the development of viable position estimation techniques
Syncrel drive less expensive and more robust compared to independent of machine parameters and operating point. The
its induction machine counterpart, recent work has focused advantage of the control algorithm in [1] is that it makes it
on investigating various position estimation techniques that possible to obtain satisfactory control of shaft speed over the
would allow the removal of shaft position sensors [1]–[7]. whole range, in contrast to those in [6] and [2], the applications
An additional motivation for the renewed interest in Syncrel of which are limited to low and high speeds, respectively.
is undoubtedly its inherent saliency and, thus, amenability to The limitations of the existing control algorithms are as
sensorless operation. follows: 1) relatively modest position estimation accuracy of
Although the sensorless Syncrel drive is of lower cost approximately 10 electrical [1], [6]; 2) low estimate update
and more mechanically robust, its control performance is rate and, hence, poor control performance [3]; 3) the use
usually compromised, and controller design is generally more of special switching procedures to force the inverter into a
complicated. The literature consistently shows that it is not desired diagnostic state, in order to carry out measurements
easy to implement sensorless control in real time, and only relevant for estimation [1], [3]; and 4) an injection of special
some authors have succeeded in achieving this [1], [2], [6]. high-frequency signals which have to be filtered to obtain the
position estimate [8].
Paper IPCSD 97–66, presented at the 1996 Industry Applications Society This paper is, in some sense, an extension of the work
Annual Meeting, San Diego, CA, October 6–10, and approved for publication
in the IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS by the Industrial Drives initiated in [4]. It presents a real-time software implementation
Committee of the IEEE Industry Applications Society. Manuscript released of a new observer-based sensorless algorithm for the maximum
for publication October 20, 1997. torque per ampere control strategy that overcomes some of
M. G. Jovanović was with the Department of Electrical and Computer
Engineering, University of Newcastle, Callaghan, NSW 2308, Australia. He the above deficiencies. It includes a parameter-dependent
is now with the School of Electrical Engineering, Electronics and Physics, technique for estimating the rotor position on-line from the
Liverpool John Moores University, Liverpool L3 3AF, U.K. measurements of two stator currents and the inverter dc-link
R. E. Betz is with the Institute of Energy Technology, Department of
Electrical Energy Conservation, Aalborg University, DK-9220 Aalborg East, voltage. The position is estimated using the saturated –
Denmark, on leave from the Department of Electrical and Computer Engi- model equations in discrete form and linear approximation
neering, University of Newcastle, Callaghan, NSW 2308, Australia. of the rate of change of current in the switching ripples. This
D. Platt is with the Department of Electrical and Computer Engineering,
University of Wollongong, Wollongong, NSW 2522, Australia. makes the control algorithm applicable throughout the entire
Publisher Item Identifier S 0093-9994(98)02563-8. speed range and under all loading conditions of the machine.
0093–9994/98$10.00  1998 IEEE
JOVANOVIĆ et al.: SENSORLESS VECTOR CONTROLLER FOR A SYNCHRONOUS RELUCTANCE MOTOR 347

A startup procedure is required to establish the initial position

of the rotor axis, after which the algorithm can be run.
The excellent performance of the controller is experimentally
verified on a 5.8-kW axially laminated Syncrel.

II. POSITION ESTIMATION TECHNIQUE

The fundamental principle of the estimation technique is
the detection of stator winding inductance variations with rotor
position, by examination of the switching ripple on the current
waveforms. It actually represents an improved version of the
estimation algorithm presented by the same authors in [4],
which is similar to that in [5]. A significant improvement is
that the saturation effects are included.
Consider conventional Park’s model equations for the
Syncrel using standard notation:

Fig. 1. Phasor diagram.

nonzero- and two zero-voltage phasors. The nonzero phasors

are displaced by radians
(4)

where is a dc-link voltage magnitude.

(1) Assuming no neutral connection, the currents of the
machine can be determined from the measured instantaneous
Developing the voltage equations in terms of currents and values of two phase currents (e.g., and using an
taking into account the saturation in the high-permeance rotor expression
axis axis), one obtains

(5)

Differentiating further (3) with , one gets

(2)

where is the induced voltage and

The value of stator resistance as measured by a simple dc
test is somewhat increased to compensate for skin effects and (6)
temperature variations. The measurements for the saturation where is the phasor for the incremental current, and and
characteristic, versus and versus are stored in a are the phasors that correspond to the current samples at the
lookup table, the values of which have been determined by beginning (after switching) and the end (before next switching)
off-line testing (see the Appendix). The axis is dominated of a particular switching interval, respectively. The parameter
by air, therefore, is assumed to be constant. All relevant in (2) is the time increment between the sampling instants.
machine parameters are summarized in the Appendix. The number of sampling intervals contained in can be
In order to solve (2) for the rotor electrical angular position accurately determined from a knowledge of the sampling rate
it is necessary to relate quantities in the rotor frame and the inverter switching times which are implicitly available,
directly to their measurable counterparts in the stator as the PWM waveform is generated by software. A simple
frame of reference. This can be achieved by applying the algorithm is implemented to achieve this.
following transformation equation, which can be derived from Note that the first and second terms on the right-hand side
the vector diagram in Fig. 1: of the first equation of (6) respectively account for change
(3) in current phasor and movement of the frame over
The linear approximation of the current derivative used in (6)
where represents either the induced voltage current , or gives reasonable accuracy in PWM drives, as is usually
terminal voltage very small in relation to the time constant of the machine.
Consider a Syncrel being fed by a space-vector-based An inspection of the experimental current ripple waveforms
pulsewidth modulation (PWM) inverter [9]. Depending on confirms this assumption, since they are virtually triangular in
the switching status of the legs, the inverter can produce six shape.
348 IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS, VOL. 34, NO. 2, MARCH/APRIL 1998

After substituting and into (6), the Such a high estimation rate has two important implications.
rotor frame components of become Firstly, further processing is possible to help eliminate erro-
neous results and obtain the most accurate value to be used in
the control procedure. This is achieved with a standard angular
(7) velocity observer [10], which allows accurate estimates of both
and to be deduced from the noisy estimates coming
where and are defined in Fig. 1. From the same figure, directly from the solution of (10). Note that the use of the
the following relations are also obvious: observer introduces another layer of parameter dependence, as
the load parameters need to be known.
Secondly, the technique is applicable at any speed, including
(8) standstill, and under all loading conditions of the machine. An
estimate is less sensitive to noise and measurement quantiza-
The initial induced voltage equations (2), together with (8), tion and, hence, more accurate for larger current ripples. At
can be now rearranged to the form low speeds, the current ripple is low, as there is no back EMF
to decrease the currents when the zero vector is applied. In
addition, when voltage is being applied from the supply, the
pulses are narrow. On the other hand, at higher speeds, the
accuracy is improved due to the back EMF induced ripple. At
either speed, however, the estimation errors are smaller for the
(9) inverter leg switching states when most of the driving voltage
(applied or induced) is in the low permeance axis, as the
These equations can be solved for
rate of change of current is higher.
For the best performance of the position estimator, the
current sampling rate should be sufficiently high, so that the
extremities of a switching cycle can be accurately determined.
A sufficient rate is clearly dependent on the switching rate
of the inverter; for high switching rates, the current sampling
(10)
system should sample faster as a particular switching voltage is
where being held for a shorted time. In addition, high sampling rates
allow more accurate determination of the current values at the
beginning and end of a switching cycle. This is important in
order to maximize the measured current ripple and becomes
increasingly important at lower speeds.
The existing A/D sampling system (see Section V-A) with
the maximum sampling rate of 130 kHz is not the best solution
in this regard. In addition, it is a multiplexed board is
sampled first, then , and, finally, This reduces the
effective current sampling rate to approximately 43 kHz, as
three channels are sampled. Furthermore, since four current
samples (two of each current) are required for estimation, the
The terminal voltages and currents in the stationary frame applied voltage pulsewidth should be at least about 38 s
(Fig. 1) used in the previous expressions are determined by (4) (corresponds to 26 kHz) in order for this to be possible. This
and (5), respectively. The corresponding incremental currents means that, at low speeds, the estimates during the applied
are defined in (6). voltage periods often cannot be used, as the voltage pulses are
It is also possible to solve for using (9), however, the narrow. Consequently, fewer position estimates are available
resultant expression is more sensitive to numerical errors, for further filtering, which together with the increased effect of
noise effects, and other real-time implementation inaccuracies. A/D quantization and noise, should create poorer estimates at
The value derived from this solution would require additional low speeds. This conjecture is confirmed by the experimental
filtering and computation, which would, in turn, compromise results presented in Section VI.
the control performance. These reasons contributed to the The fact that the sampling is not simultaneous imposes some
abandonment of this approach. difficulties in processing the current samples for use in the
The main advantage of the proposed estimation technique is estimation algorithm. To compensate for delay in sampling the
that it provides one estimate per leg switched. Therefore, over -phase current relative to current in phase, the first sample
one control interval, there is generally a number of solutions of of at the beginning of the considered switching interval
(9). Considering that the PWM algorithm being implemented is linearly extrapolated one sampling interval backward as the
has double-edged modulation [9], there are usually four esti- slope of the current ripple is known. The resultant value is
mates available, as four legs are normally switched per control then used as in (5) and has been experimentally shown to
interval. improve the estimator accuracy.
JOVANOVIĆ et al.: SENSORLESS VECTOR CONTROLLER FOR A SYNCHRONOUS RELUCTANCE MOTOR 349

Fig. 2. Sensorless algorithm.

III. SENSORLESS CONTROL ALGORITHM

The heart of the sensorless control algorithm shown in Fig. 2
is the position estimator based on the solutions to (9). The
output from this algorithm is fed into a closed-loop observer
based on the load model equations of (1) [10]. The merit
of using an observer is that both angular velocity and Fig. 3. Experimental vector controller.
position can be accurately predicted without any knowledge
of past information and, hence, without delay, which is crucial the axis, and no alignment torque would result. This potential
for high-performance control. The convergence of the control problem can be easily overcome by specifying two different
algorithm and machine operating stability are simply a matter spatial angles for , such that The previous
of appropriately tuning the observer gains, the main criteria condition is introduced just to have a reasonably high starting
being the quality of the estimates being produced by the torque while moving the rotor from its initial -axis (or -axis)
position estimation algorithm. If the estimates are known to be alignment position to a new one In order
good, then the observer feedback gain is increased, otherwise, to improve the test accuracy, the current magnitude should be
it is decreased. Clearly, this implies that gain scheduling is large enough to provide the torque sufficient to surpass bearing
required to get good estimates throughout the entire speed friction around the alignment position.
range of the machine. The real-time implementation includes a digital current pro-
The position estimator evaluates the electrical angle of the portional integral (PI) controller with circular voltage limiting
axis for each of the legs switching instants of the previous identical to that used in the vector controller. The difference is
control interval using current samples and and average that it performs its actions in the stationary frame (Fig.
dc-link voltage It then selects the best estimate , 1). The components of the desired current phasor are
i.e., the one having the least variation from the observer last used as references to the current loops, whereas the feedback
prediction The is also used to estimate the -axis currents are obtained from the average phase currents
currents for indexing the lookup tables containing the and using (5). The desired output voltages from the
and values required for calculating and then the controller are then passed into a PWM algorithm.
position estimates. The average currents and and
the corresponding -axis inductances are obtained similarly,
but using the average phase currents and IV. EXPERIMENTAL CONTROLLER DESIGN
The value and the average torque estimated using (1) A functional block diagram of the algorithmic structure
are fed as an input to the observer to generate a filtered and of the Syncrel speed vector controller is shown in Fig. 3.
relevant at the beginning of the next control interval. The and The current loops (one for each rotor axis) and angular
estimates, as well as and , are then used throughout the velocity loop are all conventional PI regulators with integrator
rest of the vector control algorithm for the control prediction. antiwindup [11]. An obvious advantage of the controller is a
How this is achieved is discussed in Section IV. possibility of controlling the machine speed with or without a
shaft position sensor. However, only the issues related to the
A. Startup Procedure sensorless control are of interest in this paper.
In order to properly initialize the observer (which is neces- The control algorithm and the startup procedure, including
sary for its initial position estimate to be accurate), the rotor the PWM waveform generation, are implemented entirely in
initial position should be found as accurately as possible. software written in C using floating-point arithmetic. This was
A simple startup procedure is carried out for this purpose. It a design decision to allow the maximum flexibility in the
utilizes a minimum reluctance principle, i.e., the tendency of experimental system. The control computer is a 90-MHz Intel
the rotor axis to align with the stator MMF phasor. The idea Pentium-processor-based PC. An advantage of this platform is
is, therefore, to set up the average current vector in a desired its low cost and the large variety of software development tools
angle with respect to the axis (Fig. 1) and then allow the available. System parameters are monitored with the integrated
rotor to take up the alignment position when it is obvious that PC screen display. The inverter hardware and encoder are
Clearly, this simple procedure will not work in interfaced to the CPU bus using two programmable I/O boards
every situation. For example, the initial could coincide with with interrupt capabilities, the control signal generation board
350 IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS, VOL. 34, NO. 2, MARCH/APRIL 1998

and A/D data conversion board. These are considered in more • Samples corresponding to the previous control interval are
detail in Section V. averaged and either the startup procedure or the sensorless
The control code itself essentially consists of the main pro- algorithm from Fig. 2 is run.
gram and two interrupt routines servicing the boards interrupts. • A simple regeneration control strategy is implemented.
The associated bit on the control board output port
A. Main Program is pulsed high to switch the regeneration transistor on
whenever the average dc-link voltage exceeds the user-
The main program carries out any operator interfacing. specified value. When the voltage drops below the lower
Before running the main loop, it initializes the software, limit, the bit is pulsed low and device is turned off.
programs up the I/O boards and Direct Memory Access • The control currents and are predicted using
(DMA) controller, generates contactors closing signals, and a simple Euler approximation of the current differentials
sets up interrupts. in (2). This and the remainder of the algorithm up to the
The main loop polls the keyboard to check for the program PWM waveform generation are disabled while carrying
stop command or the new values for the speed reference, out the startup procedure.
as well as current magnitude and angle while running the • The speed PI algorithm is executed to generate the
initial rotor position startup procedure. The user can also desired torque The current reference generator then
select and change on-line the desired control variables to be determines the minimum currents and required
displayed on the PC monitor or oscilloscope. Similarly, the for the machine to produce this torque. The optimal
transition to execution of the actual control algorithm (after versus characteristic is precomputed using Matlab and
determination of the initial rotor position) is accomplished by stored in a lookup table.
a user command. • The predicted state feedback voltages are calculated to
Another important function of the program main loop is compensate for the rotational voltages. These appear to
examining the status of the hardware trip indication bit of the the current regulators as unknown state disturbances [12].
control board port. Upon normal termination of the main loop • The current PI controllers are run to obtain the predictions
(stop signal received or hardware trip detected), the program for the voltages to be applied to the machine. These
does the usual post-interrupt procedure, resets the hardware, are fed into the space-vector-based PWM generator [9]
and opens all the contactors. together with In the case of the startup procedure, the
value is zero, as the current controller is then stator
B. A/D Board Interrupt Routine oriented.
The “DMA complete” interrupt routine is fairly straight- • The desired voltages are converted into the corre-
forward. It sets up the A/D board for the next set of A/D sponding inverter legs switching pattern by executing the
conversions and then converts the contents of the DMA PWM algorithm with overlap (dead) time compensation
buffer into floating-point numbers representing the actual [13].
measurements. These correspond to and in • The switching times are programmed into the appropriate
Figs. 2 and 3. It is the task of the user to ensure that the timers to generate the actual firing waveforms for the
samples are available at the beginning of each control interval. insulated gate bipolar transistors (IGBT’s) in the next
The maximum number of samples that allows this is 19 per control interval.
channel, i.e., 57 in total. • The function which programs D/A converters (DAC’s) on
The use of DMA means that the processor is not burdened the control board (Fig. 4) is called if the user wants some
down with handling relatively high-frequency interrupts from variables to be viewed on the oscilloscope.
the A/D card. It was found that removing these interrupts
resulted in a significant decrease in the execution time of the
control algorithm. V. PC INTERFACE BOARDS

C. Control Interrupt Routine A. A/D Board

The control interrupt routine first runs the startup procedure The 12-b A/D board is a base model of the commercial
and then the maximum torque per ampere control algorithm Data Translation DT2821 series [14]. Samples of analogue
in Fig. 3. The control rate of 2 kHz provides enough time for measurement signals from the system are transferred to a PC
both the DMA interrupt and the control interrupt routines to memory using DMA. The memory segments used for the
be executed before the next control interrupt occurs. The first DMA buffer are programmed into the 8237 DMA controller
priority of the routine is to predict the control to be applied and are initialized prior to DMA/data transfer, since the
to the machine in the next control interval and to output this controller is programmed in autoinitialization mode. This
to the inverter hardware. The control flow required to achieve is important, as it saves code execution time, because the
this objective is as follows. controller has to be initialized only once at the beginning
• A/D conversions and DMA circuitry are triggered by of the program. The detailed description of the steps for
sending out the appropriate bit pattern to the A/D board programming both the board and DMA controller can be found
control register. in [14].
JOVANOVIĆ et al.: SENSORLESS VECTOR CONTROLLER FOR A SYNCHRONOUS RELUCTANCE MOTOR 351

Fig. 4. Control signal generation board.

B. Control Board device on and the bottom off, and the other is vice versa.
The simplified functional block diagram of the control board The presence of two timers allows their gate inputs to be
is presented in Fig. 4. Its major function is to generate the connected to complementary outputs from a toggling flip-flop,
control signals for firing the inverter transistors and closing which provides a low signal into the appropriate timer upon a
and opening contactors. This is essentially achieved using five control interrupt. This design ensures that there is no timing
lots of 8254 interval timers and a single 8255A chip with skew in the generation of the PWM.
three 8-b ports. The coordination circuit for all the actions carried out by
The master timer is clocked from a local 4.9152-MHz quartz the control board is a PAL. It performs the address decoding,
crystal oscillator, which produces a master clock signal. It is accepts and processes the output signals from the timers and
programmed in a square-wave mode with the minimum count ports, and generates the resultant control signals, which are
(2) for the highest possible clock rate (2.4576 MHz) and, then transferred to the inverter hardware. The PWM waveform
hence, the best resolution. signals are also available for viewing on an oscilloscope.
The interrupt timer operates in the same mode and generates Another interface to the inverter and oscilloscopes is a
an interrupt request to the 8259 controller at the 2-kHz control 8255A programmed in the basic I/O mode. The input port bits
rate. Its output signals are also fed to a toggle flip-flop, which are used to monitor the state of the inverter power devices and
controls the gates of the PWM waveform generation timers. hardware trip condition and can be read at any time. Output
The overlap timer is also programmed to operate in a ports bits are dedicated to opening/ closing contactors, drive
continuous square-wave mode. It produces a clock signal enable signals, regeneration transistor control, gate control
which is used by a programmable array logic device (PAL) flip-flop clear signal, and an oscilloscope trigger.
for the generation of the nonoverlapping firing waveforms The bottom part of Fig. 4 is a block diagram of the
for the IGBT’s. The period of this signal (which is user interface system for the encoder and oscilloscopes to the PC
programmable) is the dead time between the top and bottom bus. The encoder subsystem connects the 10-b Gray code
devices of the same inverter leg being turned off/on. This is measurements to the 16-b data bus by means of two octal
to prevent the possibility of shoot through. buffers with common enable gates. The translation of the Gray
The drive signal generation timers are programmed in code input into its binary equivalent is done in software using
software-triggered strobe mode. They are used in pairs to a 1024-word-long lookup table (due to noise problems with a
produce the desired three-phase firing waveforms. Each pair PAL-based translator).
is associated with a single transistor of the inverter (except The DAC subsystem consists of eight 8-b DAC’s that enable
for the regeneration). One timer is used to switch the top monitoring of several software variables on an oscilloscope.
352 IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS, VOL. 34, NO. 2, MARCH/APRIL 1998

Fig. 5. Rotor position estimates and estimation errors at 1200 r/min. Fig. 6. Performance of position estimator and observer at 100 r/min.

VI. EXPERIMENTAL RESULTS

Experimental results have been generated for a 5.8-kW axi-
ally laminated Syncrel (see the Appendix) executing maximum
torque per ampere sensorless control, as shown in Fig. 3. The
machine was running unloaded at various speeds down to zero
with full voltage on the dc link 600 V).
The plots in Fig. 5 present the estimated electrical position
of the rotor obtained directly from (10) the position
after passing through an observer and the absolute
variations of these estimates from 10-b absolute encoder
measurements. The machine speed was 1200 r/min. The raw
estimates, are fairly accurate (average error is 2.4 ),
despite error spikes which have been found to be mainly due
to the noise effects, measurement quantization, and sensitivity
to parameter knowledge inaccuracies. A significant improve-
ment in accuracy is achieved by processing through the
observer. The average error is reduced to approximately 1.5 ,
with the maximum values being about 2.5 or less.
In order to demonstrate the validity and very high accuracy
of the sensorless algorithm at very low speeds, the machine
was driven at 100 r/min. The plots for this case are shown in
Fig. 6. The estimator performance, as predicted, deteriorates
with decreasing the speed. The estimates are much noisier
than at 1200 r/min, the error ripples being occasionally larger
than 50 . The average estimation error is, however, still
reasonably low (7.4 ). The effectiveness of the observer as
a low-pass filter is more than obvious from the same figure.
The accuracy of the filtered noisy used in the control
procedure is substantially better. The peak error is about 3.5
or less, with the average being 1.8 , only marginally worse
than the high-speed case.
Fig. 7 clearly illustrates the good controller performance
Fig. 7. Sensorless controller performance at low speeds.
under both the transient and steady-state conditions of the
machine. The top plot is the Syncrel response to a varying
speed reference between 100 r/min. The speed reversal this case, and for low-speed machine operation in general, the
is affected with very little overshoot. The bottom figure gains of both the speed PI regulator and the observer must be
shows the machine speed while changing the reference values lowered, as instability and divergency of the control algorithm
between 200 r/min and zero. It can be seen that the speed may be experienced due to noisy input estimates. This results
can be effectively and stably controlled, even at standstill. In in low bandwidth control and relatively slow dynamic response
JOVANOVIĆ et al.: SENSORLESS VECTOR CONTROLLER FOR A SYNCHRONOUS RELUCTANCE MOTOR 353

of the machine. It is important to note, however, that both TABLE I

speed characteristics are very smooth and accurately follow SYNCREL PARAMETERS
the desired trajectories. The similar curves can be naturally
obtained for higher speeds when the estimation accuracy is
better.

VII. CONCLUSIONS
The main contribution of this paper is the development of a
new, effective sensorless control algorithm that implements the
maximum torque per ampere strategy for the Syncrel. The high
performance of the sensorless vector controller is confirmed by
experimental results for a 5.8-kW axially laminated machine.
The advantages of the presented sensorless algorithm over the
existing ones discussed in the introduction can be summarized
as follows.
• The algorithm is applicable over the entire speed range
of the machine, including standstill.
• The rotor position is estimated on-line at the control rate,
allowing the controller to effectively replace the encoder
instantaneous measurements, with obvious implications
dc-link voltage. Torque was measured using a 100-N m torque
on high quality control.
transducer.
• The algorithm does not require the injection of any special
The load was a 30-kW dc machine with a through shaft
signals or special inverter switching techniques. Instead, it
enabling the attachment of a 10-b absolute encoder. It was fed
works using the current ripple that is inherent with PWM
from a Ward–Leonard-based dc supply, which allowed simple
voltage control.
regeneration back into the three-phase mains.
• The very high instantaneous accuracy of both the position
The Syncrel maximum torque performance and design pa-
and angular velocity estimates at either speed is achieved
rameters are summarized in Table I. As the rotor is not
using a conventional load model-based observer. The
optimally designed, the machine is capable of developing
electrical position estimation error was shown to be less
only about 5.8 kW at the rated speed and current. One of
than 2 electrical.
the major causes for this relatively modest power production
One limitation is a requirement for a simple startup pro- is the small airgap (0.48 mm as compared to 0.517 mm
cedure to determine rotor initial position before executing in [15]). Furthermore, difficulties in getting 0.5-mm grain-
the control algorithm. This is yet to be investigated, as the oriented steel laminations in Australia have imposed the use
authors believe that it can be eliminated by relying on the of standard transformer laminations, which has resulted in less
position estimation technique itself. A further limitation is steel being present in the rotor. Consequently, it has lower iron
the parameter dependence inherent with this technique. On- losses, but saturates easier, compromising a saliency ratio and
line parameter estimation techniques are being investigated. performance of the Syncrel prototype.
The fulfillment of these two objectives would further improve The total inertia constant was determined by conducting
viability of the control algorithm. a simple step-torque test, the speed response of the unloaded
Syncrel being monitored on a digital oscilloscope. The lin-
APPENDIX earized load model equation in (1) was then used to predict
EXPERIMENTAL MACHINE from a knowledge of the rate of change of angular velocity
(neglecting friction).
The test machine is an inverter-fed Syncrel having a com- The inductances were measured by running an instanta-
mercial DF132M frame size, 7.5-kW three-phase Y-connected neous flux-linkage locked rotor test [16]. The results obtained
induction machine stator, and an axially laminated rotor. The can be accurately represented by the following sixth-order
IGBT-based inverter and the rotor were both designed and polynomial:
built at the workshop of the Department of Electrical and
Computer Engineering, University of Newcastle. The rotor
was constructed based on a design from the University of
Glasgow [15].
where A. For A, is naturally unsaturated.
The 10-kW inverter uses a Mitsubishi CM50TF-24E six-
transistor power module and a CM50E3Y-24E device from
the same manufacturer for regeneration. Both component sets ACKNOWLEDGMENT
are rated at 1200 V and 50 A, with the switching frequency The authors would like to acknowledge T. Wylie, who
up to 20 kHz. Hall-effect transducers and precise potential constructed much of the experimental system hardware, and
dividers were used to measure machine currents and inverter P. McLauchlan and R. Hicks, who built the Syncrel rotor.
354 IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS, VOL. 34, NO. 2, MARCH/APRIL 1998

REFERENCES Milutin G. Jovanović received the Dipl.Eng. and

M.E.E. degrees from the University of Belgrade,
[1] M. Schroedl and P. Weinmeier, “Sensorless control of reluctance ma- Belgrade, Yugoslavia, in 1987 and 1991, respec-
chines at arbitrary operating conditions including standstill,” IEEE tively, and the Ph.D. degree from the University of
Trans. Power Electron., vol. 9, pp. 225–231, Mar. 1994. Newcastle, Newcastle, Australia, in 1997.
[2] R. Lagerquist, I. Boldea, and T. J. E. Miller, “Sensorless control of the He is currently a Lecturer in the School of Electri-
synchronous reluctance motor,” IEEE Trans. Ind. Applicat., vol. 30, pp. cal Engineering, Electronics and Physics, Liverpool
673–682, May/June 1994. John Moores University, Liverpool, U.K. His major
[3] M. S. Arefeen, M. Ehsani, and T. A. Lipo, “An analysis of the accuracy interests lie in the areas of electrical machines and
of indirect shaft sensor for synchronous reluctance motor,” IEEE Trans. drives, industrial electronics, and power systems.
Ind. Applicat., vol. 30, pp. 1202–1209, Sept./Oct. 1994. Dr. Jovanović is a member of the Industrial
[4] M. Jovanovic, R. E. Betz, and D. Platt, “Position and speed estimation Drives Committee of the IEEE Industry Applications Society.
of sensorless synchronous reluctance motor,” in Proc. IEEE PEDS’95
Conf., Singapore, Feb. 1995, pp. 844–849.
[5] Y. Q. Xiang and S. A. Nasar, “Estimation of rotor position and speed
of a synchronous reluctance motor for servodrives,” Proc. Inst. Elect.
Eng., vol. 142, no. 3, pp. 201–205, May 1995.
[6] T. Matsuo and T. A. Lipo, “Rotor position detection scheme for Robert E. Betz (M’92) received the B.E., M.E.,
synchronous reluctance motor based on current measurements,” IEEE and Ph.D. degrees from the University of Newcas-
Trans. Ind. Applicat., vol. 31, pp. 860–868, July/Aug. 1995. tle, Newcastle, Australia in 1979, 1982, and 1984,
[7] Z. I. Boldea and S. A. Nasar, “Torque vector control (tvc) of axially- respectively.
laminated anisotropic (ala) rotor reluctance synchronous motors,” Elect. He is currently a Senior Lecturer in the De-
Mach. Power Syst., vol. 19, pp. 381–398, 1991. partment of Electrical and Computer Engineering,
[8] P. L. Jansen and R. D. Lorenz, “Transducerless position and velocity University of Newcastle. His major interests are
estimation in induction and salient ac machines,” IEEE Trans. Ind. electrical machine drives, real-time operating sys-
Applicat., vol. 31, pp. 240–247, Mar./Apr. 1995. tems, and industrial electronics.
[9] H. W. VanDerBroeck, H. C. Skudelny, and G. V. Stanke, “Analysis and Dr. Betz is a member of the Industrial Drives
realization of a pulsewidth modulator based on voltage space vectors,” Committee of the IEEE Industry Applications Soci-
IEEE Trans. Ind. Applicat., vol. 24, pp. 142–150, Jan./Feb. 1988. ety.
[10] R. D. Lorenz and K. W. VanPatten, “High-resolution velocity estimation
for all-digital, ac servo drives,” IEEE Trans. Ind. Applicat., vol. 27, pp.
701–705, July/Aug. 1991.
[11] K. J. Astrom and B. Wittenmark, Computer-Controlled Systems. En-
glewood Cliffs, NJ: Prentice-Hall, 1990.
[12] R. D. Lorenz and D. B. Lawson, “Performance of feedforward current Don Platt (S’86–M’87) received the B.Sc. and B.E.
regulators for field-oriented induction machine controllers,” IEEE Trans. degrees in electrical engineering from the University
Ind. Applicat., vol. 23, pp. 597–602, July/Aug. 1987. of New South Wales, Sydney, Australia, in 1970
[13] R. B. Sepe and J. H. Lang, “Inverter nonlinearities and discrete-time and the Ph.D. degree from the University of Wol-
vector current control,” IEEE Trans. Ind. Applicat., vol. 30, pp. 62–70, longong, Wollongong, Australia, in 1988.
Jan./Feb. 1994. He has worked in several industries, including
[14] User Manual for DT2821 Series, Data Translation, Marlboro, MA, 1987. both light and heavy processing industries. He is
[15] W. L. Soong, D. A. Staton, and T. J. E. Miller, “Design of a new presently a Senior Lecturer in the Department of
axially-laminated interior permanent magnet motor,” IEEE Trans. Ind. Electrical and Computer Engineering, University of
Applicat., vol. 31, pp. 358–367, Mar./Apr. 1995. Wollongong, Wollongong, Australia. His areas of
[16] C. Cossar and T. J. E. Miller, “Electromagnetic testing of switched interest include magnetic circuits, electric machines
reluctance motors,” in Proc. ICEM, 1992, pp. 470–474. and drives, and power electronics.