of parasitic anti-resonance (which comes from the length of the

inserted third metal), we cut the RF path and insert a chip resistor
R S with a high value. We used a dielectric substrate with a relative
dielectric constant ␧ r of 3.05 and the height of 1.524 mm in order
to implement a CPW line with high Z C . The varactor, which has
capacitances of 0.5 to 9.5 pF with respect to the bias voltage of 0
to 30 V, is an Infenion BB833, denoted as C var in Figure 4. The
fabricated impedance transformer in Figure 5 has an overall di-
mension of 2.9 ⫻ 1.0 cm2.
The simulated result, shown in Figure 6, indicates that the
impedance transformer can be electronically varied from 19⍀ to
65⍀ in a 50⍀ environment at the center frequency of 900 MHz,
and it has a bandwidth of 200 MHz when the bias supply varies
from 0 to 16 V. The input impedance of an impedance transformer
has a parasitic-reactance element due to the lumped elements, such
as varactors and chip capacitors, while the control voltage varies.
This is based on the difference of the propagation constants for the
different bias conditions. However, its level is quite low within a
specific frequency band. The scattering-parameter measurements
were performed using an Agilent 8753D network analyzer over the Figure 6 Simulated and measured results with the varying bias voltages
frequency range from 0.8 to 1 GHz. Figure 6 also shows the of 0 to 16 V of the tunable impedance transformer (for 0, 2, 4, 6, 8, 10, and
16 V)
measured responses of the impedance transformer in which the
tuning range was found to be 10⍀ to 69.5⍀. This agrees well with
the simulated result, except for the minor discrepancy due to the REFERENCES
parasitic of varactors. 1. W. Bischof, Variable impedance tuner for MMICs, IEEE Microwave
Guided Wave Lett 4 (1994), 172–174.
4. CONCLUSION 2. L.-Y.V. Chen, R. Forse, D. Chase, and R.A. York, Analog tunable
matching network using integrated thin-film BST capacitors, IEEE Int
This paper has proposed a new type of variable Z C transmission Microwave Symp Dig 1 (2004), 261–264.
line and its application to a tunable impedance transformer in order 3. J. de Mingo, A. Valdovinos, A. Crespo, D. Navarro, and P. Garcia, An
to enhance the efficiency of an RF front-end. It is promising for RF electronically controlled impedance tuning network design and its
high-performance transceiver and reconfigurable antenna systems. application to an antenna input impedance automatic matching system,
The measurement of the experimental demonstrator has shown that IEEE Trans Microwave Theory Tech 52 (2004), 489 – 497.
the proposed impedance transformer has a broad tuning range with 4. H.-T. Kim, S. Jung, K. Kang, J.-H. Park, Y.-K. Kim, and Y. Kwon,
single bias voltage. This tunable impedance transformer can be Low-loss analog and digital micromachined impedance tuners at the
easily constructed by applying conventional MMIC techniques. It Ka-band, IEEE Trans Microwave Theory Tech 49 (2001), 2394 –
is especially promising for RF MEMS applications when switches 2400.
5. J. Papapolymerou, K.L. Lange, C.L. Goldsmith, A. Malczewski, and J.
are used for the digitally controlled impedance transformer.
Kleber, Reconfigurable double-stub tuners using MEMs switches for
intelligent RF frontends, IEEE Trans Microwave Theory Tech 51
ACKNOWLEDGMENT (2003), 271–278.
6. T. Vaha-Heikkila, J. Varis, J. Tuovinen, and G.M. Rebeiz, A recon-
Dr. Y.-H. Chun was supported by the Post-Doctoral Fellowship
figurable 6 –20-GHz RF MEMS impedance tuner, IEEE Int Microwave
Program of the Korea Research Foundation (KRF). Symp Dig 2 (2004), 729 –732.

© 2005 Wiley Periodicals, Inc.

DESIGN PARAMETERS OF AN
OMNIDIRECTIONAL PLANAR
MICROSTRIP ANTENNA
Randy Bancroft
Centurion Wireless Technologies
6252 West 91st Avenue
Westminster, Colorado 80031

Received 23 May 2005

ABSTRACT: The design parameters of an omnidirectional planar mi-

crostrip antenna are examined. The impedance bandwidth and radiation
efficiency increase as the element width is increased. Increasing the ele-
ment width can cause a lower-order mode to increase in frequency and
degrade the omnidirectional pattern and sidelobes, and thus creates a
beam scan from broadside. This beam scan and pattern degradation can
Figure 5 Fabricated circuit board of a tunable impedance transformer be eliminated with narrower elements or by moving the position of the

414 MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 47, No. 5, December 5 2005
 TABLE 1 Omnidirectional RN vs. N (We ⴝ 10 mm, H ⴝ 0.762
mm, ␧r ⴝ 2.6, Wm ⴝ 2 mm, Ls ⴝ Le/2, a ⴝ 0.5 mm)

N R N (HFSS) [⍀] R N (1) [⍀]

2 389.2 389.2
3 289.6 259.5
4 219.9 194.6
5 152.2 155.7
6 133.2 129.7
7 115.2 111.2

2. OMA DESCRIPTION
The geometry of a five-section (N ⫽ 5) omnidirectional microstrip
antenna (OMA) is presented in Figure 1. The bottom conductor has
three wide sections and two narrow sections. The top conductor
has three narrow sections and two wide ones. The length of the
connecting microstrip line is designated L m and the length of each
element is L e . In our examples, L e ⫽ L m . The width of the
microstrip line is W m and each element is W e . Two shorts on either
end of the antenna are at a distance L s from the edge of the upper
conductor.
The sections are arranged to have a length of approximately ␭/2
(⬇L e ) in the dielectric. If the antenna is fed with L d ⬇ 0, we note
that a wave traveling downward from the driving point of section
one toward the short circuit will have a 90° spatial phase shift, then
Figure 1 Geometry of the five-section (N ⫽ 5) omnidirectional micro-
a 180° phase shift occurs when the wave is reflected at the short
strip antenna
circuit and a 90° spatial phase shift occurs as the wave arrives back
at its origin for a 360° total phase shift, which brings the reflected
wave into phase with the upward travelling wave propagating from
shorting pins at the expense of impedance bandwidth. © 2005 Wiley the source. The symmetry of the antenna enforces this condition at
Periodicals, Inc. Microwave Opt Technol Lett 47: 414 – 418, 2005; the opposite end, as illustrated in Figure 2.
Published online in Wiley InterScience (www.interscience.wiley.com). The upper illustration in Figure 2 shows a side view of the
DOI 10.1002/mop.21187 electric field and current on a typical microstrip transmission line.
The omnidirectional microstrip antenna takes half-wavelength sec-
Key words: microstrip antenna; omnidirectional; printed antenna; de- tions of microstrip transmission line and flips them. Radiation
sign parameters occurs from the current on each edge of the wide sections. By
inverting the sections, the radiating currents are all brought into
1. INTRODUCTION phase, as required for an omnidirectional antenna.
An omnidirectional microstrip antenna, recently introduced by The driving-point location (L d ) shown in Figure 1 allows a
Bancroft and Bateman, is useful for 802.11b (2.40 –2.50 GHz) or designer to adjust the driving-point impedance from very low
802.11a (5.15–5.35 GHz) wireless applications [1, 2]. A version of values near the short to a maximum value at the junction.
the antenna with superior sidelobe performance has also been The bottom conductor of the first section has a groundplane
described [3], but no detailed discussion of the design parameters relief through which a coaxial-cable center conductor is passed and
of the basic antenna has been undertaken. This paper discusses the connected to the upper microstrip conductor. The two conductors
changes in driving-point impedance, radiation pattern, and imped- are on either side of a dielectric material of width W, length L, and
ance bandwidth for this omnidirectional microstrip antenna design. thickness H (Fig. 1).

3. DRIVING-POINT IMPEDANCE
The OMA is generally driven using a coaxial cable which has its
outer conductor soldered to the groundplane side of the lower most

TABLE 2 Omnidirectional Ld vs. NRd ⴝ 50⍀ (We ⴝ 10 mm,

H ⴝ 0.762 mm, ␧r ⴝ 2.6, Wm ⴝ 2 mm, Ls ⴝ Le/2, a ⴝ 0.5 mm)

N L d (HFSS) [mm] L d (2) [mm]

2 14.88 16.00
3 13.50 15.18
4 12.00 14.25
Figure 2 Currents on (a) microstrip transmission line and (b) flipped 5 10.50 12.70
half-wavelength sections of microstrip transmission line (N ⫽ 7). The 6 9.50 12.11
currents on the groundplane sections are in phase 7 9.00 11.31

MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 47, No. 5, December 5 2005 415
 TABLE 3 Omnidirectional Antenna Bandwidth vs. N (We ⴝ 20
mm, H ⴝ 0.762 mm, ␧r ⴝ 2.6, tan ␦ ⴝ 0.0009, Wm ⴝ 2 mm, Ls
ⴝ Le/2, a ⴝ 0.5 mm)

N Bandwidth %
2 280 MHz 11.42%
3 240 MHz 9.79%
4 190 MHz 7.75%
5 260 MHz 10.61%
6 — —

When the substrate thickness H is increased to 1.524 mm (0.060

in.) R 2 increases to 1068⍀.
In many cases, one may narrow the element width to produce
a 50⍀ driving-point location between the groundplane edge and
Figure 3 VSWR of OMA vs. N computed with HFSS (H ⫽ 0.762 mm,
the short. In situations where one is unable to sufficiently narrow
␧ r ⫽ 2.6, W e ⫽ 20 mm, W m ⫽ 2 mm)
the antenna to produce a desired driving-point impedance, the use
of a microstrip impedance transformer on the bottom section often
allows one to match the OMA.
element and whose center conductor passes through a small aper- The driving-point resonant resistance varies approximately as
ture in the copper to feed the microstrip transmission line above. the square of the cosine, which allows one to estimate the location
The driving-point impedance varies from zero at the short to a of a desired driving-point resistance using Eq. (2). R N is the
maximum at the edge of the bottom element. In many cases, this driving-point resistance with L d ⫽ 0 for an OMA with N sections.
allows a designer to determine a 50⍀ driving-point location L d . R d is the desired driving point resistance which is generally 50⍀.
An OMA design with two sections (N ⫽ 2) has the largest ␭ is the wavelength in the microstrip transmission line:
driving-point impedance at the edge of the first section (that is,

冑
L d ⫽ 0). When the antenna is at resonance we will designate this
␭ Rd
maximum edge resistance for the two section OMA as R 2 . The Ld ⫽ cos⫺1 . (2)
driving-point impedance decreases as in Eq. (1) with N ⫽ 2, 3, 2␲ RN
4, . . . , where N is the total number of sections of the OMA. This
relationship holds true for W e values which do not produce de- The change in driving-point impedance location with respect to
generate modes (as illustrated in Fig. 4): the number of elements N is illustrated using 0.762-mm (0.030-in.)
thick (H) Taconic TLX-7 substrate (␧ r ⫽ 2.6, tan ␦ ⫽ 0.0019) in
2 Table 2. Given the value of R N , Eq. (2) is useful for estimating an
RN ⫽ 䡠R . (1) approximate 50⍀ location.
N 2
4. IMPEDANCE BANDWIDTH
As N increases, the 50⍀ driving-point location moves in a
direction away from the short and in some cases a 50⍀ location The impedance bandwidth of the OMA is affected by the width of
will no longer exist between the short and the groundplane edge. the elements. As the width of the elements are increased, the
Narrowing the element width W e produces a larger element im- impedance bandwidth of the antenna also increases. Unfortunately,
pedance. Thickening the antenna substrate also produces a larger as the number of elements is increased the next lower-order mode
element impedance. The value of R 2 in Table 1 is 389.2⍀ for H of can rise in frequency and merge with the desired resonance. The
0.762 mm (0.030 in.) with a design frequency ( f 0 ) of 2.45 GHz. lower-order resonant mode has a butterfly-shaped pattern and
causes increased sidelobes and main bean squint. Figure 3 shows
the VSWR of an OMA with W e ⫽ 20 mm versus N. We can see
the lower mode increasing in frequency until, when N ⫽ 7, the
two have merged. This pattern degradation can be rectified by
narrowing the width of the elements. In Figure 4, we have the
VSWR of the same design as in Figure 3, except that W e is now
10 mm. The lower mode moves upward, but begins at a much

TABLE 4 Omnidirectional Antenna Bandwidth vs. N (We ⴝ

10 mm, H ⴝ 0.762 mm, ␧r ⴝ 2.6, Wm ⴝ 2 mm, Ls ⴝ Le/2,
a ⴝ 0.5 mm)

N Bandwidth %
2 125 MHz 5.10%
3 90 MHz 3.67%
4 85 MHz 3.47%
5 90 MHz 3.67%
6 85 MHz 3.47%
Figure 4 VSWR of OMA vs. N computed with HFSS (H ⫽ 0.762 mm, 7 80 MHz 3.26%
␧ r ⫽ 2.6, W e ⫽ 10 mm, W m ⫽ 2 mm)

416 MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 47, No. 5, December 5 2005
TABLE 5 Omnidirectional Antenna Bandwidth vs. N (We ⴝ
10 mm, H ⴝ 1.524 mm, ␧r ⴝ 2.6, Wm ⴝ 2 mm, Ls ⴝ Le/2,
a ⴝ 0.5 mm)

N Bandwidth %
2 75 MHz 3.06%
3 58 MHz 2.37%
4 63 MHz 2.57%
5 50 MHz 2.00%
6 45 MHz 1.84%
7 45 MHz 1.84%

lower frequency and by N ⫽ 7 is still isolated from the desired

resonant mode.
The bandwidth of an OMA versus the number of sections N
with wide elements (W e ⫽ 20 mm) is presented in Table 3. The
impedance bandwidth of the antenna is very large and begins to
decrease, but increases again as the lower mode merges and Figure 6 Radiation patterns of OMA vs. N computed using HFSS
extends the impedance bandwidth at the expense of pattern band- (W e ⫽ 10 mm, H ⫽ 0.762 mm, ␧ r ⫽ 2.6, f 0 ⫽ 2.45 GHz)
width. When the element widths are narrowed to 10 mm, as shown
in Table 4, we note the bandwidth for this design is approximately
3.5% for N ⬎ 2, but largest for N ⫽ 2. When the thickness of the When the thickness H of the antenna is increased to 1.524 mm
antenna is increased from 0.762 to 1.524 mm, the impedance with W e ⫽ 10 mm and other parameters, as detailed in Table 6, the
bandwidth again decreases as N increases. The bandwidth de- predicted efficiency varies from 90.7% (N ⫽ 2) to 86.0% (N ⫽ 7)
creases as the substrate thickness increases. This is detailed in in this case.
Table 5. If the thickness of the dielectric substrate is decreased to H ⫽
0.127 mm by N ⫽ 4 (0.005-in.-thick Taconic TLY-5 with ␧ r ⫽
5. GAIN AND EFFICIENCY
2.2 and tan ␦ ⫽ 0.0009), the efficiency decreases to 72.2% (⫺1.14
The computed radiation patterns for the antenna parameters of dB). This decrease in efficiency is due to increased copper losses
Table 3 (W e ⫽ 20 mm, f 0 ⫽ 2.45 GHz) for N ⫽ 2 to 7 are and has been confirmed experimentally.
presented in Figure 5. For N ⫽ 2 to 4 this antenna design has an HFSS analysis using a low-dielectric-constant commercially
omnidirectional pattern with no sidelobes. available substrate material with ␧ r ⫽ 1.35, tan ␦ ⫽ 0.005, and
The nonsymmetric sidelobes seen in Figure 5 for N ⫽ 5 to 7 H ⫽ 1.016 mm (0.040-in.-thick Arlon FoamClad 100) shows that
are caused by the merging of a lower mode, which produces a with W m ⫽ 2 and W e ⫽ 10 mm, at N ⫽ 4, the radiation efficiency
butterfly type of pattern with the desired omnidirectional mode. has dropped to 71.1% (⫺1.48 dB). Increasing W e and decreasing
Figure 6 shows that the patterns become symmetric when the W m increases the radiation efficiency, but produces a large reactive
antenna element widths are reduced to 10 mm. part in the driving-point impedance, and decreases the real part to
Table 6 reveals that the radiation efficiency of the design around 10⍀ to 15⍀. A matching network is then required for an
increases as W e increases, but this increase in radiation efficiency antenna designed with this low-dielectric-constant material. This
is at the expense of pattern symmetry. mismatch has also been observed in designs with tapered element
widths used to produce low side-lobe levels.

6. ANTENNA PHASE CORRECTION

As noted previously, when W e becomes large, a lower-frequency
mode begins to steer the main beam of the antenna toward the
direction of the driving point. As the number of sections is in-
creased, the amount of beam scanning also increases. When N ⫽
7 with the design values detailed in Table 3, the beam is scanned
10° off-axis, as shown in Figure 7.

TABLE 6 Omnidirectional Antenna Efficiency vs. N (H ⴝ

0.762 mm, ␧r ⴝ 2.6, Wm ⴝ 2 mm, Ls ⴝ Le/2, a ⴝ 0.5 mm)

N We We
20 mm 10 mm
2 96.45% 91.34%
3 95.29% 89.31%
4 94.73% 88.60%
5 94.50% 89.78%
6 94.72% 88.34%
Figure 5 Radiation patterns of OMA vs. N computed with HFSS (W e ⫽ 7 94.68% 87.61%
20 mm, H ⫽ 0.762 mm, ␧ r ⫽ 2.6, f 0 ⫽ 2.45 GHz)

MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 47, No. 5, December 5 2005 417
 DESIGN OF BROAD QUAD-BAND
PLANAR INVERTED-F ANTENNA FOR
CELLULAR/PCS/UMTS/DMB
APPLICATIONS
Hoon Park and Jaehoon Choi
Division of Electrical and Computer Engineering
Hanyang University
17 Haengdang-dong, Seongdong-gu
Seoul, 133-791, Korea

Received 30 May 2005

ABSTRACT: A novel and broad quad-band planar inverted-F antenna

(PIFA) for cellular/PCS/UMTS/DMB applications is proposed. The pro-
posed antenna consists of the main patch with a slant slit and L-shaped
patch, occupying a total volume of 15 ⫻ 44 ⫻ 8 mm3. The triple reso-
nant frequencies are obtained by changing both the length and width of
a slant slit on the main patch and a position of feed line. A broadband
characteristic is achieved by optimizing not only the length and height
of the L-shaped patch, but also the length and width of the slant slit on
the main patch, which is excited by the modified CPW-fed antenna.
Good broadside radiation patterns with reasonable gains are achieved
for all four-frequency bands of interest. © 2005 Wiley Periodicals, Inc.
Figure 7 Radiation patterns of N ⫽ 7 OMA with and without phase Microwave Opt Technol Lett 47: 418 – 421, 2005; Published online in
correction using HFSS Wiley InterScience (www.interscience.wiley.com). DOI 10.1002/mop.
21188

Key words: internal antenna; multiband antenna; planar inverted-F

The distance from the short to the edge of the first element L s antenna (PIFA); wideband antenna
and the value of L m ⫽ L e may be adjusted so as to eliminate the
undesired mode. 1. INTRODUCTION
The pattern results for the N ⫽ 7 case with f 0 ⫽ 2.45 GHz are Due to the rapid development and widespread usage of various
shown in Figure 7. The length to the shorts L s was decreased by 12 communication systems, multiband internal antennas are necessary
mm. The length of the elements L e ⫽ L m went from 34.32 to 40.66 for recent mobile handsets [1–3]. To meet the antenna requirement
mm. The peak gain increased from 6.35 to 6.78 dBi with a slight for handheld terminals, size reduction is one of the key require-
predicted efficiency change from 94.6% to 91.8%. The lower- ments, along with maintaining good multiband and wideband
frequency mode, which almost became degenerate with the desired performances. Furthermore, a small internal antenna that can be
omnidirectional mode, moved downward in the corrected design. integrated into the handset offers many advantages over conven-
tional external-monopole or helical antennas. A potential candidate
for such an antenna is the planar inverted-F antenna (PIFA) [4 –7].
7. CONCLUSION However, one of the principal disadvantages of a basic PIFA
The design parameters of an omnidirectional microstrip antenna element is its narrow bandwidth of about 4% to 12% for return loss
have been investigated. The driving-point impedance increases as less than ⫺10 dB.
the width of the elements W e is decreased at the expense of In this paper, a novel PIFA fed by a coplanar waveguide (CPW)
impedance bandwidth. As the number of sections N increase for with broadband impedance characteristic is designed to operate at
large W e , an undesired lower mode increases its frequency and the centre frequencies of 870, 2000, and 2610 MHz. CPW-fed
becomes degenerate with the desired mode. The lower mode has a antennas have the advantage of being compatible with other mi-
butterfly-shaped pattern and degrades the desired omnidirectional crowave integrated circuits. The introduction of a slant slit into the
pattern and produces a main-beam tilt and increased sidelobes. main patch generates three separate resonant modes, while the use
This beam tilt can be eliminated by decreasing the element width of a L-shaped patch provides wide bandwidth for the desired
W e or by moving the location of the shorts L s . Both methods quad-band operation and improves the matching performance at
produce a broadside beam at the expense of impedance bandwidth. the upper frequency. This antenna satisfies the 10-dB return loss
requirement to cover the cellular (824 – 894 MHz), PCS (personal
communication service, 1750 –1870 MHz), UMTS (universal mo-
bile telecommunications system, 1920 –2170 MHz), and DMB
REFERENCES (digital multimedia broadcasting, 2605–2655 MHz) services at the
1. R. Bancroft and B. Bateman, An omnidirectional planar microstrip same time.
antenna, IEEE Trans Antennas Propagat 52 (2004), 3151–3153.
2. U.S. Patent Pending, U.S. Patent Application Serial No. 60/461,689.
2. ANTENNA DESIGN
3. R. Bancroft and B. Bateman, An omnidirectional planar microstrip
antenna with low sidelobes, Microwave Opt Technol Lett 42 (2004), Figure 1 shows the proposed antenna mounted on a ground plane
68 – 69. with dimensions of 62 ⫻ 44 mm2. The antenna consists of a main
patch with a slant slit at the top layer, a ground plane at the bottom,
© 2005 Wiley Periodicals, Inc. and a L-shaped patch in between, as well as a CPW-fed structure

418 MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 47, No. 5, December 5 2005