Microstrip Antennas PDF
Microstrip Antennas PDF
Microstrip Antennas PDF
Microstrip Antennas
Microstrip Antennas
International Journal of Antennas and Propagation
Microstrip Antennas
Copyright © 2012 Hindawi Publishing Corporation. All rights reserved.
This is a focus issue published in “International Journal of Antennas and Propagation.” All articles are open access articles distributed
under the Creative Commons Attribution License, which permits unrestricted use, distribution, and reproduction in any medium, pro-
vided the original work is properly cited.
Editorial Board
M. Ali, USA Se-Yun Kim, Republic of Korea Sadasiva M. Rao, USA
Charles Bunting, USA Ahmed A. Kishk, Canada Sembiam R. Rengarajan, USA
Felipe Cátedra, Spain Tribikram Kundu, USA Ahmad Safaai-Jazi, USA
Dau-Chyrh Chang, Taiwan Byungje Lee, Republic of Korea Safieddin Safavi-Naeini, Canada
Deb Chatterjee, USA Ju-Hong Lee, Taiwan Magdalena Salazar-Palma, Spain
Z. N. Chen, Singapore L. Li, Singapore Stefano Selleri, Italy
Michael Yan Wah Chia, Singapore Yilong Lu, Singapore Krishnasamy T. Selvan, India
Christos Christodoulou, USA Atsushi Mase, Japan Zhongxiang Q. Shen, Singapore
Shyh-Jong Chung, Taiwan Andrea Massa, Italy John J. Shynk, USA
Lorenzo Crocco, Italy Giuseppe Mazzarella, Italy M. Singh Jit Singh, Malaysia
Tayeb A. Denidni, Canada Derek McNamara, Canada Seong-Youp Suh, USA
Antonije R. Djordjevic, Serbia C. F. Mecklenbräuker, Austria Parveen Wahid, USA
Karu P. Esselle, Australia Michele Midrio, Italy Yuanxun Ethan Wang, USA
Francisco Falcone, Spain Mark Mirotznik, USA Daniel S. Weile, USA
Miguel Ferrando, Spain Ananda S. Mohan, Australia Quan Xue, Hong Kong
Vincenzo Galdi, Italy P. Mohanan, India Tat Soon Yeo, Singapore
Wei Hong, China Pavel Nikitin, USA Young Joong Yoon, Korea
Hon Tat Hui, Singapore A. D. Panagopoulos, Greece Wenhua Yu, USA
Tamer S. Ibrahim, USA Matteo Pastorino, Italy Jong Won Yu, Republic of Korea
Nemai Karmakar, Australia Massimiliano Pieraccini, Italy Anping Zhao, China
Contents
Modal Resonant Frequencies and Radiation Quality Factors of Microstrip Antennas, Jan Eichler,
Pavel Hazdra, Miloslav Capek, and Milos Mazanek
Volume 2012, Article ID 490327, 9 pages
Tunable Compact UHF RFID Metal Tag Based on CPW Open Stub Feed PIFA Antenna, Lingfei Mo and
Chunfang Qin
Volume 2012, Article ID 167658, 8 pages
Some Recent Developments of Microstrip Antenna, Yong Liu, Li-Ming Si, Meng Wei, Pixian Yan,
Pengfei Yang, Hongda Lu, Chao Zheng, Yong Yuan, Jinchao Mou, Xin Lv, and Housjun Sun
Volume 2012, Article ID 428284, 10 pages
New Configurations of Low-Cost Dual-Polarized Printed Antennas for UWB Arrays, Guido Valerio,
Simona Mazzocchi, Alessandro Galli, Matteo Ciattaglia, and Marco Zucca
Volume 2012, Article ID 786791, 10 pages
Design and Analysis of Wideband Nonuniform Branch Line Coupler and Its Application in a Wideband
Butler Matrix, Yuli K. Ningsih, M. Asvial, and E. T. Rahardjo
Volume 2012, Article ID 853651, 7 pages
Isolation Improvement of a Microstrip Patch Array Antenna for WCDMA Indoor Repeater Applications,
Hongmin Lee and Jinwon Park
Volume 2012, Article ID 264618, 8 pages
Vertical Meandering Approach for Antenna Size Reduction, Li Deng, Shu-Fang Li, Ka-Leung Lau,
and Quan Xue
Volume 2012, Article ID 980252, 5 pages
A Wideband High-Gain Dual-Polarized Slot Array Patch Antenna for WiMAX Applications in 5.8 GHz,
Amir Reza Dastkhosh and Hamid Reza Dalili Oskouei
Volume 2012, Article ID 595290, 6 pages
Hindawi Publishing Corporation
International Journal of Antennas and Propagation
Volume 2012, Article ID 490327, 9 pages
doi:10.1155/2012/490327
Research Article
Modal Resonant Frequencies and Radiation Quality Factors of
Microstrip Antennas
Copyright © 2012 Jan Eichler et al. This is an open access article distributed under the Creative Commons Attribution License,
which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited.
The chosen rectangular and fractal microstrip patch antennas above an infinite ground plane are analyzed by the theory of
characteristic modes. The resonant frequencies and radiation Q are evaluated. A novel method by Vandenbosch for rigorous
evaluation of the radiation Q is employed for modal currents on a Rao-Wilton-Glisson (RWG) mesh. It is found that the
resonant frequency of a rectangular patch antenna with a dominant mode presents quite complicated behaviour including having
a minimum at a specific height. Similarly, as predicted from the simple wire model, the radiation Q exhibits a minimum too. It is
observed that the presence of out-of-phase currents flowing along the patch antenna leads to a significant increase of the Q factor.
3. The Radiation Q Factor Figure 1: Distance between nonoverlapping current elements [23].
P3 (x3 , y3 ) z
β
h13 (0,1)
A T1 (x,y,z) T1 (α ,β,γ) 0.01
P1 (x1 , y1 ) P0
P0
h23 α
0.005
r2
y h12 (0,0) (1,0) 0
P2 (x2 , y2 ) 2H
r1
(0,0) x
−0.005
y
(a) (b)
Figure 2: Self-term evaluation. (a) Original problem, (b) simplex −0.01 x
coordinates transformation [23].
0.015
0.01
f r (TCM)
4. Applications: Rectangular Patch Antenna f r (analytic equation)
Let us first concentrate on a rectangular patch antenna of Figure 4: R50 × 30 resonant frequency of the dominant TM01
dimensions L = 50 mm and W = 30 mm (further noted as mode. The dashed red curve is a quasianalytical equation from [1].
R50 × 30) placed in air at a height H above an infinite ground
plane. Only the dominant TM01 mode will be studied. The
reason for choosing a patch with L/W = / 1 is that we do not H/λres < 0.08), the resonant frequency decreases “regularly,”
have to deal with degenerated modes. and quasianalytical formulas (see, e.g., [1, 3]) based on
Using the image theory, the radiator in the XY plane the fringing field concept are valid below this range. For
at height z = H above an infinite electric ground plane is H ∼= 25 mm (H/λres ∼ = 0.188) there is absolute minimum
modelled as two patches separated by 2H. The total number of the TM01 resonant frequency. Further on, the resonant
of triangular elements is 676. In the TCM analyser, a proper frequency rises to reach its maximum for H ∼ = 40 mm
out-of-phase mode is selected (Figure 3). (H/λres ∼
= 0.51). Around this specific height the patch also
The resonant frequency of the dominant mode is shown shows the minimum of the radiation Q. The above described
as a function of height H, see Figure 4. It has been evaluated process repeats periodically. It is yet unclear to the authors
from a modal resonant condition for eigenvalues λ = as what is the physical background to the resonant frequency
2ω(W m − W e ) = 0 employing an adaptive frequency discontinuity around H/λres ∼= 0.5.
sweep for each height. The behaviour is quite peculiar, The terms 2ωW m , 2ωWe , and 2ω(W m − W e ) obtained
especially for greater heights. For low heights (H < 10 mm or from (5)–(9) and eigenvalues λ are plotted at Figure 5 for
4 International Journal of Antennas and Propagation
10
f res After deriving (18), the condition is worked-out
5
tan(k2H) = k2H, (19)
0
and the first nontrivial root of (19) could be approximated
−5
as [25]
H ∼ 3 − 1 = 0.358.
−10
= 2
(20)
1.5 1.6 1.7 1.8 1.9 2 2.1 2.2 2.3 2.4 2.5 2.6 2.7 2.8 2.9 3 λ min 8 6π
f (GHz) For sinusoidal currents on dipoles the minimum (evaluated
numerically) occurs for H = 0.36λ.
λ 2ω (We)
The minimum of the patch under study is obtained at
2ω (Wm) 2ω (Wm-We)
H ∼= 0.4λ, a value that is remarkably close to the simple
Figure 5: Reactive energies and their differences for an R50 × 30 dipole model.
patch at height of 25 mm.
4.1. Algorithm Convergence. Since no other methods for
calculating modal Q are available, Qeig is taken as a reference,
50
and the relative error percentage is defined as:
45
40 QJ − Qeig
relative error = · 100, (21)
35 Qeig
30
Q (–)
10
9 J1 J2
8
Relative error (%)
7
6
5
4
3
2
1
0 σ1 σ2
0 200 400 600 800 1000 1200 1400
Number of triangle elements
H = 1 mm (0.01 λ) H = 10 mm (0.0803 λ)
H = 2 mm (0.0185 λ) H = 20 mm (0.151 λ)
5
4.5
4 W
3.5
J1 J2
3
FBW (%)
2.5
L
2
1.5
1
0.5
0 σ1 σ2
1 1.5 2 2.5 3 3.5 4 4.5 5
H (mm)
FBW (CST)
FBW (Q eig )
0 0
Figure 8: Fractional bandwidth FBW (VSWR < 2) for a R50 × 30
patch. Figure 10: The first two characteristic modes (currents and
charges) for the SAU2 structure.
280 100
260 90
240 80 Q1
220 70
Mode no.2 Mode no.3
200 60
αn
Mode no.1
Q
180 50
160 40
30
140 Q2
20
120
10
100
1 1.25 1.5 1.75 2 2.25 2.5 2.75 3 3.25 3.5 3.75 4 0
10 12 14 16 18 20 22 24 26 28 30
f (Hz) × 109
H (mm)
(a) (b)
Figure 12: Characteristic angles (left) and radiation Q for the SAU2.
0.03 0.03
0.02 J1 J2
0.02
0.01 0.01
0 0
−0.03 −0.02 −0.01 0 0.01 0.02 0.03 −0.03 −0.02 −0.01 0 0.01 0.02 0.03
(a) (b)
0.03 0.03
0.02 σ1 0.02 σ2
0.01 0.01
0 0
−0.01 −0.01
−0.02 −0.02
−0.03 −0.03
−0.03 −0.02 −0.01 0 0.01 0.02 0.03 −0.03 −0.02 −0.01 0 0.01 0.02 0.03
(c) (d)
Figure 13: Degenerated dominant mode J1 , and J2 of the FCL2 antenna (currents and charges).
Figure 11 presents a very simple concept showing the 5.1. The FCL-2 Fractal Antenna. The second presented struc-
main current paths for the J1 and J2 modes discussed above ture is the so-called fractal clover leaf (FCL) of the second
including the mirroring effect of the infinite ground plane. It iteration, [14]. The antenna is fed by an L-probe [30] that
could be simply stated that more opposing current paths lead excites its dominant mode and is located at height H =
to significant increase in Q. 36 mm. Actually, the dominant mode is composed of two
We show detailed behaviour only for SAU2 (the situation degenerated modes J1 and J2 (Figure 13). The second higher
is similar for SAU1)—see Figure 12 that confirms high Q for mode J3 is shown at Figure 14 for completeness.
the J1 mode. Characteristic angles are calculated for H =
29 mm, the actual height for which the dual-band antenna Figure 15 shows the main current paths of these modes,
was designed [29]. and we can again deduce that the dominant mode will exhibit
International Journal of Antennas and Propagation 7
0.03 0.03
0.02 J3 0.02 σ3
0.01 0.01
0 0
−0.01
−0.02
−0.03
−0.03 −0.02 −0.01 0 0.01 0.02 0.03 −0.03 −0.02 −0.01 0 0.01 0.02 0.03
(a) (b)
Mode 1, 2 8.25
J1 + J2 6.87
5.5
4.12
2.75
1.37
= 0
−3 .6 2
−7 .2 4
−1 0 .8 7
−1 4 .4 9
−1 8 .1 2
−2 1 .7 4
Mode 3 7.32
6.1
J3 4.88
3.66
2.44
1.22
= 0
−3 .7 7
−7 .5 5
−1 1 .3 3
−1 5 .1 1
−1 8 .8 9
−2 2 .6 7
Figure 15: Schematic depiction of the dominant current paths for the dominant (J1 + J2 ) and the second higher J3 modes together with their
modal radiation patterns.
lower Q compared to J3 . This is confirmed by Figure 16—J3 Secondly, it has been observed that resonant frequency is
has more than 200x higher radiation Q. quite a complicated function of height. Unfortunately we do
not yet have any physical explanation as to why some modes
present minimum values of fr .
6. Resonant Properties of Studied Antennas Looking at Figure 18, it is clear (and interesting) that
The properties of studied antennas are summarized in this the resonant frequency behaves quite differently for low-
section. At first we observed that microstrip antenna could Q and high-Q modes. The resonant frequency of low-Q
support different kinds of modes regarding their Q factors modes is much more sensitive to the height, whereas high-Q
(see Figure 17): modes exhibit almost constant fr when the height is varied.
The proposed explanation is that the opposite currents
(a) low Q modes with the current flowing in one direc- (responsible for high Q) keep reactive fields very close to the
tion and not changing its phase (dominant modes of radiating structure so the effect of a fringing field coupled to
simple shapes like rectangular, circular patch, and so the ground plane becomes almost negligible.
forth.)
(b) high Q modes with part of the currents flowing in the 7. Conclusions
opposite direction. These modes exist even on simple
“U” shaped patch (Figure 9 left) and on complex Modal resonant properties of selected microstrip patch
(fractal) geometries. antennas have been studied with the help of characteristic
8 International Journal of Antennas and Propagation
280 3
260 2.8
Low-Q
240 2.6 modes
220 2.4
2.2
fr (GHz)
200 Qeig3 = 226.3
2
αn
180
High-Q
1.8
160 modes
Qeig1 = Qeig2 = 10.5 1.6
140
1.4
120
1.2
100
1
80 0 5 10 15 20 25 30 35
1 1.1 1.2 1.3 1.4 1.5 1.6 1.7 1.8 1.9 2 H (mm)
f (GHz) × 109
FCL2 SAU2 mode 2
Mode 1 SAU1 mode 1 R50 × 50
Mode 2 SAU1 mode 2 R50 × 30
Mode 3 SAU2 mode 1
Figure 16: Characteristic angles for the FCL2 structure at H = Figure 18: Resonant frequencies for different antennas/modes.
29 mm.
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Hindawi Publishing Corporation
International Journal of Antennas and Propagation
Volume 2012, Article ID 167658, 8 pages
doi:10.1155/2012/167658
Research Article
Tunable Compact UHF RFID Metal Tag Based on
CPW Open Stub Feed PIFA Antenna
Copyright © 2012 L. Mo and C. Qin. This is an open access article distributed under the Creative Commons Attribution License,
which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited.
For the ultrahigh frequency radio frequency identification (UHF RFID) metal tag, it always has the difficulties of compact
designing, especially for the conjugate impedance matching, low antenna gain, and fabrication or environmental detuning. In
this paper, a tunable compact UHF RFID metal tag is designed based on CPW open stub feed PIFA antenna. By changing the
length of the open stub, the impedance of the PIFA antenna could be tuned in a large scale for conjugate impedance matching. The
open stub makes it easy to tune the resonant frequency to alleviate the fabrication detuning or the environmental detuning, even
after the manufacture. Moreover, the CPW structure of the open stub feed can resist the effects of the metallic surface and increase
the antenna gain for the compact PIFA antenna. Modeling analysis and simulation are in good agreement with the measurement
results. It showed that the UHF RFID metal tag could be designed compact with good performance based on the CPW open stub
feed PIFA antenna.
for commercial UHF RFID tags manufacture, the impedance Power + query
Antenna Chip
may not be matched well between the antenna and the chip
because of the simulation error and the fabrication variabil- Tag
ity, such as the substrate permittivity difference, manufactur- Reader Za Zc
ing difference, and chip impedance difference. To get good
impedance matching and performance, some manufacturers
use automatic laser or mill machine to adjust the antenna Backscattered wave
physical structure to tune the impedance of the antenna.
For this purpose, the tag antenna should be designed to Antenna impedance Chip impedance
have an easy tuning structure, which is easy for laser milling Figure 1: Principle of back-scattered UHF RFID system.
machine to adjust and achieving good impedance matching.
Impedance tuning is also very useful to alleviate the detuning
effects due to the different metallic application environments
For UHF RFID tag, one of the most important criteria of
[2]. Besides the impedance matching and tuning, antenna performance is the read range. The maximum read range of
gain is another challenge for the compact RFID metal tag. the tag can be calculated as [18]
Compact RFID metal tags always have lower antenna gain
and shorter read range. PIFA antenna is widely used for λ Pt Gt Gr τ
the compact UHF RFID tag mountable on metallic objects r= , (1)
4π Pth
[9–11]. However, with radiation patch size reduction, the
antenna gain decreases and the impedance matching become where λ is the free space wavelength, Pt is the power
difficult. Using two PIFAs can get better antenna gains [12, transmitted by the reader, Gt is the gain of the antenna of
13] and flexible impedance matching [5, 14]. But the size the reader, Gr is the gain of the antenna of the tag, τ is the
would increase obviously. Therefore, all of these previous power transmission coefficient between the tag antenna and
works cannot fit the requirement of compact profile, easy the chip, and Pth is the threshold power of the chip. When the
tuning, and satisfied antenna gain. reader and the chip of the tag keep the same, the maximum
In this paper, a tunable compact UHF RFID metal tag read range of the UHF RFID tag is mainly determined by the
antenna with a CPW open stub feed is proposed. Through design of the tag antenna, especially the gain of the antenna
PIFA antenna, the tag was compact designed. The impedance of the tag (Gr ) and the power transmission coefficient (τ)
of the antenna could be tuned freely by changing the [2]. The power transmission coefficient τ is determined by
length of the open stub [15]. Because of the open stub the impedance matching of the chip and the antenna, which
design, the impedance matching can be tuned even after can be calculated as follows:
the fabrication of the tag. Together with the open stub, the 4Rc Ra
CPW is used to resist the effects of the metallic objects τ= , 0 ≤ τ ≤ 1, (2)
| Z c + Za | 2
[16] and improve the performance of antenna gain [17]. In
Section 2, the considerations of the UHF RFID tag design are where Zc = Rc + jXc is the impedance of the tag chip,
discussed first. Based on these considerations, the proposed Za = Ra + jXa is the impedance of the tag antenna. When
antenna is described in Section 3, with antenna structure, the impedances of the antenna and the chip are conjugate
theoretical modeling and simulation results. In Section 4, matching, the transmission coefficient τ could get the maxi-
measurement results of a prototype based on this design mum value 1 and the most energy will be transmitted from
are also provided. Finally, discussion and conclusions are the antenna to the chip when the tag is being enquired by the
presented in Section 5. reader.
Besides the gain and the impedance matching, the band-
width and the radiation pattern are also important consid-
2. UHF RFID Tag Design Considerations erations for UHF RFID tag antenna design. Wide bandwidth
makes the tag to be read in a required bandwidth and the
A typical passive RFID tag is composed of a chip and an broadside radiation pattern makes the tag to be read in a
antenna, with no internal battery. All the energy it needs wide direction scale [19, 20]. In the realistic application, the
is obtained from the electromagnetic wave transmitted by a size and shape of the tag must be designed to be embedded or
RFID reader. In a passive back-scattered UHF RFID system, attached to the target objects and have a reliable performance
the reader transmits a modulated signal with periods of un- [21–23]. And for a commercial RFID tag, the cost is also
modulated carry wave, which is received by the antenna an important requirement to be considered [24, 25]. The
of the tag. When the chip of the tag is activated by the cost of the RFID tag is a critical factor for this technology
power from the antenna, it will send back its identification to be widely used around the world. Another requirement
information by modulating the backscattered signal. The for RFID tag antenna design is the easiness for the mass
backscattered signal is modulated by switching the load production [26]. This includes the antenna manufacture,
impedance of the chip between two states [1]. Figure 1 the chip bonding, the tag package, the performance testing,
illustrates the operation of back-scattered passive UHF RFID and the frequency tuning. The frequency tuning is useful
system. for reducing the differences of chips and substrate materials,
International Journal of Antennas and Propagation 3
keeping the tags with the same performance before going to Table 1: Requirements for designing a UHF RFID tag antenna.
be used.
Requirements of UHF RFID Effects of improvement
The requirements for designing a UHF RFID tag antenna
tag antenna design Read range Cost Reliability
are concluded in Table 1. The proposed tunable compact tag
√
antenna is designed according to these requirements. (1) Good impedance matching
√
(2) High antenna gain
3. Proposed Antenna Design √ √
(3) Wide bandwidth
√ √
A tunable compact UHF RFID metal tag antenna is proposed (4) Broadside radiation pattern
√
in this paper. The PIFA antenna design makes the antenna (5) Compact size and shape
√
compact than normal microstrip antenna. With an open stub (6) Low manufacture cost
√ √ √
feed, the antenna can be conjugate impedance matched with (7) Easy for testing and tuning
the chip easily by tuning the inset depth and the open stub √ √
(8) Stable performance for use
length [15]. This metal tag antenna can be fabricated cheaply
with normal PCB (Printed Circuit Board) technology. More- CPW open stub Shorting wall
over, with the open stub as tuning structure, the working h Chip
frequency of the tag can be tuned even after the tag has been
manufactured. The antenna bandwidth, radiation pattern,
and metal stability also keep with good performance. Ls Radiation patch
Linset
3.1. Antenna Structure. The structure and dimensions of the Winset
proposed antenna are illustrated in Figure 2. It is a planar L
W
inverted F antenna with a shorting wall to reduce size. The Z
radiation patch has dimensions of W (20 mm) × L (38 mm) Y
and is printed on a FR4 substrate (εr = 4.4, tan δ = 0.02).
The dimension of the substrate is (W + 2 mm) × (L + X Ground
2 mm) × h (3 mm). The open stub feed line is inset into the
patch to decrease the input impedance of the patch [27]. The
inset structure has a length of Linset and a width of Winset Figure 2: The open stub feed PIFA antenna structure. The dimen-
(8 mm). The open stub feed line has a length of Ls and a sions of the radiation patch are (L, W) while the open stub has
width of Ws (3 mm). The chip is attached on the feed port dimensions (Ls , Ws ) and the height of the substrate (FR4) between
the patch and the metallic surface is h.
composed by the open stub line and the radiation patch. In
order to enhance the gain of the compact PIFA antenna, a
CPW structure is designed for the open stub feed line. The
antenna is attached on a 200 mm × 200 mm metal plate. The The input impedance of the open stub only has imagi-
parameters Linset and Ls are used as variables for impedance nary part and its function of line length is shown in Figure 4.
matching. It shows that the reactance of the CPW open stub feed line is
capacitive when the length is less than 0.25 wavelength and
3.2. Theoretical Modeling Analysis. The transmission line is inductive when the length is between the 0.25 wavelength
model of the antenna is shown in Figure 3. From the antenna and 0.5 wavelength. The reactance of the CPW open stub is
model, it is easy to know that the radiation patch and the a function of cotangent, which means that when the length
CPW open stub feed lines are in series. Therefore, the input of the open stub changes from 0 to 0.5 λ, the imaginary part
impedance of the feed port of the antenna can be calculated of the input impedance of the open stub changes from −∞
as to +∞. Therefore, the imaginary part of the input impedance
of the antenna can be tuned freely by the length of the open
1
Zin = Zin 2
+ Zin , (3) stub in a large scale.
1
where Zin is the input impedance of the radiation patch of the 3.3. Simulation and Optimization. In order to get a bet-
PIFA antenna, Zin 2
is the input impedance of the CPW open ter impedance matching for the antenna, Finite-Element-
stub feed line. Method (FEM) based computational simulation software
According to the basic RF circuit theory [28], the input HFSS 12 is used for the simulation and optimization. For
impedance of the open stub can be simplified as the UHF RFID tags, the chips generally have complex
impedance, whose imaginary part is large and negative
1 1 because of the rectifier and energy storage capacitor. In
2
Zin = − jZ02 = − jZ02 , (4)
tan βLs tan(2πLs /λ) order to achieve the maximum energy transfer between the
antenna and the chip, the input impedance of the antenna
where Z02 is the characteristic impedance of the CPW open and the chip should be conjugate matching. That is, the
stub feed line, β is the wave number, Ls is the length of the real part is equal, and the imaginary part is opposite. As
CPW open stub feed line. the imaginary part is much larger than the real part of the
4 International Journal of Antennas and Propagation
0
frequency of 918 MHz. However, as the imaginary part of the
impedance is much larger than the real part, the impedance
Zin
2
Figure 4: The input impedance of the CPW open stub feed line.
4. Measurement Results
Based on the above-optimized parameters, the antenna
sample was produced with an FR4 dielectric plate, as shown
impedance, the impedance matching is mainly determined in Figure 7. The chip was attached to the antenna feed port
by the imaginary part matching. So, the antenna should with the traditional bonding technology. In order to test and
be designed to have a structure easy for impedance tuning, compare the performance with the simulation results, the
especially for imaginary part tuning. As the proposed tag tag was also mounted on a 200 × 200 mm2 copper plate.
antenna in this paper is designed for the North America UHF A commercial RFID reader, CSL-461 4-Port EPC Class1
RFID bandwidth (902 MHz∼928 MHz), the tag antenna Gen2 UHF RFID Reader [29], was used to test the tag. The
should have good impedance matching at this bandwidth. bandwidth of the reader is 902 MHz∼928 MHz. The output
The chip used for the tag is the RI UHF 00001 01 UHF power of the reader can be tuned from 15 dBm to 30 dBm,
RFID chip of TI (Texas Instruments), whose impedance is with a step of 0.25 dBm. The antenna of the reader is CS-771-
9.9-j60.3 Ω at the frequency of 915 MHz. The structure of 2-R with a gain of 6 dBi. Combining the output power of the
the antenna is shown in Figure 2. In order to simulate the reader and the reader antenna gain, the maximum radiation
tag antenna on the surface of metal, the tag is simulated on power is 36 dBm (4 W EIRP). The reader and the tag are
the surface of a reference metallic plate of 200 × 200 mm2 . manufactured with the protocol of EPC Class1 Gen2 and
According to the relative permittivity of the substrate, the ISO 18000-6C. According to the tag performance parameters
length of the radiation patch (L) of the PIFA antenna is and test methods of EPCglobal, the performance of the tag
chosen as 37 mm, which makes the PIFA antenna resonant was measured based on the back-scattering method [30].
near the frequency of 915 MHz. The impedance matching The maximum read range, power bandwidth, and radiation
between the antenna and the chip is tuned by Linset and pattern were measured with the same method.
Ls . Through the back-scattering method, the best impedance
For patch antennas with the inset feed structure, increas- matching frequency (resonant frequency) of the antenna
ing the depth of the inset could decrease the input impedance could be measured. Because of the fabrication variability,
of the antenna [27]. Therefore, the length of the inset Linset the best impedance matching frequency is a little lower than
can be used to tune the real part of the antenna impedance. 915 MHz. Then, as shown in Figures 5(b) and 6(b), through
Figure 5(a) shows the resistance tuning of the proposed cutting the length of the open stub, the imaginary part of the
antenna with different inset depths (Linset ). The resistance of impedance can be decreased and the resonant frequency can
the antenna decreases with the increase of the inset depth be increased to the target working frequency. In this way, the
Linset . As we analyze above, the CPW open stub feed line can fabricated tag prototype is optimized by tuning the length
International Journal of Antennas and Propagation 5
20 150
120
15
Resistance (ohm)
Reactance (ohm)
90
10
60
Chip resistance
5
30 Chip reactance
conjugate
0 0
0.86 0.88 0.9 0.92 0.94 0.96 0.86 0.88 0.9 0.92 0.94 0.96
Frequency (GHz) Frequency (GHz)
Linset = 12 mm Linset = 15 mm Ls = 28 mm Ls = 31 mm
Linset = 13 mm Linset = 16 mm Ls = 29 mm Ls = 32 mm
Linset = 14 mm Ls = 30 mm
(a) (b)
Figure 5: Antenna input impedance tuning. (a) Input resistance curves of the antenna with different inset depths Linset (L = 37 mm, W =
20 mm, h = 3 mm, Ls = 30 mm). (b) Input reactance curves of the antenna with different open stub length Ls (L = 37 mm, W = 20 mm,
h = 3 mm, Linset = 14 mm).
100 0
37 MHz
−5
80 895 MHz 932 MHz
Reflection coefficient S11 (dB)
60 −15
(a) (b)
Figure 6: The input impedance and reflection coefficient S11 of the antenna with optimized parameters: (a) impedance, (b) reflection
coefficient S11 .
of the open stub to alleviate the detuning effects due to the same. The power bandwidth of the tag was measured in
fabrication process. Figure 9. The output power of the reader needed to read
The maximum read range of the tag in North America the tags at different frequencies was normalized with the
bandwidth is plotted in the Figure 8. The antenna has a stable minimum value. The minimum is 0 dB. From Figure 9, it is
read range in the whole North America UHF RFID band easy to calculate that the 3 dB power bandwidth is 903 MHz∼
with a max value of 4.7 meters at the frequency of 915 MHz. 927 MHz, which covers most of the North America UHF
The tested results and the theoretical values are almost the RFID bandwidth. Compared with the 3 dB bandwidth of the
6 International Journal of Antennas and Propagation
24 MHz
3
1
Figure 7: Photograph of the fabricated tag antenna.
0
904 908 912 916 920 924 928
6 Frequency (GHz)
0
904 908 912 916 920 924 928 of the open stub even after the tag has been fabricated.
Frequency (GHz) This can be used to alleviate the detuning effects of the
fabrication error and the metallic application environments.
Testing results
Simulation results
With deceasing the length of the open stub, the imaginary
part of the antenna can be reduced and the working
Figure 8: Theoretical and experimental read ranges for the open frequency can be increased. With increasing the length of the
stub feed patch antenna (EIRP = 4 W). open stub, the imaginary part of the antenna can be increased
and the working frequency can be decreased.
The testing results were in good agreement with the
simulation. This antenna has stable performance on different
reflection coefficient S11 , the 3 dB power bandwidth is a little
sizes of metallic objects. Four features can be concluded for
narrow, but the central bandwidth is almost the same. The
this antenna design as follows.
measured radiation patterns of the tag at the frequency of
915 MHz are shown in Figure 10. The antenna has nearly
broadside hemisphere radiation pattern performance at both (1) By PIFA antenna design, the size of the tag can be
E plane and H plane. Tested results agree well with the effectively reduced. The length of the PIFA antenna
simulation results. The tag was attached on the metallic is only one half of that of microstrip antenna.
plates of different size to test its metal performance. The (2) An open stub feed is used to realize the impedance
testing results are plotted in Table 2, which shows that the matching for this compact PIFA antenna. The impe-
metal tag has stable read range when it is placed on the dance matching between the antenna and the chip
surface of different metallic objects. could be achieved easily by tuning the length of
the CPW open stub feed line. And this impedance
5. Discussion and Conclusion matching method could be used with different chips
and input impedances.
A tunable compact UHF RFID metal tag based on CPW open
stub feed PIFA antenna is designed in this paper. Using CPW (3) With the CPW open stub feed line, the impedance
open stub feed line, the impedance matching and antenna matching of the tag could be tuned even after the
gain can be well designed. Moreover, because of the PIFA manufacture of the tag. This makes it suitable for
and the CPW structure, the antenna has stable performance accurate impedance matching of the UHF RFID tag
for attaching on the surface of metallic objects. The working for manufacture and different application environ-
frequency of the antenna can be tuned by milling the length ments.
International Journal of Antennas and Propagation 7
0 0
0 0
330 30 330 30
−10 −10
−30 −30
(dB)
(dB)
−40 270 90 −40 270 90
−30 −30
−20 −20
240 120 240 120
−10 −10
Figure 10: Radiation patterns of the open stub feed PIFA antenna: (a) E plane, (b) H plane.
(4) The CPW structure of the open stub feed can [7] K. H. Kim, J. G. Song, D. H. Kim, H. S. Hu, and J. H.
resist the effects of the metallic surface and increase Park, “Fork-shaped RFID tag antenna mountable on metallic
the antenna gain for the PIFA antenna, which will surfaces,” Electronics Letters, vol. 43, no. 25, pp. 1400–1402,
keep the metal tag having a stable performance for 2007.
attaching on the surface of different metallic objects. [8] H.-W. Son and S.-H. Jeong, “Wideband RFID tag antenna
for metallic surfaces using proximity-coupled feed,” IEEE
Antennas and Wireless Propagation Letters, vol. 10, pp. 377–
Acknowledgment 380, 2011.
[9] M. Hirvonen, P. Pursula, K. Jaakkola, and K. Laukkanen,
This work was supported by the major projects of the Edu- “Planar inverted-F antenna for radio frequency identification,”
cation Administration (Y200907699), Zhejiang province, Electronics Letters, vol. 40, no. 14, pp. 848–850, 2004.
China. [10] H. Kwon and B. Lee, “Compact slotted planar inverted-F RFID
tag mountable on metallic objects,” Electronics Letters, vol. 41,
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8 International Journal of Antennas and Propagation
Review Article
Some Recent Developments of Microstrip Antenna
Yong Liu, Li-Ming Si, Meng Wei, Pixian Yan, Pengfei Yang, Hongda Lu, Chao Zheng,
Yong Yuan, Jinchao Mou, Xin Lv, and Housjun Sun
Department of Electronic Engineering, School of Information and Electronics, Beijing Institute of Technology, Beijing 100081, China
Copyright © 2012 Yong Liu et al. This is an open access article distributed under the Creative Commons Attribution License,
which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited.
Although the microstrip antenna has been extensively studied in the past few decades as one of the standard planar antennas, it still
has a huge potential for further developments. The paper suggests three areas for further research based on our previous works
on microstrip antenna elements and arrays. One is exploring the variety of microstrip antenna topologies to meet the desired
requirement such as ultrawide band (UWB), high gain, miniaturization, circular polarization, multipolarized, and so on. Another
is to apply microstrip antenna to form composite antenna which is more potent than the individual antenna. The last is growing
towards highly integration of antenna/array and feeding network or operating at relatively high frequencies, like sub-millimeter
wave or terahertz (THz) wave regime, by using the advanced machining techniques. To support our points of view, some examples
of antennas developed in our group are presented and discussed.
Patch
Stripline-microstrip transition
H
0 25 50
(mm)
(a) The structure of the dual-polarized microstrip antenna array
2
Table 2: The measured gain of the quasi-Yagi antenna (unit: dBi).
H-port
V-port
2.2. Dual-Polarized Microstrip Antenna Array. The dual-
polarized antenna is highly required for the radar, electronic Figure 4: The VSWR of the dual-polarized microstrip antenna
countermeasure, and aerospace systems. It is known that the array.
microstrip antenna can easily be integrated with microwave
circuits and feeding network. Here, a novel Ku-band dual-
polarization microstrip antenna array with a mixed feeding The VSWR, radiation patterns, and the isolation between
network, that is, the slot coupled feeding (V-port) and the co- two polarizations of the proposed dual-polarized microstrip
plane feeding (H-port), is designed by our group, as shown antenna array are shown in Figures 4, 5, and 6, respectively.
in Figure 3. It is a three layers structure: top microstrip patch The results indicate that this microstrip antenna array has a
layer, middle stripline feeding network layer, and bottom good impedance matching, good radiation performance, as
coplane microstrip feeding network layer. Through proper well as very high isolation (less than −25 dB), which can be
array arrangement, very good isolation can be obtained. an idea candidate for the dual-polarized wireless systems.
4 International Journal of Antennas and Propagation
Figure 5: The radiation patterns of the dual-polarized microstrip antenna array at the center frequency.
Slot
Dielectric
Groove guide
Z
Slot array Circular polarization grid
Groove guide
Feeding network
X Y
(a) The structure of the monopulse circular-polarized DCWS antenna
array (separating view)
Parasitic patch 22
Layer 1 h1 , εr1
Layer 2 h2 , εr2
Driven patch
Layer 3 h3 , εr3 21
Upper-ground coupled slot
Layer 4 h4 , εr4
Gain (dBi)
Feeding stripline
Layer 5 h5 , εr5 20
Lower ground
(a) Schematic side view of the structure of the high
integrate broadband microstrip antenna array 19
18
15.4 15.6 15.8 16 16.2 16.4 16.6 16.8 17
Frequency (GHz)
1#
2#
−10
Gain (dB)
−20
−30
−40
−80 −60 −40 −20 0 20 40 60 80
θ (◦ )
Figure 14: The radiation pattern of the high integrate broad-band microstrip antenna array using MPCB technology at the center
frequency.
5. Conclusion
national security, space exploration and communication,
and so forth [39–46]. To realize THz transceiver system, The advantages and disadvantages of microstrip antennas are
antenna is an essential component. We often use horn anten- discussed in this paper. In particular, three areas for further
na, lens antenna, and dielectric parabolic antenna, for THz development of microstrip antennas are presented based
systems. However, they are not easy to integrate with mono- on our previous works on microstrip antenna elements and
lithic integrate circuits. Although the microstrip antenna arrays. Variety of microstrip antenna topologies and micros-
has the merits of small volume, light weight, and easy trip-antenna-based composite antenna are discussed, and
8 International Journal of Antennas and Propagation
1200
1400
Responsivity (mV/mW)
1000
Responsivity (mV/mW)
1200
1000 800
800
600
600
400
12 14 16 18 20 22 26 28 30 32 34 36 38 40 42
Frequency (GHz) Frequency (GHz)
(a) Ku-band (b) Ka-band
500 200
400
Responsivity (mV/mW)
150
Responsivity (mV/mW)
300
100
200
50
100
0 0
50 55 60 65 70 75 75 80 85 90 95 100 105 110 115
Frequency (GHz) Frequency (GHz)
(c) V-band (d) W-band
Figure 16: Frequency responses test results of the THz wave planar integrated active microstrip antenna covered by a dielectric lens.
the advanced machining techniques pushing the microstrip [6] K. R. Carver and J. W. Mink, “Microstrip antenna technology,”
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and Propagation, vol. 51, no. 3, pp. 421–429, 2003. metamaterial resonators,” in Proceedings of the International
10 International Journal of Antennas and Propagation
Research Article
New Configurations of Low-Cost Dual-Polarized Printed
Antennas for UWB Arrays
Copyright © 2012 Guido Valerio et al. This is an open access article distributed under the Creative Commons Attribution License,
which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited.
A novel class of structures is proposed to realize ultra-wide-band radiating elements for large arrays, providing dual polarization,
beam scanning, and compact and inexpensive realization based on suitable rhombic arrangements of dipoles printed on low-cost
layered substrates. In a first implementation, four rhombic shapes, orthogonally placed on the same layer, provide two orthogonal
polarizations. In a second implementation, the two polarizations are excited by two rhombic shapes printed on two different layers
in a stacked-patch-like arrangement. This latter structure leads to a better lateral shielding of the single radiating element, in order
to reduce mutual interactions among adjacent elements in array environment. The behavioral features of these antennas have
been tested with various parametric analyses. Practical aspects have been addressed such as the choice of appropriate feeding and
of commercially available dielectric layers. The resulting antennas are matched at the input ports in an extremely wide range of
frequencies (5–25 GHz), covering various microwave applications, such as aircraft surveillance, weather polarimetric radars, and
control and communications systems. Good radiating features, in terms of pattern shape and gain, are observed in a large band of
frequencies. The basic scanning performance of large and small array configurations is finally investigated.
2. Basic Structures
The two structures presented here are based on various y
arrangements of printed dipoles of different lengths forming
approximately rhombic-shaped elements (see Figures 1 and
x
2). The central patch is the longest and can be fed at its ends Port
4 z
with one or two probes; the side patches, being six or eight
depending on the structure, have smaller dimensions and are (b)
parasitic. To further increase the bandwidth, the dipoles are
Figure 1: Geometry of the radiating element under analysis in
printed on a three-layer structure, designed with low-cost
Section 3, based on four rhombic coplanar elements printed on
commercially available dielectric substrates. In the present
the top of a three-layer configuration. (a) 3D view of the antenna.
analysis, the lower and the upper layers are duroid RT 5880 (b) Upper view of the structure. Lowest substrate: thickness h =
Rogers substrates with relative permittivity εr,1 = 2.2 and 1.5 mm, relative dielectric constant εr = 2.2, loss tangent tan δ =
dielectric loss tangent tan δ1 = 9.0 · 10−4 , the middle layer 9 · 10−4 (duroid RT 5880 Rogers). Middle substrate thickness h =
is an FR4 epoxy substrate with εr,2 = 4.4 and dielectric 7.5 mm, relative dielectric constant εr = 4.4, loss tangent tan δ =
loss tangent tan δ2 = 200.0 · 10−4 . It should be pointed 200 × 10−4 (FR4 epoxy). Uppermost substrate thickness h = 2 mm,
out that, even if loss effects can become nonnegligible relative dielectric constant εr = 2.2, loss tangent tan δ = 9 · 10−4 (RT
(particularly in higher-frequency ranges); in this case the 5880 Roger). Overall unit-cell dimension 43 × 43 mm. Geometrical
choice of such inexpensive commercial substrates is mainly parameters of the patches: w1 = 1.9 mm, l1 = 17.3 mm, w2 = 2 mm,
related to the strong reduction of costs in the implementation l2 = 7 mm, w3 = 1 mm, l3 = 11.3 mm, w4 = 0.6 mm, l4 = 4 mm,
of large arrays of elements, exploiting also well-established w5 = 0.4 mm, and l5 = 2 mm (wi and li are the width and the length
of the ith patch, resp.). Diameter of the probe d = 0.86 mm.
manufacturing printed-circuit techniques of PCB [17]. A
number of other materials having similar electric parameters
but reduced losses can be chosen if higher efficiency is desired
(e.g., for the internal layer a dielectric such as TMM4, having a couple of rhombic elements, mutually rotated of 90◦ . The
a loss tangent one order of magnitude lower than FR4 can be relevant features of two possible rhombic arrangements are
employed). Some compared results concerning the influence analyzed in detail in the next two sections.
of such losses on the gain and efficiency of these antennas will
be presented next. In the following analyses, also nonideal
effects of the metalization are taken into account, considering 3. Unshielded Antenna with Coplanar
finite-conductivity strips made of copper (σ = 5.8 · 107 S/m, Dual Rhombic Elements
μr = 0.999991) having nonzero thickness (10−6 m). A single
rhombic shape mainly provides a field linearly polarized, 3.1. Structure. A fully dual-polarized radiating element can
with components related to the direction of the relevant be obtained by arranging four rhombic shapes, printed on
dipoles. The dual polarizations can thus be obtained using the top of the three-layer structure described in the previous
International Journal of Antennas and Propagation 3
x
3.2. Analysis and Results. A throughout optimization pro-
z
cedure, involving the dimensions of the patches and the
(b) positions of the feeding probes, has been carried out with
the software package “ModeFrontier” [18], with the aim of
Figure 2: Geometry of the radiating element under analysis in
Section 4, based on another rhombic element printed on the top achieving the best input matching in the desired frequency
layer, and a rhombic element printed on the second layer. (a) 3D band. The optimization has been performed at first on a
view of the antenna. (b) Upper view of the structure. Same three- single rhombic shape and then refined with the full four-
layer configuration as in Figure 1. Geometrical parameters of the rhombus structure. The algorithm used is a multiobjective
patches of the upper element: w1 = 1 mm, l1 = 29.6 mm, w2 = genetic algorithm with multisearch elitism for enhanced
0.8 mm, l2 = 10 mm, w3 = 0.6 mm, l3 = 8 mm, w4 = 0.6 mm, and robustness [19]. Its objectives were chosen as the conditions
l4 = 7 mm. The diameter of the probes feeding the upper element |Si j | < −10 dB at 5 GHz for any i, j, and the variables were all
is d = 0.24 mm. Geometrical parameters of the patches of the lower the dimensions and positions of the various patches.
element: w1 = 0.6 mm, l1 = 26.8 mm, w2 = 0.6 mm, l2 = 15.6 mm, The results shown here have been computed with the
w3 = 0.6 mm, l3 = 6 mm, w4 = 0.6 mm, and l4 = 6 mm. (wi and li time-domain solver of the high-frequency electromagnetic
are the width and the length of the ith patch, resp.) The diameter of
CAD “CST Studio 2010” [20]. The layered substrate is
the probes feeding the lower element is d = 0.4 mm. The size of the
box is 40 mm.
assumed laterally unbounded, and a single radiating element,
composed by four rhombic shapes, is considered as in
Figure 1. The input ports are modeled as coaxial cables,
fed with proper phase shifts. The feeding network providing
section, as in Figure 1. The central patch of each rhombic these shifts among the different ports is not simulated
element is fed with a probe at its outer end, and opposite here and will be object of future work. The CST model is
patches are fed with signals having the same magnitude discretized with hexahedral cells of average dimension λ/10;
and opposite sign. With this geometrical arrangement, one open conditions are placed at the side boundary of the cell,
couple of opposite rhombic elements provides one linear thus assuming a laterally unbounded substrate; extra space is
polarization, and the other couple of opposite elements, added in the top half plane in order to accurately estimate the
rotated of 90◦ , provides the orthogonal polarization. If the radiation patterns. Waveguide ports are defined at the coaxial
same signal is radiated through both the polarizations, a cables and excited with the fundamental TEM mode.
general elliptical polarization can be obtained, by suitably Based on the data sheets of the electromagnetic param-
tuning the relevant phase shifts among the input probes. eters of these materials as a function of the frequency (if
In particular, a circular polarization can be obtained by available), the dispersive effects can be taken properly into
imposing a 90◦ phase shift between adjacent elements. account in the simulations. The relevant analysis shows
The antenna is designed to work in a very wide frequency anyway that these dispersive effects do not change sensitively
range, possibly greater than C-X bands (4–12 GHz) in the the performance of our antennas in the ranges of interest.
case of interest. Its transverse dimensions of the element Due to the geometrical symmetries, the input features
are approximately 4 cm, leading to a possible compact of the antenna are described by three different scattering
implementation in array at these frequencies. Different parameters. In Figure 3, the magnitude in dB of these three
4 International Journal of Antennas and Propagation
0 0
10
30 30
−10 0
−20 −10
(dBi)
60
−20
−30
−30
−40
−40 90
−50
(a)
−60 0
10
0 5 10 15 20 25 30 30
f (GHz) 0
| S11 | (dB) −10
(dBi)
| S21 | (dB) 60
| S31 | (dB) −20
−30
Figure 3: Antenna of Figure 1. Magnitude of the scattering
coefficients of the radiating element fed by the four probes.
−40 90
Reflection coefficient |S11 | at the input port 1 (black line) and
transmission coefficients |S21 | from port 1 to port 2 (gray line) and (b)
from port 1 to port 3 (dashed black line).
Figure 4: Antenna in Figure 1. Gain in polar form on the principal
elevation plane φ = 0◦ of the single element at different frequencies
in the operational bandwidth 5–25 GHz: 5 GHz (solid black line),
parameters is shown in the frequency band considered, from 10 GHz (solid gray line), 20 GHz (dashed black line). (a) Ports 1 and
dc to 25 GHz. The reflection coefficient at one input cable 2 are fed, exciting a field along the θ direction. (b) Ports 3 and 4 are
fed, exciting a field along the ϕ direction.
is shown in black solid line (due to the symmetry of the
structure, the cable considered is arbitrary, since S11 = S22 =
S33 = S44 ). The coupling between opposite ports (e.g.,
|S21 |) is shown in gray solid line, and the coupling between 4. Shielded Antenna with Stacked
adjacent ports (e.g., |S31 |) is shown in black dashed line. The Dual Rhombic Elements
operational frequency band where the parameters are under
the conventional threshold of −10 dB is extremely wide: in 4.1. Structure. Once the antenna described in the previous
the figure, the band 5–25 GHz is well matched, but further section is introduced in an array environment, the UWB
simulations would show a good matching also beyond input performance of the array could deteriorate due
30 GHz, even though, as said, the radiation patterns tend to to strong mutual coupling among adjacent elements. To
deteriorate as the frequency increases. In Figure 4(a), the gain overcome such a problem, a lateral shielding of the single
is shown on the principal plane ϕ = 0◦ at different frequen- radiating element can be implemented with metallic walls.
cies from 5 GHz to 25 GHz if the ports 1 and 2 are fed with Such a configuration also enables us to reduce adverse effects
common amplitude and opposite phase: by symmetry, the related to possible launching of surface and leaky waves in the
electric field has only a θ-component. In Figure 4(b), the gain layered structure [11, 12]. As an example of this approach,
is shown on the same plane if the ports 3 and 4 are fed with an alternative shielded radiating element is presented here,
common amplitude and opposite phase: in this case, by sym- based on the same rhombic printed shapes described above,
metry, the electric field has only a ϕ-component. Good radi- but with elements placed on different layer interfaces.
ation features are found in the lower band of frequency, while In this case, shown in Figure 2, each couple of opposite
the pattern tends to deform at higher frequencies as expected. rhombic elements is here replaced by a single rhombus,
The features of the maximum gain are resumed in whose central larger patch is fed by two different probes at its
Table 1, considering their values with respect to the isotropic ends, carrying signals with common amplitude and opposite
radiator (dBi and the relevant beam angular locations on phase. With this original feeding structure involving two
the different planes). In the same table, maximum gain and probe elements on the opposite sides of each main strip, it is
radiation efficiencies are also presented with reference to noted that, by properly choosing the phase shift between the
the alternative choice of the TMM4 for the middle layer feeders, it is also possible to achieve a straightforward control
(with reduced losses, see Section 2 for details). As shown, of the current configuration excited on the strips, thus
acceptable results are achieved even with the lossy substrate, suitably influencing the directional features of the radiated
while significant improvements in the efficiency can be beam. In this structure, the two orthogonal polarizations are
obtained at higher frequencies with the use of a lower-loss then provided by two different rhombic elements, rotated
substrate. by 90◦ and printed on two different layers: the dipoles of
International Journal of Antennas and Propagation 5
f (GHz) Direction (ϑ) Max gain Efficiency Max gain (low loss) Efficiency (low loss)
5 ϑ = 0◦ 6.6 dBi 92.9% 7.1 dBi 99%
10 ϑ = 90◦ , ϕ = 54◦ 3.8 dBi 78.7% 4.5 dBi 96.6%
15 ϑ = 45◦ , ϕ = 22◦ 3.2 dBi 80.0% 3.7 dBi 96.8%
20 ϑ = 38◦ , ϕ = 24◦ 3.7 dBi 64.1% 4.9 dBi 92.7%
25 ϑ = 90◦ , ϕ = 9◦ 5.5 dBi 68.4% 6.2 dBi 93.3%
(dB)
−30
configuration than in the previous one, leading to an easier
input matching also in the shielded configuration. −40
Again, the dimensions of the various patches have been
optimized in order to yield the optimum input matching in −50
the frequency band 5–25 GHz. Of course, considering such a
different arrangement of the rhombic shapes, rather different −60
geometrical parameters have been reached in this second 0 5 10 15 20 25
structure as optimum values. f (GHz)
| S11 |
4.2. Analysis and Results. The results shown here have | S33 |
been obtained through full-wave simulations of the final | S13 |
optimized stacked and shielded structure again with the
time-domain solver of CST Studio. Figure 5: Antenna of Figure 2. Magnitude of the scattering
In Figure 5, the magnitude in dB of the three scattering coefficients of the radiating element fed by the four probes.
parameters of the shielded configuration is shown; also with Reflection coefficients |S11 | at the input port 1 feeding the upper
structure (black line), |S33 | at the input port 3 feeding the lower
this structure, a good input matching is reached in the very
structure (gray line), and transmission coefficient |S31 | from port
wide frequency band 5–25 GHz. The reflection coefficient 1 to port 3 (dashed black line).
at port 1 (i.e., feeding the upper rhombic shape) is shown
in solid black line, the reflection coefficient at port 3 (i.e.,
feeding the lower rhombic shape) is shown in solid gray line,
while the coupling between the rhombic shapes is shown in In Tables 2 and 3, a summary of the maximum-gain
dashed black line. values, locations, and radiation efficiencies is also given
The radiation patterns on the principal planes are shown for different frequencies, with reference to the feeding
again for various values of the frequency. In Figure 6(a), of the upper and lower element, respectively. As for the
the gain is shown on the principal plane ϕ = 0◦ , when previous unshielded antenna, maximum gain and efficiency
probes 1 and 2 feed the upper element with signals with is also presented with reference to the alternative choice of
common magnitude and 180◦ phase shift. The electric the TMM4 for the middle layer (with reduced losses, see
field is in this case polarized along the θ direction by Section 2 for details). In particular, while the lower element
symmetry. In Figure 6(b), the gain is shown on the same has a reduced efficiency at higher frequencies, its behavior
plane when the lower element is fed through probes 3 and can be substantially improved with the use of a lower-loss
4, with common magnitude and 180◦ phase shift. A dual substrate.
polarization is radiated with respect to the previous result,
the electric field being polarized along the ϕ direction. As 4.3. Array Behavior. In order to test further this type
seen, fairly regular radiation patterns are observed for both of structure, a first simple analysis has been led which
polarizations, in particular in the lower part of frequency. gives basic information on the scanning-beam directional
Since this configuration is studied with an “open add space” features for large arrays. To this aim, the radiation pattern
lateral boundary in CST, an estimation of backlobes is also of an array of 140 × 140 elements has been computed
present in the results. with an array-factor approximation, for different values of
The gain on the other principal plane ϕ = 90◦ is shown the pointing angle, depending on the selected phase shift
in Figure 7. The upper rhombic shape is fed in Figure 7(a), between adjacent elements of the array. In Figure 8(a), the
while the lower rhombic shape is fed in Figure 7(b). pattern on the plane ϕ = 0◦ at 5 GHz is shown for a
6 International Journal of Antennas and Propagation
20 0 0
30 20
30 30 30
10 10
0 0
(dBi)
60
(dBi)
−10 −10 60
−20 −20
−30 −30
−40 90 −40 90
Figure 6: Antenna in Figure 2. Gain in polar form on the principal elevation plane φ = 0◦ of the single element at different frequencies
in the operational bandwidth 5–25 GHz: 5 GHz (solid black line), 10 GHz (gray line), 20 GHz (dashed black line). (a) Ports 1 and 2 feed the
upper element, exciting a field along the θ direction. (b) Ports 3 and 4 feed the lower element, exciting a field along the ϕ direction.
0 0
20 20
30 30 30 30
10 10
0 0
(dBi)
(dBi)
−10 60 −10 60
−20 −20
−30 −30
−40 90 −40 90
Figure 7: Antenna in Figure 2. Gain in polar form on the principal elevation plane φ = 90◦ of the single element at different frequencies
in the operational bandwidth 5–25 GHz: 5 GHz (solid black line), 10 GHz (gray line), 20 GHz (dashed black line). (a) Ports 1 and 2 feed the
upper element, exciting a field along the θ direction. (b) Ports 3 and 4 feed the lower element, exciting a field along the ϕ direction.
broadside radiation when the four probes of each element in connection of the element spacing as phase shift and
are fed with common amplitude and a 90◦ shift in order frequency are varied.
to radiate a circular polarized field. The two components Further results have been obtained considering a linear
along θ and ϕ are shown in solid black line and in dashed array made of a small number of cells (3 × 1) along the x
gray line, respectively. In Figure 8(b), the same quantities direction in the reference system of Figure 2. In this case, to
are computed for a beam pointing at the elevation θ = 30◦ accurately predict the array performance, a nonapproximate
and azimuth ϕ = 0◦ . The gain at the main lobe direction is full-wave analysis has been necessary with a proper CAD
fairly regular, around 50 dBi; the side-lobe levels are rather implementation of the overall physical structure of the three
reduced (about 20 dB below the main lobe); the considered antennas. This rigorous approach allows us to to verify
phase-scanned pencil beams have half-power beamwidth of the actual scanning features of the radiated beam as a
about 0.4◦ . Effects related to grating lobes can be present function of the relevant phase shift. Representative behaviors
International Journal of Antennas and Propagation 7
f (GHz) Direction (ϑ) Max gain Efficiency Max gain (low loss) Efficiency (low loss)
5 ϑ = 0◦ 6.5 dBi 92.9% 6.9 dBi 99.3%
10 ϑ = 0◦ 6.5 dBi 56.1% 10.67 dBi 96.8%
15 ϑ = 55◦ , ϕ = 70◦ 4.2 dBi 47.3% 7.01 dBi 88.5%
20 ϑ = 50◦ , ϕ = 67◦ 2.5 dBi 40.6% 9.04 dBi 75.0%
25 ϑ = 21◦ , ϕ = 90◦ 4.2 dBi 41.5% 7.63 dBi 86.9%
f (GHz) Direction (ϑ) Max gain Efficiency Max gain (low loss) Efficiency (low loss)
5 ϑ = 0◦ 7.5 dBi 79.0% 8.4 dBi 95.8%
10 ϑ = 0◦ 8.2 dBi 37.0% 10.3 dBi 88.1%
15 ϑ = 61◦ , ϑ = 20◦ 2.5 dBi 28.0% 6.8 dBi 80.0%
20 ϑ = 0◦ 2.3 dBi 28.2% 7.8 dBi 88.6%
25 ϑ = 45◦ , ϑ = 20◦ 3.4 dBi 29.1% 7.5 dBi 80.8%
60 60
40 40
20 20
(dBi)
(dBi)
0 0
−20 −20
−40 −40
−90 −45 0 45 90 −90 −45 0 45 90
θ (◦ ) θ (◦ )
Figure 8: Array of shielded elements as in Figure 2. Gain on the principal elevation plane φ = 0◦ of a 140 × 140 array at 5 GHz. All the
ports are fed, exciting a circularly polarized field (its θ component in solid black line, its ϕ component in dashed gray line). (a) Beam pointing
at broadside (θ = 0◦ ). (b) Beam pointing at θ = 30◦ .
of radiation patterns for this small array are reported in the other cables are shown, avoiding coefficients having equal
Figure 9. In Figure 9(a), the gain on the two principal planes values due to evident geometrical symmetries. As expected,
is shown when the three upper sets of patches are fed in phase in the band of frequency investigated, a low level of coupling
(black curves) and when three lower sets of patches are fed in is found also in this 3 × 1 array. In Figure 10(b), the three
phase (gray curves). In Figure 9(b), the scanning capabilities lower sets of patches of the array are fed in phase through
of this small array are presented, showing the gain on the one of the two ports; the magnitude of the active reflection
principal plane ϕ = when the lower sets of patches are fed, coefficient at the middle cell and at one side cell is reported,
pointing their main beam along the ϕ = 0◦ plane, at θ = 10◦ , proving a good input matching also in this active array
20◦ , 30◦ , 40◦ . As expected, a fan beam is obtained, as typical configuration.
of linear arrays, having maximum gain between 12 and 9 dBi. As a last result, in Figure 11 the radiated patterns of
In Figure 10, couplings among cells are analyzed, again an array made of 3 × 3 elements is shown on both the
with reference to the feeding of the lower patches, only for the principal planes, ϕ = 0◦ , 90◦ for both the upper (Figure 11(a))
sake of brevity. In Figure 10(a), an input cable feeds the lower and lower (Figure 11(b)) patches. Again, a full-wave analysis
patch only in the central cell. The magnitude of the reflection has been performed, without recurring to simplified array-
coefficient at this port and of the coupling coefficients with factor formulations. All the cells of the array are fed in phase
8 International Journal of Antennas and Propagation
20 60
10 40
0 20
(dBi)
(dBi)
−10 0
−20 −20
−30 −40
−90 −60 −30 0 30 60 90 −90 −60 −30 0 30 60 90
θ (◦ ) θ (◦ )
(a) (b)
Figure 9: Array of 3 adjacent shielded cells as in Figure 2 at 5 GHz. (a) Gain on the principal elevation planes φ = 0◦ (solid lines) and φ = 90◦
(dashed lines). Beam pointing at broadside (θ = 0◦ , all the cells are excited in phase) by the upper patches (black lines) and by the lower
patches (gray lines). (b) The three lower sets of patches are phased in order to point at θ = 10◦ (black line), θ = 20◦ (gray line), θ = 30◦ (black
line with squares), and θ = 40◦ (gray line with squares).
0 0
−5
−10
−10
−20
−15
(dB)
(dB)
−30 −20
−25
−40
−30
−50
−35
−60 −40
5 10 15 20 25 5 10 15 20 25
f (GHz) f (GHz)
(a) (b)
Figure 10: Array of 3 adjacent shielded cells as in Figure 2 at 5 GHz. (a) Only the input port 3 feeds the lower patches in the central cell.
Magnitude of reflection coefficient (thick black line), of the coupling coefficient with a cable connected to the upper patch in the same cell
(thick gray line), and of coupling coefficients with cables in an adjacent cell (thin black lines). (b) All the three lower sets of patches as fed in
phase through port 3 (see Figure 2) and the respective translated ports in the other two cells. Magnitude of the active reflection coefficient
versus frequencies at the middle cell of the 3 × 1 array (black line) and at one side cell (gray line).
thus radiating a single beam at broadside, with gain around orientation. Attention was paid to practical implementa-
17 dBi, angular width 20◦ , and sidelobe level −15 dB with tion with inexpensive dielectrics commonly used in PCB
respect to the maximum gain. As shown in the pictures, a technology. These structures are low profile and lightweight
remarkable symmetry of the pattern is obtained on the prin- and are characterized by high modularity/scalability, which
cipal planes, also if the different polarizations are compared. makes them suitable to implement low-profile phased array
antennas of various shapes and sizes.
5. Conclusion Suitable extensive parametric analyses have been carried
out by means of advanced numerical tools as concerns the
A new class of UWB low-cost printed antennas has been pre- most efficient choice of the strip geometry configurations.
sented and optimized for dual-polarized radar applications A first design of the antenna element is made with four
in the microwave range 5–25 GHz. The basic single element coplanar rhombic elements, each fed by a probe reaching one
is composed by properly fed printed dipoles arranged in end of the central dipole. Relevant results are shown for a
rhombic configurations. Different elements can provide single element printed on a laterally unbounded substrate.
almost orthogonal polarizations, depending on their mutual An advanced alternative design has been proposed, leading
International Journal of Antennas and Propagation 9
Research Article
Design and Analysis of Wideband Nonuniform Branch Line
Coupler and Its Application in a Wideband Butler Matrix
Copyright © 2012 Yuli K. Ningsih et al. This is an open access article distributed under the Creative Commons Attribution License,
which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited.
This paper presents a novel wideband nonuniform branch line coupler. An exponential impedance taper is inserted, at the series
arms of the branch line coupler, to enhance the bandwidth. The behavior of the nonuniform coupler was mathematically analyzed,
and its design of scattering matrix was derived. For a return loss better than 10 dB, it achieved 61.1% bandwidth centered
at 9 GHz. Measured coupling magnitudes and phase exhibit good dispersive characteristic. For the 1 dB magnitude difference
and phase error within 3◦ , it achieved 22.2% bandwidth centered at 9 GHz. Furthermore, the novel branch line coupler was
implemented for a wideband crossover. Crossover was constructed by cascading two wideband nonuniform branch line couplers.
These components were employed to design a wideband Butler Matrix working at 9.4 GHz. The measurement results show that
the reflection coefficient between the output ports is better than 18 dB across 8.0 GHz–9.6 GHz, and the overall phase error is less
than 7◦ .
A =1
P1 √ P3
Z(z)/ 2
P1 P3 S 11 S 31
4.75
Zo Zo
8 12.75 14.5
S 21 S 41
3
P2 P4
P2 P4
Z(z)
Useful conversions for two-port network parameters for Se21 = − √ 1 + j ,
the even and odd modes of S11 and S21 can be defined as 2
(17)
follows [22]: Z(z)
So21 = √ 1 − j .
2
(A + B − C − D)e
Se11 = , (5) Based on (13), S31 can be expressed as follows
ΔY e
(A + B − C − D)o 1 Z(z)
So11 = , (6) S31 = − √ 1+ j − 1− j
ΔY o 2 2
2 1 Z(z)
Se21 = , (7) =− √ 1+ j−1+ j (18)
ΔY e 2 2
Z(z)
2 = −j √ .
So21 = , (8) 2
ΔY e
where Following (14), S41 nonuniform branch line coupler can be
calculating as follows
ΔY e = (A + B + C + D)e , (9)
1 Z(z)
S41 = −√ 1+ j + 1− j
ΔY = (A + B + C + D)o .
o
(10) 2 2
1 Z(z)
=− √ 1+ j +1− j (19)
Since the amplitude of the incident waves for these two 2 2
ports are ±1/2, the amplitudes of the emerging wave at each Z(z)
port of the nonuniform branch line coupler can be expressed =− √ .
2
as [22]:
From this result, both S31 and S41 nonuniform branch
1 e
line couplers have equal magnitudes of −3 dB. Therefore, due
S11 = S + So11 , (11)
2 11 to symmetry property, we also have that S11 = S22 = S33 =
1 e S44 = 0, S13 = S31 , S14 = S41 , and S21 = S34 . Therefore, the
S21 = S − So11 , (12) nonuniform branch line coupler has the following scattering
2 11
matrix in (20):
1 e
S31 = S + So21 , (13) ⎡ ⎤
2 21 0 0 j 1
1 e Z(z) ⎢
⎢0 0 1 j
⎥
⎥
S41 = S − So21 . (14) S=− √ ⎢ ⎥. (20)
2 21 2 ⎣ j 1 0 0 ⎦
1 j 0 0
Parameters even and odd modes of S11 nonuniform
branch line coupler can be expressed as (15) and (16) as
follows:
3. Fabrication and Measurement
Result of Wideband Nonuniform Branch
Z(z) −1 + j − j + 1 Line Coupler
Se11 = √ = 0, (15)
2 −1 + j + j − 1
To verify the equation, the nonuniform branch line coupler
Z(z) 1 + j − j − 1 was implemented and its S-parameter was measured. It
So11 = √ = 0. (16) was integrated on TLY substrate, which has a thickness
2 1+ j + j+1
of 1.57 mm. Figure 4 shows a photograph of a wideband
An ideal branch line coupler is designed to have zero nonuniform branch line coupler. Each branch at the series
reflection power and splits the input power in port 1 (P1 ) arm comprises an exponentially tapered microstrip line
4 International Journal of Antennas and Propagation
0
6 6.5 7 7.5 8 8.5 9 9.5 10 10.5 11 11.5 12 12.5
S-parameters magnitude (dB)
−5
−10
−15
−20
−25
Figure 8: Photograph of microstrip nonuniform crossover.
−30
Frequency (GHz)
−15
−20
200
Phase characteristic (deg)
150 −25
100 −30
50 Frequency (GHz)
0
−50 6 7 8 9 10 11 12 S (1, 1) S (3, 1)
S (2, 1) S (4, 1)
−100
−150 Figure 9: Measurement result for nonuniform crossover.
−200
Frequency (GHz)
S (4, 1)
S (3, 1)
3 dB hybrid
Cross over coupler
Figure 10: Final layout of the proposed wideband Butler Matrix
4 × 4.
P1 P2 P3 P4
−10
−4
−4 −15
−6 −20
−8 −25
−30
−10
−35
−12
S (2, 2) simulated S (2, 2) measured
S (5, 2) simulated S (5, 2) measured S (3, 3) simulated S (3, 3) measured
S (8, 2) simulated S (8, 2) measured
S (7, 2) measured S (6, 2) simulated (b) Input port 2 or 3 are excited
S (7, 2) simulated S (6, 2) measured
Figure 12: Return loss of the proposed Butler Matrix when different
(b) Input port 2 excitations ports are fed.
Frequency (GHz)
0
8 8.2 8.4 8.6 8.8 9 9.2 9.4 9.6 9.8 10
−2
Insertion loss (dB)
−4
4. Design and Fabrication of the Wideband
−6
Butler Matrix
−8
−10 Figure 7 shows the basic schematic of the 4 × 4 Butler Matrix
−12 [21]. Crossover also known as 0 dB couplers is a four-port
device and must provide for a very good matching and
S (5, 3) simulated S (5, 3) measured
isolation, while the transmitted signal should not be affected.
S (8, 3) simulated S (8, 3) measured
S (7, 3) measured S (6, 3) simulated In order to achieve wideband characteristic crossover, this
S (7, 3) simulated S (6, 3) measured paper proposes the cascade of two nonuniform branch line
couplers.
(c) Input port 3 excitations
Figure 8 shows the microstrip layout of the optimized
Frequency (GHz)
0 crossover. The crossover has a frequency bandwidth of
1 2 3 4 5 6 7 8 9 10 11
−2 1.3 GHz with VSWR = 2, which is about 22.2% of its centre
Insertion loss (dB)
50 generated beam.
40
30 P5 (◦ ) P7 (◦ ) P6 (◦ ) P8 (◦ ) β (◦ ) θ (◦ )
20 P1 45 90 135 180 45 14.4 [1L]
10 P2 135 0 225 90 −135 −48.6 [2R]
0
8 8.2 8.4 8.6 8.8 9 9.2 9.4 9.6 9.8 10 P3 90 225 0 135 135 48.6 [2L]
Frequency (GHz) P4 180 135 90 45 −45 −14.4 [1R]
Phase (port 3)-phase (port 1)
Phase (port 2)-phase (port 3)
Phase (port 4)-phase (port 2) of fabrication could contribute to reduction of the insertion
(a) Input port 1 excitation loss.
The simulated and measured results of the return loss
Frequency (GHz)
−115 at each port of the wideband 4 × 4 Butler Matrix is shown
Phase different (deg)
145
140
is excited, the phase difference was 45◦ , the direction of
135 generated beam (θ) will be 14.4◦ for 1L. It is summarized
130 in Table 1.
125
120
115 5. Conclusion
8 8.2 8.4 8.6 8.8 9 9.2 9.4 9.6 9.8 10
Frequency (GHz) A novel nonuniform branch line coupler has been
employed to achieve a wideband characteristic by expo-
Phase (port 6)-phase (port 5)
nential impedance taper technique. It is a simple design
Phase (port 6)-phase (port 5)
Phase (port 6)-phase (port 5)
without needs of using multilayer technology and this
will lead to cost reduction and design simplification. The
(c) Input port 3 excitations scattering matrix of the nonuniform branch line coupler
was derived and it was proved that the nonuniform branch
Frequency (GHz) line coupler has equal magnitude of −3 dB. Moreover, the
0 novel nonuniform branch line coupler has been employed
8 8.2 8.4 8.6 8.8 9 9.2 9.4 9.6 9.8 10
to achieve a wideband 0 dB crossover. Furthermore, these
Phase different (deg)
−10
−20
components have been implemented in the Butler Matrix
and that achieves wideband characteristics.
−30
−40
−50 References
−60
[1] T. A. Denidni and T. E. Libar, “Wide band four-port butler
matrix for switched multibeam antenna arrays,” in Proceedings
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(d) Input port 4 excitations Koukourlis, and S. Panas, “On the design of switched-beam
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beamforming techniques for scanned and multiple beam
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[4] W-D. Wirth, Radar Techniques Using Array Antennas, IEE high return losses and isolation,” in Asia Pacific Microwave
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[5] S. Y. Zheng, S. H. Yeung, W. S. Chan, and K. F. Man, “Broad- December 2009.
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[13] S. Gruszczyński, K. Wincza, and K. Sachse, “Reduced sidelobe
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[14] Y. C. Su, M. E. Bialkowski, F. C. E. Tsai, and K. H. Cheng,
“UWB switched-beam array antenna employing UWB butler
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Hindawi Publishing Corporation
International Journal of Antennas and Propagation
Volume 2012, Article ID 264618, 8 pages
doi:10.1155/2012/264618
Research Article
Isolation Improvement of a Microstrip Patch Array Antenna for
WCDMA Indoor Repeater Applications
Copyright © 2012 H. Lee and J. Park. This is an open access article distributed under the Creative Commons Attribution License,
which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited.
This paper presents the isolation improvement techniques of a microstrip patch array antenna for the indoor wideband code
division multiple access (WCDMA) repeater applications. One approach is to construct the single-feed switchable feed network
structure with an MS/NRI coupled-line coupler in order to reduce the mutual coupling level between antennas. Another approach
is to insert the soft surface unit cells near the edges of the microstrip patch elements in order to reduce backward radiation
waves. In order to further improve the isolation level, the server antenna and donor antenna are installedin orthogonal direction.
The fabricated antenna exhibits a gain over 7 dBi and higher isolation level between server and donor antennas below −70 dB at
WCDMA band.
(dB) 120 mm
0
−10
−20 4 5
S-parameter
−30
−40
120 mm
1
−50
−60
−70
0 0.5 1 1.5 2 2.5 3 3.5 4 4.5 5 3 2
Frequency (GHz)
S11 S31
S21 S41
Figure 3: Simulated S-parameter results for the MS/NRI coupled- Figure 4: Geometry of the feed network for two-by-two microstrip
line coupler of Figure 2. patch array antenna.
S41 34 mm
180 mm
180 mm
19 mm
20 mm
19 mm
coupler. When the MS/NRI coupled-line coupler was not The antenna consists of three-layer structure: an air layer
used, the scattering parameters (S25 , S34 , S52 , S43 ) between having a thickness of 1 mm, a dielectric substrate layer,
two antennas showed lower isolation. Since the patch and a stacked microstrip patch layer. The square microstrip
antennas were designed to be excited at their fundamental patches are fed by microstrip lines from perpendicular
resonant mode (TM10 ), strong coupling is produced between directions using the proximity coupled method. This switch-
two antennas that are placed parallel to the radiating edge able feed network was designed to achieve beam pattern
direction. It can be seen that the minimum isolation levels reconfigurable array antenna, which generates ±45◦ linearly
between the output ports exceed −18 dB at the aimed polarized slanted beam patterns at the Tx/Rx frequency
two switching frequencies of the proposed switchable feed bands. In order to dual slant beam, two Rx and Tx microstrip
network. patch elements were placed orthogonally. Figure 6 shows
the simulated surface current distribution at the resonant
3. Antenna Design frequency of 1.95 GHz (Tx) and 2.15 GHz (Rx). At the
frequency of 1.95 GHz, two patch antennas for Tx band
Figure 5 shows the geometry of the two-by-two microstrip are resonant, and two patch antennas for Rx-band are off-
patch array antenna using the proposed switchable feed resonant. As a result, most of the surface currents flow
network and proximity coupled square microstrip patch through the feed line for Tx-band antennas. On the other
elements. hand, at the frequency of 2.15 GHz, two patch antennas for
International Journal of Antennas and Propagation 5
20 mm 3 mm
12 mm l 3 mm
h
w
1.27 mm
(a) Parallel plate waveguide model with a cell (b) Geometry of unit cell
(dB)
−30
−40
−50
S-parameter
−60
−70
−80
1.9 1.95 2 2.05 2.1 2.15 2.2
Frequency (GHz)
w = 6.6 mm w = 7.2 mm
w = 6.8 mm w = 7.4 mm
w = 7 mm
Figure 8: Parallel plate waveguide model with different sizes of unit cells.
Rx band are resonant. The perspective view of a microstrip It depends on the physical size of the unit cell However,
patch array antenna system for the indoor WCDMA repeater the bandwidth of each stop band shows very narrow
is shown in Figure 7. The proposed repeater antenna consists characteristics due to the resonant nature of the unit cell
of a server antenna, a donor antenna, and alumina housing. and a high dielectric constant of the substrate. In order to
It occupies a volume of 180 mm × 180 mm × 20 mm. cover the bandwidth within the WCDMA frequency band,
In order to reduce the surface waves radiation from a three different sizes of unit cell array configuration structure
server antenna and a donor antenna, corner-edged via (w = l = 6.6 mm, 7.0 mm, 7.4 mm) were used in this work.
mushroom-type unit cells are formed near the edges of the In addition, a server antenna and a donor antenna backed by
upper dielectric substrate. Figure 8 shows the parallel plate the alumina housings are arranged in orthogonal direction
waveguide model with different sizes of unit cells and the in order to get higher isolation between two antennas.
results of the simulated transmission characteristics as a
function of frequency. As shown in Figure 8(b), the unit cell
consists of two parallel rectangular plates with the same size 4. Experimental Result
(w = l) and a corner-edged via. The height of the via is
1.27 mm. Inside two rectangular plates, dielectric material The photographs of the fabricated two-by-two microstrip
(relative dielectric constant = 10.2, thickness = 1.27 mm) is patch array antenna structure are shown in Figure 9. The
placed. The transmission coefficient S21 of the parallel plate switchable feeder layer and a stacked microstrip patch layer
waveguide ports without cell exhibits near −43 dB, as shown are etched on a Rogers RO3210 substrate (relative dielectric
in Figure 8(c). When the unit cell is inserted between two constant = 10.2) having different thickness of 2.54 mm
parallel plate waveguides, a stop band occurs at a certain fre- and 1.27 mm, respectively. The characteristic impedance
quency. of each of the two branch feed lines from the coaxial
6 International Journal of Antennas and Propagation
−20 Figure 10. It is noted that the resonant frequencies are barely
changed.
−40 In the higher and lower bands, the measured −10 dB
return loss bandwidths are about 84 MHz and 96 MHz,
−50 respectively. It meets the bandwidth requirement for
WCDMA (1.92–2.17 GHz) applications. Compared to the
−60
fabricated antenna without isolated soft surface unit cells,
the fabricated antenna with isolated soft surface unit cells
1.5 2 2.5 exhibits higher isolation level. When the isolated soft surface
Frequency (GHz) unit cells are used, the maximum isolation level at the
frequency of 1.94 and 2.15 GHz is −92 dB and −70 dB,
S11 W/O cell S21 W/O cell
respectively. The measured far-field radiation patterns in
S11 with cell S21 with cell
the x-y plane (θ = 0◦ ) and y-z plane (φ = 0◦ ) at the
Figure 10: Measured return loss and isolation characteristics of frequency of 1.94 and 2.15 GHz are shown, respectively, in
two-by-two microstrip patch array antenna. Figure 11. It shows linear polarized radiation patterns, and
the main direction of the radiated power was changed due
International Journal of Antennas and Propagation 7
0 0
10 10
5 330 30 5 330 30
0 0
−5 −5
−10 300 60 −10 300 60
−15 −15
−20 −20
−25 −25
−30 270 90 −30 270 90
−25 −25
−20 −20
−15 −15
−10 120 −10 120
240 240
−5 −5
0 0
5 5
210 150 210 150
10 10
180 180
(a) x-y plane (1.94 GHz) (b) y-z plane (1.94 GHz)
0 0
10 10
330 30 330 30
5 5
0 0
−5
−5 300 300
60 −10 60
−10
−15 −15
−20 −20
−25 −25
−30 270 90 −30 270 90
−25 −25
−20 −20
−15
−15
−10
240 120 −10 240 120
−5
−5
0 0
5 5
210 150 210 150
10 10
180 180
Co-pol Co-pol
Cross-pol Cross-pol
Power sum Power sum
(c) x-y plane (2.15 GHz) (d) y-z plane (2.15 GHz)
to the placement of two patches. The main beam direction switchable feed network structure with MS/NRI coupled-
was slanted about −45◦ for the lower band (Tx) and +45◦ line coupler for higher isolation level and (2) the isolated
for the higher band (Rx). The measured peak gain/radiation soft surface unit cells structure for reducing the surface
efficiency for Tx/Rx band was 7.1 dBi/77% and 8.9 dBi/80%, waves. Both structures have been discussed in the paper
respectively. through proper numerical simulation. In order to improve
the isolation further, the server antenna and donor antenna
for an indoor repeater system were placed orthogonally.
5. Conclusion As a result, the fabricated server and donor antennas have
small separation of 20 mm and exhibit higher isolation level.
The new techniques for the isolation improvement of a Experimental results shows that the maximum isolation level
microstrip patch array antenna have been presented. The at the frequency of 1.94 and 2.15 GHz is −92 dB and −70 dB,
two main techniques presented here are (1) the single-feed respectively. The proposed techniques can be easily used for
8 International Journal of Antennas and Propagation
Acknowledgment
This research was supported by the Basic Science Research
Program through the National Research Foundation of Korea
(NRF) funded by the Ministry of Education, Science and
Technology (no. 2010-0011646).
References
[1] Y. J. Park, A. Herschlein, and W. Wiesbeck, “A photonic
bandgap (PBG) structure for guiding and suppressing surface
waves in millimeter-wave antennas,” IEEE Transactions on
Microwave Theory and Techniques, vol. 49, no. 10, pp. 1854–
1859, 2001.
[2] D. Sievenpiper, L. Zhang, R. F. J. Broas, N. G. Alexöpolous, and
E. Yablonovitch, “High-impedance electromagnetic surfaces
with a forbidden frequency band,” IEEE Transactions on
Microwave Theory and Techniques, vol. 47, no. 11, pp. 2059–
2074, 1999.
[3] P. S. Kildal and A. Kishk, “EM modeling of surfaces with stop or
go characteristics—artificial magnetic conductors and soft and
hard surfaces,” Applied Computational Electromagnetics Society
Journal, vol. 18, no. 1, pp. 32–40, 2003.
[4] P. S. Kildal, A. A. Kishk, and S. Maci, “Special issue on artificial
magnetic conductors, soft/hard surfaces, and other complex
surfaces,” IEEE Transactions on Antennas and Propagation, vol.
53, no. 1, part 1, pp. 2–7, 2005.
[5] H. M. Lee, “Pattern reconfigurable microstrip patch array
antenna using switchable feed-network,” in Proceedings of the
Asia-Pacific Microwave Conference (APMC ’10), pp. 2017–2020,
December 2010.
[6] H. M. Lee and J. K. Kim, “Front-to-back ratio improvement
of a microstrip patch antenna using an isolated soft surface
structure,” in Proceedings of the European Microwave Conference
(EuMC ’09), pp. 385–388, October 2009.
[7] J. H. Kim and H. M. Lee, “Backward wave reduction of a
microstrip patch antenna using dual-band isolated soft surface
structures,” in Proceedings of the IEEE International Symposium
on Antennas and Propagation Society (AP-S ’10), pp. 1–4, July
2010.
[8] R. Islam and G. V. Eleftheriades, “A planar metamaterial co-
directional coupler that couples power backwards,” in Proceed-
ings of the IEEE MTT-S International Microwave Symposium
Digest, vol. 1, pp. 321–324, June 2003.
Hindawi Publishing Corporation
International Journal of Antennas and Propagation
Volume 2012, Article ID 681431, 5 pages
doi:10.1155/2012/681431
Research Article
Series-Fed Microstrip Array Antenna with Circular Polarization
Tuan-Yung Han
Department of Computer and Communication Engineering, Chienkuo Technology University, Chang-Hua City 500, Taiwan
Copyright © 2012 Tuan-Yung Han. This is an open access article distributed under the Creative Commons Attribution License,
which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited.
This study proposes a novel 2 × 2 array antenna design with broadband and circularly-polarized (CP) operation. The proposed
design uses a simple series-fed network to increase the CP bandwidth without requiring one-by-one adjustment of each array
element or a complex feed network. Selecting the appropriate spacing between each array element allows the proposed array
antenna to generate CP radiation with a low axial ratio. Experimental results based on a prototype show that this 2 × 2 microstrip
array antenna achieves a wide 3 dB axial ratio bandwidth of more than 10%. Simulated data are also provided to confirm the
measured results.
1. Introduction than that for the CP array element type. This is because
the process of adjusting the CP performance of each array
Due to their attractive features, such as low profile, light element can be avoided. However, the resulting compound
weight, and ease of manufacturing using printed circuit tech- array gain is 3 dB lower than the array using the CP element.
niques, microstrip array antennas have come into high Sequential rotation of radiating elements can increase the
demand for satellite communication applications. To gen- input impedance bandwidth and polarization purity and
erate circularly-polarized (CP) radiation from a microstrip achieve a good symmetrical radiation pattern [6]. However, a
array antenna, previous research recommends a simple sequential rotation feed network must provide a delay line to
design method that uses a corporate-fed network [1] to excite allow various feed point with different phases to connect to
each array element simultaneously. To allow each array ele- each array element. This results in a relatively complicated
ment to exhibit equal power amplitude and phase distribu- circuit layout compared to the traditional corporate- or
tion, most corporate-fed networks employ transmission lines series-fed network. Several studies indicate that a corporate-
(of the same length) and power dividers. However, the CP fed array antenna using sequential rotation techniques [7, 8]
bandwidth of an array antenna design with a corporate- can improve the bandwidth of a corporate-fed array an-
fed network is usually limited to that of a single array ele- tenna and enhance the purity of polarization. However, dis-
ment. Thus, previous studies propose the method of apply- advantages such as a complex feed network still exist, creating
ing sequential rotation techniques to the feed network to the possibility of producing multiple reflections between the
improve the CP bandwidth of an array antenna effectively element and the feed.
[2–4]. Due to the sequentially rotated structure of the feed This study proposes a novel 2 × 2 CP microstrip array
network (with a single feed point), the phases of the four antenna design using series-fed lines. Experimental results
array elements (2 × 2 array) are usually orientated sequen- show that the proposed array design and one using sequen-
tially at 0◦ , 90◦ , 180◦ , and 270◦ . This offers an impedance tial rotation-fed network achieve similar levels of CP perfor-
bandwidth approximately three times wider than that of a mances, including CP bandwidth and peak gain. Since the
single array element. In this design, the designated array proposed array is fed by a simple series-fed network, there
element can be either linearly polarized (LP) [5] or CP type. is no need for an elaborate phase shifting or power dividing
If an LP array element is used, the design process of the array circuit. This study presents both simulated and measured re-
antenna (with sequential rotation technique) will be easier sults for the proposed design.
2 International Journal of Antennas and Propagation
Unit: mm
x
Circular patch
16
Ground plane
d
11 0◦ 90◦
10.5
Ring slot
13.5 1
3
80 150
3 11
80 150
(a) (b)
Figure 1: Geometry of the studied microstrip antennas. (a) Single ring-slot-coupled microstrip antenna. (b) 2 × 2 CP microstrip array
antenna.
2. Array Element Structure were generated using IE3D software. Experimental results
indicate that the CP operating bandwidth of the prototype
Figure 1(a) illustrates the geometry of a single aperture- (also defined as 3 dB axial ratio) is approximately 150 MHz
coupled microstrip antenna. Two orthogonal modes of the (5.1%) with respect to the center frequency measured at
antenna can be excited in series by a microstrip feed line 2950 MHz. Since the prototype is microstrip-fed, it can be
through the coupling of the annular-ring slot in the ground arranged in an array element manner.
plane. Choosing appropriate length (optimum at 11 mm) for
the open stub in the feed line allows the antenna to generate 3. New Array Design and Results
good CP radiation. A previous study analyzes the parameters
and design procedure of this aperture-coupled microstrip Figure 1(b) depicts the proposed CP microstrip array anten-
antenna [9]. A prototype was first fabricated according to na. Four ring-slot-coupled microstrip antenna elements were
the dimensions revealed in Figure 1(a). Figure 2 shows the excited through a series-fed network. Simulation results sug-
measured results along with the simulated results, which gest that the element spacing, d, is the dominant parameter
International Journal of Antennas and Propagation 3
0 10
9
5
8
10 7
15 6
5
20 4
25 3
2
30 1
35 0
2.6 2.7 2.8 2.9 3 3.1 3.2 3.3
40 Frequency (GHz)
2.6 2.7 2.8 2.9 3 3.1 3.2 3.3
Frequency (GHz) measured
simulated
(a) (b)
Figure 2: Measured and simulated results for the single CP ring-slot-coupled microstrip antenna. (a) Return loss. (b) Axial ratio.
90◦
180◦
Feed 0◦ Feed
point point
270◦
(a) (b)
Figure 3: Structure of the reference antennas. (a) Corporate feed network, Array B. (b) Sequential rotation feed networks, Array C.
to determine the CP performance of the proposed array, and totypes. Figure also presents simulated results for Array A,
the optimum value is approximately 73 mm for the studied indicating that they agree with the experimental results.
structure (d ∼ 0.76λ0 , where λ0 is referred to the center fre- These results confirm that all of the tested array prototypes
quency of the CP operating bandwidth). To confirm the have good impedance matching within their CP operating
simulated results, a prototype (Array A) was constructed bandwidths. In addition, the CP bandwidth (4.7%) of Array
based on the dimensions presented in Figure 1(b). Two other B is almost the same as that of its array element, and the
prototypes, Arrays B and C, using the corporate and sequen- CP bandwidth (13.8%) of Array C is about three times that
tial rotation feed networks, respectively, were also con- of Array B. As for Array A, the CP bandwidth centered at
structed as references (Figure 3). Except for different feeding 3110 MHz is approximately 11.5% which is slightly less than
networks, the three array prototypes were designed with the that of Array C.
same structure, antenna dimensions, and element spacing. Figure 5 plots the measured radiation patterns of Array
Due to the simplicity of series-fed network design, antenna A at 3110 MHz, revealing good left-hand CP (LHCP) radia-
engineers can ignore the process of adjusting the CP char- tions in the broadside direction. The main beam tilts slightly
acteristics of each array element. However, the complex feed (about 3 degrees) to the left side in the x-z plane. This might
network designs in sequential rotational-fed and corporate- be because the array elements are not fed with an exactly
fed array antennas necessitate tuning the CP performance of equal power level. Nevertheless, the measured peak gain of
each array element. Array A is approximately 12 dBi, which is only 0.3 dB lower
Figures 4(a) and 4(b) present the measured return loss than Array C. This small difference might be due to the feed-
and axial ratio against frequency for the respective array pro- ing phase errors of each element. Table 1 summarizes the ex-
4 International Journal of Antennas and Propagation
10
0
9
5
8
10 7
Return loss (dB)
Figure 4: Measured results for the three array prototypes; d = 73 mm. (a) Return loss. (b) Axial ratio.
20 20
Magnitude (dB)
Magnitude (dB)
10 10
0 0
−10 −10
−20 −20
−30 −30
−40 −40
180 150 120 90 60 30 0 −30 −60 −90 −120 −150 −180 180 150 120 90 60 30 0 −30 −60 −90 −120 −150 −180
Angle (deg) Angle (deg)
LHCP
RHCP
(a) (b)
Figure 5: Radiation patterns of Array A measured at 3110 MHz. (a) x-z plane. (b) y-z plane.
Table 1: Experimental results for the studied single element and 4. Conclusions
array prototypes.
This study presents a 2 × 2 circularly polarized microstrip
3 dB axial-ratio CP center Peak gain array antenna using a series-fed network. Experimental re-
bandwidth (MHz, %) frequency (MHz) (dBi) sults indicate that this array has a broad CP operating band-
Single element 150, 5.1 2950 7 width and an acceptable antenna gain. Moreover, the pro-
Array A 360, 11.5 3110 12 posed design is relatively simple compared to the traditional
Array B 140, 4.7 2980 12.4 array antenna, which uses a corporate or sequential rotation
Array C 410, 13.8 2965 12.3 feed network.
References
[1] P. S. Hall and C. M. Hall, “Coplanar corporate feed effects in
perimental results of CP performance for all three array pro- microstrip patch array design,” IEE Proceedings H, vol. 135, no.
totypes. These results show that the CP bandwidth of Array 3, pp. 180–186, 1988.
A is approximately 6.8% larger than that of Array B, and [2] J. W. Baik, K. J. Lee, W. S. Yoon, T. H. Lee, and Y. S. Kim,
2.3% smaller than that of Array C. Although array A demon- “Circularly polarised printed crossed dipole antennas with
strates a slightly narrower bandwidth than array C, its peak broadband axial ratio,” Electronics Letters, vol. 44, no. 13, pp.
785–786, 2008.
gain measured in the boresight direction is almost similar [3] M. Elhefnawy and W. Ismail, “A microstrip antenna array for
(compared to Arrays B and C) at approximately 12 dBi. Array indoor wireless dynamic environments,” IEEE Transactions on
A also possesses a very simple feed network structure that Antennas and Propagation, vol. 57, no. 12, pp. 3998–4002, 2009.
does not require other circuit features, such as phase shifting [4] R. Caso, A. Buffi, M. Rodriguez Pino, P. Nepa, and G.
or power dividing circuits. Thus, Array A is a preferred can- Manara, “A novel dual-feed slot-coupling feeding technique for
didate compared to Array C if a simple feed network struc- circularly polarized patch arrays,” IEEE Antennas and Wireless
ture is required. Propagation Letters, vol. 9, pp. 183–186, 2010.
International Journal of Antennas and Propagation 5
Research Article
Vertical Meandering Approach for Antenna Size Reduction
Copyright © 2012 Li Deng et al. This is an open access article distributed under the Creative Commons Attribution License, which
permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited.
A novel vertical meandering technique to reduce the lateral size of a planar printed antenna is presented. It is implemented by
dividing a conventional spiral patch into a different number of segments and placing them on different sides of the microwave
substrate with vias as the connections. To confirm the validity of this technique, measured electrical performance and radiation
characteristics of five antennas with different numbers of segments are compared. The smallest antenna is reduced in size by 84%
when compared with the conventional printed spiral antenna.
Via
Spiral-shaped
patch
w
w
a a
r2 r1 r1 y
Ground plane y r2 x
Ground plane
Feeding probe
x Feeding probe
L2
L2
b
b
Bottom side Bottom side
Top side Top side
(a) (b)
Bottom side
z
y
x
Feeding probe
Ground plane
(c)
Figure 1: (a) Top view of the conventional printed spiral antenna. (b) Top view of the novel printed spiral antenna. (c) 3D view of the novel
structure.
2. Antenna Structure the microwave substrate. Finally, vertical conducting vias are
used to connect the corners of the segments on the top
Figure 1(a) shows the geometry of a conventional printed and bottom layers together (as is well known, the vias are
spiral antenna. It consists of a small rectangular ground plane used widely in PCB for connections between layers, it is easy
etched on the bottom side of a F4B substrate, which has to fabricate). For maximization of the size reduction, the
thickness of 2 mm and dielectric constant of 2.65. A single vertical vias should be staggered arranged. Staggered vias at
spiral-shaped patch and a 50 Ω microstrip line are printed the cross-corners can make the distance of two adjacent vias
on the top side of the same substrate. This line is coaxial larger than center arranged vias. Thus, staggered arranged
fed by a 50 Ω SMA (SubMiniature version A) connector vias have smaller coupling capacitances than center arranged
underneath the ground plane. The feeding probe has radius vias; then the effect of cancellation to the inductances is
of 0.5 mm. In order to reduce the lateral size of this antenna, smaller than center arranged vias too. Therefore, for the
a novel technique is applied. This technique is implemented whole antenna with the same length, the staggered vias at
by separating the conventional spiral patch in Figure 1(a) the cross-corners can have larger inductances than center
into a different number of segments (denoted by N). Then, arranged vias, and larger inductance leads to larger size
half of these segments are moved to the bottom side of reduction. The proposed antenna structure is depicted in
International Journal of Antennas and Propagation 3
X-pol φ = 90
0 0
10 10
330 30 330 30
0 0
X-pol φ = 0 X-pol φ = 0
−10 −10
−40 −40
Co-pol φ = 90
270 90 270 90
Co-pol φ = 0 Co-pol φ = 90
Co-pol φ = 0
120 240 120
240
210 150
210 150
180 180
(a) (b)
0
0 10
10 330 30
330 30 0
0
−10
Co-pol φ = 90 −10 X-pol φ = 90
300 −20 60
300 −20 60
−30
−30
X-pol φ = 0
−40
−40
270 90
270 90
X-pol φ = 0
Co-pol φ = 0 X-pol φ = 90
Figure 5: (a) Measured radiation patterns with N = 1. (b) Measured radiation patterns with N = 16. (c) Measured radiation patterns with
N = 32. (d) Measured radiation patterns with N = 64.
spiral antennas with a different number of segments are Research and Development Program of China (863 Program,
constructed, tested, and analyzed. According to the results no. 2006AA04A106).
achieved, it is evident that the size of the antenna can be
significantly reduced by increasing its number of segments.
References
Acknowledgments [1] J. Rashed and C. T. Tai, “A new class of resonant antennas,”
IEEE Transactions on Antennas and Propagation, vol. 39, no. 9,
The authors wish to thank Dr. Ji Li and Weijun Hong for pp. 1428–1430, 1991.
their helpful comments on the work presented in this paper. [2] J. M. Kim and J. G. Yook, “Compact mender-type slot
This work was supported in part by Shenzhen Science and antennas,” in Proceedings of the IEEE Antennas and Propagation
Technology Planning Project for the Establishment of Key Society International Symposium, vol. 2, pp. 724–727, Boston,
Laboratory in 2009 (CXB 200903090021A) and Hi-Tech Mass, USA, July 2001.
International Journal of Antennas and Propagation 5
Research Article
Microstrip Patch Antenna Bandwidth Enhancement Using
AMC/EBG Structures
Copyright © 2012 R. C. Hadarig et al. This is an open access article distributed under the Creative Commons Attribution License,
which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited.
A microstrip patch antenna with bandwidth enhancement by means of artificial magnetic conductor (AMC)/electromagnetic
band-gap structure (EGB) is presented. The electrical characteristics of the embedded structure are evaluated using MoM
simulations. The manufactured prototypes are characterized in terms of return loss, gain, and radiation pattern measurements
in an anechoic chamber.
W p = 65.52 mm
35.44 mm
L p = 81.9 mm
26.15 mm
(a) (b) (c)
Figure 1: Manufactured prototypes: (a) Patch antenna, (b) Patch antenna-EBG, (c) Planar AMC.
the authors to operate at 2.48 GHz. Then, the patch anten- is represented in Figure 2. The AMC resonant frequency is
na is placed above the AMC. This combination will be hence- 2.48 GHz, and the AMC operation bandwidth is approxi-
forth referred as Patch antenna-AMC. Secondly, the AMC mately 130 MHz (5.24%) (see Figure 2(b)). The structure ex-
structure is modified to act as an EBG at a frequency close hibits several advantages such as uniplanar feature since nei-
to the patch antenna resonance frequency. Finally the EBG is ther multilayer substrate no via holes are required, simpli-
combined with the patch antenna on the same layer, resulting fying the implementation and reducing its costs.
in a design with a uniplanar feature and reduced cost. This
combination will be henceforth referred as patch antenna- 2.2. EBG Characterization. The periodic structure can be
EBG. Return loss, gain, and radiation patterns of the three characterized as EBG using the suspended strip method [21,
prototypes (all having the same dimensions) are analyzed 22] (See Figure 3). A suspended strip line over the 4 × 4 cell
based on measurements in an anechoic chamber. arrangement is used to test the transmission response of the
electromagnetic waves. The strip height is 0.02 λ. The struc-
2. Microstrip Antenna Design ture will block the transmission of power along the strip line
for frequencies within the band gap region and a notice-
The microstrip patch antenna is a narrow band design. In
able reduction in S21 can be observed at a certain frequency
this work, the patch antenna suitable for RFID applications
band. The band-gap of the EBG lattice is designed to
at 2.48 GHz is designed using ROGER3010 substrate with
be adjacent to the frequency band of the patch antenna,
a thickness of 1.27 mm, relative dielectric permittivity εr =
so that when integrating the two structures on the same
10.2, and loss tangent of 0.0023.The geometry of the patch
layer, their resonances couple each other, and, as a result, a
antenna with its dimensions is shown in Figure 1(a). The
wider bandwidth will be generated without disturbing other
characteristic impedance of the transmission line is 50 Ω.
characteristics of the patch antenna such as the radiation
The antenna design has been carried out by a set of method-
pattern. The dimension of the unit cell in the case of EBG
of-moments (MoM) simulations with commercial software
characterization is W2 = 16.38 mm. The simulated band gap
[19]. From Figure 4, it can be extracted that the simulated
of the EBG structure closer to the patch antenna bandwidth
operating bandwidth of the patch antenna is 20 MHz.
is 45 MHz around 2.5 GHz (see Figure 4).
2.1. AMC Characterization. An adaptation of the AMC
previously designed by the authors [20] is carried out shifting 3. Microstrip Patch Antenna Combined
the resonant frequency to 2.48 GHz. Based on the Bloch- with AMC/EBG
Floquet theory and on the finite element method (FEM), a
single cell of the lattice with periodic boundary conditions 3.1. Patch Antenna Placed above the AMC Structure. A 4 ×
(PBCs) on its four sides is simulated in order to obtain the 5 cells planar AMC structure is placed as patch antenna
frequency band where the periodic structure acts as an AMC. ground plane [23] (see Figure 5) in order to analyze if the
The phase of the reflection coefficient on the AMC sur- antenna’s bandwidth and the radiation properties can be im-
face is computed using a uniform incident plane wave (see proved. The antenna is fixed to the AMC structure (see
Figure 2(a)). Depending on the unit cell geometry together Figure 1(c)) by a 0.1 mm double-sided nonconducting ad-
with the substrate’s thickness and relative dielectric permit- hesive tape. The microstrip patch antenna bandwidth is
tivity, the resonant frequency and the bandwidth of the struc- 20 MHz whereas the AMC operation bandwidth is 130 MHz,
ture can be tuned. The unit cell dimensions are W × W = having each one the same resonance frequency, 2.48 GHz,
16.93 × 16.93 mm2 and its geometry exhibits four sym- and the same dimensions. However, for combining the two
metry planes. The simulated reflection phase of normally structures, the antenna’s ground plane has been removed and
incident plane wave on the AMC surface versus frequency is placed above the AMC. As a consequence the antenna’s
International Journal of Antennas and Propagation 3
Wave port
−90
Figure 2: AMC unit cell: (a) reflection phase simulation setup, (b) phase of the reflection coefficient on the AMC surface.
Metal 0
(cooper)
−5
−10
|Si j | (dB)
−15
−20
−25
−30
2.4 2.44 2.48 2.52 2.56 2.6 2.64
Suspended Frequency (GHz)
line
S11 patch antenna (MoM simulation)
S21 EBG-suspended line structure (MoM simulation)
Figure 3: Schematic of suspended line above EBG surface (top
view). Figure 4: Resonances to be coupled in order to achieve bandwidth
enhancement.
resonance frequency decreases due to capacitive effects for 3.2. Patch Antenna Surrounded by the EBG Structure. In
those frequencies within the AMC bandwidth. A resonance order to suppress the surface waves and to increase the band-
is obtained in the AMC bandwidth and outside this band width by means of coupled resonators effect, the EBG lattice
the antenna behaves as if its substrate thickness had doubled. is arranged around the patch, forming a uniplanar design
Merging both effects, the combined structure resonates in a [24]. As it has been already mentioned in Section 2.2, the
bandwidth wider than the microstrip patch antenna alone, resonance frequency of both structures (patch antenna and
but narrower than the AMC bandwidth. As disadvantage, EBG structure) is mutually influenced, and depending on the
the thickness of the combined structure is increased. If the frequency difference between them and the unit-cell arrange-
dielectric substrate thickness of the patch antenna doubles, ment around the patch antenna, the resulting resonance fre-
the resulting bandwidth (30 MHz) is narrower compared to quency changes. The frequencies included on the patch
the bandwidth of patch antenna-EBG and patch antenna- antenna’s bandwidth are adjacent to the ones included on
AMC prototypes. the lower band gap. The selected EBG arrangement with
4 International Journal of Antennas and Propagation
−5
−10
−15
|S11 | (dB)
ROGER3010 −20
−25
−30
AMC BW
−35
2.35 2.4 2.42 2.45 2.5 2.55
Frequency (GHz)
Patch antenna
Patch antenna-EBG
ROGER3010
Patch antenna-AMC
Figure 5: Patch antenna-AMC prototype layout. Figure 7: Measurement comparison between the prototypes: patch
antenna, patch antenna-EBG, and patch antenna-AMC.
0
−5 for the dielectric substrate, or even more likely due to manu-
−10 facturing tolerances.
In the case of placing the antenna above the AMC surface
−15 the antenna resonance frequency is shifted downwards to
|S11 | (dB)
−20 2.43 GHz (see Figure 7) due to the capacitive effects that are
generated between the two combined structures. Also, as the
−25
AMC structure has wider bandwidth than the patch anten-
−30 na, the resulting prototype bandwidth increases to 46 MHz,
−35 meaning a 100% broader bandwidth (see Figure 7).
When the patch antenna is surrounded by one row of
−40
2.4 2.42 2.44 2.46 2.48 2.5 2.52 2.54 EBG cells the bandwidth increases 50% (see Figure 7) due
Frequency (GHz) to the property of coupling the frequency bands of the two
structures composing the prototype. It is remarkable that
Patch antenna (measurement) this 50% bandwidth improvement is achieved increasing
Patch antenna (MoM simulation) neither the prototype size nor the thickness. The percentage
Patch antenna-EBG (measurement)
bandwidth comparison of the three prototypes is presented
Patch antenna-EBG (MoM simulation)
in Table 1.
Figure 6: Simulation and measurement comparison between the Measured radiation pattern cuts in the E and H planes
prototypes: patch antenna and patch antenna-EBG. of each manufactured prototype are presented in Figure 8.
The patch antenna prototype exhibits copolarization-cross-
polarization (CP-XP) ratio better than 25 dB (see Table 2),
whereas for the patch antenna-EBG prototype the (CP-XP)
respect to the antenna is a tradeoff between performance and ratio and the directivity are even increased. In measurements
size. The dimensions of the final structure (Figure 1(b)) are the gain of the patch antenna (4.59 dB) is preserved when
W p = 65.52 mm and L p = 81.90 mm. the antenna is surrounded by one row of EBG cells (Table 1).
From the simulation results, using EGB structures around
4. Results the patch antenna its radiation efficiency increases, due to
surface wave suppression property. However from measure-
Prototypes of the patch antenna, patch antenna placed above ment results it can be concluded that for this specific ar-
the AMC surface, and patch antenna surrounded by the EBG rangement, the radiation efficiency is preserved (while im-
cells have been manufactured using laser micromachining. proving bandwidth). The difference between simulations and
The return losses of each manufactured prototype have been measurements relies on the fact that the simulation method
measured. As it can be observed in Figure 6 the measured implemented by Momentum considers infinite dielectric
operating bandwidth of the patch antenna is 23 MHz. The under the finite EBG metallization but also the difference
difference in bandwidth between simulations (20 MHz) and could be attributable to misalignments in the anechoic
measurement (23 MHz) results could be due to the fact that chamber. Radiation pattern properties of the patch antenna-
the commercial MoM software considers infinite extension AMC prototype show a (CP-XP) ratio inferior to the other
International Journal of Antennas and Propagation 5
−10 dB
−10 dB
150 30 150 30
−20 dB
−20 dB
−30 dB
−30 dB
180 0 180 0
(a) (b)
Figure 8: Measured radiation patterns of the prototypes: Patch antenna, patch antenna-EBG, and patch antenna-AMC.
Prototype Bandwidth (MHz) Directivity (dB) Gain (dB) Radiation efficiency (%)
Meas. Meas. Sim. Meas. Sim. Meas. Sim.
Patch antenna 23 (0.93%) 7.33 5.95 4.59 4.29 53.21 68.23
Patch antenna-EBG 34 (1.37%) 7.50 6.84 4.61 5.56 51.40 74.47
Patch antenna-AMC 46 (1.90%) 6.72 8.52 0 0.79 21.28 16.86
two prototypes and a gain close to 0 dB. As the AMC does not Table 2: CP-XP ratio comparison.
have the ability to suppress the surface waves and the fact that
CP-XP ratio CP-XP ratio
the thicker the substrate, the stronger are the surface waves, Prototype
(E plane, dB) (H plane, dB)
the gain of the patch antenna-AMC prototype does not
improve. Also as the CP-XP ratio is worst for patch antenna- Patch antenna 25.81 25.04
AMC than for the other prototypes, part of the energy could Patch antenna-EBG 30.43 28.79
be radiated in other polarizations and backwards (as it is Patch antenna-AMC 13.85 9.43
shown in Figure 8). In order to improve the gain, a gap be-
tween the antenna and the AMC surface could be used but
this is technologically less advantageous. In addition, the
microstrip patch antenna’s gain and directivity can be in- microstrip patch antenna in order to characterize their joint
creased when more rows or/and columns surround the performance. The prototypes have been manufactured and
prototype, so a trade-off between performance and size must characterized based on measurements in an anechoic cham-
be taken (the higher the number of unit cells in a periodic ar- ber. From the measurements of the two resulting prototypes,
rangement, the closer its behavior to an infinite EBG struc- patch antenna-AMC and patch antenna-EBG, it can be
ture). concluded that both prototypes improve the bandwidth of
the patch antenna. Due to the surface wave effect of the EBG
5. Conclusions structure, the patch antenna-EBG prototype shows better
radiation properties increasing neither the prototype size
Bandwidth enhancement of microstrip patch antenna in the nor the thickness. All the prototypes presented are robust,
RFID SHF 2.48 GHz band has been presented. Two different compact and do not require via holes, being compatible with
structures (AMC/EBG) have been combined with the same planar fabrication technology.
6 International Journal of Antennas and Propagation
Research Article
High-Performance Computational Electromagnetic Methods
Applied to the Design of Patch Antenna with EBG Structure
Copyright © 2012 R. C. Hadarig et al. This is an open access article distributed under the Creative Commons Attribution License,
which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited.
In this contribution High-Performance Computing electromagnetic methods are applied to the design of a patch antenna
combined with EBG structure in order to obtain bandwidth enhancement. The electrical characteristics of the embedded structure
(patch antenna surrounded by EBG unit cells) are evaluated by means of method of moment technique (MoM) whereas for
designing the unit cell, the finite element method (FEM) together with the Bloch-Floquet theory is used. The manufactured
prototypes are characterized in terms of return loss and radiation pattern in an anechoic chamber.
65.52 mm Conductor w
ground plane h
w
Metal
(cooper)
35.44 mm
26.15 mm Unit
81.9 mm
cell
(a) (b)
−5
−10
|Si j | (dB)
−15
−20
−25
−30
2.4 2.44 2.48 2.52 2.56 2.6 2.64
Frequency (GHz)
−5
−10
|S11 | (dB)
−15
−20
−25
−30
−35
2.42 2.44 2.46 2.48 2.5 2.52 2.54 2.56
Frequency (GHz)
Figure 4: Simulation and measurement comparison between the patch antenna and the patch antenna-EBG prototypes.
one row EBG lattice has been carried out (Figure 1(b)). In frequency steps is 63 minutes in a 2-core Intel Xeon X5560
FEM, due to the air-box size when small frequencies are 48 GB RAM (equivalent to 16 threads execution) server.
involved (as in this case), the electric size of the problem to A prototype of the Patch antenna-EBG has been manu-
be solved is rather big. A proper mesh should take at least factured using laser micromachining. The return losses of the
10 (generally 20) tetrahedra per wavelength. Depending on manufactured prototype have been measured and compared
the prototype’s physical size, this could make the matrix of to those of the microstrip patch antenna (Figure 4). The
the linear equation system to become dense, which is not principle of achieving bandwidth enhancement is based
desirable in FEM and generally leads to longer computational on coupling the resonances of the structures involved. As
times and increased memory requirements. However the the patch antenna resonates at adjacent frequency band
matrix in MoM is dense, so this is not a problem, and thus compared to the band-gap of the EBG lattice, a signifi-
this could be a better choice in general. The disadvantage cant bandwidth enhancement of the prototype combining
of MoM is related to dielectric managements as they are the two structures (Patch antenna-EBG) is obtained. As
considered infinite sized. Using MoM the mesh has been shown in Figure 4 the resulting bandwidth (34 MHz) of
defined so that 20 cells per wavelength at 3 GHz are taken the Patch antenna-EBG is 50% wider than the microstrip
which leads to 2460 rectangular cells and 6246 triangular cells patch antenna’s bandwidth (23 MHz). Radiation pattern
with a matrix size of 12365. After applying mesh reduction measurements of the prototypes have been carried out
a matrix size of 5832 results. The simulation time for 81 in anechoic chamber to complete their characterization.
4 International Journal of Antennas and Propagation
120 60 120 60
−10 −10
−30 −30
180 0 180 0
Table 1: Comparison between the two prototypes. there are also other possible approaches, time domain based
such as (Finite-difference time-domain) FDTD which could
Prototype Bandwidth (MHz) Directivity (dB) Gain (dB)
also be used.
Patch antenna 23 7.33 4.59
There was reported a 50% increase of the initial band-
Patch antenna-EBG 34 7.50 4.61 width. The patch antenna-EBG prototype presented in
this paper has several advantages such as planar feature,
compact size, and low dielectric losses. Neither via holes
Radiation pattern cuts in the E and H planes of each nor multilayer substrates are required, simplifying practical
manufactured prototype are plotted in Figure 5. Using an implementation and reducing its cost.
EBG structure to surround the patch antenna, the directivity
increases due to the surface wave suppression (Table 1).
In the case of placing the patch antenna in a frequency Acknowledgments
range outside the band-gap of the EBG structure, MoM This work has been supported by the Ministerio de Ciencia
simulations show that the unit cells have no influence in the e Innovación of Spain/FEDER under projects TEC2008-
radiation properties or in the bandwidth. 01638/TEC (INVEMTA) and CONSOLIDER-INGENIO
CSD2008-00068 (TERASENSE), by the Gobierno del Prin-
5. Conclusions cipado de Asturias (PCTI)/FEDER-FSE under projects
EQUIP08-06, FC09-COF09-12, EQUIP10-31, and PC10-06
Bandwidth enhancement of microstrip patch antenna by (FLEXANT), by grant BP10-039, and by Cátedra Telefónica-
means of EBG structure for RFID SHF 2.48 GHz band Universidad de Oviedo.
has been presented. Using High-Performance computing
electromagnetic methods the electrical characteristics of the
resonant unit cell and the patch antenna have been evaluated References
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simulated and measured results are in good agreement due Antenna Design Handbook, Artech House, Boston, Mass, USA,
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FEM, can be used once the AMC is designed using FEM with and design of dual band high directivity EBG resonator
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[3] E. Rajo-Iglesias, L. Inclán-Sánchez, and O. Quevedo-Teruel, [19] M. E. de Cos, Y. Álvarez, R. C. Hadarig, and F. Las-Heras,
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Microwave Engineering Series, Cambridge, University Press, compact electromagnetic-bandgap (EBG) structure and its
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Hindawi Publishing Corporation
International Journal of Antennas and Propagation
Volume 2012, Article ID 595290, 6 pages
doi:10.1155/2012/595290
Application Article
A Wideband High-Gain Dual-Polarized Slot Array Patch Antenna
for WiMAX Applications in 5.8 GHz
Copyright © 2012 A. R. Dastkhosh and H. Dalili Oskouei. This is an open access article distributed under the Creative Commons
Attribution License, which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is
properly cited.
A low-cost, easy-to-fabricate, wideband and high-gain dual-polarized array antenna employing an innovative microstrip slot patch
antenna element is designed and fabricated. The design parameters of the antenna are optimized using commercial softwares
(Microwave Office and Zeland IE3D) to get the suitable S-parameters and radiation patterns. Finally, the simulation results are
compared to the experimental ones and a good agreement is demonstrated. The antenna has an approximately bandwidth of 14%
(5.15–5.9 GHz) which covers Worldwide Interoperability Microwave Access (WiMAX)/5.8. It also has the peak gain of 26 dBi for
both polarizations and high isolation between two ports over a wide bandwidth.
1. Introduction This antenna has high isolation between the two ports over
a wide bandwidth more than 14%. Furthermore, this high-
Recently, microstrip patch antennas are one of the most gain (25.5 dBi) array antenna has dual polarization with
commonly used antenna types due to many advantages such a minimum half-power beamwidth (HPBW) (vertical: 7◦ ;
as light weight, low fabrication costs, planar configuration, horizontal: 6◦ ). The impedance characteristics, radiation
and capability to integrate with microwave integrated cir- pattern, return loss, and isolation between two ports for the
cuits. Thus, the patch antennas are very suitable for vari- designed array are analyzed, simulated, and optimized using
ous applications such as wireless communication systems, Microwave Office and Zeland IE3D softwares. Also, S11 , S21 ,
cellular phones, satellite communication systems, and radar and radiation pattern are measured and compared to the
systems [1–6]. Due to their inherent features they are found simulated ones.
attractive for applications in broadband networks. WiMAX
is a standard-based wireless technology for broadband
networks providing high data rate communication by using 2. Configuration of Element Antenna
low-cost equipment. The access points in this network are
usually built with large physical spacing. Therefore, the Microstrip patch antennas can be excited by different types
high-gain antenna is necessary to execute the long distance of feeds. In order to achieve the desired performances of
transmission with a lower power. WiMAX has three allocated WiMAX antenna, an aperture coupled feed is used because of
frequency bands called low band (2.5 GHz to 2.8 GHz), its good characteristics such as wide operational bandwidth
middle band (3.2 GHz to 3.8 GHz), and high band (5.2 GHz and shielding of the radiation patches. Moreover, an aperture
to 5.8 GHz). In this work, the low-cost microstrip slot array coupled feed yields better gain and radiation pattern for
antenna (8 × 8) is designed, simulated, and fabricated for a dual-polarized antenna aimed for wireless applications
operation in the frequency band of 5.15 GHz to 5.9 GHz. [7–12]. An exploded view of the dual-polarized microstrip
In each antenna element, two rectangular slots are used for antenna and a simplified equivalent circuit model for an
coupling the microstrip feed lines to the radiating patch. aperture coupled microstrip antenna are shown in Figure 1.
2 International Journal of Antennas and Propagation
Ypatch
n1
Yap
(a) (b)
Circular patch Rohacell
hr
h
Spacer
Ground Dielectric
(c)
Figure 1: Configuration of the proposed dual-polarized aperture coupled circular patch antenna; (a) 3D view, (b) simplified equivalent
circuit model of an aperture coupled microstrip antenna, and (c) 2D view Rohacell: εr = 1.06, hr = 12 mm; substrate: εr f = 4.5, h f =
0.762 mm, h = 5.9 mm; vertical and horizontal apertures’ dimensions or feed slot (La × Wa ): 14 × 2 mm.
Rohacell
(radome)
Metal
plate
(a) (b)
λ0 /4
λ0 /4
50 Ω
70.7 Ω
ρN
RL
ρ2 ρN
50 Ω 70.7 Ω ρ1
50 Ω
Z0 ΓIN ρ0 Z1 Z2 ZN
λ/4
(c) (d)
Figure 2: (a) 3D view of 8 × 8 array antenna with its ground plane. Rohacell (bottom: circular patches): εr = 1.06, hr = 12 mm; substrate
(top: slots, bottom: feed lines): εr f = 4.5, h f = 0.762 mm. (b) Feed structure of array antenna. (c) Quarter-wave matching transformer.
(d) N-section λ/4 transformer.
International Journal of Antennas and Propagation 3
−5 9
−10
−15
8.5
Gain (dB)
−20
S (dB)
8
−25
−30
7.5
−35
−40 7
5 5.2 5.4 5.6 5.8 6 5 5.2 5.4 5.6 5.8 6
Frequency (GHz) Frequency (GHz)
S11 S12 Vertical
S21 S22 Horizontal
(a) (b)
−20 0
−5
−30
−10
−40
−15
S (dB)
S (dB)
−50 −20
−25
−60
−30
−70
−35
5 5.2 5.4 5.6 5.8 6
5 5.2 5.4 5.6 5.8 6
Frequency (GHz) Frequency (GHz)
S12 and S21 (Measurement) Measurement (S11) Simulation (S22)
S12 and S21 (Simulation) Measurement (S22) Simulation (S11)
(c) (d)
Figure 3: (a) Return loss and isolation versus frequency of one element of dual-polarized antenna element. (b) Gain versus frequency of one
element of dual polarized antenna element. (c) Isolation. (d) Return loss versus frequency of 8 × 8 array antenna.
Table 1: Wideband dual-polarized patch antenna specification. the bandwidth, and a radome. The input impedance of the
antenna at the center of the slot is given by [13, 14]
Frequency range 5.15–5.9 GHz
Peak gain 26 dBi n22
Zin = 2 − jZ0m cot βm Ls , (1)
Horizontal beamwidth 6◦ n1 Ypatch + Yap
Vertical beamwidth 7◦
where Ypatch is the patch admittance and Yap is the aperture
Front/back ratio Better than 28 dB admittance (Figure 1(b)). Z0m , βm , and Ls are the microstrip
Vertical: −11 dB (center frequency) line parameters in this equation. Also the coupling of the
SLL
Horizontal: −14 dB (center frequency) patch to the microstrip line is described by a transformer
Polarization (Dual) vertical or horizontal [14]. The dimensions of the element antenna such as slots,
VSWR 1.9 : 1 (max) feed lines, circular patch, and spaces between them are
Impedance 50 Ohms optimized with the use of IE3D to achieve best radiation
Mechanical Specification Length = width: 44 cm; depth: 4 cm characteristics, wide impedance bandwidth, and high iso-
lation between two ports. The optimized element antenna
has a circular patch with 11.89 mm radius positioned at
the bottom side of Rohacell. Furthermore, two 50 ohms
The antenna consists of only one substrate (Rogers TMM 4 microstrip feed lines (W = 1.5 mm, LV = 15 mm, and LH =
with dielectric constant εr = 4.5), an air layer for enhancing 23 mm) at the bottom side of the substrate (Rogers TMM 4
4 International Journal of Antennas and Propagation
25 33
32
20
31
15
30
10
29
5 28
5 5.2 5.4 5.6 5.8 6 5 5.2 5.4 5.6 5.8 6
Frequency (GHz) Frequency (GHz)
Vertical Vertical
Horizontal Horizontal
(a) (b)
Figure 4: (a) Simulated front-to-back ratio versus frequency of 8 × 8 array antenna without plate at the back of antenna. (b) Measured
front-to-back ratio versus frequency of 8 × 8 array antenna with metal plate at the back of antenna.
27 26
26 25
24
Gain (dB)
25
Gain (dB)
23
24
22
23
21
5 5.2 5.4 5.6 5.8 6
22
5 5.2 5.4 5.6 5.8 6 Frequency (GHz)
Frequency (GHz) Vertical
Horizontal
Vertical
Horizontal
(a) (b)
Figure 5: Gain versus frequency of 8 × 8 array antenna: (a) simulated and (b) measured.
0 0
−10 −10
−20 −20
−30 −30
−40 −40
−50 −50
−60 −60
−90 −50 0 50 90 −90 −50 0 50 90
θ θ
H plane H plane
E plane E plane
(a) (b)
Figure 6: Simulated antenna far-field radiation pattern at 5.5 GHz: (a) vertical and (b) horizontal.
0 0
−10 −10
Normalized gain (dB)
−20 −20
−30 −30
−40 −40
−50 −50
−60 −60
−90−80 −60 −40 −20 0 20 40 60 80 90 −90 −50 0 50 90
θ θ
H plane H plane
E plane E plane
(a) (b)
Figure 7: Measured antenna far-field radiation pattern at 5.5 GHz: (a) vertical and (b) horizontal.
transformer is usually another transmission line with the antenna size, its gain, beamwidth, side lobe level, and front-
desired characteristic impedance (Figure 2(d)). The spaces to-back ratio are summarized in Table 1.
between elements are set at 50 mm for better radiation
characteristics. The simulated and measured return loss (S11 ) 4. Conclusions
and isolation (S21 ) of 8 × 8 dual-polarized microstrip patch
slot array antenna are illustrated in Figures 3(c) and 3(d). This paper has reported the design of a low-cost high-gain
Furthermore, the metal plate at the back of array antenna dual-polarized patch array antenna for WiMAX applications
reduces the front-to-back ratio about −20 dB, as can be in the 5.15–5.9 GHz frequency band. The antenna has an
seen in Figure 4. Likewise, the gain of the array antenna in approximately bandwidth of 14% and the peak gain of 26 dBi
different frequencies is demonstrated in Figure 5. Moreover, for both polarizations. The design has been achieved with
the simulated and measured E and H plane far-field radiation the use of commercial software packages AWR Microwave
patterns of the array antenna at center frequency are shown Office and Zeland IE3D. The design process aimed at best
in Figures 6 and 7. Finally, all vital parameters such as return losses and fine quality radiation characteristics over
6 International Journal of Antennas and Propagation
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